19-2373; Rev 0; 4/02 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller The MAX1953/MAX1954/MAX1957 is a family of versatile, economical, synchronous current-mode, pulse-width modulation (PWM) buck controllers. These step-down controllers are targeted for applications where cost and size are critical. The MAX1953 operates at a fixed 1MHz switching frequency, thus significantly reducing external component size and cost. Additionally, excellent transient response is obtained using less output capacitance. The MAX1953 operates from low 3V to 5.5V input voltage and can supply up to 10A of output current. Selectable current limit is provided to tailor to the external MOSFETs’ on-resistance for optimum cost and performance. The output voltage is adjustable from 0.8V to 0.86VIN. With the MAX1954, the drain-voltage range on the highside FET is 3V to 13.2V and is independent of the supply voltage. It operates at a fixed 300kHz switching frequency and can be used to provide up to 25A of output current with high efficiency. The output voltage is adjustable from 0.8V to 0.86VHSD. The MAX1957 features a tracking output voltage range of 0.4V to 0.86VIN and is capable of sourcing or sinking current for applications such as DDR bus termination and PowerPC™/ASIC/DSP core supplies. The MAX1957 operates from a 3V to 5.5V input voltage and at a fixed 300kHz switching frequency. The MAX1953/MAX1954/MAX1957 provide a COMP pin that can be pulled low to shut down the converter in addition to providing compensation to the error amplifier. An input undervoltage lockout (ULVO) is provided to ensure proper operation under power-sag conditions to prevent the external power MOSFETs from overheating. Internal digital soft-start is included to reduce inrush current. The MAX1953/MAX1954/MAX1957 are available in tiny 10-pin µMAX packages. Applications Features ♦ Low-Cost Current-Mode Controllers ♦ Fixed-Frequency PWM ♦ MAX1953 1MHz Switching Frequency Small Component Size, Low Cost Adjustable Current Limit ♦ MAX1954 3V to 13.2V Input Voltage 25A Output Current Capability 93% Efficiency 300kHz Switching Frequency ♦ MAX1957 Tracking 0.4V to 0.86VIN Output Voltage Range Sinking and Sourcing Capability of 3A ♦ Shutdown Feature ♦ All N-Channel MOSFET Design for Low Cost ♦ No Current-Sense Resistor Needed ♦ Internal Digital† Soft-Start ♦ Thermal Overload Protection ♦ Small 10-Pin µMAX Package Ordering Information PART TEMP RANGE PIN-PACKAGE MAX1953EUB -40°C to +85°C 10 µMAX MAX1954EUB -40°C to +85°C 10 µMAX MAX1957EUB -40°C to +85°C 10 µMAX Pin Configurations TOP VIEW Printers and Scanners Graphic Cards and Video Cards PCs and Servers Microprocessor Core Supply Low-Voltage Distributed Power ILIM 1 COMP 10 BST 2 MAX1953EUB 9 LX FB 3 8 DH GND 4 7 PGND IN 5 6 DL Telecommunications and Networking µMAX †Patent Pending PowerPC is a trademark of Motorola, Inc. Pin Configurations continued at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX1953/MAX1954/MAX1957 General Description MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller ABSOLUTE MAXIMUM RATINGS IN, FB to GND...........................................................-0.3V to +6V LX to BST..................................................................-6V to +0.3V BST to GND ............................................................-0.3V to +20V DH to LX ....................................................-0.3V to (VBST + 0.3V) DL, COMP to GND.......................................-0.3V to (VIN + 0.3V) HSD, ILIM, REFIN to GND ........................................-0.3V to 14V PGND to GND .......................................................-0.3V to +0.3V IDH, IDL ................................................................±100mA (RMS) Continuous Power Dissipation (TA = +70°C) (derate 5.6mW/°C above +70°C) ..................................444mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = 5V, VBST - VLX = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER CONDITIONS Operating Input Voltage Range HSD Voltage Range MAX1954 only (Note 2) Quiescent Supply Current VFB = 1.5V, no switching MIN MAX UNITS 3.0 TYP 5.5 V 3.0 13.2 V 1 2 mA Standby Supply Current (MAX1953/ MAX1957) VIN = VBST = 5.5V, COMP = GND 220 350 µA Standby Supply Current (MAX1954) VIN = VBST = 5.5V, VHSD = 13.2V, COMP = GND 220 350 µA Undervoltage Lockout Trip Level Rising and falling VIN, 3% hysteresis 2.78 2.95 V 0.86 x VIN V Output Voltage Adjust Range (VOUT) 2.50 0.8 ERROR AMPLIFIER FB Regulation Voltage TA = 0°C to +85°C (MAX1953/MAX1954) 0.788 0.8 0.812 TA = -40°C to +85°C (MAX1953/MAX1954) 0.776 0.8 0.812 VREFIN - 8mV VREFIN VREFIN + 8mV 70 110 160 µS 5 500 nA 5 MAX1957 only Transconductance FB Input Leakage Current VFB = 0.9V REFIN Input Bias Current VREFIN = 0.8V, MAX1957 only FB Input Common-Mode Range 500 nA -0.1 1.5 V 1.5 V REFIN Input Common-Mode Range MAX1957 only -0.1 Current-Sense Amplifier Voltage Gain Low ILIM = GND (MAX1953 only) 5.67 6.3 6.93 V/V 3.15 3.5 3.85 V/V Current-Sense Amplifier Voltage Gain 2 V VILIM = VIN or ILIM = open (MAX1953 only) MAX1954/MAX1957 _______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller (VIN = 5V, VBST - VLX = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER ILIM Input Impedance Current-Limit Threshold MIN TYP MAX UNITS MAX1953 only CONDITIONS 50 125 200 kΩ VPGND - VLX, ILIM = GND (MAX1953 only) 85 105 125 VPGND - VLX, ILIM = open (MAX1953 only) 190 210 235 VPGND - VLX, ILIM = IN (MAX1953 only) 290 320 350 VPGND – VLX (MAX1954/MAX1957 only) 190 210 235 mV OSCILLATOR Switching Frequency Maximum Duty Cycle Minimum Duty Cycle MAX1953 0.8 1 1.2 MHz MAX1954/MAX1957 240 300 360 kHz Measured at DH 86 % 89 96 MAX1953, measured at DH 15 18 MAX1954/MAX1957, measured at DH 4.5 5.5 % SOFT-START MAX1953 Soft-Start Period 4 MAX1954/MAX1957 ms 3.4 FET DRIVERS DH On-Resistance, High State 2 3 Ω DH On-Resistance, Low State 1.5 3 Ω DL On-Resistance, High State 2 3 Ω DL On-Resistance, Low State 0.8 2 Ω LX, BST Leakage Current VBST = 10.5V, VLX = VIN = 5.5V, MAX1953/MAX1957 20 µA LX, BST, HSD Leakage Current VBST = 18.7V, VLX = 13.2V, VIN = 5.5V VHSD = 13.2V (MAX1954 only) 30 µA THERMAL PROTECTION Thermal Shutdown Rising temperature Thermal Shutdown Hysteresis 160 °C 15 °C SHUTDOWN CONTROL COMP Logic Level Low 3V < VIN < 5.5V COMP Logic Level High 3V < VIN < 5.5V COMP Pullup Current 0.25 V 100 µA 0.8 V Note 1: Specifications to -40°C are guaranteed by design and not production tested. Note 2: HSD and IN are externally connected for applications where VHSD < 5.5V. _______________________________________________________________________________________ 3 MAX1953/MAX1954/MAX1957 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (TA = +25°C, unless otherwise noted.) MAX1954 EFFICIENCY vs. LOAD CURRENT VOUT = 2.5V 90 80 EFFICIENCY (%) 85 VIN = 5V 75 70 65 60 80 VOUT = 1.7V 70 60 50 VOUT = 2.5V CIRCUIT OF FIGURE 1 55 50 1 0.1 0.1 1 70 60 VIN = 5V CIRCUIT OF FIGURE 3 40 10 1 0.1 LOAD CURRENT (A) MAX1953 OUTPUT VOLTAGE vs. LOAD CURRENT MAX1954 EFFICIENCY vs. LOAD CURRENT VOUT = 1.8V OUTPUT VOLTAGE (V) 90 85 80 75 70 65 MAX1953 toc05 2.60 MAX1953 toc04 100 EFFICIENCY (%) VOUT = 1.25V LOAD CURRENT (A) LOAD CURRENT (A) 95 80 50 VIN = 5V CIRCUIT OF FIGURE 2 40 10 90 EFFICIENCY (%) 90 100 MAX1953 toc02 VIN = 3.3V 95 100 MAX1953 toc01 100 MAX1957 EFFICIENCY vs. LOAD CURRENT MAX1953 toc03 MAX1953 EFFICIENCY vs. LOAD CURRENT EFFICIENCY (%) 2.55 VIN = 5V 2.50 VIN = 3.3V 2.45 60 VIN = 12V CIRCUIT OF FIGURE 4 55 50 0 5 10 15 CIRCUIT OF FIGURE 1 2.40 0 25 20 0.5 1.0 1.5 2.5 3.0 LOAD CURRENT (A) MAX1954 OUTPUT VOLTAGE vs. LOAD CURRENT MAX1954 OUTPUT VOLTAGE vs. LOAD CURRENT 1.75 OUTPUT VOLTAGE (V) 2.50 VHSD = VIN = 5V 2.45 MAX1953 toc07 1.80 MAX1953 toc06 2.55 1.70 VHSD = VIN = 5V 1.65 1.60 2.40 CIRCUIT OF FIGURE 2 CIRCUIT OF FIGURE 2 1.55 2.35 0 1 2 3 4 LOAD CURRENT (A) 4 2.0 LOAD CURRENT (A) 2.60 OUTPUT VOLTAGE (V) MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller 5 6 0 1 2 3 4 5 LOAD CURRENT (A) _______________________________________________________________________________________ 6 10 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller 1.25 VIN = 5V MAX1953 toc10 1.74 2.55 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 1.30 1.76 MAX1953 toc09 2.60 MAX1953 toc08 1.35 ILOAD = 3A 2.50 ILOAD = 0 1.72 ILOAD = 0 1.70 ILOAD = 5A 1.68 2.45 1.20 1.66 -1 0 1 2 1.64 2.40 3.0 3 3.5 4.0 4.5 5.0 MAX1954 OUTPUT VOLTAGE vs. INPUT VOLTAGE 2.51 4.5 5.0 5.5 MAX1957 OUTPUT VOLTAGE vs. INPUT VOLTAGE 1.29 MAX1953 toc11 2.52 4.0 INPUT VOLTAGE (V) INPUT VOLTAGE (V) LOAD CURRENT (A) 1.27 OUTPUT VOLTAGE (V) ILOAD = 0 2.50 2.49 3.5 3.0 5.5 ILOAD = 5A 2.48 MAX1953 toc12 -2 OUTPUT VOLTAGE (V) ILOAD = 0 1.25 1.23 ILOAD = 3A 1.21 2.47 CIRCUIT OF FIGURE 3 CIRCUIT OF FIGURE 2 2.46 1.19 3.5 4.0 4.5 5.0 3.0 3.5 4.0 4.5 5.0 INPUT VOLTAGE (V) MAX1953 FREQUENCY vs. INPUT VOLTAGE MAX1954/MAX1957 FREQUENCY vs. INPUT VOLTAGE VOUT = 2.5V 1.06 5.5 INPUT VOLTAGE (V) 320 310 FREQUENCY (kHz) 1.04 TA = -40°C 1.02 1.00 VOUT = 1.25V 315 5.5 MAX1953 toc14 3.0 MAX1953 toc13 -3 CIRCUIT OF FIGURE 2 CIRCUIT OF FIGURE 1 CIRCUIT OF FIGURE 3 1.15 FREQUENCY (MHz) OUTPUT VOLTAGE (V) MAX1954 OUTPUT VOLTAGE vs. INPUT VOLTAGE MAX1953 OUTPUT VOLTAGE vs. INPUT VOLTAGE MAX1957 OUTPUT VOLTAGE vs. LOAD CURRENT TA = -40°C 305 300 TA = +25°C 295 290 285 TA = +85°C 0.98 TA = +85°C 280 TA = +25°C 275 0.96 270 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) _______________________________________________________________________________________ 5 MAX1953/MAX1954/MAX1957 Typical Operating Characteristics (continued) (TA = +25°C, unless otherwise noted.) MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller Typical Operating Characteristics (continued) (TA = +25°C, unless otherwise noted.) MAX1954 LOAD TRANSIENT MAX1953 LOAD TRANSIENT MAX1953 toc16 MAX1953 toc15 VOUT AC-COUPLED 100mV/div VOUT AC-COUPLED 100mV/div 3A ILOAD 1.5A 5A 2.5A ILOAD CIRCUIT OF FIGURE 1 400µs/div 400µs/div MAX1953 NO-LOAD SWITCHING WAVEFORMS MAX1957 LOAD TRANSIENT MAX1953 toc18 MAX1953 toc17 VOUT AC-COUPLED 50mV/div 3A ILOAD ILX 2A/div LX 5V/div DL 5V/div DH 5V/div -3A 400µs/div 6 2µs/div _______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953 SHORT-CIRCUIT SWITCHING WAVEFORMS MAX1953 FULL-LOAD SWITCHING WAVEFORMS MAX1954/MAX1957 NO-LOAD SWITCHING WAVEFORMS MAX1953 toc20 MAX1953 toc19 ILX 2A/div LX 5V/div DL 5V/div DH 5V/div MAX1953 toc21 ILX 5A/div LX 5V/div DL 5V/div ILX 2A/div LX 10V/div DL 5V/div DH 10V/div 5V/div DH 2µs/div 2µs/div 4µs/div MAX1954/MAX1957 SHORT-CIRCUIT SWITCHING WAVEFORMS MAX1954/MAX1957 FULL-LOAD SWITCHING WAVEFORMS MAX1953 toc22 ILX MAX1953 toc23 2A/div ILX 5A/div LX 10V/div LX 10V/div DL 5V/div DL 5V/div DH 10V/div DH 10V/div 4µs/div 4µs/div _______________________________________________________________________________________ 7 MAX1953/MAX1954/MAX1957 Typical Operating Characteristics (continued) (TA = +25°C, unless otherwise noted.) Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957 Pin Description PIN MAX1953 8 MAX1954 MAX1957 NAME FUNCTION 1 — — ILIM ILIM Sets the Current-Limit Threshold for the Low-Side N-Channel MOSFET, as well as the Current-Sense Amplifier Gain. Connect to IN for 320mV, leave floating for 210mV, or connect to GND for 105mV current-limit threshold. — 1 — HSD HSD Senses the Voltage at the Drain of the High-Side N-Channel MOSFET. Connect to the high-side MOSFET drain using a Kelvin connection. — — 1 REFIN REFIN Sets the FB Regulation Voltage. Drive REFIN with the desired FB regulation voltage using an external resistor-divider. Bypass to GND with a 0.1µF capacitor. 2 2 2 COMP Compensation and Shutdown Control Pin. Connect an RC network to compensate control loop. Drive to GND to shut down the IC. 3 3 3 FB 4 4 4 GND Feedback Input. Regulates at VFB = 0.8V (MAX1953/MAX1954) or REFIN (MAX1957). Connect FB to a resistor-divider to set the output voltage (MAX1953/MAX1954). Connect to output through a decoupling resistor (MAX1957). Ground 5 5 5 IN Input Voltage (3V to 5.5V). Provides power for the IC. For the MAX1953/MAX1957, IN serves as the current-sense input for the highside MOSFET. Connect to the drain of the high-side MOSFET (MAX1953/MAX1957). Bypass IN to GND close to the IC with a 0.22µF (MAX1954) capacitor. Bypass IN to GND close to the IC with 10µF and 4.7µF in parallel (MAX1953/MAX1957) capacitors. Use ceramic capacitors. 6 6 6 DL Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET. Swings from PGND to VIN. 7 7 7 PGND 8 8 8 DH High-Side Gate-Drive Output. Drives the high-side MOSFET. DH is a floating driver output that swings from VLX to VBST. 9 9 9 LX Master Controller Current-Sense Input. Connect LX to the junction of the MOSFETs and inductor. LX is the reference point for the current limit. 10 10 10 BST Boost Capacitor Connection for High-Side Gate Driver. Connect a 0.1µF ceramic capacitor from BST to LX and a Schottky diode to IN. Power Ground. Connect to source of the synchronous rectifier close to the IC. _______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller IN THERMAL LIMIT MAX1953 MAX1954 MAX1957 UVLO HSD (MAX1954 ONLY) SLOPE COMPENSATION SHUTDOWN COMPARATOR 0.5V BST COMP ERROR AMPLIFIER FB DH PWM CONTROL CIRCUITRY CURRENTSENSE CIRCUITRY GND LX IN DL REFIN (MAX1957 ONLY) REFERENCE AND SOFT-START DAC PGND CLOCK SHORT-CIRCUIT CURRENT-LIMIT CIRCUITRY CURRENT-LIMIT COMPARATOR ILIM (MAX1953 ONLY) Typical Operating Circuit INPUT 3V TO 5.5V IN ILIM BST DH MAX1953 LX OUTPUT 0.8V TO 0.86VIN COMP DL PGND GND FB _______________________________________________________________________________________ 9 MAX1953/MAX1954/MAX1957 Functional Diagram MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller Detailed Description The MAX1953/MAX1954/MAX1957 are single-output, fixed-frequency, current-mode, step-down, PWM, DCDC converter controllers. The MAX1953 switches at 1MHz, allowing the use of small external components for small applications. Table 1 lists suggested components. The MAX1954 switches at 300kHz for higher efficiency and operates from a wider range of input voltages. Figure 1 is the MAX1953 typical application circuit. The MAX1953/MAX1954/MAX1957 are designed to drive a pair of external N-channel power MOSFETs in a synchronous buck topology to improve efficiency and cost compared with a P-channel power MOSFET topology. The on-resistance of the low-side MOSFET is used for short-circuit current-limit sensing, while the high-side MOSFET on-resistance is used for current-mode feedback and current-limit sensing, thus eliminating the need for current-sense resistors. The MAX1953 has three selectable short-circuit current-limit thresholds: 105mV, 210mV, and 320mV. The MAX1954 and MAX1957 have 210mV fixed short-circuit current-limit thresholds. The MAX1953/MAX1954/MAX1957 accept input voltages from 3V to 5.5V. The MAX1954 is configured with a high-side drain input (HSD) allowing an extended input voltage range of 3V to 13.2V that is independent of the input supply (Figure 2). The MAX1957 is tailored for tracking output voltage applications such as DDR bus termination supplies, referred to as VTT. It utilizes a resistor-divider network connected to REFIN to keep the 1/2 ratio tracking between VTT and VDDQ (Figure 3). The MAX1957 can source and sink up to 3A. Figure 4 shows the MAX1954 20A circuit. DC-DC Converter Control Architecture The MAX1953/MAX1954/MAX1957 step-down converters use a PWM, current-mode control scheme. An internal transconductance amplifier establishes an integrated error voltage. The heart of the PWM controller is an openloop comparator that compares the integrated voltagefeedback signal against the amplified current-sense signal plus the slope compensation ramp, which are summed into the main PWM comparator to preserve inner-loop stability and eliminate inductor staircasing. At each rising edge of the internal clock, the high-side MOSFET turns on until the PWM comparator trips or the maximum duty cycle is reached. During this on-time, current ramps up through the inductor, storing energy in a magnetic field and sourcing current to the output. The current-mode feedback system regulates the peak inductor current as a function of the output voltage error signal. The circuit acts as a switch-mode transconductance amplifier and pushes the output LC filter pole normally found in a voltage-mode PWM to a higher frequency. During the second half of the cycle, the high-side MOSFET turns off and the low-side MOSFET turns on. The inductor releases the stored energy as the current ramps down, providing current to the output. The output capacitor stores charge when the inductor current exceeds the required load current and discharges when the inductor current is lower, smoothing the voltage across the load. Under overload conditions, when the inductor current exceeds the selected current-limit (see the Current Limit Circuit section), the high-side MOSFET is not turned on at the rising clock edge and the low-side MOSFET remains on to let the inductor current ramp down. The MAX1953/MAX1954/MAX1957 operate in a forcedPWM mode. As a result, the controller maintains a constant switching frequency, regardless of load, to allow for easier postfiltering of the switching noise. Table 1. Suggested Components DESIGNATION 10 MAX1953 MAX1954 MAX1957 20A CIRCUIT C1 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.22µF, 10V X7R CER Kemet C0603C224M8RAC 3 x 22µF, 6.3V X5R CER Taiyo Yuden JMK316BJ226ML 0.22µF, 10V X7R CER Kemet C0603C224M8RAC C2 0.1µF, 50V X7R CER Taiyo Yuden UMK107BJ104KA 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.1µF, 50V X7R CER Taiyo Yuden UMK107BJ104KA 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG C3 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 0.1µF, 50V X7R CER Taiyo Yuden UMK107BJ104KA 270µF, 2V SP Polymer Panasonic EEFUEOD271R 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957 Table 1. Suggested Components (continued) DESIGNATION MAX1953 MAX1954 180µF, 4V SP Polymer Panasonic EEFUEOG181R MAX1957 20A CIRCUIT C4 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 270µF, 2V SP Polymer Panasonic EEFUEOD271R 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG C5 4.7µF, 6.3V X5R CER Taiyo Yuden JMK212BJ475MG — 270µF, 2V SP Polymer Panasonic EEFUEOD271R 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG C6 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG — 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG 10µF, 6.3V X5R CER Taiyo Yuden JMK212BJ106MG C7 — — 4.7µF, 6.3V X5R CER Taiyo Yuden JMK212BJ475MG 0.1µF, 50V X7R CER Taiyo Yuden UMK107BJ104KA C8 — — 0.1µF, 50V X7R CER Taiyo Yuden UMK107BJ104KA 270µF, 2V SP polymer Panasonic EEFUEOD271R C9-C13 — — C14 — — CC 270pF, 10V X7R CER Kemet C0402C271M8RAC Cf — — 270µF, 2V SP polymer Panasonic EEFUEOD271R 1500pF, 50V X7R CER Murata GRM39X7R152K50 — 1000pF, 10V X7R CER Kemet C0402C102M8RAC 470pF, 50V X7R CER Murata GRM39X7R471K50 560pF, 10V X7R CER Kemet C0402C561M8RAC 47pF, 10V C0G CER Kemet C0402C470K8GAC 68pF, 50V COG CER Murata GRM39COG680J50 15pF, 10V C0G CER Kemet C0402C150K8GAC D1 Schottky diode Central Semiconductor CMPSH1-4 Schottky diode Central Semiconductor CMPSH1-4 Schottky diode Central Semiconductor CMPSH1-4 Schottky diode Central Semiconductor CMPSH1-4 L1 1µH 3.6A Toko 817FY-1R0M 2.7µH 6.6A Coilcraft DO3316-272HC 2.7µH 6.6A Coilcraft DO3316-272HC 0.8µH 27.5A Sumida CEP125U-0R8 Dual MOSFET 20V 5A Fairchild FDS6898A Dual MOSFET 20V Fairchild FDS6890A Dual MOSFET 20V Fairchild FDS6898A N-channel 30V International Rectifier IRF7811W N1-N2 N3-N4 — — — N-channel 30V Siliconix Si4842DY R1 16.9kΩ 1% 9.09kΩ 1% 2kΩ 1% 10kΩ 1% R2 8.06kΩ 1% 8.06kΩ 1% 2kΩ 1% 8.06kΩ 1% R3 RC 10kΩ 5% 33kΩ 5% 62kΩ 5% 51.1kΩ 5% 270kΩ 5% ______________________________________________________________________________________ 11 MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller VIN 3V TO 5.5V C6 10µF C1 10µF C5 4.7µF IN ILIM D1 N1 BST DH RC 33kΩ MAX1953 L1 1µH C2 0.1µF LX COMP CC 270pF VOUT 2.5V AT 3A R1 16.9kΩ DL C3 10µF C4 10µF PGND GND R2 8.06Ω FB Figure 1. Typical Application Circuit for the MAX1953 VIN 3V TO 5.5V C2 10µF C1 0.22µF VHSD 5.5V TO 13.2V D1 IN HSD RC 62kΩ BST DH MAX1954 DL Cf 47pF C3 0.1µF L1 2.7µH VOUT 1.7V AT 3A LX COMP CC 1000pF N1 R1 9.09kΩ C4 180µF PGND GND FB R2 8.06kΩ Figure 2. Typical Application Circuit for the MAX1954 12 ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MAX1953/MAX1954/MAX1957 VIN 3V TO 5.5V C6 10µF C1 3 ✕ 22µF C7 4.7µF RC 51.1kΩ COMP VDDQ IN N1 BST Cf 68pF CC 470pF D1 DH R1 2kΩ MAX1957 L1 2.7µH C2 0.1µF LX REFIN DL C8 0.1µF R2 2kΩ VTT = 1/2 VDDQ C14 1500pF R3 10kΩ C3 270µF C4 270µF C5 270µF PGND GND FB Figure 3. Typical Application Circuit for the MAX1957 VHSD 10.8V TO 13.2V VIN 3V TO 5.5V C2 10µF C3 10µF C4 10µF C5 10µF C6 10µF D1 C1 0.22µF HSD IN RC 270kΩ BST DH MAX1954 LX COMP CC 560pF DL Cf 15pF N1 N2 L1 0.8µH C7 0.1µF N3 VOUT 1.8V AT 20A N4 R1 10kΩ C8 270µF C9 270µF C10 270µF C11 270µF C12 270µF C13 270µF PGND GND FB R2 8.06kΩ Figure 4. 20A Circuit ______________________________________________________________________________________ 13 MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller Current-Sense Amplifier Synchronous Rectifier Driver (DL) The MAX1953/MAX1954/MAX1957s’ current-sense circuit amplifies (AV = 3.5 typ) the current-sense voltage (the high-side MOSFET’s on-resistance (RDS(ON)) multiplied by the inductor current). This amplified currentsense signal and the internal-slope compensation signal are summed (VSUM) together and fed into the PWM comparator’s inverting input. The PWM comparator shuts off the high-side MOSFET when V SUM exceeds the integrated feedback voltage (VCOMP). Synchronous rectification reduces conduction losses in the rectifier by replacing the normal Schottky catch diode with a low-resistance MOSFET switch. The MAX1953/MAX1954/MAX1957 use the synchronous rectifier to ensure proper startup of the boost gatedriver circuit and to provide the current-limit signal. The DL low-side waveform is always the complement of the DH high-side drive waveform. A dead-time circuit monitors the DL output and prevents the high-side MOSFET from turning on until DL is fully off, thus preventing cross-conduction or shoot-through. In order for the dead-time circuit to work properly, there must be a lowresistance, low-inductance path from the DL driver to the MOSFET gate. Otherwise, the sense circuitry in the MAX1953/MAX1954/MAX1957 can interpret the MOSFET gate as OFF when gate charge actually remains. The dead time at the other edge (DH turning off) is determined through gate sensing as well. Current-Limit Circuit The current-limit circuit employs a lossless current-limiting algorithm that uses the low-side and high-side MOSFETs’ on-resistances as the sensing elements. The voltage across the high-side MOSFET is monitored for current-mode feedback, as well as current limit. This signal is amplified by the current-sense amplifier and is compared with a current-sense voltage. If the currentsense signal is larger than the set current-limit voltage, the high-side MOSFET turns off. Once the high-side MOSFET turns off, the low-side MOSFET is monitored for current limit. If the voltage across the low-side MOSFET (RDS(ON) ✕ IINDUCTOR) does not exceed the shortcircuit current limit, the high-side MOSFET turns on normally. In this condition, the output drops smoothly out of regulation. If the voltage across the low-side MOSFET exceeds the short-circuit current-limit threshold at the beginning of each new oscillator cycle, the MAX1953/MAX1954/MAX1957 do not turn on the highside MOSFET. In the case where the output is shorted, the low-side MOSFET is monitored for current limit. The low-side MOSFET is held on to let the current in the inductor ramp down. Once the voltage across the low-side MOSFET drops below the short-circuit current-limit threshold, the high-side MOSFET is pulsed. Under this condition, the frequency of the MAX1953/MAX1954/ MAX1957 appears to decrease because the on-time of the low-side MOSFET extends beyond a clock cycle. The actual peak output current is greater than the short-circuit current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the low-side MOSFET on-resistance, inductor value, input voltage, and output voltage. The short-circuit current-limit threshold is preset for the MAX1954/MAX1957 at 210mV. The MAX1953, however, has three options for the current-limit threshold: connect ILIM to IN for a 320mV threshold, connect ILIM to GND for 105mV, or leave floating for 210mV. 14 High-Side Gate-Drive Supply (BST) Gate-drive voltage for the high-side switch is generated by a flying capacitor boost circuit (Figure 5). The capacitor between BST and LX is charged from the VIN supply up to VIN, minus the diode drop while the lowside MOSFET is on. When the low-side MOSFET is switched off, the stored voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller then closes an internal switch between BST and DH to turn the high-side MOSFET on. Undervoltage Lockout If the supply voltage at IN drops below 2.75V, the MAX1953/MAX1954/MAX1957 assume that the supply voltage is too low to make valid decisions, so the UVLO circuitry inhibits switching and forces the DL and DH gate drivers low. After the voltage at IN rises above 2.8V, the controller goes into the startup sequence and resumes normal operation. Startup The MAX1953/MAX1954/MAX1957 start switching when the voltage at IN rises above the UVLO threshold. However, the controller is not enabled unless all four of the following conditions are met: • VIN exceeds the 2.8V UVLO threshold. • The internal reference voltage exceeds 92% of its nominal value (VREF > 1 V). • The internal bias circuitry powers up. • The thermal overload limit is not exceeded. ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller Setting the Output Voltage IN To set the output voltage for the MAX1953/MAX1954, connect FB to the center of an external resistor-divider connected between the output to GND (Figures 1 and 2). Select R2 between 8kΩ and 24kΩ, and then calculate R1 by: BST DH MAX1953 MAX1954 MAX1957 LX V R1 = R2 × OUT − 1 VFB DL where VFB = 0.8V. R1 and R2 should be placed as close to the IC as possible. Figure 5. DH Boost Circuit Once these conditions are met, the step-down controller enables soft-start and starts switching. The soft-start circuitry gradually ramps up to the feedback-regulation voltage in order to control the rate-of-rise of the output voltage and reduce input surge currents during startup. The soft-start period is 1024 clock cycles (1024/fS, MAX1954/MAX1957) or 4096 clock cycles (4096/f S, MAX1953) and the internal soft-start DAC ramps the voltage up in 64 steps. The output reaches regulation when soft-start is completed, regardless of output capacitance and load. Shutdown The MAX1953/MAX1954/MAX1957 feature a low-power shutdown mode. Use an open-collector transistor to pull COMP low to shut down the IC. During shutdown, the output is high impedance. Shutdown reduces the quiescent current (IQ) to approximately 220µA. Thermal Overload Protection Thermal overload protection limits total power dissipation in the MAX1953/MAX1954/MAX1957. When the junction temperature exceeds TJ = +160°C, an internal thermal sensor shuts down the device, allowing the IC to cool. The thermal sensor turns the IC on again after the junction temperature cools by 15°C, resulting in a pulsed output during continuous thermal overload conditions. For the MAX1957, connect FB directly to the output through a decoupling resistor of 10kΩ to 21kΩ (Figure 3). The output voltage is then equal to the voltage at REFIN. Again, this resistor should be placed as close to the IC as possible. Determining the Inductor Value There are several parameters that must be examined when determining which inductor is to be used. Input voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor current ripple to DC load current. A higher LIR value allows for a smaller inductor, but results in higher losses and higher output ripple. A good compromise between size, efficiency, and cost is an LIR of 30%. Once all of the parameters are chosen, the inductor value is determined as follows: L= ( VOUT × VIN − VOUT ( ) ) VIN × fS × ILOAD MAX × LIR where fS is the switching frequency. Choose a standard value close to the calculated value. The exact inductor value is not critical and can be adjusted in order to make trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but they also increase the output ripple and reduce the efficiency due to higher peak currents. By contrast, higher inductor values increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower AC current levels. For any area-restricted applications, find a low-core loss inductor having the lowest possible DC resistance. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 300kHz. ______________________________________________________________________________________ 15 MAX1953/MAX1954/MAX1957 Design Procedures MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller The chosen inductor’s saturation current rating must exceed the expected peak inductor current (IPEAK). Determine IPEAK as: LIR IPEAK = ILOAD(MAX ) + × ILOAD(MAX ) 2 Setting the Current Limit The MAX1953/MAX1954/MAX1957 use a lossless current-sense method for current limiting. The voltage drops across the MOSFETs created by their on-resistances are used to sense the inductor current. Calculate the current-limit threshold as follows: VCS = 0.8V A CS where ACS is the gain of the current-sense amplifier. ACS is 6.3 for the MAX1953 when ILIM is connected to GND and 3.5 for the MAX1954/MAX1957, and for the MAX1953 when ILIM is connected to IN or floating. The 0.8V is the usable dynamic range of COMP (VCOMP). Initially, the high-side MOSFET is monitored. Once the voltage drop across the high-side MOSFET exceeds VCS, the high-side MOSFET is turned off and the low-side MOSFET is turned on. The voltage across the low-side MOSFET is then monitored. If the voltage across the lowside MOSFET exceeds the short-circuit current limit, a short-circuit condition is determined and the low-side MOSFET is held on. Once the monitored voltage falls below the short-circuit current-limit threshold, the MAX1953/MAX1954/MAX1957 switch normally. The shortcircuit current-limit threshold is fixed at 210mV for the MAX1954/ MAX1957 and is selectable for the MAX1953. When selecting the high-side MOSFET, use the following method to verify that the MOSFET’s RDS(ON) is sufficiently low at the operating junction temperature (TJ): RDS(ON)N1 ≤ 0.8V A CS × IPEAK The voltage drop across the low-side MOSFET at the valley point and at ILOAD(MAX) is: LIR VVALLEY = RDS(ON) × (ILOAD(MAX) − × ILOAD(MAX ) ) 2 where RDS(ON) is the maximum value at the desired maximum operating junction temperature of the MOS- 16 FET. A good general rule is to allow 0.5% additional resistance for each °C of MOSFET junction temperature rise. The calculated VVALLEY must be less than VCS. For the MAX1953, connect ILIM to GND for a shortcircuit current-limit voltage of 105mV, to VIN for 320mV or leave ILIM floating for 210mV. MOSFET Selection The MAX1953/MAX1954/MAX1957 drive two external, logic-level, N-channel MOSFETs as the circuit switch elements. The key selection parameters are: • On-Resistance (RDS(ON)): The lower, the better. • Maximum Drain-to-Source Voltage (VDSS): Should be at least 20% higher than the input supply rail at the high side MOSFET’s drain. • Gate Charges (Qg, Qgd, Qgs): The lower, the better. For a 3.3V input application, choose a MOSFET with a rated RDS(ON) at VGS = 2.5V. For a 5V input application, choose the MOSFETs with rated RDS(ON) at VGS ≤ 4.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET (N1) that has conduction losses equal to switching loss at the nominal input voltage and output current. The selected low-side and highside MOSFETs (N2 and N1, respectively) must have RDS(ON) that satisfies the current-limit setting condition above. For N2, make sure that it does not spuriously turn on due to dV/dt caused by N1 turning on, as this would result in shoot-through current degrading the efficiency. MOSFETs with a lower Qgd/Qgs ratio have higher immunity to dV/dt. For proper thermal management design, the power dissipation must be calculated at the desired maximum operating junction temperature, T J(MAX). N1 and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore, major losses are the channel conduction loss (PN2CC) and the body diode conduction loss (PN2DC): USE RDS(ON)AT TJ(MAX) V PN2CC = (1 − OUT ) × I2LOAD × RDS(ON) VIN PN2DC = 2 × ILOAD × VF × tDT × fS where VF is the body diode forward-voltage drop, tdt is the dead time between N1 and N2 switching transitions, and fS is the switching frequency. ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller V PN1CC = OUT × I2 LOAD × RDS(ON) USE RDS(ON) AT TJ(MAX) V IN Q + QGD PN2SW = VIN × ILOAD × GS × fS IGATE ( ) where IGATE is the average DH driver output current capability determined by: IGATE ≅ 1 VIN × 2 RDH + RGATE where RDH is the high-side MOSFET driver’s on-resistance (3Ω max) and RGATE is the internal gate resistance of the MOSFET (~ 2Ω): PN1DR = QG × VGS × fS × RGATE RGATE + RDH where VGS ~ VIN. In addition to the losses above, allow about 20% more for additional losses due to MOSFET output capacitances and N2 body diode reverse recovery charge dissipated in N1 that exists, but is not well defined in the MOSFET data sheet. Refer to the MOSFET data sheet for the thermal-resistance specification to calculate the PC board area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. The minimum load current must exceed the high-side MOSFET’s maximum leakage current over temperature if fault conditions are expected. Input Capacitor The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit’s switching. The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = ILOAD × VOUT × (VIN − VOUT ) VIN I RMS has a maximum value when the input voltage equals twice the output voltage (VIN = 2 x VOUT), where IRMS(MAX) = ILOAD/2. Ceramic capacitors are recom- mended due to their low ESR and ESL at high frequency, with relatively low cost. Choose a capacitor that exhibits less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability. Output Capacitor The key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor’s ESR, and the voltage drop across the ESL caused by the current into and out of the capacitor: VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR, ESL, and output capacitance is: VRIPPLE(ESR) = IP−P × ESR VRIPPLE(C) IP−P 8 × COUT × fS V VRIPPLE(ESL) = IN ESL L V −V OUT × VOUT IP−P = IN fS × L VIN where IP-P is the peak-to-peak inductor current (see the Determining the Inductor Value section). These equations are suitable for initial capacitor selection, but final values should be chosen based on a prototype or evaluation circuit. As a general rule, a smaller current ripple results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output voltage ripple decreases with larger inductance, and increases with higher input voltages. Ceramic capacitors are recommended for the MAX1953 due to its 1MHz switching frequency. For the MAX1954/ MAX1957, using polymer, tantalum, or aluminum electrolytic capacitors is recommended. The aluminum electrolytic capacitor is the least expensive; however, it has higher ESR. To compensate for this, use a ceramic capacitor in parallel to reduce the switching ripple and noise. For reliable and safe operation, ensure that the capacitor’s voltage and ripple-current ratings exceed the calculated values. ______________________________________________________________________________________ 17 MAX1953/MAX1954/MAX1957 N1 operates as a duty-cycle control switch and has the following major losses: the channel conduction loss (PN1CC), the voltage and current overlapping switching loss (PN1SW), and the drive loss (PN1DR). MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller The MAX1953/MAX1954/MAX1957s’ response to a load transient depends on the selected output capacitors. In general, more low-ESR output capacitance results in better transient response. After a load transient, the output voltage instantly changes by ESR ✕ ∆ILOAD. Before the controller can respond, the output voltage deviates further, depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on its closed-loop bandwidth. With a higher bandwidth, the response time is faster, preventing the output voltage from further deviation from its regulating value. Compensation Design The MAX1953/MAX1954/MAX1957 use an internal transconductance error amplifier whose output compensates the control loop. The external inductor, highside MOSFET, output capacitor, compensation resistor, and compensation capacitors determine the loop stability. The inductor and output capacitors are chosen based on performance, size, and cost. Additionally, the compensation resistor and capacitors are selected to optimize control-loop stability. The component values shown in the Typical Application Circuits (Figures 1 through 4) yield stable operation over the given range of input-to-output voltages and load currents. The controller uses a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor. The MAX1953/ MAX1954/MAX1957 use the voltage across the highside MOSFET’s on-resistance (RDS(ON)) to sense the inductor current. Current-mode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor, resulting in a smaller phase shift and requiring less elaborate error-amplifier compensation. A simple single-series RC and CC is all that is needed to have a stable high bandwidth loop in applications where ceramic capacitors are used for output filtering. For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and ESR is lower than the desired close loop crossover frequency. Another compensation capacitor should be added to cancel this ESR zero. The basic regulator loop may be thought of as a power modulator, output feedback divider, and an error amplifier. The power modulator has DC gain set by gmc x RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its equivalent series resistance (RESR). 18 Below are equations that define the power modulator: GMOD = gmc × RLOAD × (fS × L) RLOAD + fS × L ( ) where RLOAD = VOUT/IOUT(MAX), and gmc = 1/(ACS ✕ RDS(ON)), where ACS is the gain of the current-sense amplifier and RDS(ON) is the on-resistance of the highside power MOSFET. ACS is 6.3 for the MAX1953 when ILIM is connected to GND, and 3.5 for the MAX1954/ MAX1957 and for the MAX1953 when ILIM is connected to VIN or floating. The frequencies at which the pole and zero due to the power modulator occur are determined as follows: fpMOD = fzMOD = 1 R LOAD × fS × L + RESR 2π × COUT × R f L + × LOAD S ( ( ) ) 1 2π × COUT × RESR The feedback voltage-divider used has a gain of GFB = VFB/VOUT, where VFB is equal to 0.8V. The transconductance error amplifier has DC gain, GEA(DC) = gm ✕ RO. RO is typically 10MΩ. A dominant pole is set by the compensation capacitor (C C ), the amplifier output resistance (RO), and the compensation resistor (RC). A zero is set by the compensation resistor (RC) and the compensation capacitor (CC). There is an optional pole set by Cf and RC to cancel the output capacitor ESR zero if it occurs before crossover frequency (fC): 1 2 π × CC × (RO + RC ) 1 fzEA = 2π × C C × R C 1 fpEA = 2π × C f × R C fpdEA = The crossover frequency (fC) should be much higher than the power modulator pole f pMOD . Also, the crossover frequency should be less than 1/5 the switching frequency: f fpMOD << fC < S 5 ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller MANUFACTURER COMPONENT PHONE Diode 631-435-1110 Central Semiconductor WEBSITE www.centralsemi.com Coilcraft Inductors 800-322-2645 www.coilcraft.com Fairchild MOSFETs 800-341-0392 www.fairchildsemi.com Kemet Capacitors 864-963-6300 www.kemet.com Panasonic Capacitors 714-373-7366 www.panasonic.com Taiyo Yuden Capacitors 408-573-4150 www.t-yuden.com Inductors 800-745-8656 www.toko.com Toko so the loop-gain equation at the crossover frequency is: GEA ( fC ) × GMOD( fC ) × VFB =1 VOUT Applications Information See Table 2 for suggested manufacturers of the components used with the MAX1953/MAX1954/MAX1957. PC Board Layout Guidelines For the case where fzESR is greater than fc: GEA ( fC ) = gmEA × RC and GMOD( fC ) = gmc × fpMOD RLOAD × (fs × L) × RLOAD + (fs × L) fC then RC is calculated as: RC = VOUT gmEA × VFB × GMOD( fC ) where gmEA = 110µS. The error amplifier compensation zero formed by RC and CC should be set at the modulator pole fpMOD. CC is calculated by: VOUT CC = IOUT(MAX) VOUT IOUT(MAX) × (fS × L) × + (fS × L) COUT RC As the load current decreases, the modulator pole also decreases. However, the modulator gain increases accordingly, and the crossover frequency remains the same. For the case where fzESR is less than fC, add another compensation capacitor Cf from COMP to GND to cancel the ESR zero at fzESR. Cf is calculated by: 1 Cf = 2π × RC × fzESR Figure 6 illustrates a numerical example that calculates RC and CC values for the typical application circuit of Figure 1 (MAX1953). Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout: 1) Place decoupling capacitors as close to IC pins as possible. Keep separate power ground plane (connected to pin 7) and signal ground plane (connected to pin 4). 2) Input and output capacitors are connected to the power ground plane; all other capacitors are connected to the signal ground plane. 3) Keep the high current paths as short as possible. 4) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for recommended copper area. 5) Ensure all feedback connections are short and direct. Place the feedback resistors as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP). 7) Place the high-side MOSFET as close as possible to the controller and connect IN (MAX1953/MAX1957) or HSD (MAX1954) and LX to the MOSFET. 8) Use very short, wide traces (50mils to 100mils wide if the MOSFET is 1in from the device). Chip Information TRANSISTOR COUNT: 2930 PROCESS: BiCMOS ______________________________________________________________________________________ 19 MAX1953/MAX1954/MAX1957 Table 2. Suggested Manufacturers MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller VOUT = 2.5V IOUT(MAX) = 3A COUT = 20µF L = 1µH RESR = 0.0025Ω gmEA = 110µS A VCS = 6.3A RDS(ON) = 0.013Ω 1 = 12.21S A VCS × RDS(ON) fS = 1MHz VOUT 2.5V RLOAD = = = 0.833Ω IOUT(MAX) 3A gmc = fpMOD = 1 R LOAD × fS × L 2π × COUT × R LOAD fS × L ( 1 fzESR = ( ) ) + RESR ) = 1 = 17.42kHz 0.833Ω × 1MHz × 1µH + 0.0025Ω 2π × 20µF × 0.833Ω + 1MHz × 1µH ( ) 1 = 3.2MHz 2π × COUT × RESR 2 π × 20µF × .0025Ω Pick the crossover frequency (fC ) at < 1/ 5 the switching frequency (fS ). We choose 100kHz < fzESR, so CF is not needed. The power modulator gain at fC is : GMOD(fC ) = gmc × = fpMOD 0.833Ω × (1MHz × 1µH) RLOAD × (fS × L) 17.42kHz × = 0.967 × = 12.21S × 0.833Ω + (1MHz × 1µH) 100kHz RLOAD (fS × L) fC then : RC = VOUT gmEA × VFB × GMOD(fC ) = 2.5V 110µS × 0.8V × .937 ≈ 33kΩ And : VOUT CC = IOUT(MAX) VOUT IOUT(MAX) × (fS × L) 2.5V × + (fS × L) COUT RC = 3A 2.5V 3A × (1MHz × 1µH) + (1MHz × 1µH) × 20µF 33kΩ ≈ 270pF Figure 6. Numerical Example to Calculate RC and CC Values of the Typical Operating Circuit of Figure 1 (MAX1953) 20 ______________________________________________________________________________________ Low-Cost, High-Frequency, Current-Mode PWM Buck Controller TOP VIEW HSD 1 COMP 2 MAX1954EUB 10 BST REFIN 1 9 COMP LX FB 3 8 DH GND 4 7 PGND IN 5 6 DL µMAX 10 BST 2 MAX1957EUB 9 LX FB 3 8 DH GND 4 7 PGND IN 5 6 DL µMAX ______________________________________________________________________________________ 21 MAX1953/MAX1954/MAX1957 Pin Configurations (continued) Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.) e 10LUMAX.EPS MAX1953/MAX1954/MAX1957 Low-Cost, High-Frequency, Current-Mode PWM Buck Controller 4X S 10 INCHES 10 H ÿ 0.50±0.1 0.6±0.1 1 1 0.6±0.1 BOTTOM VIEW TOP VIEW D2 MILLIMETERS MAX DIM MIN A 0.043 A1 0.006 0.002 A2 0.030 0.037 D1 0.116 0.120 D2 0.114 0.118 E1 0.116 0.120 E2 0.114 0.118 H 0.187 0.199 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S α 0∞ 6∞ MAX MIN 1.10 0.15 0.05 0.75 0.95 3.05 2.95 2.89 3.00 2.95 3.05 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0∞ 6∞ E2 GAGE PLANE A2 c A b D1 A1 α E1 L L1 FRONT VIEW SIDE VIEW PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE, 10L uMAX/uSOP APPROVAL DOCUMENT CONTROL NO. 21-0061 REV. I 1 1 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 22 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.