CS5132 CS5132 Dual Output CPU Buck Controller Description Features CS5132 dual output provides the industry’s most highly integrated solution, minimizing external component count, total solution size, and cost. The CS5132 is specifically designed to power Intel’s Pentium® II processor and includes the following features: 5 bit DAC and fixed 1.23V reference, Power-Good output, hiccup mode overcurrent protection, adaptive voltage positioning, and overvoltage protection. The CS5132 will operate over an 8.4V to 14V range and is available in 24 lead surface mount package. The CS5132 is a dual output CPU power supply controller. It contains a synchronous dual NFET buck controller utilizing the V2TM control method to achieve the fastest possible transient response and best overall regulation. The CS5132 also contains a second nonsynchronous NFET buck controller. These synchronous and non-synchronous buck regulators are designed to power the core and I/O logic of the latest high performance CPUs. The CS5132 incorporates many additional features required to ensure the proper operation and protection of the CPU and power system. The Application Diagram C6-C11 1200µF x 6 10V C3-C5 20 1200µF x 3 10V 23 FS70VSJ-03 16 VID2 GATE(H) Q3 PCB TRACE (FreeCurrent Sensing Element) FS70VSJ-03 L2 VID3 3 17 Q2 COMP2 6.6mΩ C12 0.1µF C22 100pF GATE 51Ω R1 510Ω 10K VFFB2 12 MBRD835L C18-C21 11 D1 1200µF 10V x4 14 PWRGD 0.1µF LGnd 9 COFF1 5 4 PGnd C15 680pF C14 0.1µF 5 bit DAC with 1% Tolerance ■ Hiccup Mode Overcurrent Protection ■ 65ns adaptive FET Non-Overlap Time ■ Non-Synchronous Switching Regulator Controller (VI/O) ■ Single N-Channel MOSFET buck design ■ Adjustable Output with 2% Tolerance ■ System Power Management Power-Good Output Monitors VCORE Switching Regulator Output 0.01µF 100Ω R10 2K 1% VID2 1 24 VID1 VID3 2 23 VID0 VID4 3 22 PWRGD COFF1 4 21 OVP COMP1 5 20 VCC1 VOUT1 6 19 GATEL VFB1 7 18 PGND VFFB1 8 17 GATEH LGND 9 16 VCC2 VFFB2 10 15 GATE VFB2 11 14 COFF2 VOUT2 12 13 COMP2 R9 18 R3 R4 1.18K 1% 200ns Transient Loop Response ■ Control Topology 24L SO Wide 21 OVP ■ TM C13 0.1µF 22 COMP1 7 VFB1 R6 510Ω COFF2 V2 8 VOUT1 VFB2 ■ Package Options R8 6 VOUT2 C17 R2 510Ω VFFB1 Dual N-Channel MOSFET Synchronous Buck Design R5 510Ω R7 10 3.5µH ■ OVP Signal Monitors VCORE Switching Regulator Output FS70VSJ-03 GATE(L) VID4 15 VCC(CORE) 1200µF x 8 10V 13 2 C23-C30 1.2µH 19 1 3.3mΩ L1 VID0 VCC1 VCC2 24 VID1 +3.3V (VI/O) PCB TRACE (Free Current Sensing Element) Q1 C1 1µF C2 1µF +5V Synchronous Switching Regulator Controller (VCORE) Pentium® II System VCORE and VI/O Controlled by a Single IC +5V +12V +12V ■ 10K C25 C16 390pF 5V/12V to 2V/16A for Pentium®II VCC(CORE) , 5V/12V to 3.3V/8A for VI/O V2 is a trademark of Switch Power, Inc. Pentium is a registered trademark of Intel Corporation. ON Semiconductor 2000 South County Trail, East Greenwich, RI 02818 Tel: (401)885–3600 Fax: (401)885–5786 N. American Technical Support: 800-282-9855 Web Site: www.cherry–semi.com June, 1999 - Rev. 2 1 CS5132 Absolute Maximum Ratings Pin Symbol VMAX Pin Name VMIN ISOURCE ISINK VCC1 IC Logic and Low Side Driver Power Input 16V -0.3V N/A VCC2 IC High Side Drivers Power Input 16V -0.3V N/A COMP1, COMP2 Compensation Pins for the VCORE and VI/O error amplifiers. 6V -0.3V 1mA 1.5A Peak 200mA DC 3A Peak 400mA DC 5mA 6V -0.3V 1mA 1mA VFB1, VOUT1, VID0-4, VCORE Voltage Feedback Input Pin, VOUT2, VFB2, VFFB1, VCORE Output Voltage Sense Pin, VFFB2 Voltage ID DAC Input Pins, VI/O Output Voltage Sense Pin, VI/O Voltage Feedback Input Pin, VCORE PWM comparator Fast Feedback Pin, VI/O PWM comparator Fast Feedback Pin. COFF1, COFF2 Off-Time Pins for the VCORE and VI/O regulators 6V -0.3V 1mA 50mA GATE(H), GATE High-Side FET Drivers for the VCORE and VI/O regulators. 16V -0.3V 1.5A Peak 200mA DC 1.5A Peak 200mA DC GATE(L) Low-Side FET Driver 16V -0.3V PWRGD Power-Good Output 6V -0.3V 1.5A Peak 200mA DC 1mA 1.5A Peak 200mA DC 30mA OVP Overvoltage Protection 15V -0.3V 30mA 1mA PGnd Power Ground 0V 0V 3A Peak 400mA DC N/A LGnd Logic Ground 0V 0V 40mA N/A Operating Junction Temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 125°C Lead Temperature Soldering: Reflow (SMD styles only) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60 sec max. above 183°C, 230°C Peak Storage Temperature Range, TS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -65 to 150°C ESD Susceptibility . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Package Pin Description PACKAGE PIN # PIN SYMBOL 23,24,1,2,3 VIDO – VID4 20 VCC1 17 18 19 16 GATE(H) PGnd GATE(L) VCC2 15 21 GATE OVP 22 PWRGD FUNCTION Voltage ID DAC inputs. These pins are internally pulled up to 5.65V if left open. VID4 selects the DAC range. When VID4 is high (logic one), the Error Amp reference range is 2.125V to 3.525V with 100mV increments. When VID4 is low (logic zero), the Error amp reference voltage is 1.325V to 2.075V with 50mV increments. Input power supply pin for the internal circuitry, and low side gate driver. Decouple with filter capacitor to PGnd. High side switch FET driver pin for VCORE section. Power ground for VCORE and VI/O section. Low side synchronous FET driver pin. Input power supply pin for on-board high side gate drivers. Decouple with filter capacitor to PGnd. High side switch FET driver pin for VI/O section. Overvoltage protection pin. Goes high when overvoltage condition is detected on VFB1. Power-Good Output. Open collector output drives low when VFB1 is out of regulation. 2 CS5132 Package Pin Description: continued PACKAGE PIN # PIN SYMBOL 14 COFF2 13 COMP2 12 11 10 VOUT2 VFB2 VFFB2 9 7 LGnd VFB1 6 5 VOUT1 COMP1 4 COFF1 8 VFFB1 FUNCTION Off-Time Capacitor Pin. A capacitor from this pin to LGnd sets the off time for the non-synchronous regulator (VI/O). VI/O section error amp output. PWM comparator inverting input. A capacitor to LGnd provides error amp compensation. VI/O section current limit comparator inverting input. VI/O section error amp inverting feedback input. VI/O PWM comparator fast feedback non-inverting input. VI/O section current limit comparator non-inverting input. Logic ground. VCORE section error amp inverting input, PWRGD and OVP comparator input. VCORE section current limit comparator inverting input. VCORE section error amp output. VCORE section PWM comparator inverting input. A capacitor to LGnd provides error amp compensation. Off-Time Capacitor Pin. A capacitor from this pin to LGnd sets the off time for the synchronous regulator (VCORE). VCORE section PWM comparator fast feedback non-inverting input. VCORE section current limit comparator non-inverting input. Block Diagram VFFB1 COMP1 COFF1 - + 1.06V VFB1 PWM COMP1 + + + EA1 Current Limit1 - PGnd Non-overlap Logic Q Fault Latch1 + - 0.25V GATE(H) Off Time1 One Shot Discharge Comparator + - VID0 VID1 VID2 VID3 VID4 UVLO R - + VCC2 - 86mV VOUT1 VCC1 GATE(L) S DAC PGnd COFF2 - PWM COMP2 GATE + + Off Time2 One Shot 1.10V - PGnd + - VFFB2 VFB2 + + EA2 + - 0.25V + Discharge Comparator 86mV Q + - 1.23V PWRGD - R Fault Latch2 VCC1 Current Limit2 S OVP COMP2 LGnd 3 PGnd + + - VOUT2 CS5132 Electrical Characteristics: 0°C < TA < 70°C; 0°C < TJ < 125°C; VOUT2 ≤ 3.5V, 9V ≤ VCC1 ≤ 14V, 9V ≤ VCC2 ≤ 14V; 2.0V DAC Code (VID4= VID3 = VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = CGATE = 3.3nF, COFF = 390pF; Unless otherwise stated. PARAMETER TEST CONDITIONS ■ VCORE Switching Regulator Error Amplifier VFB1 Bias Current VFB1 = 0V COMP1 Source Current COMP1 = 1.2V to 3.6V; VFB1 = 1.9 V COMP1 Sink Current COMP1=1.2V; VFB1 =2.1V; Open Loop Gain CCOMP1 = 0.1µF Unity Gain Bandwidth CCOMP1 = 0.1µF PSRR @ 1kHz CCOMP1 = 0.1µF ■ Voltage Identification DAC Accuracy (all codes) VID4 VID3 VID2 VID1 VID0 1 0 0 0 0 1 0 0 0 1 1 0 0 1 0 1 0 0 1 1 1 0 1 0 0 1 0 1 0 1 1 0 1 1 0 1 0 1 1 1 1 1 0 0 0 1 1 0 0 1 1 1 0 1 0 1 1 0 1 1 1 1 1 0 0 1 1 1 0 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 1 1 0 0 1 0 0 0 0 1 0 1 0 0 1 1 0 0 0 1 1 1 0 1 0 0 0 0 1 0 0 1 0 1 0 1 0 0 1 0 1 1 0 1 1 0 0 0 1 1 0 1 0 1 1 1 0 0 1 1 1 1 1 1 1 1 1 Line Regulation Input Threshold Measure VFB1 = COMP1, 25°C ≤ TJ≤ 125°C, VCC1 = VCC2 = 12V MIN TYP MAX UNIT -1.0 15 30 0.1 30 60 80 20 70 1.0 60 120 µA µA µA dB kHz dB -1.0 3.489 3.390 3.291 3.192 3.093 2.994 2.895 2.796 2.697 2.598 2.499 2.400 2.301 2.202 2.103 2.054 2.004 1.955 1.905 1.856 1.806 1.757 1.707 1.658 1.608 1.559 1.509 1.460 1.410 1.361 1.311 1.225 9V ≤ VCC1 ≤ 14V VID4, VID3, VID2, VID1, VID0 4 1.00 1.0 3.525 3.425 3.325 3.225 3.125 3.025 2.925 2.825 2.725 2.625 2.525 2.425 2.325 2.225 2.125 2.075 2.025 1.975 1.925 1.875 1.825 1.775 1.725 1.675 1.625 1.575 1.525 1.475 1.425 1.375 1.325 1.250 0.01 1.25 3.560 3.459 3.358 3.257 3.156 3.055 2.954 2.853 2.752 2.651 2.550 2.449 2.348 2.247 2.146 2.096 2.045 1.995 1.944 1.894 1.843 1.793 1.742 1.692 1.641 1.591 1.540 1.490 1.439 1.389 1.338 1.275 2.40 % V V V V V V V V V V V V V V V V V V V V V V V V V V V V V V V V %/V V PARAMETER Input Pull-up Resistance Pull-up Voltage ■ GATE(H) and GATE(L) High Voltage at 100mA Low Voltage at 100mA Rise Time Fall Time GATE(H) to GATE(L) Delay GATE(L) to GATE(H) Delay GATE pull-down TEST CONDITIONS VID4, VID3, VID2, VID1, VID0 MIN 25 5.48 ■ PWM Comparator 1 PWM Comparator Offset Voltage 0V ≤ VFFB1 ≤ 3.5V Transient Response VFFB1 = 0 to 3.5V VFFB1 Bias Current 0.2V ≤ VFFB1 ≤ 3.5V VCOFF1 = 1.5V VCOFF1 = 1.5V ■ Power-Good Output PWRGD Sink Current PWRGD Upper Threshold PWRGD Lower Threshold PWRGD Output Low Voltage VFB1 = 1.7V, VPWRGD = 5V % of nominal DAC code % of nominal DAC code VFB1 = 1.7V, IPWRGD = 500µA UNIT 100 5.82 kΩ V 1.2 1.0 40 40 65 65 50 2.1 1.5 80 80 100 100 115 V V ns ns ns ns kΩ 77 0.2 -7.0 100 86 0.25 0.1 800 101 0.3 7.0 2500 mV V µA µA 0.95 1.06 200 0.1 1.18 300 7.0 V ns µA 1.0 1.6 550 25 2.3 µs µA mA 0.5 5 -12 4 8.5 -8.5 0.2 15 12 -5 0.3 mA % % V -7.0 ■ COFF1 Off-Time Charge Current Discharge Current MAX 50 5.65 Measure VCC1/2 –GATE(L)/(H) Measure GATE(L)/(H) 1.6V < GATE(H)/(L) < (VCC1/2 – 2.5V) (VCC1/2 – 2.5V) > GATE(L)/(H) > 1.6V GATE(H)<2V, GATE(L)>2V 30 GATE(L)<2V, GATE(H)>2V 30 Resistance to PGnd (Note 1) 20 ■ VCORE Overcurrent Protection OVC Comparator Offset Voltage 0V < VOUT1 ≤ 3.5V Discharge Threshold Voltage VOUT1 Bias Current 0.2V ≤ VOUT1 ≤ 3.5V OVC Latch Discharge Current VCOMP = 1V TYP ■ Overvoltage Protection (OVP) Output OVP Source Current OVP = 1V OVP Threshold % of nominal DAC code OVP Pull-up Voltage IOVP = 1mA, VCC1 - VOVP 1 5 10 8.5 1.1 25 12 1.5 mA % V ■ VI/O Switching Regulator Error Amplifier VFB2 Bias Current VFB2 = 0V COMP2 Source Current COMP2 = 1.2V to 3.6V; VFB2 = 1V COMP2 Sink Current COMP2=1.2V; VFB2=1.4V; Open Loop Gain CCOMP2 = 0.1µF -1.0 15 30 0.1 30 60 80 1.0 60 120 µA µA µA dB 5 CS5132 Electrical Characteristics: 0°C < TA < 70°C; 0°C < TJ < 125°C; VOUT2 ≤ 3.5V, 9V ≤ VCC1 ≤ 14V, 9V ≤ VCC2 ≤ 14V; 2.0V DAC Code (VID4= VID3 = VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = CGATE = 3.3nF, COFF = 390pF; Unless otherwise stated. CS5132 Electrical Characteristics: 0°C < TA < 70°C; 0°C < TJ < 125°C; VOUT2 3.5V, 9V VCC1 14V, 9V VCC2 14V; 2.0V DAC Code (VID4= VID3 = VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = CGATE = 3.3nF, COFF = 390pF; Unless otherwise stated. PARAMETER TEST CONDITIONS ■ VI/O Switching Regulator Error Amplifier continued Unity Gain Bandwidth CCOMP2 = 0.1µF PSRR @ 1kHZ CCOMP2 = 0.1µF Reference Voltage VFB2=COMP2 ■ GATE High Voltage at 100mA Low Voltage at 100mA Rise Time Fall Time GATE pull-down Measure VCC2 –GATE Measure GATE 1.6V < GATE < (VCC2 – 2.5V) (VCC2 – 2.5V) > GATE > 1.6V Resistance to PGnd ■ VI/O Overcurrent Protection OVC2 Comparator Offset Voltage 0V < VOUT2 ≤ 3.5V Discharge Threshold Voltage VOUT2 Bias Current 0.2V ≤ VOUT2 ≤ 3.5V OVC2 Latch Discharge Current ■ PWM Comparator 2 PWM Comparator Offset Voltage 0V ≤ VFFB2 ≤ 3.5V Transient Response VFFB2 = 0 to 3.5V VFFB2 Bias Current 0.2V ≤ VFFB2 ≤ 3.5V ■ COFF2 Off-Time Charge Current Discharge Current MIN 6 UNIT 1.205 1.255 KHz dB V 20 1.2 1.0 40 40 50 2.1 1.5 80 80 115 V V ns ns kΩ 77 0.2 -7.0 100 86 0.25 0.1 800 101 0.3 7.0 2500 mV V µA µA 0.99 1.10 200 0.1 1.22 300 7.0 V ns µA 1.0 1.6 550 25 2.3 µs µA mA 7.9 7.6 0.15 8.4 8.1 0.30 13 6 8.9 8.6 0.60 20 9 V V V mA mA VCOFF2 = 1.5V VCOFF2 = 1.5V Note 1: Guaranteed by design, not 100% tested in production. MAX 20 70 1.230 -7.0 ■ General Electrical Specifications VCC Monitor Start Threshold All Outputs On All Outputs Off VCC Monitor Stop Threshold Hysteresis Start - Stop VCC1 Supply Current No Load on GATE(L) VCC2 Supply Current No Loads on GATE(H) and GATE TYP Line and load regulation are drastically improved because there are two independent voltage loops. A voltage mode controller relies on a change in the error signal to compensate for a deviation in either line or load voltage. This change in the error signal causes the output voltage to change corresponding to the gain of the error amplifier, which is normally specified as line and load regulation. A current mode controller maintains fixed error signal under deviation in the line voltage, since the slope of the ramp signal changes, but still relies on a change in the error signal for a deviation in load. The V2TM method of control maintains a fixed error signal for both line and load variation, since the ramp signal is affected by both line and load. Theory Of Operation V2TM Control Method The V2TM method of control uses a ramp signal that is generated by the ESR of the output capacitors. This ramp is proportional to the AC current through the main inductor and is offset by the value of the DC output voltage. This control scheme inherently compensates for variation in either line or load conditions, since the ramp signal is generated from the output voltage itself. This control scheme differs from traditional techniques such as voltage mode, which generates an artificial ramp, and current mode, which generates a ramp from inductor current. PWM Comparator Constant Off-Time To minimize transient response, the CS5132 uses a Constant Off-Time method to control the rate of output pulses. During normal operation, the Off-Time of the high side switch is terminated after a fixed period, set by the COFF capacitor. Every time the VFFB pin exceeds the COMP pin voltage an Off-Time is initiated. To maintain regulation, the V2TM Control Loop varies switch On-Time. The PWM comparator monitors the output voltage ramp, and terminates the switch On-Time. Constant Off-Time provides a number of advantages. Switch duty Cycle can be adjusted from 0 to 100% on a pulse-by pulse basis when responding to transient conditions. Both 0% and 100% Duty Cycle operation can be maintained for extended periods of time in response to Load or Line transients. + GATE(H) C GATE(L) – VFFB Ramp Signal Output Voltage Feedback VFB Error Amplifier COMP Error Signal – E + Reference Voltage Figure 1: V2TM Control Diagram. The V2TM control method is illustrated in Figure 1. The output voltage is used to generate both the error signal and the ramp signal. Since the ramp signal is simply the output voltage, it is affected by any change in the output regardless of the origin of that change. The ramp signal also contains the DC portion of the output voltage, which allows the control circuit to drive the main switch to 0% or 100% duty cycle as required. A change in line voltage changes the current ramp in the inductor, affecting the ramp signal, which causes the V2TM control scheme to compensate the duty cycle. Since the change in inductor current modifies the ramp signal, as in current mode control, the V2TM control scheme has the same advantages in line transient response. A change in load current will have an affect on the output voltage, altering the ramp signal. A load step immediately changes the state of the comparator output, which controls the main switch. Load transient response is determined only by the comparator response time and the transition speed of the main switch. The reaction time to an output load step has no relation to the crossover frequency of the error signal loop, as in traditional control methods. The error signal loop can have a low crossover frequency, since transient response is handled by the ramp signal loop. The main purpose of this ‘slow’ feedback loop is to provide DC accuracy. Noise immunity is significantly improved, since the error amplifier bandwidth can be rolled off at a low frequency. Enhanced noise immunity improves remote sensing of the output voltage, since the noise associated with long feedback traces can be effectively filtered. Programmable Output The CS5132 is designed to provide two methods for programming the output voltage of the power supply. A five bit on board digital to analog converter (DAC) is used to program the output voltage within two different ranges. The first range is 2.125V to 3.525V in 100mV steps, the second is 1.325V to 2.075V in 50mV steps, depending on the digital input code. If all five bits are left open, the CS5132 enters adjust mode. In adjust mode, the designer can choose any output voltage by using resistor divider feedback to the VFB pin, as in traditional controllers. The CS5132 is specifically designed to meet or exceed Intel’s Pentium® II specifications. Error Amplifier An inherent benefit of the V2TM control topology is that there is no large bandwidth requirement on the error amplifier design. The reaction time to an output load step has no relation to the crossover frequency, since transient response is handled by the ramp signal loop. The main purpose of this”slow”feedback loop is to provide DC accuracy. Noise immunity is significantly improved, since the error amplifier bandwidth can be rolled off at a low frequency. Enhanced noise immunity improves remote sensing of the output voltage, since the noise associated with long feedback traces can be effectively filtered. The COMP pin is the output of the error amplifier and a capacitor to LGnd compensates the error amplifier loop. Additionally, through the built-in offset on the PWM Comparator noninverting input, the COMP pin provides the hiccup timing for the Over-Current Protection, the soft start function that 7 CS5132 Application Information CS5132 Application Information: continued the 1.06V PWM comparator offset, the regulator output will softstart normally (see Figure 5). Because the start-up circuitry depends on the current sense function, a current sense resistor should always be used. minimizes inrush currents during regulator power-up, and switcher output enable. Start-up The CS5132 provides a controlled start-up of regulator output voltage and features Programmable Soft Start implemented through the Error Amp and external Compensation Capacitor. This feature, combined with overcurrent protection, prevents stress to the regulator power components and overshoot of the output voltage during start-up. As Power is applied to the regulator, the CS5132 Undervoltage Lockout circuit (UVL) monitors the ICs supply voltage (VCC) which is typically connected to the +12V output of the AC-DC power supply. The UVL circuit prevents the NFET gates from being activated until VCC exceeds the 8.4V (typ) threshold. Hysteresis of 300mV (typ) is provided for noise immunity. The Error Amp Capacitor connected to the COMP pin is charged by a 30µA current source. This capacitor must be charged to 1.06V (typ) so that it exceeds the PWM comparator’s offset before the V2 PWM control loop will permit switching to occur. When VCC has exceeded 8.4V and COMP has charged to 1.06V, the upper Gate driver (GATE(H)) is activated, turning on the upper FET. This causes current to flow through the output inductor and into the output capacitors and load according to the following equation: Start-up @ VCC > 8.4V Figure 2: Normal Start-up (2ms/div). Channel 1 - Regulator Output Voltage (1V/div) Channel 2 - COMP Pin (1V/div) Channel 3 - VCC (10V/div) Channel 4 - Regulator Input Voltage (5V/div) T L GATE(H) and the upper NFET remain on and inductor current ramps up until the initial pulse is terminated by either the PWM control loop or the overcurrent protection. This initial pulse of in-rush current minimizes start-up time, but does not overstress the regulator’s power components. The PWM comparator will terminate the initial pulse if the regulator output exceeds the voltage on the COMP pin minus the 1.06V PWM comparator offset prior to the drop across the current sense resistor exceeding the current limit threshold. In this case, the PWM control loop has achieved regulation and the initial pulse is then followed by a constant off time as programmed by the COFF capacitor. The COMP capacitor will continue to slowly charge and regulator output voltage will follow it, less the 1.06V PWM offset, until it achieves the voltage programmed by the DAC’s VID input. The Error Amp will then source or sink current to the COMP cap as required to maintain the correct regulator DC output voltage. Since the rate of increase of the COMP pin voltage is typically set much slower than the regulator’s slew capability, inrush current, output voltage, and duty cycle all gradually increase from zero. (See Figures 2, 3, and 4.) If the voltage across the Current Sense resistor generates a voltage difference between the VFFB and VOUT pins that exceeds the OVC Comparator Offset Voltage (86mV typical), the Fault latch is set. This causes the COMP pin to be quickly discharged, turning off GATE(H) and the upper NFET since the voltage on the COMP pin is now less than the 1.06V PWM comparator offset. The Fault latch is reset when the voltage on the COMP decreases below the Discharge threshold voltage (0.25V typical). The COMP capacitor will again begin to charge, and when it exceeds I = (VIN – VOUT) x Start-up @ VCC > 8.4V Initial Pulse until VOUT > COMP - PWM Offset Figure 3: Normal Start-up showing initial pulse followed by Soft Start (20µs/div). Channel 1 - Regulator Output Voltage (0.2V/div) Channel 2 – Inductor Switching Node (5V/div) Channel 3 - VCC (10V/div) Channel 4 - Regulator Input Voltage (5V/div) 8 ILIM = Current Limit Threshold; ILOAD = Load Current during start-up; COUT = Total Output Capacitance. CS5132 Application Information: continued Normal Operation During Normal operation, Switch Off-Time is constant and set by the COFF capacitor. Switch On-Time is adjusted by the V2TM Control loop to maintain regulation. This results in changes in regulator switching frequency, duty cycle, and output ripple in response to changes in load and line. Output voltage ripple will be determined by inductor ripple current and the ESR of the output capacitors Duty Cycle = VOUT / VIN 0.27V / 3.54V = 7% ≈ 5.2% Transient Response The CS5132 V2TM Control Loop’s 200ns reaction time provides unprecedented transient response to changes in input voltage or output current. Pulse-by-pulse adjustment of duty cycle is provided to quickly ramp the inductor current to the required level. Since the inductor current cannot be changed instantaneously, regulation is maintained by the output capacitor(s) during the time required to slew the inductor current. Overall load transient response is further improved through a feature called “Adaptive Voltage Positioning”. This technique pre-positions the output voltage to reduce total output voltage excursions during changes in load. Holding tolerance to 1% allows the error amplifiers reference voltage to be targeted +25mV high without compromising DC accuracy. A “Droop Resistor”, implemented through a PC board trace, connects the Error Amps feedback pin (VFB) to the output capacitors and load and carries the output current. With no load, there is no DC drop across this resistor, producing an output voltage tracking the Error amps, including the +25mV offset. When the full load current is delivered, a 50mV drop is developed across this resistor. This results in output voltage being offset 25mV low. The result of Adaptive Voltage Positioning is that additional margin is provided for a load transient before reaching the output voltage specification limits. When load current suddenly increases from its minimum level, the output is pre-positioned +25mV. Conversely, when load current suddenly decreases from its maximum level, the output is prepositioned -25mV. For best Transient Response, a combination of a number of high frequency and bulk output capacitors are usually used. Figure 4: Pulse-by-Pulse Regulation during Soft Start (2µs/div). Channel 1 - Regulator Output Voltage (0.2V/div) Channel 2 – Inductor Switching Node (5V/div) Channel 3 - VCC (10V/div) Channel 4 - Regulator Input Voltage (5V/div) OCP @ VCC > 8.5V Soft Start @ COMP > 1.06V Figure 5: Start-up with COMP pre-charged to 2V (2ms/div). Channel 1 - Regulator Output Voltage (1V/div) Channel 2 - COMP Pin (1V/div) Channel 3 - VCC (10V/div) Channel 4 - Regulator Input Voltage (5V/div) Slope Compensation The V2TM control method uses a ramp signal, generated by the ESR of the output capacitors, that is proportional to the ripple current through the inductor. To maintain regulation, the V2TM control loop monitors this ramp signal, through the PWM comparator, and terminates the switch on-time. The stringent load transient requirements of modern microprocessors require the output capacitors to have very low ESR. The resulting shallow slope presented to the PWM comparator, due to the very low ESR, can lead to pulse width jitter and variation caused by both random or synchronous noise. When driving large capacitive loads, the COMP must charge slowly enough to avoid tripping the CS5132 overcurrent protection. The following equation can be used to ensure unconditional start-up. I ICHG –I < LIM LOAD CCOMP COUT where ICHG = COMP Source Current (30µA typical); CCOMP = COMP Capacitor value (0.1µF typical); 9 CS5132 Application Information: continued Adding slope compensation to the control loop, avoids erratic operation of the PWM circuit, particularly at lower duty cycles and higher frequencies, where there is not enough ramp signal, and provides a more stable switchpoint. The scheme that prevents that switching noise prematurely triggers the PWM circuit consists of adding a positive voltage slope to the output of the Error Amplifier (COMP pin) during an off-time cycle. The circuit that implements this function for the synchronous regulator section (VCC(CORE)) is shown in Figure 6. 5 COMP1 CCOMP CS5132 R2 R1 C1 19 GATE(L) To Synchronous FET Figure 6: Small RC filter provides the proper voltage ramp at the beginning of each on-time cycle. The ramp waveform is generated through a small RC filter that provides the proper voltage ramp at the beginning of each on-time cycle. The resistors R1 and R2 in the circuit of Figure 6 form a voltage divider from the GATE(L) output, superimposing a small artificial ramp on the output of the error amplifier. A similar approach can be used also for the non-synchronous regulator section (VI/O) as shown in Figure 7. In this case, the slope compensation signal is generated directly from the GATE output, through the ac coupling capacitor C1, at the beginning of each on-cycle. It is important that in both circuits, the series combination R1/R2 is high enough in resistance not to load down and negatively affect the slew rate on the GATE(L) and GATE pins. 15 To VI/O Power Switch GATE CS5132 R1 13 Protection and Monitoring Features Over-Current Protection A loss-less hiccup mode current limit protection feature is provided, requiring only the COMP capacitor to implement. The CS5132 provides overcurrent protection by sensing the current through a “Droop” resistor, using an internal current sense comparator. The comparator compares the voltage drop across the “Droop” resistor to an internal reference voltage of 86mV (typical). If the voltage drop across the “Droop” resistor exceeds this threshold, the current sense comparator allows the fault latch to be set. This causes the regulator to stop switching. During this over current condition, the CS5132 stays off for the time it takes the COMP pin capacitor to discharge to its lower 0.25V threshold. As soon as the COMP pin reaches 0.25V, the Fault latch is reset (no overcurrent condition present) and the COMP pin is charged with a 30µA current source to a voltage 1.06V greater than the VFFB voltage. Only at this point the regulator attempts to restart normally. The CS5132 will operate initially with a duty cycle whose value depends on how low the VFFB voltage was during the overcurrent condition (whether hiccup mode was due to excessive current or hard short). This protection scheme minimizes thermal stress to the regulator components, input power supply, and PC board traces, as the over current condition persists. Upon removal of the overload, the fault latch is cleared, allowing normal operation to resume. Overvoltage Protection Overvoltage protection (OVP) is provided as result of the normal operation of the V2TM control topology and requires no additional external components. The control loop responds to an overvoltage condition within 200ns, causing the top MOSFET to shut off, disconnecting the regulator from its input voltage. This results in a “crowbar” action to clamp the output voltage and prevents damage to the load. The regulator will remain in this state until the overvoltage condition ceases or the input voltage is pulled low. Additionally, a dedicated Overvoltage protection (OVP) output pin (pin 21) is provided in the CS5132. The OVP signal will go high (overvoltage condition), if the output voltage (VCC(CORE)) exceeds the regulation voltage by 8.5% of the voltage set by the particular DAC code. The OVP pin can source up to 25mA of current that can be used to drive an SCR to crowbar the power supply. COMP2 CCOMP2 C1 R2 Figure 7: Slope compensation for the non-synchronous regulator section (VI/O). Power-Good Circuit The Power-Good pin (pin 22) is an open-collector signal consistent with TTL DC specifications. It is externally pulled up, and is pulled low (below 0.3V) when the regulator output voltage typically exceeds ± 8.5% of the nominal output voltage. Maximum output voltage deviation before Power-Good is pulled low is ± 12%. Output Enable On/off control of the regulator outputs can be implemented by pulling the COMP pins low. It is required to pull the COMP pins below the 1.06V PWM comparator offset voltage in order to disable switching on the GATE drivers. 10 Output Capacitor Discharge During Transient - 10mV CS5132-based Dual Output Buck Regulator Design Example Maximum allowable ESR is: Step 1: Define Specification 0.08V 15A = 5.3mΩ. ESR = Input Voltage from “silver box” power supply • 5V ±5% for conversion to output voltage • 12V ±5% for NFET Gate Voltage and circuit bias The ESR for a 1200µF/10V Sanyo capacitor type GX is 44mΩ per capacitor. Output Voltages • 2.0V @ 16A for VCC(CORE) • 3.3V@ 8A for VI/O • 5% Overall Voltage accuracy (load, line, temperature, ripple) • 2% DC & 5% AC Voltage Accuracy • < 2% Output Ripple Voltage • 15A Load Step @ 20A /µs - VCC(CORE) • 7A Load Step @ 5A/µs - VI/O 44 Number of Capacitors = 5.3 ≅ 8. Output voltage deviation due to ESR: ∆V = 15A × 5.5mΩ = 82mV. Thermal Management • 0 to 50° C ambient temperature range • Component junction temperatures within manufacturer’s specified ratings at full load & TA(MAX) The ESL is calculated from ∆I −6 ESL = ∆V × ∆t = 0.01V × 1 × 10 = 0.5nH. 20 ∆I It is estimated that a 10 × 12 mm Aluminum Electrolytic capacitor has approximately 4nH of package inductance. In this case we have eight (8) capacitors in parallel for a total capacitor ESL: Step 2: Determine Output Capacitors ESL = These components must be selected and placed carefully to yield optimal results. Capacitors should be chosen to provide acceptable ripple on the regulator output voltage. Key specifications for input capacitors are their ripple rating, while ESR is important for output capacitors. For best transient response, a combination of low value/high frequency and bulk capacitors placed close to the load will be required. ) = 0.5nH. The change in capacitor voltage during the transient is: ∆VC = ∆I × tTR COUT , where tTR is the output voltage transient response time. We choose tTR = 6µs: ∆VC = The voltage transient during the load step is ( 8 ∆V = ESL × ∆I = 0.5nH × 20A = 10mV. 1µs ∆t The load transients have slew rates of up to 20A /µs, while the voltage drop during a transient must be kept to less than 100mV. The output capacitors must hold the output voltage within these limits since the inductor current can not change with the required slew rate. The output capacitors must therefore have a very low ESL and ESR. ∆VOUT = ∆IOUT × 4nΗ Output voltage deviation due to ESL: Step 2a: For the 2V Output (VCC(CORE)) tTR + ESR + COUT 20A , µs = ∆t Components • Low cost is top priority. • Surface mount when possible • Small footprint important • Component Ratings determined at 80% of Maximum Load ESL ∆t 44 8 = 5.5mΩ. Total ESR = 15A × 6µs 8 × 1200µF = 9mV. Total change in output voltage as a result of an increase in load current of a 15A step with a 20A/µs slew rate is: , ∆VOUT = ( 82mV + 10mV + 9mV ) = 101mV. where tTR = output voltage transient response time. The total change in output voltage is divided as follows: Step 2b: For the 3.3V Output (VI/O) ESR - 80mV The VI/O load transients have slew rates of 5A/µs, while the voltage drop during a transient must be kept to less ESL - 10mV 11 CS5132 Application Information: continued CS5132 Application Information: continued than ±165mV. Repeating step 2a, we select four (4) 1200µF/10V Sanyo GX output capacitors. Calculate Inductor Value: L= (VIN - VOUT) tTR ∆I Step 3: Duty Cycle, Switching Frequency, TON & TOFF Duty Cycle ≈ VOUT / VIN. ∆IL= Step 3a: Calculate On-Time for 2V Output = 0.40 200kHz = 2µs ∆VOUT Total ESR IL(PEAK) = IOUT + - TON = 5µs - 2µs = 3µs. FSW Period × (1-D) = 3980 5µs × 0.6 3980 = 750pF. 0.66 D TON = = = 3.3µs FSW 200kHz ∆IL= 1 - TON = 5µs – 3.3µs = 1.7µs. FSW Select COFF2 to be 390pF. Step 4: Output Inductor The inductor should be selected based on its inductance, current capability, and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. There are many factors to consider in selecting the inductor including: cost, efficiency, EMI and ease of manufacture. The inductor must be able to handle the peak current at the switching frequency without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. There are a variety of materials and types of magnetic cores that could be used for this application. Among them are: ferrites, molypermalloy cores (MPP), amorphous and powdered iron cores. We will use a powdered iron core. Iron powdered cores are very suitable due to their high saturation flux density and have low loss at high frequencies, a distributed gap and exhibit very low EMI. = 7.3A, L = 12.4A. ( ∆I2 ) = 16A - ( 7.3A 2 ) L (VIN - VOUT) × D FSW × L = (5V - 2V) × 0.4 200kHz × 1.2µH = 5A. The maximum inductor peak current becomes: IL(PEAK) = 16A + 5A = 16A + 2.5A = 18.5A. 2 The inductor valley current becomes: Calculate Off-Time: TOFF = 5.5mΩ The selected 1.2µH inductor yields the following ripple current: A standard COFF1 capacitance value of 680pF can be used. The 3980 factor is a characteristic of the CS5132. Step 3b: Calculate On-Time for 3.3V Output 40mV = 19.6A, ( ∆I2 )= 16A + ( 7.3A 2 ) IL(VALLEY) = IOUT - Select the COFF1 capacitor in order to set the Off-Time: COFF1 = = which corresponds to the following maximum Inductor Peak and Valley currents: Calculate Off-Time: 1 =1.2µH. 15A The maximum allowable Inductor Ripple Current for a 2% ripple on the 2V output is: Select 200kHz Switching Frequency (FSW). TOFF = 3V × 6µs = 15A ∆VOUT = 2% × 2V = 40mV D = 3.3V / 5V = 66% for 3.3V output. D FSW (5V-2V) × 6µs Step 4a: Select 2% Ripple on 2V Output D = 2.0V / 5V = 40% for 2V output. TON = = IL(VALLEY) = 16A - 5A = 16A - 2.5A = 13.5A. 2 The above values are well within the maximum allowable inductor peak and valley currents for a 2% output voltage ripple. Select Toroid Powdered Iron Core, low cost, low core losses at 200kHz, low EMI. Select XFMRS Inc, XF0016-VO4 1.2µH inductor with RDC = 0.003Ω typical, 0.008Ω maximum. Step 4b: Select 2% Ripple on 3.3V Output Repeating Step 4a for the 3.3V output, we find 3.5µH is a suitable value for this output. Step 5: Input Capacitors These components must be selected and placed carefully to yield optimal results. Capacitors should be chosen to provide acceptable ripple on the input supply lines. Key specifications for input capacitors are their ripple rating. Step 5a: VCC(CORE) Buck Regulator Input Capacitors The input capacitor CIN should also be able to handle the 12 input RMS current IIN(RMS). CIN discharges during the ontime. Step 6: Power MOSFETs FET Basics The use of the MOSFET as a power switch is propelled by two reasons: 1) Its very high input impedance and 2) Its very fast switching times. The electrical characteristics of a MOSFET are considered to be those of a perfect switch. Control and drive circuitry power is therefore reduced. Because the input impedance is so high, it is voltage driven. The input of the MOSFET acts as if it were a small capacitor, which the driving circuit must charge at turn on. The lower the drive impedance, the higher the rate of rise of VGS, and the faster the turn- on time. Power dissipation in the switching MOSFET consists of 1) conduction losses, 2) leakage losses, 3) turn-on switching losses, 4) turn-off switching losses, and 5) gate-transitions losses. The latter three losses are proportional to frequency. For the conducting power dissipation rms values of current and resistance are used for true power calculations. The fast switching speed of the MOSFET makes it indispensable for high-frequency power supply applications. Not only are switching power losses minimized, but the maximum usable switching frequency is considerably higher. Switching time is independent of temperature. Also, at higher frequencies, the use of smaller and lighter components (transformer, filter choke, filter capacitor) reduces overall component cost while using less space for more efficient packaging at lower weight. The MOSFET has purely capacitive input impedance. No DC current is required. It is important to keep in mind the drain current of the FET has a negative temperature coefficient. Increase in temperature causes higher on-resistance and greater leakage current. For switching circuits, VDS(ON) should be low to minimize power dissipation at a given ID, and VGS should be high to accomplish this. MOSFET switching times are determined by device capacitances, stray capacitances, and the impedance of the gate drive circuit. Thus the gate driving circuit must have high momentary peak current sourcing and sinking capability for switching the MOSFET. The input capacitance, output capacitance and reverse-transfer capacitance also increase with increased device current rating. Two considerations complicate the task of estimating switching times. First, since the magnitude of the input capacitance, CISS, varies with VDS, the RC time constant determined by the gate-drive impedance and CISS changes during the switching cycle. Consequently, computation of the rise time of the gate voltage by using a specific gatedrive impedance and input capacitance yields only a rough estimate. The second consideration is the effect of the "Miller" capacitance, CRSS, which is referred to as Cdg in the following discussion. For example, when a device is on, VDS is fairly small and VGS is about 12V. Cdg is charged to VDS(ON) - VGS, which is a negative potential if the drain is considered the positive electrode. When the drain is "off", Cdg is charged to quite a different potential. In this case the voltage across Cdg is a positive value since the potential from gate-to-source is near zero volts and VDS is essentially the drain supply voltage. During turn-on and turn-off, The discharge current is given by: ICINDISRMS = (IL(PEAK)2 + (IL(PEAK) × IL(VALLEY)) + IL(VALLEY2) × D = 10.2A. 3 CIN charges during the off-time, the average current through the capacitor over one switching cycle is zero: D ICIN(CH) = ICIN(DIS) × ICIN(CH) = 10.2A × , 1-D 0.4 = 6.8A. (1-0.4) So the total Input RMS current is: ICIN(RMS) = (ICIN(DIS)2 × D) +(ICIN(CH)2 × (1-D)), ICIN(RMS) = (10.22 × 0.4) + (6.82 (× 0.6)) = 8.3A . The number of input capacitors required is given by: NCIN = ICIN(RMS) . IRIPPLE For Sanyo capacitors type GX: 1200µF/10V, IRIPPLE = 1.25A. Hence, NCIN = 8.3 = 6.6. 1.25 The number of input capacitors can be rounded off to 6. Calculate the Input Capacitor Ripple Voltage: VRMS = IRMS × Total ESR = 8.3A × 7.3mΩ = 60mV. Calculate the Input Capacitor Power Loss: PCIN = IRMS2 × Total ESR = 0.504W. Step 5b: VI/O Buck Regulator Input Capacitors Repeating for the 3.3V output, we select 3 GX 1200µF/10V capacitors. 13 CS5132 Application Information: continued CS5132 Application Information: continued these large swings in gate-to-drain voltage tax the current sourcing and sinking capabilities of the gate drive. In addition to charging and discharging CGS, the gate drive must also supply the displacement current required by Cdg (IGATE = Cdg dVdg/dt). Unless the gate-drive impedance is very low, the VGS waveform commonly plateaus during rapid changes in the drain-to-source voltage. The most important aspect of FET performance is the Static Drain-To-Source On-Resistance (RDSON), which effects regulator efficiency and FET thermal management requirements. The On- Resistance determines the amount of current a FET can handle without excessive power dissipation that may cause overheating and potentially catastrophic failure. As the drain current rises, especially above the continuous rating, the On-Resistance also increases. Its positive temperature coefficient is between +0.6%/C and +0.85 %/C. The higher the On-Resistance the larger the conduction loss is. Both logic level and standard FETs can be used. The reference designs derive gate drive from the 12V supply which is generally available in most computer systems and utilizes logic level FETs. Multiple FETs may be paralleled to reduce losses and improve efficiency and thermal management. Voltage applied to the FET gates depends on the application circuit used. Both upper and lower gate driver outputs are specified to drive to within 1.5V of ground when in the low state and to within 2V of their respective bias supplies when in the high state. In practice, the FET gates will be driven rail-to-rail due to overshoot caused by the capacitive load they present to the controller IC. We select Mitsubishi’s FS70VSJ-03 (D2 package): 30V withstand voltage; RDSON = 8mΩ; ΘJA = 40°C/W; Total Gate Charge = 50nC. Step 6b: Similar calculations apply for the 3.3V output. Step 6a: For the 2V Output Upper (Switching) FET Step 6c: Synchronous FET ( 2V Output) Calculate the 2V Output’s Maximum RMS Current through the Switch: Calculate Switch Conduction Losses: PSW(ON) = 5V × 16A × 60× 10-9 6 × 5 × 10-6 = 0.16W. Switch Off Losses: PSW(OFF) = VIN × IOUT × TFALL 6T , TFALL = 160ns, (from Mitsubishi FS70VSJ-03 switching characteristics performance curves): PSW(OFF) = 5V × 16A × 160 × 10-9 = 0.43W. 6 × 5 × 10-6 Upper FET Total Losses = Switching Conduction Losses + Switch On Losses + Switch Off Losses: PFETH(TOTAL) = 0.83W + 0.16W + 0.43W = 1.42W. Calculate Maximum NFET Switch Junction Temperature: TJ = TA + [(PFETH(TOTAL)) × ΘJA ], TJ = 50C + (1.412W) × 40°C/W = 107°C. Calculate the Gate Driver Losses: PGATE(H) = Q × VGATE × FSW = 50nC × 12V × 200KHz = 120mW. PRMS = IRMS2 × RDSON = [IOUT2 × (1-D)] × RDSON IRMS(H) = = [16A2 × 0.6] × 8mΩ = 1.22W. (IL(PEAK)2 + (IL(PEAK) × IL(VALLEY)) + IL(VALLEY2) × D = 10.2A. 3 Calculate Switch Conduction Losses: PRMS = IRMS2 × RDSON = 10.2A2 × 8mΩ = 0.83W. The synchronous MOSFET has no switching losses, except for losses in the internal body diode, because it turns on into near zero voltage conditions. The MOSFET body diode will conduct during the non-overlap time and the resulting power dissipation (neglecting reverse recovery losses) can be calculated as follows: Calculate Switching Losses: Switch On Losses: PSW(ON) = (VIN × IOUT × TRISE) 6T PSW = VSD × ILOAD × non-overlap time × switching frequency. , From the Mitsubishi FS70VSJ-03 source-drain diode forward characteristics curve, VSD = 0.8V: PSW = 0.8V × 16A × 65ns × 200kHz, TRISE = 60ns, PSW = 0.16W. (from Mitsubishi FS70VSJ-03 switching characteristics performance curves): 1 T= = 5µs, FSW 14 Lower (Synchronous) FET Total Losses = Switch Conduction Losses + Body Diode Losses: PFETL(TOTAL) = 1.27W + 0.16W =1.43W. Calculate Maximum NFET Switch Junction Temperature: TJ = TA + [(PFETL(TOTAL) ) × ΘJA ], where ID = average drive current; QGATE(X) = total gate charge for each MOSFET; FSW1, FSW2 = switching frequencies for the synchronous and non-synchronous sections respectively. The power dissipation for the IC when VCC1 = VCC2 = VCC is: TJ = 50C + (1.43W) × 40°C/W = 107°C. Calculate the Gate Driver Losses: PGATE(L) = Q × VGATE × FSW = 50nC × 12V × 200KHz = 120mW. PD = ICC × VCC + ID × VCC, where ICC = quiescent supply current of the IC (both from VCC1 and VCC2). For the design example in question, PD = 19mA × 12V + 0.12W + 0.12W + 0.12W = 0.59W. Step 7: Free Wheeling Schottky Diode (3.3V Output) The four most application-important characteristics of a Schottky are: 1. Forward voltage drop; 2. Reverse leakage current; 3. Reverse blocking voltage; 4. Maximum permissible junction temperature. We calculate the average Schottky current: IAVG = IOUT × (1−D) = 8Α × 0.34 = 2.72Α. We select the Motorola MBRD835L rated at 8A, with 35V DC blocking voltage and 0.51Vforward voltage drop. Neglecting reverse losses, the power dissipation is due to the conduction loss only and can be computed as follows: PSCHOTTKY = VF × IAVG, where VF = maximum instantaneous forward voltage; PSCHOTTKY = 0.51V × 2.72Α = 1.39W. Calculate maximum Schottky junction temperature: TJ = TA + [(PSCHOTTKY ) × ΘJA ], TJ = 50C + (1.39W × 80°C/W) = 161°C. Proper heatsinking (copper pad under Schottky) will be required to reduce Schottky TJ below +125°C. Step 8: IC Power Dissipation The power dissipation on the IC varies with the MOSFETs used, VCC and the CS5132 operating frequency. This power dissipation is typically dominated by the average gate charge current for the MOSFETs. The average current is approximately: ID = (QGATE(H) + QGATE(L)) × FSW1 + QGATE × FSW2, The junction temperature of the IC is primarily a function of the PCB layout, since most of the heat is removed through the traces connected to the pins of the IC. “Droop” Resistor for Adaptive Voltage Positioning and Current Limit Adaptive voltage positioning is used to help keep the output voltage within specification during load transients. To implement adaptive voltage positioning a “Droop Resistor” must be connected between the output inductor and output capacitors and load. This resistor carries the full load current and should be chosen so that both DC and AC tolerance limits are met. An embedded PC trace resistor has the distinct advantage of near zero cost implementation. However, this droop resistor can vary due to three reasons: 1) the sheet resistivity variation caused by variation in the thickness of the PCB layer; 2) the mismatch of L/W; and 3) temperature variation. 1) Sheet Resistivity For one ounce copper, the thickness variation is typically 1.26 mil to 1.48 mil. Therefore the error due to sheet resistivity is: 1.48 - 1.26 = ±8%. 1.37 2) Mismatch due to L/W The variation in L/W is governed by variations due to the PCB manufacturing process. The error due to L/W mismatch is typically 1%. 3) Thermal Considerations Due to I2 × R power losses the surface temperature of the droop resistor will increase causing the resistance to increase. Also, the ambient temperature variation will contribute to the increase of the resistance, according to the formula: R = R20 [1+ α20(Τ−20)], 15 CS5132 Application Information: continued CS5132 Application Information: continued where R20 = resistance at 20°C; α= Current Limit The current limit setpoint has to be higher than the normal full load current. Attention has to be paid to the current rating of the external power components as these are the first to fail during an overload condition. The MOSFET continuous and pulsed drain current rating at a given case temperature has to be accounted for when setting the current limit trip point. Temperature curves on MOSFET manufacturers’ data sheets allow the designer to determine the MOSFET drain current at a particular VGS and TJ (junction temperature). This, in turn, will assist the designer to set a proper current limit, without causing device breakdown during an overload condition. For future “CPUs” the full load will be 16A. The internal current sense comparator current limit voltage limits are: 77mV < VTH < 101mV. Also, there is a 21% total variation in RSENSE as discussed in the previous section. We compute the value of the current sensing element (embedded PCB trace) for the minimum current limit setpoint: RSENSE(MIN) = RSENSE(TYP) × 0.79, 0.00393 ; °C T= operating temperature; R = desired droop resistor value. For temperature T = 50°C, the % R change = 12%. Droop Resistor Tolerance Tolerance due to sheet resistivity variation ±8% Tolerance due to L/W error 1% Tolerance due to temperature variation 12% Total tolerance for droop resistor 21% In order to determine the droop resistor value the nominal voltage drop across it at full load has to be calculated. This voltage drop has to be such that the output voltage at full load is above the minimum DC tolerance spec: VDROOP(TYP) = VDAC(MIN)-VDC(MIN) 1+RDROOP(TOLERANCE) . Example: for a 450MHz Pentium ®II, the DC accuracy spec is 1.93 < VCC(CORE) < 2.07V, and the AC accuracy spec is 1.9V < VCC(CORE) < 2.1V. The CS5132 DAC output voltage is +2.004V < VDAC < +2.045V. In order not to exceed the DC accuracy spec, the voltage drop developed across the resistor must be calculated as follows: VDROOP(TYP) = = RSENSE(MAX) = RSENSE(TYP) × 1.21, RSENSE(MAX) = We select, [VDAC(MIN)-VDC (MIN)] 1+RDROOP(TOLERANCE) +2.004V-1.93V 1.21 VTH(MIN) 77mV = = 4.8mΩ. ICL(MIN) 16A RSENSE(TYP) = 3.3mΩ. We calculate the range of load currents that will cause the internal current sense comparator to detect an overload condition. = 61mV. With the CS5132 DAC accuracy being 1%, the internal error amplifier’s reference voltage is trimmed so that the output voltage will be 25mV high at no load. With no load, there is no DC drop across the resistor, producing an output voltage tracking the error amplifier output voltage, including the offset. When the full load current is delivered, a drop of -50mV is developed across the resistor. Therefore, the regulator output is pre-positioned at 25mV above the nominal output voltage before a load turn-on. The total voltage drop due to a load step is ∆V-25mV and the deviation from the nominal output voltage is 25mV smaller than it would be if there was no droop resistor. Similarly at full load the regulator output is pre-positioned at 25mV below the nominal voltage before a load turn-off. the total voltage increase due to a load turn-off is ∆V-25mV and the deviation from the nominal output voltage is 25mV smaller than it would be if there was no droop resistor. This is because the output capacitors are pre-charged to a value that is either 25mV above the nominal output voltage before a load turn-on or, 25mV below the nominal output voltage before a load turnoff . Obviously, the larger the voltage drop across the droop resistor (the larger the resistance), the worse the DC and load regulation, but the better the AC transient response. 16 Nominal Current Limit Setpoint From the overcurrent detection data in the electrical characteristics table: VTH(TYP) = 86mV, ICL(NOM) = VTH(TYP) = 86mV = 26A. RSENSE(NOM) 3.3mΩ Maximum Current Limit Setpoint From the overcurrent detection data in the electrical characteristics table: VTH(MAX) = 101mV, ICL(MAX)= = VTH(MAX) VTH(MAX) = RSENSE(MIN) RSENSE(NOM) × 0.79 101mV = 38.7A. 3.3mΩ × 0.79 where R = droop resistor value; ρ = 0.71786mΩ-mil (1 oz. copper); L = droop resistor length; W = droop resistor width. Therefore, the range of load currents that will cause the internal current sense comparator to detect an overload condition through a 3.3mΩ embedded PCB trace is: 19.3A < ICL < 38.7A, with 26A being the nominal overload condition. Design Rules for Using a Droop Resistor The basic equation for laying an embedded resistor is: RAR = ρ × L A or R = ρ × L (W × t) RDROOP = 3.3mΩ. 3.3mΩ = 0.71786mΩ-mil × , where A= W × t = cross-sectional area; ρ= the copper resistivity (µΩ-mil); L= length (mils); W = width (mils); t = thickness (mils). L . 201 mils × 1.37 mils Hence, L = 1265 mils = 1.265 in. In layouts where it is impractical to lay out a droop resistor in a straight line 1265 mils long, the embedded PCB trace can be “snaked” to fit within the available space. Thermal Management For most PCBs the copper thickness, t, is 35µm (1.37 mils) for one ounce copper; ρ = 717.86µΩ-mil. For a CPU load of 16A the resistance needed to create a 50mV drop at full load is: RDROOP = Thermal Considerations for Power MOSFETs and Diodes In order to maintain good reliability, the junction temperature of the semiconductor components should be kept to a maximum of 150°C or lower. The thermal impedance (junction to ambient) required to meet this requirement can be calculated as follows: 50mV 50mV = = 3.1mΩ. IOUT 16A Thermal Impedance = The resistivity of the copper will drift with the temperature according to the following guidelines: ∆R = 12% @ TA = +50°C; ∆R = 34% @TA = +100°C. A heatsink may be added to TO-220 components to reduce their thermal impedance. A number of PC board layout techniques such as thermal vias and additional copper foil area can be used to improve the power handling capability of surface mount components. Droop Resistor Length, Width, and Thickness The minimum width and thickness of the droop resistor should primarily be determined on the basis of the currentcarrying capacity required, and the maximum permissible droop resistor temperature rise. PCB manufacturer design charts can be used in determining current- carrying capacity and sizes of etched copper conductors for various temperature rises above ambient. For single conductor applications, such as the use of the droop resistor, PCB design charts show that for a droop resistor with a required current-carrying capacity of 16A, and a 45°C temperature rise above ambient, the recommended cross section is 275 mil2. EMI Management As a consequence of large currents being turned on and off at high frequency, switching regulators generate noise as a consequence of their normal operation. When designing for compliance with EMI/EMC regulations, additional components may be added to reduce noise emissions. These components are not required for regulator operation and experimental results may allow them to be eliminated. The input filter inductor may not be required because bulk filter and bypass capacitors, as well as other loads located on the board will tend to reduce regulator di/dt effects on the circuit board and input power supply. Placement of the power component to minimize routing distance will also help to reduce emissions. W × t = 275 mil2, where W = droop resistor width; t = droop resistor thickness. For 1oz. copper, t= 1.37 mils, therefore W = 201 mils = 0.201 in. R=ρ× TJ(MAX) - TA . Power L , W×t 17 CS5132 Application Information: continued CS5132 Application Information: continued Layout Guidelines When laying out the CPU buck regulator on a printed circuit board, the following checklist should be used to ensure proper operation of the CS5132. 1) Rapid changes in voltage across parasitic capacitors and abrupt changes in current in parasitic inductors are major concerns for a good layout. 8) Use fewer, but larger output capacitors, keep the capacitors clustered, and use multiple layer traces with heavy copper to keep the parasitic resistance low. 9) Place the switching MOSFET as close to the +5V input capacitors as possible. 10) Place the output capacitors as close to the load as possible. 11) Place the VFFB,VOUT filter resistors (510Ω) in series with the VFFB and VOUT pins as close as possible to the pins. 2) Keep high currents out of sensitive ground connections. Avoid connecting the IC Gnd between the source of the lower FET and the input capacitor Gnd. 12) Place the COFF and COMP capacitors as close as possible to the COFF and COMP pins. 3) Avoid ground loops as they pick up noise. Use star or single point grounding. 13) Place the current limit filter capacitors between the VFFB and VOUT pins, as close as possible to the pins. 4) For high power buck regulators on double-sided PCBs a single ground plane (usually the bottom) is recommended. 14) Connect the filter components of the following pins: VFB, VFFB, VOUT, COFF, and COMP to the LGnd pin with a single trace, and connect this local LGnd trace to the output capacitor Gnd. 5) Even though double sided PCBs are usually sufficient for a good layout, four-layer PCBs are the optimum approach to reducing susceptibility to noise. Use the two internal layers as the power and Gnd planes, the top layer for the high current connections and component vias, and the bottom layer for the noise sensitive traces. 15) The “Droop” Resistor (embedded PCB trace) has to be wide enough to carry the full load current. 16) Place the VCC bypass capacitors as close as possible to the VCC pins and connect them to PGnd. 6) Keep the inductor switching node small by placing the output inductor, switching and synchronous FETs close together. 7) The MOSFET gate traces to the IC must be as short, straight, and wide as possible. Ideally, the IC has to be placed right next to the MOSFETs. 18 CS5132 Package Specification PACKAGE THERMAL DATA PACKAGE DIMENSIONS IN mm (INCHES) Thermal Data RΘJC typ RΘJA typ D Lead Count Metric Max Min 15.60 15.20 24L SO Wide English Max Min .614 .598 24L SO Wide 16 80 °C/W °C/W Surface Mount Wide Body (DW); 300 mil wide 7.60 (.299) 7.40 (.291) 10.65 (.419) 10.00 (.394) 0.51 (.020) 0.33 (.013) 1.27 (.050) BSC 2.49 (.098) 2.24 (.088) 1.27 (.050) 0.40 (.016) 2.65 (.104) 2.35 (.093) 0.32 (.013) 0.23 (.009) D REF: JEDEC MS-013 0.30 (.012) 0.10 (.004) Ordering Information Part Number CS5132GDW24 CS5132GDWR24 ON Semiconductor and the ON Logo are trademarks of Semiconductor Components Industries, LLC (SCILLC). ON Semiconductor reserves the right to make changes without further notice to any products herein. For additional information and the latest available information, please contact your local ON Semiconductor representative. Description 24L SO Wide 24L SO Wide (tape & reel) 19 © Semiconductor Components Industries, LLC, 2000 Notes