CS5151 CS5151 CPU 4-Bit Nonsynchronous Buck Controller Features Description The CS5151 is a 4-bit nonsynchronous N-Channel buck controller. It is designed to provide unprecedented transient response for today’s demanding high-density, high-speed logic. The regulator operates using a proprietary control method, which allows a 100ns response time to load transients. The CS5151 is designed to operate over a 4.25-16V range (VCC) using 12V to power the IC and 5V as the main supply for conversion. The CS5151 is specifically designed to power Pentium® processors with MMX™ Technology and other high performance core logic. It includes the following features: on board, 4-bit DAC, short circuit protection, 1.0% output tolerance, VCC monitor, and programmable soft start capability. The CS5151 is upwards compatible with the 5-bit CS5156, allowing the mother board designer the capability of using either the CS5151 or the CS5156 with no change in layout. The CS5151 is available in 16 pin surface mount and DIP packages. ■ N-Channel Design ■ Excess of 1MHz Operation ■ 100ns Transient Response ■ 4-Bit DAC ■ Upward Compatible with 5-Bit CS5155/5156 and Adjustable CS5120/5121 ■ 30ns Gate Rise/Fall Times ■ 1% DAC Accuracy ■ 5V & 12V Operation ■ Remote Sense ■ Programmable Soft Start ■ Lossless Short Circuit Protection Application Diagram ■ VCC Monitor Switching Power Supply for core logic - Pentium® processor with MMX™ Technology 12V 5V VCC1 VCC2 VID0 VID0 VID1 VID2 VID1 VID2 VID3 VID3 2µH 2.1V to 3.5V @ 13A Package Options CS5151 3 MBR735 16 Lead SO Narrow & PDIP 1200µF/16V x 5 AlEl COFF PGnd 330pF SS LGnd VFFB VID0 VID1 VFB 1 COMP LGnd VID2 VFB COMP 0.33µF ■ Overvoltage Protection IRL3103 VGATE 1,2 0.1µF ■ Current Sharing 1200µF/16V x 3 AlEl 0.1µF ■ Adaptive Voltage Positioning ■ V2™ Control Topology VID3 3.3k VCC1 NC SS 100pF NC COFF PGnd VGATE VFFB VCC2 V2 is a trademark of Switch Power, Inc. Pentium is a registered trademark and MMX is a trademark of Intel Corporation. Cherry Semiconductor Corporation 2000 South County Trail, East Greenwich, RI 02818 Tel: (401)885-3600 Fax: (401)885-5786 Email: [email protected] Web Site: www.cherry-semi.com Rev. 1/5/99 1 A ® Company CS5151 Absolute Maximum Ratings Pin Name Max Operating Voltage Max Current VCC1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .25mA DC/1.5A peak VCC2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20mA DC/1.5A peak SS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-100µA COMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .200µA VFB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA COFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA VFFB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.2µA VID0 - VID3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-50µA VGATE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V/-0.3V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100mA DC/1.5A peak LGnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .25mA PGnd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100mA DC/1.5A peak Operating Junction Temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .°0 to 150°C Lead Temperature Soldering Wave Solder (through hole styles only) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .10 sec. max, 260°C peak Reflow (SMD styles only) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .60 sec. max above 183°C, 230°C peak Storage Temperature Range, TS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-65° to 150°C ESD Susceptibility . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2kV Electrical Characteristics: 0°C < TA < +70°C; 0°C < TJ < +85°C; 8V < VCC1 < 14V; 5V < VCC2 < 14V; DAC Code: VID2 = VID1 = VID0 = 1; VID3 = 0; CVGATE = 1nF; COFF = 330pF; CSS = 0.1µF, unless otherwise specified. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT ■ Error Amplifier VFB Bias Current Open Loop Gain Unity Gain Bandwidth COMP SINK Current COMP SOURCE Current COMP CLAMP Current COMP High Voltage COMP Low Voltage PSRR VFB = 0V 1.25V < VCOMP < 4V; Note 1 Note 1 VCOMP = 1.5V; VFB = 3V; VSS > 2V VCOMP = 1.2V; VFB = 2.7V; VSS = 5V VCOMP = 0V; VFB = 2.7V VFB = 2.7V; VSS = 5V VFB =3V 8V < VCC1 < 14V @ 1kHz; Note 1 ■ VCC1 Monitor Start Threshold Stop Threshold Hysteresis Output switching Output not switching Start-Stop 3.75 3.70 3.90 3.85 50 4.05 4.00 V V mV VID0, VID1, VID2, VID3 VID0, VID1, VID2, VID3 1.00 25 4.85 1.25 50 5.00 2.40 100 5.15 1.0 V kΩ V % 1.2315 2.1186 2.2176 2.3166 2.4156 2.5146 2.6136 2.7126 2.8116 2.9106 3.0096 1.2440 2.1400 2.2400 2.3400 2.4400 2.5400 2.6400 2.7400 2.8400 2.9400 3.0400 1.2564 2.1614 2.2624 2.3634 2.4644 2.5654 2.6664 2.7674 2.8684 2.9694 3.0704 ■ DAC Input Threshold Input Pull Up Resistance Pull Up Voltage Accuracy VID3 VID2 VID1 VID0 1 1 1 1 1 1 1 0 1 1 0 1 1 1 0 0 1 0 1 1 1 0 1 0 1 0 0 1 1 0 0 0 0 1 1 1 0 1 1 0 0 1 0 1 50 500 0.4 30 0.4 4.0 60 0.3 60 3000 2.5 50 1.0 4.3 160 85 Measure VFB = VCOMP, 25°C ≤ TJ ≤ 85°C 2 1.0 8.0 70 1.6 5.0 600 µA dB kH mA µA mA V mV dB V V V V V V V V V V V VID0 = 1; VID3 = 0; CVGATE = 1nF; COFF = 330pF; CSS = 0.1µF, unless otherwise specified. PARAMETER ■ DAC: continued VID3 VID2 VID1 0 1 0 0 0 1 0 0 1 0 0 0 0 0 0 TEST CONDITIONS VID0 0 1 0 1 0 ■ VGATE Out SOURCE Sat at 100mA Out SINK Sat at 100mA Out Rise Time Out Fall Time Shoot-Through Current VGATE Resistance VGATE Schottky TYP MAX UNIT 3.1086 3.2076 3.3066 3.4056 3.5046 3.1400 3.2400 3.3400 3.4400 3.5400 3.1714 3.2724 3.3734 3.4744 3.5754 V V V V V 1.2 1.0 30 30 V V ns ns mA kΩ mV 20 50 600 2.0 1.5 50 50 50 100 800 1.6 25 1.0 0.50 0.9 3.3 100 3.3 0.95 1.0 2.5 5.0 200 6.0 1.10 1.1 3.0 ms ms % V V V VFFB = 0 to 5V to VGATE = 9V to 1V; VCC1 = VCC2 = 12V VFFB = 0V 100 125 ns ■ Supply Current ICC1 ICC2 Operating ICC1 Operating ICC2 No Switching No Switching VFB = COMP = VFFB VFB = COMP = VFFB 8.5 1.6 8 2 13.5 3.0 13 5 mA mA mA mA ■ COFF Normal Charge Time Extension Charge Time Discharge Current VFFB = 1.5V; VSS = 5V VSS = VFFB = 0 COFF to 5V; VFB >1V 1.0 5.0 5.0 1.6 8.0 2.2 11.0 µs µs mA VFB = VCOMP; VFFB = 2V; Record VGATE Pulse High Duration VFFB = 0V 10 30 50 µs 35 50 65 % ■ Soft Start (SS) Charge Time Pulse Period Duty Cycle COMP Clamp Voltage VFFB SS Fault Disable High Threshold ■ PWM Comparator Transient Response VFFB Bias Current ■ Time Out Timer Time Out Time Fault Mode Duty Cycle Measure VCC2 – VGATE Measure VGATE – VPGnd; 1V < VGATE < 9V; VCC1 = VCC2 = 12V 9V > VGATE > 1V; VCC1 = VCC2 = 12V Note 1 Resistor to LGnd LGnd to VGATE @ 10mA MIN (Charge Time/Pulse Period) × 100 VFB = 0V; VSS = 0 VGATE = Low Note 1: Guaranteed by design, not 100% tested in production. 3 0.3 µA CS5151 Electrical Characteristics: 0°C < TA < +70°C; 0°C < TJ < +85°C; 8V < VCC1 < 14V; 5V < VCC2 < 14V; DAC Code: VID2 = VID1 = CS5151 Package Pin Description PACKAGE PIN # PIN SYMBOL FUNCTION 16L SO Narrow & PDIP 1,2,3,4 VID0 – VID3 Voltage ID DAC input pins. These pins are internally pulled up to 5V providing logic ones if left open. The DAC range is 2.14V to 3.54V with 100mV increments. VID0 - VID3 select the desired DAC output voltage. Leaving all 4 DAC input pins open results in a DAC output voltage of 1.244V, allowing for adjustable output voltage, using a traditional resistor divider. 5 SS Soft Start Pin. A capacitor from this pin to LGnd in conjunction with internal 60µA current source provides soft start function for the controller. This pin disables fault detect function during Soft Start. When a fault is detected, the soft start capacitor is slowly discharged by internal 2µA current source setting the time out before trying to restart the IC. Charge/discharge current ratio of 30 sets the duty cycle for the IC when the regulator output is shorted. 6, 12 NC No connection. 7 COFF A capacitor from this pin to ground sets the time duration for the on board one shot, which is used for the constant off time architecture. 8 VFFB Fast feedback connection to the PWM comparator. This pin is connected to the regulator output. The inner feedback loop terminates on time. 9 VCC2 Boosted power for the gate driver. 10 VGATE MOSFET driver pin capable of 1.5A peak switching current. 11 PGnd High current ground for the IC. The MOSFET driver is referenced to this pin. Input capacitor ground and the anode of the Schottky diode should be tied to this pin. 13 VCC1 Input power for the IC. 14 LGnd Signal ground for the IC. All control circuits are referenced to this pin. 15 COMP Error amplifier compensation pin. A capacitor to ground should be provided externally to compensate the amplifier. 16 VFB Error amplifier DC feedback input. This is the master voltage feedback which sets the output voltage. This pin can be connected directly to the output or a remote sense trace. 4 CS5151 Block Diagram VCC2 VCC1 - VCC1 Monitor Comparator 5V + - 3.90V 3.85V 0.7V + 2µA Q S Q PGnd FAULT FAULT Latch SS High Comparator - VID0 VID2 R FAULT + 60µA SS VID1 VGATE SS Low Comparator 4 BIT DAC + VID3 2.5V Error Amplifier - PWM Comparator - VFB Slow Feedback COMP VFFB R Maximum On-Time Timeout + S Fast Feedback Extended Off-Time Timeout - Off-Time Timeout GATE = OFF Q PWM Latch Normal Off-Time Timeout GATE = ON Q COFF One Shot S + LGnd COFF R Q VFFB Low Comparator 1V PWM COMP Time Out Timer (30µs) Edge Triggered Applications Information The V2™ control method is illustrated in Figure 1. The output voltage is used to generate both the error signal and the ramp signal. Since the ramp signal is simply the output voltage, it is affected by any change in the output regardless of the origin of that change. The ramp signal also contains the DC portion of the output voltage, which allows the control circuit to drive the main switch to 0% or 100% duty cycle as required. A change in line voltage changes the current ramp in the inductor, affecting the ramp signal, which causes the V2™ control scheme to compensate the duty cycle. Since the change in inductor current modifies the ramp signal, as in current mode control, the V2™ control scheme has the same advantages in line transient response. A change in load current will have an affect on the output voltage, altering the ramp signal. A load step immediately changes the state of the comparator output, which controls the main switch. Load transient response is determined only by the comparator response time and the transition speed of the main switch. The reaction time to an output load step has no relation to the crossover frequency of the error signal loop, as in traditional control methods. The error signal loop can have a low crossover frequency, since transient response is handled by the ramp signal loop. The main purpose of this ‘slow’ feedback loop is to provide DC accuracy. Noise immunity is significantly improved, since the error amplifier bandwidth can be rolled off at a low frequency. Enhanced noise immunity improves remote sens- Theory of Operation V2™ Control Method The V2™ method of control uses a ramp signal that is generated by the ESR of the output capacitors. This ramp is proportional to the AC current through the main inductor and is offset by the value of the DC output voltage. This control scheme inherently compensates for variation in either line or load conditions, since the ramp signal is generated from the output voltage itself. This control scheme differs from traditional techniques such as voltage mode, which generates an artificial ramp, and current mode, which generates a ramp from inductor current. PWM Comparator + VGATE C – Ramp Signal VFFB VFB Error Amplifier COMP Error Signal Output Voltage Feedback – E + Reference Voltage Figure 1: V2™ Control Diagram 5 CS5151 Applications Information: continued ing of the output voltage, since the noise associated with long feedback traces can be effectively filtered. Line and load regulation are drastically improved because there are two independent voltage loops. A voltage mode controller relies on a change in the error signal to compensate for a deviation in either line or load voltage. This change in the error signal causes the output voltage to change corresponding to the gain of the error amplifier, which is normally specified as line and load regulation. A current mode controller maintains fixed error signal under deviation in the line voltage, since the slope of the ramp signal changes, but still relies on a change in the error signal for a deviation in load. The V2™ method of control maintains a fixed error signal for both line and load variation, since the ramp signal is affected by both line and load. approximately equal to the maximum on time, resulting in a 50% duty cycle. Then, the Gate pin will drive high, and the cycle repeats. When regulator output voltage achieves the 1V level present at the COMP pin, regulation has been achieved and normal off time will ensue. The PWM comparator terminates the switch on time, with off time set by the COFF capacitor. The V2™ control loop will adjust switch duty cycle as required to ensure the regulator output voltage tracks the output of the error amplifier. The soft start and COMP capacitors will charge to their final levels, providing a controlled turn on of the regulator output. Regulator turn on time is determined by the COMP capacitor charging to its final value. Its voltage is limited by the soft start COMP clamp and the voltage on the soft start pin (see Figures 2 and 3). Constant Off Time To maximize transient response, the CS5151 uses a constant off time method to control the rate of output pulses. During normal operation, the off time of the high side switch is terminated after a fixed period, set by the COFF capacitor. To maintain regulation, the V2™ control loop varies switch on time. The PWM comparator monitors the output voltage ramp, and terminates the switch on time. Constant off time provides a number of advantages. Switch duty cycle can be adjusted from 0 to 100% on a pulse by pulse basis when responding to transient conditions. Both 0% and 100% duty cycle operation can be maintained for extended periods of time in response to load or line transients. PWM slope compensation to avoid sub-harmonic oscillations at high duty cycles is avoided. Switch on time is limited by an internal 30µs timer, minimizing stress to the power components. Trace 1 - Regulator Output Voltage (1V/div.) Trace 2 - Inductor Switching Node (2V/div.) Trace 3 - 12V input (VCC1 and VCC2) (5V/div.) Trace 4 - 5V Input (1V/div.) Programmable Output The CS5151 is designed to provide two methods for programming the output voltage of the power supply. A four bit on board digital to analog converter (DAC) is used to program the output voltage from 2.14V to 3.54V in 100mV steps, depending on the digital input code. If all four bits are left open, the CS5151 enters adjust mode. In adjust mode, the designer can choose any output voltage by using resistor divider feedback to the VFB and VFFB pins, as in traditional controllers. The CS5151 is specifically designed to be upwards compatible with the CS5156, which uses a five bit DAC code. Figure 2: CS5151 demonstration board startup in response to increasing 12V and 5V input voltages. Extended off time is followed by normal off time operation when output voltage achieves regulation to the error amplifier output. Start Up Until the voltage on the VCC1 supply pin exceeds the 3.9V monitor threshold, the soft start and gate pins are held low. The FAULT latch is reset (no Fault condition). The output of the error amplifier (COMP) is pulled up to 1V by the comparator clamp. When the VCC1 pin exceeds the monitor threshold, the Gate output is activated, and the soft start capacitor begins charging. The Gate output will remain on, enabling the NFET switch, until terminated by either the PWM comparator, or the maximum on time timer. If the maximum on time is exceeded before the regulator output voltage achieves the 1V level, the pulse is terminated. The Gate pin drives low for the duration of the extended off time. This time is set by the time out timer and is Trace 1 - Regulator Output Voltage (1V/div.) Trace 3 - COMP Pin (error amplifier output) (1V/div.) Trace 4 - Soft Start Pin (2V/div.) Figure 3: CS5151 demonstration board startup waveforms. 6 If the input voltage rises quickly, or the regulator output is enabled externally, output voltage will increase to the level set by the error amplifier output more rapidly, usually within a couple of cycles (see Figure 4). Trace1 - Regulator Output Voltage (10V/div.) Trace 2 - Inductor Switching Node (5V/div.) Figure 6: Peak-to-peak ripple on VOUT = 2.8V, IOUT = 13A (heavy load). Trace 1 - Regulator Output Voltage (5V/div.) Trace 2 - Inductor Switching Node (5V/div.) Transient Response The CS5151 V2™ control loop’s 100ns reaction time provides unprecedented transient response to changes in input voltage or output current. Pulse by pulse adjustment of duty cycle is provided to quickly ramp the inductor current to the required level. Since the inductor current cannot be changed instantaneously, regulation is maintained by the output capacitor(s) during the time required to slew the inductor current. Overall load transient response is further improved through a feature called “adaptive voltage positioning”. This technique pre-positions the output capacitor’s voltage to reduce total output voltage excursions during changes in load. Holding tolerance to 1% allows the error amplifier’s reference voltage to be targeted +40mV high without compromising DC accuracy. A “droop resistor“, implemented through a PC board trace, connects the error amplifier’s feedback pin (VFB) to the output capacitors and load and carries the output current. With no load, there is no DC drop across this resistor, producing an output voltage tracking the error amplifier’s, including the +40mV offset. When the full load current is delivered, an 80mV drop is developed across this resistor. This results in output voltage being offset -40mV low. The result of adaptive voltage positioning is that additional margin is provided for a load transient before reaching the output voltage specification limits. When load current suddenly increases from its minimum level, the output capacitor is pre-positioned +40mV. Conversely, when load current suddenly decreases from its maximum level, the output capacitor is pre-positioned -40mV (see Figures 7, 8, and 9). For best transient response, a combination of a number of high frequency and bulk output capacitors are usually used. Figure 4: CS5151 demonstration board enable startup waveforms. Normal Operation During normal operation, switch off time is constant and set by the COFF capacitor. Switch on time is adjusted by the V2™ control loop to maintain regulation. This results in changes in regulator switching frequency, duty cycle, and output ripple in response to changes in load and line. Output voltage ripple will be determined by inductor ripple current working into the ESR of the output capacitors (see Figures 5 and 6). Trace 1 - Regulator Output Voltage (10V/div.) Trace 2 - Inductor Switching Node (5V/div.) Figure 5: Peak-to-peak ripple on VOUT = 2.8V, IOUT = 0.5A (light load). 7 CS5151 Applications Information: continued CS5151 Applications Information: continued If the maximum on time is exceeded while responding to a sudden increase in load current, a normal off time occurs to prevent saturation of the output inductor. Trace1 - Regulator Output Voltage (1V/div.) Trace 2 - Inductor Switching Node (5V/div.) Trace 3 - Output Current (13 to 0.5 Amps) (20V/div.) Figure 9: CS5151 demonstration board response to 13A load turn off (output set for 2.8V). V2™ control topology immediately connects inductor to ground, providing 0% duty cycle. Regulation is achieved in less than 10µs. Trace 1 - Regulator Output Voltage (1V/div.) Trace 3 - Regulator Output Current (20V/div.) Figure 7: CS5151 demonstration board response to a 0.5 to 13A load pulse (output set for 2.8V). Protection and Monitoring Features VCC1 Monitor To maintain predictable startup and shutdown characteristics an internal VCC1 monitor circuit is used to prevent the part from operating below 3.75V minimum startup. The VCC1 monitor comparator provides hysteresis and guarantees a 3.70V minimum shutdown threshold. Short Circuit Protection A lossless hiccup mode short circuit protection feature is provided, requiring only the soft start capacitor to implement. If a short circuit condition occurs (VFFB < 1V), the VFFB low comparator sets the FAULT latch. This causes the MOSFET to shut off, disconnecting the regulator from its input voltage. The soft start capacitor is then slowly discharged by a 2µA current source until it reaches its lower 0.7V threshold. The regulator will then attempt to restart normally, operating in its extended off time mode with a 50% duty cycle, while the soft start capacitor is charged with a 60µA charge current. If the short circuit condition persists, the regulator output will not achieve the 1V low VFFB comparator threshold before the soft start capacitor is charged to its upper 2.5V threshold. If this happens the cycle will repeat itself until the short is removed. The soft start charge/discharge current ratio sets the duty cycle for the pulses (2µA/60µA = 3.3%), while actual duty cycle is half that due to the extended off time mode (1.65%). This protection feature results in less stress to the regulator Trace 1 - Regulator Output Voltage (1V/div.) Trace 2 - Inductor Switching Node (5V/div.) Trace 3 - Output Current (0.5 to 13 Amps) (20V/div.) Figure 8: CS5151 demonstration board response to 13A load turn on (output set for 2.8V). Upon completing a normal off time, the V2™ control loop immediately connects the inductor to the input voltage, providing 100% duty cycle. Regulation is achieved in less than 20µs. 8 CS5151 Applications Information: continued components, input power supply, and PC board traces than occurs with constant current limit protection (see Figures 10 and 11). If the short circuit condition is removed, output voltage will rise above the 1V level, preventing the FAULT latch from being set, allowing normal operation to resume. External Output Enable Circuit On/off control of the regulator can be implemented through two additional discrete components (see Figure 12). This circuit operates by pulling the soft start pin high, and the VFFB pin low, emulating a short circuit condition. 5V MMUN2111T1 (SOT-23) 5 SS CS5151 8 V FFB IN4148 Shutdown Input Figure 12: Implementing shutdown with the CS5151. Trace 4 - 5V Supply Voltage (2V/div.) Trace 3 - Soft Start Timing Capacitor (1V/div.) Trace 2 - Inductor Switching Node (2V/div.) External Power Good Circuit An optional Power Good signal can be generated through the use of four additional external components (see Figure 15). The threshold voltage of the Power Good signal can be adjusted per the following equation: Figure 10: CS5151 demonstration board hiccup mode short circuit protection. Gate pulses are delivered while the soft start capacitor charges, and cease during discharge. VPower Good = (R1 + R2) × 0.65V R2 This circuit provides an open collector output that drives the Power Good output to ground for regulator voltages less than VPower Good. 5V R3 10k R1 10k PN3904 VOUT CS5151 Power Good PN3904 R2 6.2k Trace 4 = 5V from PC Power Supply (2V/div.) Trace 2 = Inductor Switching Node (2V/div.) Figure 11: Startup with regulator output shorted. Figure 13: Implementing Power Good with the CS5151. Overvoltage Protection Overvoltage protection (OVP) is provided as result of the normal operation of the V2™ control topology and requires no additional external components. The control loop responds to an overvoltage condition within 100ns, causing the MOSFET to shut off, disconnecting the regulator from its input voltage. 9 CS5151 Applications Information: continued Channel 3 = VGATE M1= VGATE - 5VIN Channel 2 = Inductor Switching Node Trace 3 = 12V Input (VCC1) and VCC2) (10V/div.) Trace 4 = 5V Input (2V/div.) Trace 1 = Regulator Output Voltage (1V/div.) Trace 2 = Power Good Signal (2V/div.) Figure 15: CS5151 gate drive waveforms depicting rail to rail swing. Figure 14: CS5151 demonstration board during power up. Power Good signal is activated when output voltage reaches 1.70V. The most important aspect of MOSFET performance is RDSON, which effects regulator efficiency and MOSFET thermal management requirements. The power dissipated by the MOSFET and the Schottky diode may be estimated as follows; Switching MOSFET: Power = ILOAD2 × RDSON × duty cycle Selecting External Components The CS5151 can be used with a wide range of external power components to optimize the cost and performance of a particular design. The following information can be used as general guidelines to assist in their selection. NFET Power Transistors Both logic level and standard MOSFETs can be used. The reference designs derive gate drive from the 12V supply which is generally available in most computer systems and use logic level MOSFETs. A charge pump may be easily implemented to support 5V only systems. Multiple MOSFETs may be paralleled to reduce losses and improve efficiency and thermal management. Voltage applied to the MOSFET gate depends on the application circuit used. The gate driver output is specified to drive to within 1.5V of ground when in the low state and to within 2V of its bias supply when in the high state. In practice, the MOSFET gate will be driven rail to rail due to overshoot caused by the capacitive load it presents to the controller IC. For the typical application where VCC1 = VCC2 = 12V and 5V is used as the source for the regulator output current, the following gate drive is provided; Schottky diode: Power = VFORWARD × ILOAD × (1 - duty cycle) VOUT + VFORWARD Duty Cycle = V + V IN FORWARD - (ILOAD × RDSON OF SWITCH FET) Off Time Capacitor (COFF) The COFF timing capacitor sets the regulator off time: TOFF = COFF × 4848.5 When the VFFB pin is less than 1V, the current charging the COFF capacitor is reduced. The extended off time can be calculated as follows: TOFF = COFF × 24,242.5. Off time will be determined by either the TOFF time, or the time out timer, whichever is longer. The preceding equations for duty cycle can also be used to calculate the regulator switching frequency and select the VGATE = 12V - 5V = 7V (see Figure 15). 10 COFF timing capacitor: COFF = Thermal Impedance = Period × (1 - duty cycle) , 4848.5 A heatsink may be added to TO-220 components to reduce their thermal impedance. A number of PC board layout techniques such as thermal vias and additional copper foil area can be used to improve the power handling capability of surface mount components. where: Period = 1 switching frequency EMI Management As a consequence of large currents being turned on and off at high frequency, switching regulators generate noise as a consequence of their normal operation. When designing for compliance with EMI/EMC regulations, additional components may be added to reduce noise emissions. These components are not required for regulator operation and experimental results may allow them to be eliminated. The input filter inductor may not be required because bulk filter and bypass capacitors, as well as other loads located on the board will tend to reduce regulator di/dt effects on the circuit board and input power supply. Placement of the power component to minimize routing distance will also help to reduce emissions. “Droop” Resistor for Adaptive Voltage Positioning Adaptive voltage positioning is used to reduce output voltage excursions during abrupt changes in load current. Regulator output voltage is offset +40mV when the regulator is unloaded, and -40mV at full load. This results in increased margin before encountering minimum and maximum transient voltage limits, allowing use of less capacitance on the regulator output (see Figure 7). To implement adaptive voltage positioning, a “droop” resistor must be connected between the output inductor and output capacitors and load. This is normally implemented by a PC board trace of the following value: RDROOP = TJUNCTION(MAX) - TAMBIENT Power 80mV IMAX Adaptive voltage positioning can be disabled for improved DC regulation by connecting the VFB pin directly to the load using a separate, non-load current carrying circuit trace. 2µH Input and Output Capacitors These components must be selected and placed carefully to yield optimal results. Capacitors should be chosen to provide acceptable ripple on the input supply lines and regulator output voltage. Key specifications for input capacitors are their ripple rating, while ESR is important for output capacitors. For best transient response, a combination of low value/high frequency and bulk capacitors placed close to the load will be required. 33Ω 2µH 1200µF x 3/16V + 1000pF Figure 16: Filter components Figure 17: Input Filter Layout Guidelines 1. Place 12V filter capacitor next to the IC and connect capacitor ground to pin 11 (PGnd). 2. Connect pin 11 (PGnd) with a separate trace to the ground terminals of the 5V input capacitors. 3. Place fast feedback filter capacitor next to pin 8 (VFFB) and connect its ground terminal with a separate, wide trace directly to pin 14 (LGnd). 4. Connect the ground terminals of the Compensation capacitor directly to the ground of the fast feedback filter capacitor to prevent common mode noise from effecting the PWM comparator. 5. Place the output filter capacitor(s) as close to the load as possible and connect the ground terminal to pin 14 (LGnd). Output Inductor The inductor should be selected based on its inductance, current capability, and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. Thermal Management Thermal Considerations for Power MOSFETs and Diodes In order to maintain good reliability, the junction temperature of the semiconductor components should be kept to a maximum of 150°C or lower. The thermal impedance (junction to ambient) required to meet this requirement can be calculated as follows: 6. To implement adaptive voltage positioning, connect both slow and fast feedback pins 16 (VFB) and 8 (VFFB) to the regulator output right at the inductor terminal. Connect inductor to the output capacitors via a trace with the following resistance: 11 CS5151 Applications Information: continued CS5151 Applications Information: continued RTRACE = 80mV IMAX This causes the output voltage to be +40mV with no load, and -40mV with a full load, improving regulator transient response. This trace must be wide enough to carry the full output current. (Typical trace is 1.0 inch long, 0.17 inch wide). Care should be taken to minimize any additional losses after the feedback connection point to maximize regulation. 7. If DC regulation is to be optimized (at the expense of degraded transient regulation), adaptive voltage positioning can be disabled by connecting to VFB pin directly to the load with a separate trace (remote sense). 8. Place 5V input capacitors close to the switching MOSFET. Route gate drive signal VGATE (pin 10) with a trace that is a minimum of 0.025 inches wide. VCC 0.1µF To the negative terminal of the input capacitors 15 11 1.0µF VCOMP 8 100pF VFFB 5 SOFTSTART OFF TIME To the negative terminal of the output capacitors Figure 18: Layout Guidelines 12 CS5151 Additional Application Circuits 5V 3.3V 12V 0.1µF MBRS 120 1µF MBRS120 + 1µF MBRS120 + 100µF/10V x 3 Tantalum Si9410 Si4410DY 1µF VCC2 VCC1 VCC1 VGATE 3µH 3.3V/10A VID1 VID2 2 CS5151 COFF SS 0.1µF 3.3k 0.33µF + 100µF/10V x 3 Tantalum 5V MBRS120 + 1µF 100µF/10V x 3 Tantalum Si4410 VCC2 VGATE 3µH Remote Sense 3.3V/10A VID0 VID1 VID2 VID3 VFB 10Ω CS5151 2 + 100µF/10V x 3 Tantalum MBR1535CT COFF 1,3 330pF SS 0.1µF PGnd COMP LGnd 0.33µF VFFB VFFB 3.3k 100pF Figure 21: 3.3V to 2.5V/7A converter with 12V bias. Figure 19: 5V to 3.3V/10A converter. VCC1 PGnd COMP LGnd 100pF MBRS120 1,3 SS VFFB 3.3k 100pF 2.5V/7A + MBR1535CT COFF 0.33µF 1µF VFB 330pF VFB LGnd 0.1µF CS5151 2 PGnd COMP MBRS 120 5µH 1,3 330pF 0.1µF VGATE VID3 MBR1535CT VID3 VCC2 VID0 VID0 VID1 VID2 33µF/25V x 3 Tantalum Connect to other circuits for current sharing Figure 20: 5V to 3.3V/10A converter with current sharing. 13 100µF/10V x 2 Tantalum CS5151 Package Specification PACKAGE THERMAL DATA PACKAGE DIMENSIONS IN mm (INCHES) D Lead Count Metric Max Min 10.00 9.80 19.69 18.67 16L SO Narrow 16L PDIP Thermal Data English Max Min .394 .386 .775 .735 RΘJC RΘJA 16L SO Narrow 28 115 typ typ 16L PDIP 42 80 ˚C/W ˚C/W Surface Mount Narrow Body (D); 150 mil wide 4.00 (.157) 3.80 (.150) 6.20 (.244) 5.80 (.228) 0.51 (.020) 0.33 (.013) 1.27 (.050) BSC 1.75 (.069) MAX 1.57 (.062) 1.37 (.054) 1.27 (.050) 0.40 (.016) 0.25 (.010) 0.19 (.008) D 0.25 (0.10) 0.10 (.004) REF: JEDEC MS-012 Plastic DIP (N); 300 mil wide 7.11 (.280) 6.10 (.240) 8.26 (.325) 7.62 (.300) 1.77 (.070) 1.14 (.045) 2.54 (.100) BSC 3.68 (.145) 2.92 (.115) .356 (.014) .203 (.008) 0.39 (.015) MIN. .558 (.022) .356 (.014) D REF: JEDEC MS-001 Some 8 and 16 lead packages may have 1/2 lead at the end of the package. All specs are the same. Ordering Information Part Number CS5151GD16 CS5151GDR16 CS5151GN16 Rev. 1/5/99 Description 16L SO Narrow 16L SO Narrow (tape & reel) 16L PDIP Cherry Semiconductor Corporation reserves the right to make changes to the specifications without notice. Please contact Cherry Semiconductor Corporation for the latest available information. 14 © 1999 Cherry Semiconductor Corporation