a High Output Current Differential Driver AD815 FEATURES Flexible Configuration Differential Input and Output Driver or Two Single-Ended Drivers High Output Power Power Package 26 dBm Differential Line Drive for ADSL Application 40 V p-p Differential Output Voltage, RL = 50 ⍀ 500 mA Minimum Output Drive/Amp, RL = 5 ⍀ Thermally Enhanced SOIC 400 mA Minimum Output Drive/Amp, RL = 10 ⍀ Low Distortion –66 dB @ 1 MHz THD, R L = 200 ⍀, VOUT = 40 V p-p 0.05% and 0.45ⴗ Differential Gain and Phase, RL = 25 ⍀ (6 Back-Terminated Video Loads) High Speed 120 MHz Bandwidth (–3 dB) 900 V/s Differential Slew Rate 70 ns Settling Time to 0.1% Thermal Shutdown APPLICATIONS ADSL, HDSL and VDSL Line Interface Driver Coil or Transformer Driver CRT Convergence and Astigmatism Adjustment Video Distribution Amp Twisted Pair Cable Driver PRODUCT DESCRIPTION The AD815 consists of two high speed amplifiers capable of supplying a minimum of 500 mA. They are typically configured as a differential driver enabling an output signal of 40 V p-p on ± 15 V supplies. This can be increased further with the use of a TOTAL HARMONIC DISTORTION – dBc –40 –50 VS = 615V G = +10 VOUT = 40V p-p FUNCTIONAL BLOCK DIAGRAM 15-Lead Through-Hole SIP (Y) and Surface-Mount DDPAK(VR) TAB IS +VS 15 NC 14 NC 13 NC 12 11 NC 10 –IN2 9 8 OUT2 +VS AD815 +IN2 7 –VS 6 OUT1 5 –IN1 4 3 +IN1 2 NC 1 NC NC NC = NO CONNECT REFER TO PAGE 3 FOR 24-LEAD SOIC PACKAGE coupling transformer with a greater than 1:1 turns ratio. The low harmonic distortion of –66 dB @ 1 MHz into 200 Ω combined with the wide bandwidth and high current drive make the differential driver ideal for communication applications such as subscriber line interfaces for ADSL, HDSL and VDSL. The AD815 differential slew rate of 900 V/µs and high load drive are suitable for fast dynamic control of coils or transformers, and the video performance of 0.05% and 0.45° differential gain and phase into a load of 25 Ω enable up to 12 back-terminated loads to be driven. Three package styles are available, and all work over the industrial temperature range (–40°C to +85°C). Maximum output power is achieved with the power package available for through-hole mounting (Y) and surface-mounting (VR). The 24-lead SOIC (RB) is capable of driving 26 dBm for full rate ADSL with proper heat sinking. +15V –60 1/2 AD815 100V –70 –80 AMP1 RL = 50V (DIFFERENTIAL) 499V RL = 200V (DIFFERENTIAL) –90 VIN = 4Vp-p 110V G = +10 RL 120V VD = 40Vp-p VOUT = 40Vp-p 499V –100 –110 100 R1 = 15V 100V 1k 10k 100k FREQUENCY – Hz 1M 1/2 AD815 10M Total Harmonic Distortion vs. Frequency R2 = 15V AMP2 1:2 TRANSFORMER –15V Subscriber Line Differential Driver REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 AD815–SPECIFICATIONS (@ T = +25ⴗC, V = ⴞ15 V dc, R A Model DYNAMIC PERFORMANCE Small Signal Bandwidth (–3 dB) Bandwidth (0.1 dB) Differential Slew Rate Settling Time to 0.1% NOISE/HARMONIC PERFORMANCE Total Harmonic Distortion Input Voltage Noise Input Current Noise (+IIN) Input Current Noise (–IIN) Differential Gain Error Differential Phase Error DC PERFORMANCE Input Offset Voltage S FB = 1 k⍀ and RLOAD = 100 ⍀ unless otherwise noted) Conditions VS Min G = +1 G = +1 G = +2 G = +2 VOUT = 20 V p-p, G = +2 10 V Step, G = +2 ± 15 ±5 ± 15 ±5 ± 15 ± 15 100 90 f = 1 MHz, RLOAD = 200 Ω, VOUT = 40 V p-p f = 10 kHz, G = +2 (Single Ended) f = 10 kHz, G = +2 f = 10 kHz, G = +2 NTSC, G = +2, RLOAD = 25 Ω NTSC, G = +2, RLOAD = 25 Ω AD815A Typ Max 120 110 40 10 900 70 MHz MHz MHz MHz V/µs ns ± 15 ± 5, ± 15 ± 5, ± 15 ± 5, ± 15 ± 15 ± 15 –66 1.85 1.8 19 0.05 0.45 dBc nV/√Hz pA/√Hz pA/√Hz % Degrees ±5 ± 15 5 10 ±5 ± 15 20 0.5 0.5 ± 5, ± 15 10 10 ± 5, ± 15 2 ± 5, ± 15 10 800 TMIN – TMAX Input Offset Voltage Drift Differential Offset Voltage TMIN – TMAX Differential Offset Voltage Drift –Input Bias Current TMIN – TMAX +Input Bias Current TMIN – TMAX Differential Input Bias Current TMIN – TMAX ± 5, ± 15 Open-Loop Transresistance TMIN – TMAX INPUT CHARACTERISTICS Differential Input Resistance Output Current1, 2 VR, Y RB-24 Short Circuit Current Output Resistance MATCHING CHARACTERISTICS Crosstalk POWER SUPPLY Operating Range3 Quiescent Current 2 4 5 90 150 5 5 75 100 5.0 mV mV mV µV/°C mV mV mV µV/°C µA µA µA µA µA µA MΩ MΩ ± 15 ± 15 ±5 ± 5, ± 15 ± 5, ± 15 57 80 ± 15 ±5 ± 15 ± 15 11.0 1.1 21 22.5 11.7 1.8 23 24.5 ±V ±V ±V ±V RLOAD = 10 Ω ± 15 ±5 ± 15 ± 15 ± 15 500 350 400 750 400 500 1.0 13 mA mA mA A Ω f = 1 MHz ± 15 –65 dB ±5 ± 15 ±5 ± 15 ± 5, ± 15 23 30 TMIN – TMAX TMIN – TMAX Single Ended, RLOAD = 25 Ω Differential, RLOAD = 50 Ω TMIN – TMAX RLOAD = 5 Ω TMIN – TMAX TMIN – TMAX Power Supply Rejection Ratio 8 15 30 7 15 1.4 13.5 3.5 65 100 Differential Input Capacitance Input Common-Mode Voltage Range Common-Mode Rejection Ratio Differential Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Voltage Swing 1.0 0.5 ± 15 +Input –Input Units TMIN – TMAX –55 –66 ± 18 30 40 40 55 MΩ Ω pF ±V ±V dB dB V mA mA mA mA dB NOTES 1Output current is limited in the 24-lead SOIC package to the maximum power dissipation. See absolute maximum ratings and derating curves. 2See Figure 12 for bandwidth, gain, output drive recommended operation range. 3Observe derating curves for maximum junction temperature. Specifications subject to change without notice. –2– REV. B AD815 ABSOLUTE MAXIMUM RATINGS 1 MAXIMUM POWER DISSIPATION Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V Total Internal Power Dissipation2 Plastic (Y and VR) . . 3.05 Watts (Observe Derating Curves) Small Outline (RB) . . 2.4 Watts (Observe Derating Curves) Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . ± VS Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V Output Short Circuit Duration . . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves Can Only Short to Ground Storage Temperature Range Y, VR and RB Package . . . . . . . . . . . . . . . –65°C to +125°C Operating Temperature Range AD815A . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C Lead Temperature Range (Soldering, 10 sec) . . . . . . . +300°C The maximum power that can be safely dissipated by the AD815 is limited by the associated rise in junction temperature. The maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature of the plastic, about 150°C. Exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175°C for an extended period can result in device failure. The AD815 has thermal shutdown protection, which guarantees that the maximum junction temperature of the die remains below a safe level, even when the output is shorted to ground. Shorting the output to either power supply will result in device failure. To ensure proper operation, it is important to observe the derating curves and refer to the section on power considerations. NOTES 1 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air with 0 ft/min air flow: 15-Lead Through-Hole and Surface Mount: θJA = 41°C/W; 24-Lead Surface Mount: θJA = 52°C/W. It must also be noted that in high (noninverting) gain configurations (with low values of gain resistor), a high level of input overdrive can result in a large input error current, which may result in a significant power dissipation in the input stage. This power must be included when computing the junction temperature rise due to total internal power. PIN CONFIGURATION 24-Lead Thermally-Enhanced SOIC (RB-24) 24 NC NC 2 23 NC NC 3 22 NC NC 4 21 NC 5 THERMAL HEAT TABS +VS* AD815 6 20 19 TOP VIEW 7 (Not to Scale) 18 8 MAXIMUM POWER DISSIPATION – Watts NC 1 14 THERMAL HEAT TABS +VS* 17 TJ = 1508C 13 θJA = 168C/W SOLDERED DOWN TO COPPER HEAT SINK (STILL AIR = 0FT/MIN) AD815 AVR, AY 12 11 10 9 8 7 6 5 4 3 θJA = 418C/W (STILL AIR = 0FT/MIN) NO HEAT SINK AD815 AVR, AY θJA = 528C/W +IN1 9 16 +IN2 –IN1 10 15 –IN2 1 OUT1 11 14 OUT2 0 –50 –40 –30 –20 –10 0 10 20 30 40 50 60 AMBIENT TEMPERATURE – 8C –VS 12 13 +VS NC = NO CONNECT 2 (STILL AIR = 0 FT/MIN) NO HEAT SINK AD815ARB-24 70 80 90 Plot of Maximum Power Dissipation vs. Temperature *HEAT TABS ARE CONNECTED TO THE POSITIVE SUPPLY. ORDERING GUIDE Model Temperature Range Package Description Package Option AD815ARB-24 AD815ARB-24-REEL AD815AVR AD815AY AD815AYS AD815-EB –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C 24-Lead Thermally Enhanced SOIC 24-Lead Thermally Enhanced SOIC 15-Lead Surface Mount DDPAK 15-Lead Through-Hole SIP with Staggered Leads and 90° Lead Form 15-Lead Through-Hole SIP with Staggered Leads and Straight Lead Form Evaluation Board RB-24 RB-24 VR-15 Y-15 YS-15 CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD815 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. B –3– WARNING! ESD SENSITIVE DEVICE AD815–Typical Performance Characteristics AD815 36 34 10 5 0 5 10 15 SUPPLY VOLTAGE – 6Volts VS = 65V 24 30 60 NO LOAD 20 RL = 50V (DIFFERENTIAL) RL = 25V (SINGLE-ENDED) 40 20 10 0 5 10 15 SUPPLY VOLTAGE – 6Volts 40 30 15 VS = 65V 20 10 27 24 21 2 10 4 6 8 10 12 SUPPLY VOLTAGE – 6Volts 16 14 SIDE A, B 0 INPUT BIAS CURRENT – mA 20 TA = +258C Figure 5. Total Supply Current vs. Supply Voltage DIFFERENTIAL OUTPUT VOLTAGE – Volts p-p 50 25 100 30 0 60 VS = 615V 80 18 0 20 Figure 2. Output Voltage Swing vs. Supply Voltage 30 0 20 40 60 JUNCTION TEMPERATURE – 8C 33 TOTAL SUPPLY CURRENT – mA 80 –20 Figure 4. Total Supply Current vs. Temperature DIFFERENTIAL OUTPUT VOLTAGE – V p-p SINGLE-ENDED OUTPUT VOLTAGE – V p-p 26 18 –40 20 40 0 SINGLE-ENDED OUTPUT VOLTAGE – Volts p-p 28 22 Figure 1. Input Common-Mode Voltage Range vs. Supply Voltage 5 30 20 0 10 VS = 615V 32 15 SUPPLY CURRENT – mA COMMON-MODE VOLTAGE RANGE – 6Volts 20 +I B VS = 615V, 65V –10 –20 VS = 65V –30 SIDE B –I B –40 SIDE A –50 SIDE B –60 SIDE A –70 –I B VS = 615V 0 0 10 100 1k 10k LOAD RESISTANCE – (Differential – V) (Single-Ended – V/2) –80 –40 Figure 3. Output Voltage Swing vs. Load Resistance –20 0 20 40 60 JUNCTION TEMPERATURE – 8C 80 100 Figure 6. Input Bias Current vs. Temperature –4– REV. B AD815 0 80 –2 60 –4 40 RTI OFFSET – mV INPUT OFFSET VOLTAGE – mV TA = 258C VS = 65V –6 –8 VS = 615V VS = 610V 20 VS = 65V 0 VIN 1/2 f = 0.1Hz 100V AD815 VOUT –20 –10 49.9V VS = 615V –12 –40 –14 –40 –60 –2.0 –1.6 –1.2 1kV 0 20 40 60 JUNCTION TEMPERATURE – 8C –20 80 100 Figure 7. Input Offset Voltage vs. Temperature RL= 5V 1kV 0 –0.8 –0.4 0.4 0.8 LOAD CURRENT – Amps 1.2 1.6 2.0 Figure 10. Thermal Nonlinearity vs. Output Current Drive 750 CLOSED-LOOP OUTPUT RESISTANCE – V SHORT CIRCUIT CURRENT – mA VS = 615V 700 SOURCE 650 600 SINK 550 500 450 –60 –40 –20 0 20 40 60 80 100 JUNCTION TEMPERATURE – 8C 120 100 10 VS = 65V VS = 615V 1 0.1 0.01 140 30k Figure 8. Short Circuit Current vs. Temperature 100k 300k 1M 3M 10M FREQUENCY – Hz 30M 100M 300M Figure 11. Closed-Loop Output Resistance vs. Frequency RTI OFFSET – mV 10 DIFFERENTIAL OUTPUT VOLTAGE – V p-p 15 VS = 610V TA = 258C RL = 25V VS = 65V VS = 615V 5 0 VIN 1/2 f = 0.1Hz 100V AD815 VOUT –5 49.9V –10 1kV 1kV –15 –20 RL= 25V RL = 100V 30 RL = 50V 20 RL = 25V 10 RL = 1V 0 –16 –12 –8 –4 0 4 VOUT – Volts 8 12 16 0 20 Figure 9. Gain Nonlinearity vs. Output Voltage REV. B TA = 258C VS = ±15V 40 2 4 10 6 8 FREQUENCY – MHz 12 14 Figure 12. Large Signal Frequency Response –5– AD815 100 100 120 TRANSIMPEDANCE 10 NONINVERTING INPUT CURRENT NOISE INPUT VOLTAGE NOISE 1 10 100 1k FREQUENCY – Hz 100 500 80 0 –50 60 –100 50 –150 40 –200 30 –250 100 Figure 13. Input Current and Voltage Noise vs. Frequency 80 TOTAL HARMONIC DISTORTION – dBc COMMON-MODE REJECTION – dB 10k 100k 1M FREQUENCY – Hz 10M 100M –40 VS = 615V 70 SIDE B 60 SIDE A 50 562V 562V VOUT VIN 30 562V 1/2 AD815 562V 20 10 10k 100k 1M FREQUENCY – Hz –50 –60 –70 –80 1k 10k 100k FREQUENCY – Hz 1M 10M Figure 17. Total Harmonic Distortion vs. Frequency 10 VS = 615V G = +2 RL = 100V –30 –40 –PSRR –50 +PSRR –60 –70 –80 –90 –100 0.01 RL = 200V (DIFFERENTIAL) –100 OUTPUT SWING FROM ±V TO 0 – Volts –20 RL = 50V (DIFFERENTIAL) –90 0 –10 VS = 615V G = +10 VOUT = 40V p-p –110 100 100M 10M Figure 14. Common-Mode Rejection vs. Frequency PSRR – dB 1k Figure 16. Open-Loop Transimpedance vs. Frequency 90 40 PHASE 70 1 100k 10k 100 90 PHASE – Degrees 10 TRANSIMPEDANCE – dB INVERTING INPUT CURRENT NOISE CURRENT NOISE – pA/ √ Hz VOLTAGE NOISE – nV/ √ Hz 110 8 1% 0.1% 6 GAIN = +2 VS = 615V 4 2 0 –2 –4 –6 1% 0.1% –8 –10 0.1 1 10 FREQUENCY – MHz 100 300 0 Figure 15. Power Supply Rejection vs. Frequency 20 60 40 70 SETTLING TIME – ns 80 100 Figure 18. Output Swing and Error vs. Settling Time –6– REV. B AD815 700 1400 5 1200 500 1000 G = +2 400 800 300 600 200 400 100 200 0 OPEN-LOOP TRANSRESISTANCE – MV 600 DIFFERENTIAL SLEW RATE – V/ms SINGLE-ENDED SLEW RATE – V/ms (PER AMPLIFIER) G = +10 0 0 5 10 15 OUTPUT STEP SIZE – V p-p 20 4 SIDE B Figure 19. Slew Rate vs. Output Step Size +TZ SIDE A 2 –TZ SIDE B 1 0 –40 25 SIDE A 3 –20 0 20 40 60 JUNCTION TEMPERATURE – 8C 100 Figure 22. Open-Loop Transresistance vs. Temperature –85 15 VS = 615V VS = 615V SIDE B –80 14 OUTPUT SWING – Volts +PSRR SIDE A PSRR – dB 80 –75 –70 SIDE A SIDE B –65 RL = 150V +VOUT | –VOUT | 13 +VOUT RL = 25V 12 | –VOUT | 11 –PSRR –60 –40 –20 0 20 40 60 JUNCTION TEMPERATURE – 8C 80 10 –40 100 Figure 20. PSRR vs. Temperature –20 0 20 40 60 JUNCTION TEMPERATURE – 8C 80 100 Figure 23. Single-Ended Output Swing vs. Temperature 27 –74 –73 26 OUTPUT SWING – Volts CMRR – dB –72 –71 –70 –69 –CMRR VS = 615V RL = 50V 25 –VOUT +VOUT 24 –68 –66 –40 23 +CMRR –67 –20 0 20 40 60 JUNCTION TEMPERATURE – 8C 80 22 –40 100 0 20 40 60 JUNCTION TEMPERATURE – 8C 80 100 Figure 24. Differential Output Swing vs. Temperature Figure 21. CMRR vs. Temperature REV. B –20 –7– AD815 2 3 4 5 6 7 8 9 2 BACK TERMINATED LOADS (75V) 0.010 0.005 0.000 –0.005 –0.010 –0.015 –0.020 –0.025 –0.030 PHASE GAIN GAIN 1 2 PHASE 3 4 5 6 7 8 9 0.12 0.10 0.08 G = +2 0.06 RF = 1kV 0.04 NTSC 0.02 0.00 –0.02 –0.04 10 11 –3 –4 B VIN –0.3 –5 100V –0.4 VOUT –6 65V 49.9V –0.5 499V 499V 100V –7 –8 –0.6 1 10 FREQUENCY – MHz 100 –9 300 Figure 28. Bandwidth vs. Frequency, G = +2 1 VS = 615V G = +2 RF = 499V VS = 615V, 65V VIN = 400mVrms RL = 100V SIDE B –50 –60 –70 SIDE A –80 –90 –100 –110 0.03 A A –0.2 NORMALIZED OUTPUT VOLTAGE – dB CROSSTALK – dB –40 –2 B –0.1 –10 –30 –1 0 –0.7 0.1 Figure 25. Differential Gain and Differential Phase (per Amplifier) –20 615V 0.1 0 NORMALIZED FREQUENCY RESPONSE – dB 1 65V NORMALIZED FLATNESS – dB GAIN 1 615V DIFF PHASE – Degrees PHASE 0.5 0.4 0.3 G = +2 RF = 1kV 0.2 0.1 NTSC 0.0 –0.1 –0.2 –0.3 10 11 DIFF PHASE – Degrees DIFF GAIN – % DIFF GAIN – % 6 BACK TERMINATED LOADS (25V) 0.04 0.03 0.02 0.01 0.00 –0.01 –0.02 –0.03 –0.04 0.1 1 10 FREQUENCY – MHz 100 0 SIDE A SIDE B –1 –2 –3 VIN 100V 49.9V –5 124V Figure 26. Output-to-Output Crosstalk vs. Frequency 499V 100V –6 –7 0.1 300 VOUT –4 1 10 FREQUENCY – MHz 100 300 Figure 29. –3 dB Bandwidth vs. Frequency, G = +5 2 1 VS = 615V VIN = 0 dBm SIDE B OUTPUT VOLTAGE – dB 0 100 90 SIDE A –1 –2 –3 VIN 100V VOUT –4 –5 49.9V 562V 10 100V 0% –6 5V –7 –9 0.1 1 10 FREQUENCY – MHz 100 1ms 300 Figure 30. 40 V p-p Differential Sine Wave, RL = 50 Ω, f = 100 kHz Figure 27. –3 dB Bandwidth vs. Frequency, G = +1 –8– REV. B AD815 RF 562V 10mF +15V 0.1mF 0.1mF RS 8 8 1/2 AD815 100V VIN 7 PULSE GENERATOR 50V 1/2 AD815 0.1mF 50V Figure 35. Test Circuit, Gain = 1 + R F /RS G = +1 RF = 698V RL = 100V G = +5 RF = 562V RL = 100V RS = 140V SIDE A SIDE B SIDE B 5V 20ns 100ns Figure 36. 20 V Step Response, G = +5 Figure 32. 500 mV Step Response, G = +1 SIDE A RL = 100V 10mF –15V TR/TF = 250ps Figure 31. Test Circuit, Gain = +1 100mV 0.1mF 7 PULSE GENERATOR –15V SIDE A 100V VIN RL = 100V 10mF TR/TF = 250ps 10mF +15V G = +1 RF = 562V RL = 100V 562V 10mF +15V SIDE B 0.1mF 562V 8 VIN PULSE GENERATOR 55V 1/2 AD815 100V 0.1mF 7 TR/TF = 250ps RL = 100V 10mF –15V 1V 20ns Figure 33. 4 V Step Response, G = +1 Figure 37. Test Circuit, Gain = –1 SIDE A SIDE A G = +1 RF = 562V RL = 100V SIDE B SIDE B 2V 100mV 50ns Figure 34. 10 V Step Response, G = +1 REV. B G = –1 RF = 562V RL = 100V 20ns Figure 38. 500 mV Step Response, G = –1 –9– AD815 Choice of Feedback and Gain Resistors SIDE A The fine scale gain flatness will, to some extent, vary with feedback resistance. It therefore is recommended that once optimum resistor values have been determined, 1% tolerance values should be used if it is desired to maintain flatness over a wide range of production lots. Table I shows optimum values for several useful configurations. These should be used as starting point in any application. G = –1 RF = 562V RL = 100V SIDE B Table I. Resistor Values 1V 20ns G= Figure 39. 4 V Step Response, G = –1 THEORY OF OPERATION The AD815 is a dual current feedback amplifier with high (500 mA) output current capability. Being a current feedback amplifier, the AD815’s open-loop behavior is expressed as transimpedance, ∆VO/∆I–IN , or TZ . The open-loop transimpedance behaves just as the open-loop voltage gain of a voltage feedback amplifier, that is, it has a large dc value and decreases at roughly 6 dB/octave in frequency. 562 499 499 499 1k ⬁ 499 499 125 110 As to be expected for a wideband amplifier, PC board parasitics can affect the overall closed-loop performance. Of concern are stray capacitances at the output and the inverting input nodes. If a ground plane is to be used on the same side of the board as the signal traces, a space (5 mm min) should be left around the signal lines to minimize coupling. T Z (S ) VO =G× VIN T Z (S ) + G × RIN + RF POWER SUPPLY BYPASSING Adequate power supply bypassing can be critical when optimizing the performance of a high frequency circuit. Inductance in the power supply leads can form resonant circuits that produce peaking in the amplifier’s response. In addition, if large current transients must be delivered to the load, then bypass capacitors (typically greater than 1 µF) will be required to provide the best settling time and lowest distortion. A parallel combination of 10.0 µF and 0.1 µF is recommended. Under some low frequency applications, a bypass capacitance of greater than 10 µF may be necessary. Due to the large load currents delivered by the AD815, special consideration must be given to careful bypassing. The ground returns on both supply bypass capacitors as well as signal common must be “star” connected as shown in Figure 41. where: RF RG R IN = 1/gM ≈ 25 Ω G = 1+ RF RG RIN RG (⍀) PRINTED CIRCUIT BOARD LAYOUT CONSIDERATIONS Since RIN is proportional to 1/gM, the equivalent voltage gain is just TZ × gM , where the gM in question is the transconductance of the input stage. Using this amplifier as a follower with gain, Figure 40, basic analysis yields the following result: RN +1 –1 +2 +5 +10 RF (⍀) VOUT +VS VIN +IN Figure 40. Current Feedback Amplifier Operation RF Recognizing that G × RIN << RF for low gains, it can be seen to the first order that bandwidth for this amplifier is independent of gain (G). RG (OPTIONAL) +OUT RF –OUT Considering that additional poles contribute excess phase at high frequencies, there is a minimum feedback resistance below which peaking or oscillation may result. This fact is used to determine the optimum feedback resistance, RF. In practice parasitic capacitance at the inverting input terminal will also add phase in the feedback loop, so picking an optimum value for RF can be difficult. –IN –VS Figure 41. Signal Ground Connected in “Star” Configuration Achieving and maintaining gain flatness of better than 0.1 dB at frequencies above 10 MHz requires careful consideration of several issues. –10– REV. B AD815 There are three major noise and offset terms to consider in a current feedback amplifier. For offset errors refer to the equation below. For noise error the terms are root-sum-squared to give a net output error. In the circuit below (Figure 42), they are input offset (VIO) which appears at the output multiplied by the noise gain of the circuit (1 + RF/RG), noninverting input current (IBN × RN) also multiplied by the noise gain, and the inverting input current, which when divided between RF and R G and subsequently multiplied by the noise gain always appear at the output as IBI × RF. The input voltage noise of the AD815 is less than 2 nV/√Hz. At low gains though, the inverting input current noise times RF is the dominant noise source. Careful layout and device matching contribute to better offset and drift specifications for the AD815 compared to many other current feedback amplifiers. The typical performance curves in conjunction with the equations below can be used to predict the performance of the AD815 in any application. Figure 44 gives the relationship between output voltage swing into various loads and the power dissipated by the AD815 (PIN). This data is given for both sine wave and square wave (worst case) conditions. It should be noted that these graphs are for mostly resistive (phase < ± 10°) loads. When the power dissipation requirements are known, Equation 1 and the graph on Figure 45 can be used to choose an appropriate heat sinking configuration. RL = 50V f = 1kHz 4 SQUARE WAVE SINE WAVE PIN – Watts DC ERRORS AND NOISE 3 RL = 100V 2 RL = 200V 1 R R VOUT = VIO × 1 + F ± IBN × RN × 1 + F ± IBI × RF RG RG 10 RF I BN RN 40 Figure 44. Total Power Dissipation vs. Differential Output Voltage I BI RG 20 30 VOUT – Volts p-p Normally, the AD815 will be soldered directly to a copper pad. Figure 45 plots θJA against size of copper pad. This data pertains to copper pads on both sides of G10 epoxy glass board connected together with a grid of feedthroughs on 5 mm centers. VOUT Figure 42. Output Offset Voltage POWER CONSIDERATIONS The 500 mA drive capability of the AD815 enables it to drive a 50 Ω load at 40 V p-p when it is configured as a differential driver. This implies a power dissipation, PIN, of nearly 5 watts. To ensure reliability, the junction temperature of the AD815 should be maintained at less than 175°C. For this reason, the AD815 will require some form of heat sinking in most applications. The thermal diagram of Figure 43 gives the basic relationship between junction temperature (TJ) and various components of θ JA. TJ = TA + PIN θJA This data shows that loads of 100 ohms or less will usually not require any more than this. This is a feature of the AD815’s 15-lead power SIP package. An important component of θJA is the thermal resistance of the package to heatsink. The data given is for a direct soldered connection of package to copper pad. The use of heatsink grease either with or without an insulating washer will increase this number. Several options now exist for dry thermal connections. These are available from Bergquist as part # SP600-90. Consult with the manufacturer of these products for details of their application. Equation 1 35 θ A (JUNCTION TO TJ 30 DIE MOUNT) θJA – 8C/W AD815AVR, AY (θJC = 28C/W) θ B (DIE MOUNT TO CASE) TA θ A + θ B = θ JC CASE TJ PIN θ CA θ JC θ JA 20 15 TA WHERE: PIN = DEVICE DISSIPATION TA = AMBIENT TEMPERATURE TJ = JUNCTION TEMPERATURE θ JC = THERMAL RESISTANCE – JUNCTION TO CASE θ CA = THERMAL RESISTANCE – CASE TO AMBIENT 10 0 0.5k 1k 1.5k 2k 2.5k COPPER HEAT SINK AREA (TOP AND BOTTOM) – mm2 Figure 45. Power Package Thermal Resistance vs. Heat Sink Area Figure 43. A Breakdown of Various Package Thermal Resistances REV. B 25 –11– AD815 Other Power Considerations There are additional power considerations applicable to the AD815. First, as with many current feedback amplifiers, there is an increase in supply current when delivering a large peak-to-peak voltage to a resistive load at high frequencies. This behavior is affected by the load present at the amplifier’s output. Figure 12 summarizes the full power response capabilities of the AD815. These curves apply to the differential driver applications (e.g., Figure 49 or Figure 53). In Figure 12, maximum continuous peak-to-peak output voltage is plotted vs. frequency for various resistive loads. Exceeding this value on a continuous basis can damage the AD815. The AD815 is equipped with a thermal shutdown circuit. This circuit ensures that the temperature of the AD815 die remains below a safe level. In normal operation, the circuit shuts down the AD815 at approximately 180°C and allows the circuit to turn back on at approximately 140°C. This built-in hysteresis means that a sustained thermal overload will cycle between power-on and power-off conditions. The thermal cycling typically occurs at a rate of 1 ms to several seconds, depending on the power dissipation and the thermal time constants of the package and heat sinking. Figures 46 and 47 illustrate the thermal shutdown operation after driving OUT1 to the + rail, and OUT2 to the – rail, and then short-circuiting to ground each output of the AD815. The AD815 will not be damaged by momentary operation in this state, but the overload condition should be removed. resistor should be placed in series with each output. See Figure 48. This circuit can deliver 800 mA into loads of up to 12.5 Ω. 499V 499V +15V 0.1mF 5 100V 10mF 8 1V 1/2 AD815 6 4 50V 499V 499V RL 10 100V 1V 1/2 AD815 11 9 7 0.1mF 10mF –15V Figure 48. Parallel Operation for High Current Output Differential Operation Various circuit configurations can be used for differential operation of the AD815. If a differential drive signal is available, the two halves can be used in a classic instrumentation configuration to provide a circuit with differential input and output. The circuit in Figure 49 is an illustration of this. With the resistors shown, the gain of the circuit is 11. The gain can be changed by changing the value of RG. This circuit, however, provides no common-mode rejection. OUT 1 +15V 100 90 +IN 0.1mF 100V 4 OUT 2 8 1/2 AD815 5 VIN 200ms 5V Figure 46. OUT2 Shorted to Ground, Square Wave Is OUT1, RF = 1 kΩ, R G = 222 Ω RG 100V RL 10 –IN OUT 1 6 RF 499V 10 0% 10mF 100V 11 VOUT RF 499V 1/2 AD815 OUT 2 9 7 0.1mF 10mF –15V 100 90 Figure 49. Fully-Differential Operation OUT 1 Creating Differential Signals If only a single ended signal is available to drive the AD815 and a differential output signal is desired, several circuits can be used to perform the single-ended-to-differential conversion. OUT 2 10 0% 5V 5ms Figure 47. OUT1 Shorted to Ground, Square Wave Is OUT2, RF = 1 kΩ, R G = 222 Ω Parallel Operation To increase the drive current to a load, both of the amplifiers within the AD815 can be connected in parallel. Each amplifier should be set for the same gain and driven with the same signal. In order to ensure that the two amplifiers share current, a small One circuit to perform this is to use a dual op amp as a predriver that is configured as a noninverter and inverter. The circuit shown in Figure 50 performs this function. It uses an AD826 dual op amp with the gain of one amplifier set at +1 and the gain of the other at –1. The 1 kΩ resistor across the input terminals of the follower makes the noise gain (NG = 1) equal to the inverter’s. The two outputs then differentially drive the inputs to the AD815 with no common-mode signal to first order. –12– REV. B AD815 +15V +15V +15V 0.1mF 0.1mF 100V 4 3 1kV 1/2 AD826 VIN 1/2 AD815 8 1 5 2 10mF 8 6 RF 499V 1kV 5 10 1/2 AD826 4 7 100V 0.1mF 11 10 9 AMP 2 7 11 0.1mF –15V +15V 6 5 1kV 50V 200V RL 1kV 10 100V 11 1/2 AD815 9 10mF –15V Figure 51. Differential Driver with Transformer Input Direct Single-Ended-to-Differential Conversion Two types of circuits can create a differential output signal from a single-ended input without the use of any other components than resistors. The first of these is illustrated in Figure 52. REV. B This circuit can work at various gains with proper resistor selection. But in general, in order to change the gain of the circuit, at least two resistor values will have to be changed. In addition, the noise gain of the two op amps in this configuration will always be different by one, so the bandwidths will not match. Each of the AD815’s op amps is configured as a unity gain follower by the feedback resistors (RA). Each op amp output also drives the other as a unity gain inverter via the two RB s, creating a totally symmetrical circuit. 7 0.1mF When the + input of Amp 1 is driven with a signal, the same signal appears at the – input of Amp 1. This signal serves as an input to Amp 2 configured for a gain of –5, (–RF2/R G). Thus the two outputs move in opposite directions with the same gain and create a balanced differential signal. A second circuit that has none of the disadvantages mentioned in the above circuit creates a differential output voltage feedback op amp out of the pair of current feedback op amps in the AD815. This circuit, drawn in Figure 53, can be used as a high power differential line driver, such as required for ADSL (asymmetrical digital subscriber loop) line driving. 10mF 8 1/2 AD815 9 7 Amp 1 has its + input driven with the input signal, while the + input of Amp 2 is grounded. Thus the – input of Amp 2 is driven to virtual ground potential by its output. Therefore Amp 1 is configured for a noninverting gain of five, (1 + RF1/RG), because RG is connected to the virtual ground of Amp 2’s – input. One advantage of using a transformer is its ability to provide isolation between circuit sections and to provide good commonmode rejection. The disadvantages are that transformers have no dc response and can sometimes be large, heavy, and expensive. This circuit is shown in Figure 51. 50V 1/2 AD815 Figure 52. Direct Single-Ended-to-Differential Conversion Another means for creating a differential signal from a singleended signal is to use a transformer with a center-tapped secondary. The center tap of the transformer is grounded and the two secondary windings are connected to obtain opposite polarity signals to the two inputs of the AD815 amplifiers. The bias currents for the AD815 inputs are provided by the center tap ground connection through the transformer windings. 0.1mF RF2 499V –15V Figure 50. Differential Driver with Single-Ended Differential Converter 4 VOUT 10mF –15V 100V RL RF 499V 1/2 AD815 6 RF1 402V RG 100V RL 1kV 6 8 1/2 AD815 5 RG 100V 1kV 4 AMP 1 If the + input to Amp 2 is grounded and a small positive signal is applied to the + input of Amp 1, the output of Amp 1 will be driven to saturation in the positive direction and the output of Amp 2 driven to saturation in the negative direction. This is similar to the way a conventional op amp behaves without any feedback. –13– AD815 ~20pF Twelve Channel Video Distribution Amplifier RF 499V The high current of the AD815 enables it to drive up to twelve standard 75 Ω reverse terminated video loads. Figure 54 is a schematic of such an application. +15V RI 499V VIN 0.1mF VCC 4 50V 1/2 AD815 AMP1 The input video signal is terminated in 75 Ω and applied to the noninverting inputs of both amplifiers of the AD815. Each amplifier is configured for a gain of two to compensate for the divide-by-two feature of each cable termination. Six separate 75 Ω resistors for each amplifier output are used for the cable back termination. In this manner, all cables are relatively independent of each other and small disturbances on any cable will not have an effect on the other cables. (OPTIONAL) 10mF 8 6 5 RA 499V 250 (50V) (OPTIONAL) RB 499V 10 AMP2 1/2 AD815 11 7 100V RB 499V RA 499V When driving six video cables in this fashion, the load seen by each amplifier output is resistive and is equal to 150 Ω/6 or 25 Ω. The differential gain is 0.05% and the differential phase is 0.45°. 50V 9 VCC +15V 10mF 0.1mF 0.1mF –15V 499V 10mF 12 3 75V Figure 53. Single-Ended-to-Differential Driver 499V If a resistor (RF) is connected from the output of Amp 2 to the + input of Amp 1, negative feedback is provided which closes the loop. An input resistor (RI) will make the circuit look like a conventional inverting op amp configuration with differential outputs. The inverting input to this dual output op amp becomes Pin 4, the positive input of Amp 1. 5 8 6 100V 4 12 3 VIDEO OUT TO 75V CABLES AD815 VIDEO IN 75V 100V The gain of this circuit from input to either output will be ± RF/ RI. Or the single-ended-to-differential gain will be 2 × RF/RI. 11 9 10 The differential outputs can be applied to the primary of a transformer. If each output can swing ± 10 V, the effective swing on the transformer primary is 40 V p-p. The optional capacitor can be added to prevent any dc current in the transformer due to dc offsets at the output of the AD815. 499V 7 499V 10mF 0.1mF –15V Figure 54. AD815 Video Distribution Amp Driving 12 Video Cables C1 B2 +15V TP4 TP3 B1 TP2 B3 +15V –15V J1 C2 0.1mF R1 4 R19 R5 1/2 AD815 5 T1 C3 10mF 8 J5 R3 2 C6 R7 6 R8 U1 R6 R17 R4 1 J7 C13 2 10 C9 –15V J4 2 3 R15 2 3 R20 5 R22 3 R16 TP1 1 7 1 R2 J2 4 1 C10 0.1mF R14 9 6 11 8 12 JP1 J3 R10 11 R9 R21 R12 7 1/2 AD815 10 U1 C11 10mF R13 J6 9 R11 R18 Figure 55. AD815 Evaluation Board Schematic –14– REV. B AD815 Figure 56. AD815 AVR Evaluation Board Assembly Drawing Figure 58. AD815 AVR Evaluation Board Layout (Solder Side) Figure 57. AD815 AVR Evaluation Board Layout (Component Side) REV. B –15– AD815 OUTLINE DIMENSIONS 15-Lead Through-Hole SIP with Staggered Leads and 90ⴗ Lead Form (Y-15) 8° 0° 0.079 (2.006) DIA 2 PLACES 0.600 (15.24) BSC 0.146 (3.70) 0.138 (3.50) 0.063 (1.60) 0.057 (1.45) 0.394 (10.007) 1 15 PIN 1 0.024 (0.61) 0.014 (0.36) 0.798 (20.27) 0.778 (19.76) 0.079 (2.006) DIA 2 PLACES 0.798 (20.27) 0.778 (19.76) 0.666 ±0.006 (16.916 ±0.152) LONG LEAD 0.024 (0.61) 0.014 (0.36) 0.182 (4.62) 0.172 (4.37) 0.182 (4.62) 0.172 (4.37) 0.100 (2.54) BSC 0.671 ±0.006 (17.043 ±0.152) SHORT 0.080 (2.03) LEAD 0.065 (1.65) 2 PLACES 0.042 (1.066) TYP 0.691 ±0.010 (17.551 ±0.254) 0.766 ±0.010 (19.456 ±0.254) 0.791 ±0.010 (20.091 ±0.254) PIN 1 0.694 (17.63) 0.684 (17.37) 0.426 (10.82) 0.416 (10.57) 0.080 (2.03) 0.065 (1.65) 2 PLACES 15 0.152 (3.86) 0.148 (3.76) 0.137 (3.479) TYP 0.088 (2.24) 0.068 (1.72) 0.694 (17.63) 0.684 (17.37) 0.137 (3.479) 0.516 TYP (13.106) 0.042 (1.066) TYP 1 0.110 (2.79) BSC 0.063 (1.60) 0.057 (1.45) 0.394 (10.007) 0.426 (10.82) 0.416 (10.57) 0.110 (2.79) 0.152 (3.86) BSC 0.148 (3.76) 0.516 (13.106) 15-Lead Surface Mount DDPAK (VR-15) C2106a–0–12/99 Dimensions shown in inches and (mm). 0.209 ±0.010 (5.308 ±0.254) SEATING PLANE 0.031 (0.79) SEATING 0.024 (0.60) PLANE 0.050 (1.27) BSC 0.031 (0.79) 0.024 (0.60) 0.700 (17.78) BSC 24-Lead Thermally Enhanced SOIC (RB-24) 15-Lead Through-Hole SIP with Staggered Leads and Straight Lead Form (YS-15) PIN 1 0.1043 (2.65) 0.0926 (2.35) 0.0291 (0.74) x 45° 0.0098 (0.25) 0.063 (1.60) 0.057 (1.45) 0.394 (10.007) 0.137 (3.48) TYP 0.042 (1.07) TYP 1 15 0.080 (2.03) 0.065 (1.65) 2 PLACES PIN 1 0.0118 (0.30) 0.0040 (0.10) 0.0500 (1.27) BSC 8° 0.0201 (0.51) 0° SEATING 0.0125 (0.32) 0.0130 (0.33) PLANE 0.0091 (0.23) 0.0500 (1.27) 0.0157 (0.40) 0.700 (17.78) BSC 0.079 (2.007) DIA 2 PLACES 0.601 ±0.010 (15.265 0.710 (18.03) ±0.254) 0.690 (17.53) LONG LEAD 0.627 ±0.010 (15.926 ±0.254) SHORT LEAD 0.024 (0.61) 0.014 (0.36) 0.798 (20.27) 0.778 (19.76) 0.176 (4.47) 0.150 (3.81) 0.169 0.200 (4.29) (5.08) BSC BSC PRINTED IN U.S.A. 12 0.694 (17.63) 0.684 (17.37) 0.426 (10.82) 0.416 (10.57) 1 0.4193 (10.65) 0.3937 (10.00) 13 0.2992 (7.60) 0.2914 (7.40) 24 0.110 0.152 (3.86) (2.79) BSC 0.148 (3.76) 0.516 (13.106) 0.6141 (15.60) 0.5985 (15.20) 0.182 (4.62) 0.172 (4.37) 0.050 (1.27) BSC –16– SEATING PLANE 0.031 (0.79) 0.024 (0.60) REV. B