LTC3406AB 1.5MHz, 600mA Synchronous Step-Down Regulator in ThinSOT DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ High Efficiency: Up to 96% 600mA Output Current 2.5V to 5.5V Input Voltage Range 1.5MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 200μA ±2% 0.6V Reference Shutdown Mode Draws <1μA Supply Current Internal Soft-Start Limits Inrush Current Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM Package APPLICATIONS ■ ■ ■ ■ ■ ■ Cellular Telephones Satellite and GPS Receivers Wireless and DSL Modems Digital Still Cameras Media Players Portable Instruments The LTC®3406AB is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current with no load is 200μA, dropping to <1μA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3406AB ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery run time in portable systems. PWM pulse skipping mode operation provides very low output ripple voltage for noise sensitive applications. Refer to LTC3406A for applications that require Burst Mode® operation. Switching frequency is internally set at 1.5MHz, allowing the use of small surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.6V feedback reference voltage. The LTC3406AB is available in a low profile (1mm) ThinSOT package. , LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 6580258. TYPICAL APPLICATION Efficiency vs Load Current 100 2.2μH VIN 4.7μF CER SW 22pF LTC3406AB RUN GND 10μF CER VFB 619k 309k 3406AB TA09 VOUT = 1.8V 90 1.8V, 600mA VOUT 80 EFFICIENCY (%) VIN 70 60 50 40 30 20 10 0.1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000 3406B TA14 3406abfa 1 LTC3406AB ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) TOP VIEW Input Supply Voltage ....................................– 0.3V to 6V RUN, VFB Voltages .......................................–0.3V to VIN SW Voltage (DC) ........................... – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) (Note 7)................................................................800mA N-Channel Switch Sink Current (DC) (Note 7) .....800mA Peak SW Sink and Source Current (Note 7) .............1.3A Operating Temperature Range (Note 2) ...– 40°C to 85°C Junction Temperature (Notes 3, 6) ....................... 125°C Storage Temperature Range...................– 65°C to 150°C Lead Temperature (Soldering, 10 sec) .................. 300°C RUN 1 5 VFB GND 2 SW 3 4 VIN S5 PACKAGE 5-LEAD PLASTIC TSOT-23 TJMAX = 125°C, θJA = 250°C/W, θJC = 90°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3406ABES5#PBF LTC3406ABES5#TRPBF LTCXZ 5-Lead Plastic TSOT-23 –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER IVFB Feedback Current CONDITIONS MIN TYP ● VFB Regulated Feedback Voltage (Note 4) ● ΔVFB Reference Voltage Line Regulation VIN = 2.5V to 5.5V (Note 4) ● IPK Peak Inductor Current VIN = 3V, VFB = 0.5V Duty Cycle < 35% VLOADREG Output Voltage Load Regulation VIN Input Voltage Range IS Input DC Bias Current Active Mode Shutdown (Note 5) VFB = 0.63V VRUN = 0V, VIN = 5.5V fOSC Oscillator Frequency VFB = 0.6V RPFET RDS(ON) of P-Channel FET ISW = 100mA RNFET RDS(ON) of N-Channel FET ISW = –100mA ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V tSOFTSTART Soft-Start Time VFB from 10% to 90% Full-scale 0.5880 0.75 MAX UNITS ±30 nA 0.6 0.6120 0.04 0.4 %/V 1 1.25 A 0.5 ● 2.5 1.2 0.6 % 5.5 V 300 1 μA μA 1.5 1.8 MHz 0.23 0.35 Ω 0.21 0.35 Ω ±0.01 ±1 μA 0.9 1.2 ms 200 0.1 ● V 3406abfa 2 LTC3406AB ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER VRUN RUN Threshold ● RUN Leakage Current ● IRUN CONDITIONS Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3406ABE is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3406AB: TJ = TA + (PD)(250°C/W) MIN TYP MAX UNITS 0.3 1 1.5 V ±0.01 ±1 μA Note 4: The LTC3406AB is tested in a proprietary test mode that connects VFB to the output of the error amplifier. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note7: Limited by long term current density considerations. TYPICAL PERFORMANCE CHARACTERISTICS (From Front Page Figure Except for the Resistive Divider Resistor Values) Efficiency vs Load Current 100 90 90 80 80 70 70 EFFICIENCY (%) EFFICIENCY (%) Efficiency vs Input Voltage 100 60 50 40 VOUT = 1.2V 60 50 40 30 30 IL = 100mA IL = 600mA IL = 10mA 20 10 VOUT = 1.8V 0 3 2 5 4 INPUT VOLTAGE (V) 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 0 0.1 6 3406B G01 3406B G02 Reference Voltage vs Temperature Efficiency vs Load Current 100 90 0.615 VOUT = 2.5V REFERENCE VOLTAGE (V) EFFICIENCY (%) 70 60 50 40 30 10 0 0.1 VIN = 3.6V 0.610 80 20 1000 1 100 10 OUTPUT CURRENT (mA) VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000 3406B G03 0.605 0.600 0.595 0.590 0.585 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 3406AB G21 3406abfa 3 LTC3406AB TYPICAL PERFORMANCE CHARACTERISTICS (From Front Page Figure Except for the Resistive Divider Resistor Values) Oscillator Frequency vs Temperature 1.55 1.50 1.45 1.40 1.35 50 25 0 75 TEMPERATURE (°C) 100 125 1.812 1.50 1.45 1.40 1.35 1.30 1.796 1.792 1.784 1.20 1.780 2.0 2.5 3.0 3.5 4.0 4.5 5.0 INPUT VOLTAGE (V) 5.5 RDS(ON) (Ω) 0.30 0.20 Dynamic Supply Current 300 VIN = 2.7V VIN = 3.6V 0.25 VIN = 4.2V 0.20 0.15 0.10 MAIN SWITCH SYNCHRONOUS SWITCH 4 3 5 2 INPUT VOLTAGE (V) MAIN SWITCH SYNCHRONOUS SWITCH 0.05 0 –50 –25 0.10 1 6 7 0 50 75 25 TEMPERATURE (°C) 100 VOUT = 1.2V ILOAD = 0A 250 200 150 100 50 0 125 2 3 2.5 4.5 5 3.5 4 INPUT VOLTAGE (V) 5.5 3406B G26 3406B G25 Dynamic Supply Current vs Temperature Switch Leakage vs Input Voltage 140 VIN = 3.6V VOUT = 1.2V ILOAD = 0A 120 6 3406B G27 Switch Leakage vs Temperature 300 1000 MAIN SWITCH SYNCHRONOUS SWITCH MAIN SWITCH SYNCHRONOUS SWITCH 900 RUN = 0V 150 100 SWITCH LEAKAGE (pA) SWITCH LEAKAGE (nA) 800 200 100 80 60 40 700 600 500 400 300 200 50 0 –50 –25 600 400 3406B G24 DYNAMIC SUPPLY CURRENT (μA) 0.30 0 200 OUTPUT CURRENT (mA) RDS(ON) vs Input Voltage 0.35 0.15 VIN = 2.7V VIN = 3.6V VIN = 4.2V 0 6.0 0.40 0.35 RDS(0N) (Ω) 1.800 1.788 RDS(ON) vs Input Voltage DYNAMIC SUPPLY CURRENT (μA) 1.804 3406B G07 0.40 250 1.808 1.25 3406B G22 0.25 VOUT = 1.8V 1.816 1.55 OUTPUT VOLTAGE (V) VIN = 3.6V 1.30 –50 –25 Output vs Load Current 1.820 1.60 OSCILLATOR FREQUENCY (MHz) OSCILLATOR FREQUENCY (MHz) 1.60 Oscillator Frequency vs Supply Voltage 20 50 25 75 0 TEMPERATURE (°C) 100 125 3406B G28 0 –50 –25 100 0 50 25 75 0 TEMPERATURE (°C) 100 125 3406B G29 0 1 3 4 2 INPUT VOLTAGE (V) 5 6 3406B G30 3406abfa 4 LTC3406AB TYPICAL PERFORMANCE CHARACTERISTICS (From Front Page Figure Except for the Resistive Divider Resistor Values) Start-Up from Shutdown Load Step Load Step VOUT 200mV/DIV VOUT 200mV/DIV IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV RUN 2V/DIV VOUT 1V/DIV ILOAD 500mA/DIV VIN = 3.6V 400μs/DIV VOUT = 1.8V ILOAD = 600mA (3Ω RES) 3406B G31 VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 0mA TO 600mA Load Step VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 50mA TO 600mA Load Step VOUT 200mV/DIV VOUT 200mV/DIV IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 100mA TO 600mA 3406B G32 3406B G34 3406B G33 Discontinuous Operation SW (2V/DIV) VOUT 20mV/DIV AC COUPLED IL 200mA/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 200mA TO 600mA 3406B G35 VIN = 3.6V VOUT = 1.8V ILOAD = 25mA 500ns/DIV 3406B G36 PIN FUNCTIONS RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1μA supply current. Do not leave RUN floating. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2μF or greater ceramic capacitor. VFB (Pin 5): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. GND (Pin 2): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. 3406abfa 5 LTC3406AB FUNCTIONAL DIAGRAM SLOPE COMP OSC OSC 4 VIN FREQ SHIFT – – + 5 0.6V + – EA S Q R Q RS LATCH VIN RUN SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW 0.6V REF SHUTDOWN + 1 5Ω + ICOMP VFB IRCMP 2 GND – 3406AB BD OPERATION (Refer to Functional Diagram) Main Control Loop The LTC3406AB uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. When the load current increases, it causes a slight decrease in the feedback voltage, FB, relative to the 0.6V reference, which in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. The main control loop is shut down by grounding RUN, resetting the internal soft-start. Re-enabling the main control loop by pulling RUN high activates the internal soft-start, which slowly ramps the output voltage over approximately 0.9ms until it reaches regulation. Pulse Skipping Mode Operation At light loads, the inductor current may reach zero or reverse on each pulse. The bottom MOSFET is turned off by the current reversal comparator, IRCMP, and the switch voltage will ring. This is discontinuous mode operation, and is normal behavior for the switching regulator. At very light loads, the LTC3406AB will automatically skip pulses in pulse skipping mode operation to maintain output regulation. Refer to the LTC3406A data sheet if Burst Mode operation is preferred. 3406abfa 6 LTC3406AB OPERATION (Refer to Functional Diagram) Dropout Operation Slope Compensation and Inductor Peak Current As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles >40%. However, the LTC3406AB uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. An important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3406AB is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). 3406abfa 7 LTC3406AB APPLICATIONS INFORMATION The basic LTC3406AB application circuit is shown on the front page. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Table 1. Representative Surface Mount Inductors PART NUMBER VALUE (μH) DCR (Ω MAX) MAX DC CURRENT (A) SIZE W × L × H (mm3) Sumida CDRH3D16 1.5 2.2 3.3 4.7 0.043 0.075 0.110 0.162 1.55 1.20 1.10 0.90 3.8 × 3.8 × 1.8 Sumida CMD4D06 2.2 3.3 4.7 0.116 0.174 0.216 0.950 0.770 0.750 3.5 × 4.3 × 0.8 Panasonic ELT5KT 3.3 4.7 0.17 0.20 1.00 0.95 4.5 × 5.4 × 1.2 Murata LQH32CN 1.0 2.2 4.7 0.060 0.097 0.150 1.00 0.79 0.65 2.5 × 3.2 × 2.0 Inductor Selection For most applications, the value of the inductor will fall in the range of 1μH to 4.7μH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ΔIL = 240mA (40% of 600mA). ΔIL = ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ VIN ⎠ ( f )(L ) ⎝ CIN and COUT Selection (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 720mA rated inductor should be enough for most applications (600mA + 120mA). For better efficiency, choose a low DC-resistance inductor. Inductor Core Selection Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3406AB requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3406AB applications. In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX ⎡⎣ VOUT ( VIN − VOUT ) ⎤⎦ VIN 1/22 This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). 3406abfa 8 LTC3406AB APPLICATIONS INFORMATION Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ΔVOUT is determined by: ⎛ 1 ⎞ ΔVOUT ≅ ΔIL ⎜ ESR + 8 fCOUT ⎟⎠ ⎝ where f = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Output Voltage Programming In the adjustable version, the output voltage is set by a resistive divider according to the following formula: ⎛ R2 ⎞ VOUT = 0.6 V ⎜ 1+ ⎟ ⎝ R1⎠ (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 1. 0.6V ≤ VOUT ≤ 5.5V R2 VFB LTC3406AB R1 GND Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3406AB’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this 3406AB F03 Figure 1. Setting the LTC3406AB Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. 3406abfa 9 LTC3406AB APPLICATIONS INFORMATION Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3406AB circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 2. VIN = 3.6V POWER LOSS (W) 0.1 0.01 0.0001 0.1 RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. 1 0.001 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: VOUT = 1.2V VOUT = 1.8V VOUT = 2.5V 10.0 100.0 1.0 OUTPUT CURRENT (mA) 1000.0 3406B F08 Figure 2. Power Lost vs Load Current 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3406AB does not dissipate much heat due to its high efficiency. But, in applications where the LTC3406AB is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3406AB from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. 3406abfa 10 LTC3406AB APPLICATIONS INFORMATION The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3406AB in dropout at an input voltage of 2.7V, a load current of 600mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.27Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 97.2mW For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.0972)(250) = 94.3°C which is below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ΔILOAD • ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10μF capacitor charging to 3.3V would require a 250μs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3406AB. These items are also illustrated graphically in Figures 3 and 4. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace, the VOUT trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does CIN connect to VIN as closely aspossible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node, SW, away from the sensitive VFB node. 5. Keep the (–) plates of CIN and COUT and the IC ground, as close as possible. 3406abfa 11 LTC3406AB APPLICATIONS INFORMATION 1 RUN VFB LTC3406AB 2 5 R2 R1 GND – COUT VOUT 3 + L1 SW VIN CFWD 4 CIN + VIN – BOLD LINES INDICATE HIGH CURRENT PATHS 3406AB F05a Figure 3. LTC3406AB Layout Diagram VIA TO VIN R1 CFWD LTC3406AB L1 VIA TO VOUT R2 PIN 1 VOUT VIN SW COUT CIN GND 3406AB F06a Figure 4. LTC3406AB Suggested Layout Design Example As a design example, assume the LTC3406AB is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.6A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using Equation (1), L= ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ VIN ⎠ ( f )( ΔIL ) ⎝ (3) Substituting VOUT = 2.5V, VIN = 4.2V, ΔIL = 240mA and f = 1.5MHz in Equation (3) gives: L= 2.5V ⎛ 2.5V ⎞ 1− = 2.81μH 1.5MHz(240mA) ⎜⎝ 4.2V ⎟⎠ A 2.2μH inductor works well for this application. For best efficiency choose a 720mA or greater inductor with less than 0.2Ω series resistance. CIN will require an RMS current rating of at least 0.3A≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will satisfy this requirement. 3406abfa 12 LTC3406AB APPLICATIONS INFORMATION For the feedback resistors, choose R1 = 316k. R2 can then be calculated from Equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1= 1000k ⎝ 0.6 ⎠ Figure 5 shows the complete circuit along with its efficiency curve. (4) 100 90 4 † CIN 4.7μF CER VIN SW 3 2.2μH* 22pF VFB RUN GND 2 70 COUT** 10μF CER LTC3406AB 1 80 2.5V, 600mA VOUT 5 1M *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † 3406AB TA09a TAIYO YUDEN JMK212BJ475MG 316k EFFICIENCY (%) VIN VOUT = 2.5V 60 50 40 30 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 0 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3406B G03 Load Step Load Step VOUT 200mV/DIV VOUT 100mV/DIV IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20μs/DIV VOUT = 2.5V ILOAD = 200mA TO 450mA 3406B F10 VIN = 3.6V 20μs/DIV VOUT = 2.5V ILOAD = 300mA TO 600mA 3406B F16 Figure 5 3406abfa 13 LTC3406AB TYPICAL APPLICATIONS Single Li-Ion 1.2V/600mA Regulator for High Efficiency and Small Footprint VIN † CIN 4.7μF CER SW 3 VFB RUN GND 2 1.2V, 600mA VOUT 22pF COUT** 10μF CER LTC3406AB 1 100 2.2μH* 5 301k *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG 3406AB TA09b 301k 90 VOUT = 1.2V 80 70 EFFICIENCY (%) VIN Efficiency vs Load Current 60 50 40 30 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 0 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3406B G02 Load Step Load Step VOUT 200mV/DIV VOUT 100mV/DIV IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20μs/DIV VOUT = 1.2V ILOAD = 200mA TO 500mA 3406B F12 VIN = 3.6V 20μs/DIV VOUT = 1.2V ILOAD = 300mA TO 600mA 3406B F14 3406abfa 14 LTC3406AB PACKAGE DESCRIPTION S5 Package 5-Lead Plastic SOT-23 (Reference LTC DWG # 05-08-1633 Rev B) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 1.90 BSC S5 TSOT-23 0302 REV B 3406abfa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3406AB RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3406/LTC3406B 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converters 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20μA, ISD <1μA, ThinSOT Package LTC3407/LTC3407-2 Dual 600mA/800mA (IOUT), 1.5MHz/2.25MHz, Synchronous Step-Down DC/DC Converters 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD <1μA, MS10E, DFN Packages LTC3410/LTC3410B 300mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converters 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26μA, ISD <1μA, SC70 Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60mA, ISD <1μA, MS10, DFN Packages LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60μA, ISD <1μA, TSSOP-16E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V to 5.5V, IQ = 25μA, ISD <1μA, MS10, DFN Packages LTC3530 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 1.8V to 5.5V, VOUT(MIN) = 1.8V to 5.25V, IQ = 40μA, ISD <1μA, MS10, DFN Packages LTC3531/LTC3531-3/ 200mA (IOUT), 1.5MHz, Synchronous Buck-Boost DC/DC Converters LTC3531-3.3 95% Efficiency, VIN: 1.8V to 5.5V, VOUT(MIN) = 2V to 5V, IQ = 16μA, ISD <1μA, ThinSOT, DFN Packages LTC3532 500mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.4V to 5.5V, VOUT(MIN) = 2.4V to 5.25V, IQ = 35μA, ISD <1μA, MS10, DFN Packages LTC3542 500mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 26μA, ISD <1μA, 2mm × 2mm DFN Package LTC3544/LTC3544B Quad 300mA + 2 x 200mA + 100mA 2.25MHz, Synchronous Step-Down DC/DC Converters 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 70μA, ISD <1μA, 3mm × 3mm QFN Package LTC3547/LTC3547B Dual 300mA 2.25MHz, Synchronous Step-Down DC/DC Converters 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD <1μA, 2mm × 3mm DFN Package LTC3548/LTC3548-1/ Dual 400mA and 800mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converters LTC3548-2 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD <1μA, MS10E, DFN Packages LTC3560 800mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 16μA, ISD <1μA, ThinSOT Package LTC3561 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 240μA, ISD <1μA, DFN Package 3406abfa 16 Linear Technology Corporation LT 0907 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2007