Final Electrical Specifications LTC1877 High Efficiency Monolithic Synchronous Step-Down Regulator U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LTC ®1877 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 10µA and drops to < 1µA in shutdown. The 2.65V to 10V input voltage range makes the LTC1877 ideally suited for both single and dual Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. High Efficiency: Up to 95% Very Low Quiescent Current: Only 10µA During Operation 600mA Output Current at VIN = 5V 2.65V to 10V Input Voltage Range 550kHz Constant Frequency Operation Synchronizable from 400kHz to 700kHz Selectable Burst ModeTM Operation/ Pulse Skipping Mode No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle 0.8V Reference Allows Low Output Voltages Shutdown Mode Draws < 1µA Supply Current ±2% Output Voltage Accuracy Current Mode Control for Excellent Line and Load Transient Response Overcurrent and Overtemperature Protected Available in 8-Lead MSOP Package Switching frequency is internally set at 550kHz, allowing the use of small surface mount inductors and capacitors. For noise sensitive applications the LTC1877 can be externally synchronized from 400kHz to 700kHz. Burst Mode operation is inhibited during synchronization or when the SYNC/MODE pin is pulled low, preventing low frequency ripple from interfering with audio circuitry. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.8V feedback reference voltage. The LTC1877 is available in a space saving 8-lead MSOP package. Lower input voltage applications (less than 7V abs max) should refer to the LTC1878 data sheet. U APPLICATIONS ■ ■ ■ ■ ■ ■ Cellular Telephones Wireless Modems Personal Information Appliances Portable Instruments Distributed Power Systems Battery-Powered Equipment , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U TYPICAL APPLICATION Efficiency vs Output Current 100 95 High Efficiency Step-Down Converter 7 10µF** CER 6 1 2 SW SYNC VIN 5 20pF RUN ITH GND + VOUT† 3.3V 47µF*** LTC1877 VFB 3 887k VIN = 10V 85 80 75 VIN = 7.2V VIN = 5V 70 65 280k 220pF VIN = 3.6V 90 10µH* EFFICIENCY (%) VIN 2.65V TO 10V May 2000 60 4 VOUT = 3.3V L = 10µH Burst Mode OPERATION 55 *TOKO D62CB A920CY-100M **TAIYO-YUDEN CERAMIC LMK325BJ106MN ***SANYO POSCAP 6TPA47M † VOUT CONNECTED TO VIN FOR 2.65V < VIN < 3.3V 50 0.1 1877 TA01 1.0 100 10 OUTPUT CURRENT (mA) 1000 1877 • TA02 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 1 LTC1877 W U PACKAGE/ORDER INFORMATION U W W W (Note 1) Input Supply Voltage (VIN).........................– 0.3V to 12V ITH, PLL LPF Voltage ................................– 0.3V to 2.7V RUN, VFB Voltages ...................................... – 0.3V to VIN SYNC/MODE Voltage .................................. – 0.3V to VIN SW Voltage ................................... – 0.3V to (VIN + 0.3V) P-Channel MOSFET Source Current (DC) ........... 800mA N-Channel MOSFET Sink Current (DC) ............... 800mA Peak SW Sink and Source Current ........................ 1.5A Operating Ambient Temperature Range Extended Commercial (Note 2) ........... – 40°C to 85°C Junction Temperature (Note 3) ............................ 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U ABSOLUTE MAXIMUM RATINGS ORDER PART NUMBER TOP VIEW RUN ITH VFB GND 1 2 3 4 8 7 6 5 PLL LPF SYNC/MODE VIN SW MS8 PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 150°C/ W LTC1877EMS8 MS8 PART MARKING LTLU Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 5V unless otherwise specified. SYMBOL IVFB PARAMETER Feedback Current CONDITIONS (Note 4) ● MIN TYP 4 MAX 30 UNITS nA VFB Regulated Output Voltage (Note 4) 0°C ≤ TA ≤ 85°C (Note 4) –40°C ≤ TA ≤ 85°C ● 0.784 0.74 0.8 0.8 0.816 0.84 V V ∆VOVL = VOVL – VFB ● 20 50 110 mV 0.05 0.15 %/V 0.1 – 0.1 0.5 – 0.5 % % 10 V 230 10 0 350 15 1 µA µA µA 550 80 605 kHz kHz 700 kHz 10 –10 20 –20 µA µA ∆VOVL Output Overvoltage Lockout ∆VFB Reference Voltage Line Regulation VIN = 2.65V to 10V (Note 4) VLOADREG Output Voltage Load Regulation Measured in Servo Loop; VITH = 0.9V to 1.2V Measured in Servo Loop; VITH = 1.6V to 1.2V VIN Input Voltage Range IQ Input DC Bias Current Pulse Skipping Mode Burst Mode Operation Shutdown (Note 5) 2.65V < VIN < 10V, VSYNC/MODE = 0V, IOUT =0A VSYNC/MODE = VIN, IOUT =0A VRUN = 0V, VIN = 10V fOSC Oscillator Frequency VFB = 0.8V VFB = 0V fSYNC SYNC Capture Range IPLL LPF Phase Detector Output Current Sinking Capability Sourcing Capability fPLLIN < fOSC fPLLIN > fOSC RPFET RDS(ON) of P-Channel MOSFET ISW = 100mA 0.65 0.85 Ω RNFET RDS(ON) of N-Channel MOSFET ISW = –100mA 0.75 0.95 Ω 2 ● ● ● 2.65 495 400 ● ● 3 –3 LTC1877 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 5V unless otherwise specified. SYMBOL IPK PARAMETER Peak Inductor Current CONDITIONS VFB = 0.7V, Duty Cycle < 35% MIN 0.8 ILSW SW Leakage VRUN = 0V, VSW = 0V or 8.5V, VIN = 8.5V VSYNC/MODE SYNC/MODE Threshold ISYNC/MODE SYNC/MODE Leakage Current VRUN RUN Threshold IRUN RUN Input Current Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1877 is guaranteed to meet specified performance from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. ● 0.2 ● 0.2 TYP 1.0 MAX 1.25 UNITS A ±0.01 ±1 µA 1.0 1.5 V ±0.01 ±1 µA 0.7 1.5 V ±0.01 ±1 µA Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1877EMS8: TJ = TA + (PD)(150°C/W) Note 4: The LTC1877 is tested in a feedback loop which servos VFB to the balance point for the error amplifier (VITH = 1.2V). Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. U W TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Output Current Efficiency vs Input Voltage ILOAD = 100mA I LOAD = 10mA 95 90 80 85 70 ILOAD = 300mA 80 ILOAD = 1mA 75 70 ILOAD = 0.1mA 65 60 2 6 8 4 INPUT VOLTAGE (V) VIN = 3.6V 90 VIN = 7.2V 85 VIN = 7.2V VIN = 3.6V 50 40 PULSE SKIPPING MODE Burst Mode OPERATION 10 10 12 1877 • G01 0 0.1 VOUT = 2.5V L = 10µH 1.0 100 10 OUTPUT CURRENT (mA) 1000 1877 • G02 L = 15µH L = 10µH 80 75 70 65 20 50 0 60 30 VOUT = 2.5V L = 10µH Burst Mode OPERATION 55 Efficiency vs Output Current 95 90 EFFICIENCY (%) EFFICIENCY (%) 100 EFFICIENCY (%) 100 60 55 0.1 VIN = 10V VOUT = 3.3V Burst Mode OPERATION 1.0 100 10 OUTPUT CURRENT (mA) 1000 1877 • G03 3 LTC1877 U W TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature Efficiency vs Output Current Oscillator Frequency vs Temperature 605 0.814 95 VIN = 5V VIN = 3.6V 90 EFFICIENCY (%) VIN = 10V VIN = 5V 75 70 65 60 50 0.1 0.799 0.794 2.52 585 2.51 50 25 75 0 TEMPERATURE (°C) 100 565 555 545 535 525 PULSE SKIPPING MODE VIN = 4.2V L = 10µH 200 600 400 LOAD CURRENT (mA) 0 300 PULSE SKIPPING MODE VOUT = 1.8V 150 100 50 100 125 1877 • G10 3 4 5 6 7 INPUT VOLTAGE (V) 8 9 10 PULSE SKIPPING MODE 200 150 100 50 Burst Mode OPERATION BURST MODE OPERATION 0 50 25 75 0 TEMPERATURE (°C) 2 VIN = 5V 250 200 SUPPLY CURRENT (µA) DC SUPPLY CURRENT (µA) 1.2 SYNCHRONOUS SWITCH MAIN SWITCH 1 DC Supply Current vs Temperature 250 0.6 0 1877 • G09 DC Supply Current vs Input Voltage 1.4 RDS(ON) (Ω) 0 800 1877 • G08 RDS(ON) vs Temperature 0.2 –50 –25 MAIN SWITCH 0.2 1877 • G07 0.4 0.6 0.4 2.43 VIN = 5V SYNCHRONOUS SWITCH 0.8 2.46 12 125 1.0 2.47 2.44 0.8 100 RDS(ON) vs Input Voltage 2.48 505 VIN = 3V 25 50 75 0 TEMPERATURE (°C) 1.2 2.49 2.45 1.0 –25 1877 • G06 2.50 515 4 495 –50 125 RDS(ON) (Ω) 575 OUTPUT VOLTAGE (V) OSCILLATOR FREQUENCY (kHz) 595 10 535 Output Voltage vs Load Current 2.53 6 8 4 SUPPLY VOLTAGE (V) 545 1877 • G05 605 2 555 505 Oscillator Frequency vs Supply Voltage 0 565 515 1877 • G04 495 575 525 0.784 –50 –25 1000 1.0 100 10 OUTPUT CURRENT (mA) 0.804 0.789 L = 10µH VOUT = 2.5V 55 585 FREQUENCY (kHz) VIN = 7.2V 80 REFERENCE VOLTAGE (V) 0.809 85 VIN = 5V 595 0 2 6 4 INPUT VOLTAGE (V) 8 10 1877 • G11 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 1877 G19 LTC1877 U W TYPICAL PERFORMANCE CHARACTERISTICS Switch Leakage vs Temperature 16 VIN = 10V RUN = 0V 1000 MAIN SWITCH 800 600 400 SYNCHRONOUS SWITCH SW 5V/DIV 12 10 6 4 IL 200mA/DIV 2 50 25 75 0 TEMPERATURE (°C) 100 VOUT 20mV/DIV AC COUPLED SYNCHRONOUS SWITCH 8 200 0 –50 –25 RUN = 0V 14 SWITCH LEAKAGE (nA) SWITCH LEAKAGE (nA) 1200 Burst Mode Operation Switch Leakage vs Input Voltage 1400 MAIN SWITCH 0 125 2 0 1877 • G12 4 6 INPUT VOLTAGE (V) 8 10 1877 • G13 VIN = 5V VOUT = 1.5V CIN = 10µF 10µs/DIV L = 10µH COUT = 47µF ILOAD = 50mA 1877 • G14 Load Step Response Start-Up from Shutdown RUN 5V/DIV VOUT 50mV/DIV AC COUPLED VOUT 1V/DIV IL 500mA/DIV IL 500mA/DIV ITH 1V/DIV 50µs/DIV VIN = 5V VOUT = 1.5V L = 10µH VIN = 5V VOUT = 1.5V L = 10µH CIN = 10µF COUT = 47µF ILOAD = 500mA 1877 • G15 40µs/DIV CIN = 10µF COUT = 47µF ILOAD = 50mA TO 500mA PULSE SKIPPING MODE 1877 • G16 Load Step Response Load Step Response VOUT 50mV/DIV AC COUPLED VOUT 50mV/DIV AC COUPLED IL 500mA/DIV IL 500mA/DIV 10µs/DIV ITH 1V/DIV ITH 1V/DIV VIN = 5V VOUT = 1.5V L = 10µH 40µs/DIV CIN = 10µF COUT = 47µF ILOAD = 50mA TO 500mA Burst Mode OPERATION 1877 • G17 VIN = 5V VOUT = 1.5V L = 10µH 20µs/DIV CIN = 10µF COUT = 47µF ILOAD = 200mA TO 500mA PULSE SKIPPING MODE 1877 • G18 5 LTC1877 U U U PIN FUNCTIONS RUN (Pin 1): Run Control Input. Forcing this pin below 0.4V shuts down the LTC1877. In shutdown all functions are disabled drawing <1µA supply current. Forcing this pin above 1.2V enables the LTC1877. Do not leave RUN floating. ITH (Pin 2): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is from 0.5V to 1.9V. VFB (Pin 3): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. GND (Pin 4): Ground Pin. SW (Pin 5): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 6): Main Supply Pin. Must be closely decoupled to GND, Pin 4. SYNC/MODE (Pin 7): External Clock Synchronization and Mode Select Input. To synchronize with an external clock, apply a clock with a frequency between 400kHz and 700kHz. To select Burst Mode operation, tie to VIN. Grounding this pin selects pulse skipping mode. Do not leave this pin floating. PLL LPF (Pin 8): Output of the Phase Detector and Control Input of Oscillator. Connect a series RC lowpass network from this pin to ground if externally synchronized. If unused, this pin may be left open. W FUNCTIONAL DIAGRA U U VIN BURST DEFEAT PLL LPF Y Y = “0” ONLY WHEN X IS A CONSTANT “1” X 8 SLOPE COMP SYNC/MODE 7 0.8V VCO OSC 0.6V 3 VFB – 6 VIN FREQ SHIFT + – + 0.45V – EA gm = 0.5m Ω 0.8V REF EN SLEEP – + VIN S Q R Q RS LATCH 6Ω + ICOMP BURST VIN SLEEP 2 ITH VIN RUN 1 – + VREF 0.8V SWITCHING LOGIC AND BLANKING CIRCUIT ANTI SHOOTTHRU 5 SW – OVDET SHUTDOWN + + 0.85V IRCMP 4 GND – 1877 BD 6 LTC1877 U OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC1877 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch is controlled by the voltage on the ITH pin, which is the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the ITH voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. Comparator OVDET guards against transient overshoots > 6.25% by turning the main switch off and keeping it off until the fault is removed. Burst Mode Operation The LTC1877 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. To enable Burst Mode operation, simply tie the SYNC/MODE pin to VIN or connect it to a logic high (VSYNC/MODE > 1.5V). To disable Burst Mode operation and enable PWM pulse skipping mode, connect the SYNC/ MODE pin to GND. In this mode, the efficiency is lower at light loads, but becomes comparable to Burst Mode operation when the output load exceeds 50mA. The advantage of pulse skipping mode is lower output ripple and less interference to audio circuitry. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 250mA, even though the voltage at the ITH pin indicates a lower value. The voltage at the ITH pin drops when the inductor’s average current is greater than the load requirement. As the ITH voltage drops below approximately 0.45V, the BURST comparator trips, causing the internal sleep line to go high and forces off both power MOSFETs. The ITH pin is then disconnected from the output of the EA amplifier and parked a diode voltage above ground. In sleep mode, both power MOSFETs are held off and a majority of the internal circuitry is partially turned off, reducing the quiescent current to 10µA. The load current is now being supplied solely from the output capacitor. When the output voltage drops, the ITH pin reconnects to the output of the EA amplifier and the top MOSFET is again turned on and this process repeats. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 80kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has ample time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 550kHz (or the synchronized frequency) when VFB rises above 0.3V. Frequency Synchronization A phase-locked loop (PLL) is available on the LTC1877 to allow the internal oscillator to be synchronized to an external source connected to the SYNC/MODE pin. The output of the phase detector at the PLL LPF pin operates over a 0V to 2.4V range corresponding to 400kHz to 700kHz. When locked, the PLL aligns the turn-on of the top MOSFET to the rising edge of the synchronizing signal. When the LTC1877 is clocked by an external source, Burst Mode operation is disabled; the LTC1877 then operates in PWM pulse skipping mode. In this mode, when the output load is very low, current comparator ICOMP may remain tripped for several cycles and force the main switch to stay off for the same number of cycles. Increasing the output load slightly allows constant frequency PWM operation to resume. This mode exhibits low output ripple as well as low audio noise and reduced RF interference while providing reasonable low current efficiency. Frequency synchronization is inhibited when the feedback voltage VFB is below 0.6V. This prevents the external clock from interfering with the frequency foldback for shortcircuit protection. 7 LTC1877 U OPERATIO Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the internal P-channel MOSFET and the inductor. Low Supply Operation The LTC1877 is designed to operate down to an input supply voltage of 2.65V although the maximum allowable output current is reduced at this low voltage. Figure 1 shows the reduction in the maximum output current as a function of input voltage for various output voltages. VOUT = 2.5V MAX OUTPUT CURRENT (mA) 1000 VOUT = 1.5V 800 VOUT = 5V 600 VOUT = 3.3V 400 200 L = 10µH 0 0 2 4 6 VIN (V) 8 10 Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. As a result, the maximum inductor peak current is reduced for duty cycles > 40%. This is shown in the decrease of the inductor peak current as a function of duty cycle graph in Figure 2. MAXIMUM INDUCTOR PEAK CURRENT (mA) 1200 Another important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases. Therefore, the user should calculate the power dissipation when the LTC1877 is used at 100% duty cycle with a low input voltage (see Thermal Considerations in the Applications Information section). 1100 VIN = 5V 1000 900 800 700 600 12 0 1877 • F01 20 60 40 DUTY CYCLE (%) 80 100 1877 F02 Figure 1. Maximum Output Current vs Input Voltage Figure 2. Maximum Inductor Peak Current vs Duty Cycle U W U U APPLICATIONS INFORMATION The basic LTC1877 application circuit is shown on the first page. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Value Calculation The inductor selection will depend on the operating frequency of the LTC1877. The internal nominal frequency is 550kHz, but can be externally synchronized from 400kHz to 700kHz. 8 The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. However, operating at a higher frequency generally results in lower efficiency because of increased internal gate charge losses. The inductor value has a direct effect on ripple current. The ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT. LTC1877 U W U U APPLICATIONS INFORMATION V 1 ∆IL = VOUT 1 − OUT ( f)(L) VIN (1) Accepting larger values of ∆IL allows the use of low inductance, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ∆IL = 0.4(IMAX). The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 250mA. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy, or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Kool Mµ (from Magnetics, Inc.) is a very good, low loss core material for toroids with a “soft” saturation characteristic. Molypermalloy is slightly more efficient at high (>200kHz) switching frequencies but quite a bit more expensive. Toroids are very space efficient, especially when you can use several layers of wire, while inductors Kool Mµ is a registered trademark of Magnetics, Inc. wound on bobbins are generally easier to surface mount. New designs for surface mount inductors are available from Coiltronics, Coilcraft, Dale and Sumida. CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX [VOUT (VIN − VOUT )]1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple ∆VOUT is determined by: 1 ∆VOUT ≅ ∆IL ESR + 8fC OUT where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. For the LTC1877, the general rule for proper operation is: COUT required ESR < 0.25Ω The choice of using a smaller output capacitance increases the output ripple voltage due to the frequency dependent term but can be compensated for by using capacitor(s) of very low ESR to maintain low ripple voltage. The ITH pin compensation components can be opti- 9 LTC1877 U W U U APPLICATIONS INFORMATION ESR is a direct function of the volume of the capacitor. Manufacturers such as Taiyo Yuden, AVX, Sprague, Kemet and Sanyo should be considered for high performance capacitors. The POSCAP solid electrolytic capacitor available from Sanyo is an excellent choice for output bulk capacitors due to its low ESR/size ratio. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. When using tantalum capacitors, it is critical that they are surge tested for use in switching power supplies. A good choice is the AVX TPS series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include KEMET T510 and T495 series and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the harmonics of the VCO center frequency. The PLL hold-in range ∆fH is equal to the capture range, ∆fH = ∆fC = ±150kHz. The output of the phase detector is a pair of complementary current sources charging or discharging the external filter network on the PLL LPF pin. The relationship between the voltage on the PLL LPF pin and operating frequency is shown in Figure 4. A simplified block diagram is shown in Figure 5. If the external frequency (VSYNC/MODE) is greater than 550kHz, the center frequency, current is sourced continuously, pulling up the PLL LPF pin. When the external 800 OSCILLATOR FREQUENCY (kHz) mized to provide stable high performance transient response regardless of the output capacitor selected. Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: VOUT R2 = 0.8V 1 + R1 (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 3. 700 600 500 400 300 0 0.4 0.8V ≤ VOUT ≤ 10V 0.8 1.2 VPLL LPF (V) 1.6 2.0 1877 • F04 Figure 4. Relationship Between Oscillator Frequency and Voltage at PLL LPF Pin R2 VFB LTC1877 RLP R1 PHASE DETECTOR GND 2.4V CLP 1877 F03 PLL LPF Figure 3. Setting the LTC1877 Output Voltage Phase-Locked Loop and Frequency Synchronization The LTC1877 has an internal voltage-controlled oscillator and phase detector comprising a phase-locked loop. This allows the top MOSFET turn-on to be locked to the rising edge of an external frequency source. The frequency range of the voltage-controlled oscillator is 400kHz to 700kHz. The phase detector used is an edge sensitive digital type 10 SYNC/ MODE DIGITAL PHASE/ FREQUENCY DETECTOR VCO 1877 F05 Figure 5. Phase-Locked Loop Block Diagram LTC1877 U W U U APPLICATIONS INFORMATION The loop filter components CLP and RLP smooth out the current pulses from the phase detector and provide a stable input to the voltage controlled oscillator. The filter component’s CLP and RLP determine how fast the loop acquires lock. Typically RLP = 10k and CLP is 2200pF to 0.01µF. When not synchronized to an external clock, the internal connection to the VCO is disconnected. This disallows setting the internal oscillator frequency by a DC voltage on the VPLL LPF pin. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC1877 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 6. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. 1 0.1 POWER LOST (W) frequency is less than 550kHz, current is sunk continuously, pulling down the PLL LPF pin. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the PLL LPF pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable operating point the phase comparator output is high impedance and the filter capacitor CLP holds the voltage. 0.01 VIN = 4.2V L = 10µH VOUT = 1.5V VOUT = 2.5V VOUT = 3.3V Burst Mode OPERATION 0.001 0.0001 0.00001 0.1 1 10 100 LOAD CURRENT (mA) 1000 1877 F06 Figure 6. Power Lost vs Load Current 11 LTC1877 U W U U APPLICATIONS INFORMATION Thermal Considerations Checking Transient Response In most applications the LTC1877 does not dissipate much heat due to its high efficiency. But, in applications where the LTC1877 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The internal compensation provides adequate compensation for most applications. But if additional compensation is required, the ITH pin can be used for external compensation using RC, CC1 as shown in Figure 7. (The 220pF capacitor, CC2, is typically needed for noise decoupling.) To avoid the LTC1877 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = T A + T R where TA is the ambient temperature. As an example, consider the LTC1877 in dropout at an input voltage of 3V, a load current of 500mA, and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.9Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 0.225W For the MSOP package, the θJA is 150°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.225)(150) = 104°C which is below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). 12 A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1877. These items are also illustrated graphically in the layout diagram of Figure 7. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1877 signal ground consists of the resistive divider, the optional compensation network (RC and CC1) and CC2. The power ground consists of the (–) plate of CIN, the (–) plate of COUT and Pin 4 of the LTC1877 U U W U APPLICATIONS INFORMATION CC2 LTC1877 OPTIONAL RC 1 CC1 2 3 4 RUN ITH VFB GND PLL LPF SYNC/MODE R2 V 8 7 BOLD LINES INDICATE HIGH CURRENT PATHS 6 IN SW 5 + L1 + R2 + COUT VOUT + CIN VIN R1 – – 1877 F07 Figure 7. LTC1877 Layout Diagram LTC1877. The power ground traces should be kept short, direct and wide. The signal ground and power ground should converge to a common node in a starground configuration. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and signal ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node SW away from sensitive small signal nodes. Design Example As a design example, assume the LTC1877 is used in a single lithium-ion battery-powered cellular phone application. The input voltage will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.3A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), L= V 1 VOUT 1 − OUT ( f)(∆IL ) VIN (3) Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL=120mA and f = 550kHz in equation (3) gives: L= 2.5V 2.5V 1 − = 15.3µH 550kHz(120mA) 4.2V A 15µH inductor works well for this application. For best efficiency choose a 1A inductor with less than 0.25Ω series resistance. CIN will require an RMS current rating of at least 0.15A at temperature and COUT will require an ESR of less than 0.25Ω. In most applications, the requirements for these capacitors are fairly similar. For the feedback resistors, choose R1 = 412k. R2 can then be calculated from equation (2) to be: V R2 = OUT – 1 R1 = 875.5k; use 887k 0.8 Figure 8 shows the complete circuit along with its efficiency curve. 13 LTC1877 U U W U APPLICATIONS INFORMATION VIN 2.7V TO 4.2V 2 RUN ITH 8 PLL LPF 85 7 SYNC/MODE LTC1877 3 4 VFB VIN GND SW 6 5 VIN = 3.0V 90 10µF** CER EFFICIENCY (%) 220pF 1 95 15µH* VOUT 2.5V + 47µF*** 80 VIN = 3.6V VIN = 4.2V 75 70 65 60 887k VOUT = 2.5V L = 15µH 55 412k 20pF 50 0.1 1877 F08a *SUMIDA CD54-150 **TAIYO YUDEN CERAMIC LMK325BJ106MN ***SANYO POSCAP 6TPA47M 1.0 10 100 OUTPUT CURRENT (mA) 1877 • F08b Figure 8. Single Lithium-Ion to 2.5V/0.3A Regulator from Design Example U TYPICAL APPLICATIONS Dual Lithium-Ion to 2.5V/0.6A Regulator Using All Ceramic Capacitors 1 2 RUN PLL LPF ITH 220pF SYNC/MODE 8 4 VFB VIN GND SW VIN ≤ 8.4V 7 CIN** 10µF CER LTC1877 3 6 5 VOUT 2.5V/0.6A 15µH* 20pF 887k COUT*** 22µF CER 412k *SUMIDA CD54-150 **TAIYO YUDEN CERAMIC LMK325BJ106MN ***TAIYO YUDEN CERAMIC JMK325BJ226MM 1877 TA03 4-to 6-Cell NiCd/NiMH to 1.8V/0.6A Regulator Using All Ceramic Capacitors 1 2 RUN ITH 220pF PLL LPF SYNC/MODE 8 7 4 VFB VIN GND SW 6 5 VOUT 1.8V/0.6A 10µH* 20pF *TOKO D62CB A920CY-100M **TAIYO YUDEN CERAMIC LMK325BJ106MN ***TAIYO YUDEN CERAMIC JMK325BJ226MM 14 VIN ≤ 9V CIN** 10µF CER LTC1877 3 1000 887k COUT*** 22µF CER 698k 1877 TA04 LTC1877 U TYPICAL APPLICATIONS Externally Synchronized 2.5V/0.6A Regulator Using all Ceramic Capacitors 2 RUN ITH 220pF PLL LPF SYNC/MODE 8 7 LTC1877 3 4 VFB VIN GND SW VIN 2.85V TO 10V VIN 4V TO 10V 0.01µF 1 Low Noise 2.5V/0.3A Regulator EXT CLOCK 700kHz 1 CIN** 10µF CER 10k 2 RUN ITH 220pF VOUT 2.5V /0.6A 5 10µH* 20pF 4 VFB VIN GND SW 6 5 VOUT 2.5V/0.3A 15µH* 20pF COUT*** 22µF CER 1877 TA05 U PACKAGE DESCRIPTION CIN** 10µF CER 7 887k 412k *TOKO D62CB A920CY-100M **TAIYO YUDEN CERAMIC LMK325BJ106MN ***TAIYO YUDEN CERAMIC JMK325BJ226MM SYNC/MODE 8 LTC1877 3 6 PLL LPF 887k COUT*** 47µF 6.3V 412k *SUMIDA CD54-150 **TAIYO YUDEN CERAMIC LMK325BJ106MN ***SANYO POSCAP 6TPA47M 1877 TA06 Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 ± 0.004* (3.00 ± 0.102) 0.040 ± 0.006 (1.02 ± 0.15) 0.007 (0.18) 0.034 ± 0.004 (0.86 ± 0.102) 8 7 6 5 0° – 6° TYP 0.021 ± 0.006 (0.53 ± 0.015) SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) TYP 0.006 ± 0.004 (0.15 ± 0.102) * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE 0.118 ± 0.004** (3.00 ± 0.102) 0.192 ± 0.004 (4.88 ± 0.10) 1 2 3 4 MSOP (MS8) 1197 15 LTC1877 U TYPICAL APPLICATION Single Lithium-Ion to 3.3V/0.3A Regulator 1 2 RUN ITH 220pF PLL LPF SYNC/MODE 10µF** 8 + – VIN Li-Ion BATTERY 3V TO 4.2V 7 LTC1877 3 4 VFB VIN GND SW 6 5 VOUT 3.3V/0.25A 10µH* 20pF 887k 47µF*** 280k *TOKO D62CB A920CY-100M **TAIYO YUDEN CERAMIC LMK325BJ106MN ***SANYO POSCAP 6TPA47M 1877 TA07 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1174/LTC1174-3.3 LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Monolithic Switching Regulators, IOUT to 450mA, Burst Mode Operation LTC1265 1.2A, High Efficiency Step-Down DC/DC Converter Constant Off-Time, Monolithic, Burst Mode Operation LT®1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators High Frequerncy, Small Inductor, High Efficiency LTC1436/LTC1436-PLL High Efficiency, Low Noise, Synchronous Step-Down Converters 24-Pin Narrow SSOP LTC1474/LTC1475 Low Quiescent Current Step-Down DC/DC Converters Monolithic, IOUT to 250mA, IQ = 10µA, 8-Pin MSOP LTC1504A Monolithic Synchronous Step-Down Switching Regulator Low Cost, Voltage Mode IOUT to 500mA, VIN from 4V to 10V LTC1622 Low Input Voltage Current Mode Step-Down DC/DC Controller High Frequency, High Efficiency, 8-Pin MSOP LTC1626 Low Voltage, High Efficiency Step-Down DC/DC Converter Monolithic, Constant Off-Time, IOUT to 600mA, Low Supply Voltage Range: 2.5V to 6V LTC1627 Monolithic Synchronous Step-Down Switching Regulator Constant Frequency, IOUT to 500mA, Secondary Winding Regulation, VIN from 2.65V to 8.5V LTC1701 Monolithic Current Mode Step-Down Switching Regulator Constant Off-Time, IOUT to 500mA, 1MHz Operation, VIN from 2.5V to 5.5V LTC1707 Monolithic Synchronous Step-Down Switching Regulator 1.19V VREF Pin, Constant Frequency, IOUT to 600mA, VIN from 2.65V to 8.5V LTC1735 High Efficiency, Synchronous Step-Down Converter 16-Pin SO and SSOP, VIN up to 36V, Fault Protection LTC1772 Low Input Voltage Current Mode Step-Down DC/DC Controller 550kHz, 6-Pin SOT-23, IOUT up to 5A, VIN from 2.2V to 10V LTC1878 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, VIN up to 6V, IQ = 10µA, IOUT to 600mA 16 Linear Technology Corporation 1877i LT/TP 0500 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 2000