LTC3405 1.5MHz, 300mA Synchronous Step-Down Regulator in ThinSOT U FEATURES DESCRIPTIO ■ The LTC ®3405 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 20µA and drops to <1µA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3405 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ High Efficiency: Up to 96% Very Low Quiescent Current: Only 20µA During Operation 300mA Output Current at VIN = 3V 2.5V to 5.5V Input Voltage Range 1.5MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle 0.8V Reference Allows Low Output Voltages Shutdown Mode Draws < 1µA Supply Current ±2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM Package Switching frequency is internally set at 1.5MHz, allowing the use of small surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.8V feedback reference voltage. The LTC3405 is available in a low profile (1mm) ThinSOT package. U APPLICATIO S ■ ■ ■ ■ ■ For new designs, refer to the LTC3405A data sheet. For fixed 1.5V and 1.8V output versions, refer to the LTC3405A-1.5/LTC3405A-1.8 data sheet. Cellular Telephones Personal Information Appliances Wireless and DSL Modems Digital Still Cameras MP3 Players Portable Instruments , LTC and LT are registered trademarks of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. Protected by U.S. Patents, including 6580258, 5481178. U ■ TYPICAL APPLICATIO 100 95 4 † CIN 2.2µF CER VIN SW 3 22pF LTC3405 1 6 4.7µH** VOUT* 3.3V + RUN VFB MODE GND 2 5 COUT†† 33µF 887k 280k 3405 F01a *VOUT CONNECTED TO VIN FOR 2.7V < VIN < 3.3V **MURATA LQH3C4R7M34 † TAIYO YUDEN LMK212BJ225MG †† AVX TPSB336K006R0600 90 EFFICIENCY (%) VIN 2.7V TO 5.5V VIN = 3.6V 85 80 VIN = 4.2V 75 VIN = 5.5V 70 65 60 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3405 F01b Figure 1a. High Efficiency Step-Down Converter Figure 1b. Efficiency vs Load Current 3405fa 1 LTC3405 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Supply Voltage .................................. – 0.3V to 6V MODE, RUN, VFB Voltages ......................... – 0.3V to VIN SW Voltage .................................. – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 400mA N-Channel Switch Sink Current (DC) ................. 400mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................ 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW 6 MODE RUN 1 GND 2 5 VFB SW 3 4 VIN LTC3405ES6 S6 PACKAGE 6-LEAD PLASTIC TSOT-23 S6 PART MARKING TJMAX = 125°C, θJA = 250°C/ W LTXQ Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN IVFB Feedback Current IPK Peak Inductor Current VIN = 3V, VFB = 0.7V, Duty Cycle < 35% 375 VFB Regulated Feedback Voltage (Note 4) ● 0.784 ∆VOVL ∆Output Overvoltage Lockout ∆VOVL = VOVL – VFB ● 20 ∆VFB Reference Voltage Line Regulation VIN = 2.5V to 5.5V (Note 4) ● VLOADREG Output Voltage Load Regulation VIN Input Voltage Range IS Input DC Bias Current Pulse Skipping Mode Burst Mode® Operation Shutdown (Note 5) VFB = 0.7V, Mode = 3.6V, ILOAD = 0A VFB = 0.83V, Mode = 0V, ILOAD = 0A VRUN = 0V, VIN = 4.2V fOSC Oscillator Frequency VFB = 0.8V VFB = 0V RPFET RDS(ON) of P-Channel FET ISW = 100mA RNFET RDS(ON) of N-Channel FET ISW = –100mA ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V VRUN RUN Threshold ● IRUN RUN Leakage Current ● VMODE MODE Threshold ● IMODE MODE Leakage Current ● TYP MAX UNITS ±30 nA 500 625 mA 0.8 0.816 50 80 mV 0.04 0.4 %/V ● V 0.5 ● ● 2.5 1.2 0.3 0.3 % 5.5 V 300 20 0.1 400 35 1 µA µA µA 1.5 210 1.8 MHz kHz 0.7 0.85 Ω 0.6 0.90 Ω ±0.01 ±1 µA 1 1.5 V ±0.01 ±1 µA 1.5 2 V ±0.01 ±1 µA Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3405E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3405: TJ = TA + (PD)(250°C/W) Note 4: The LTC3405 is tested in a proprietary test mode that connects VFB to the output of the error amplifier. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. 3405fa 2 LTC3405 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure1a Except for the Resistive Divider Resistor Values) Efficiency vs Input Voltage 100 100 IOUT = 100mA 80 IOUT = 10mA IOUT = 1mA IOUT = 250mA 80 75 70 65 VIN = 4.2V 60 VIN = 3.6V 50 VIN = 4.2V 40 50 2.5 10 3.5 4.0 4.5 INPUT VOLTAGE (V) 3.0 VIN = 4.2V 70 5.0 5.5 50 VOUT = 1.8V 0 0.1 3405 G02 3405 G04 Oscillator Frequency vs Temperature Reference Voltage vs Temperature 0.814 100 1.70 VIN = 3.6V VIN = 3.6V 60 VIN = 4.2V 1.60 FREQUENCY (MHz) REFERENCE VOLTAGE (V) EFFICIENCY (%) 80 70 VIN = 3.6V 1.65 0.809 VIN = 2.7V 0.804 0.799 0.794 40 0.1 VOUT = 1.3V 1 100 10 OUTPUT CURRENT (mA) 50 25 75 0 TEMPERATURE (°C) 100 1.45 125 1.30 –50 –25 Oscillator Frequency vs Supply Voltage 1.834 1.7 1.824 1.5 1.4 125 RDS(ON) vs Input Voltage 1.2 1.1 Burst Mode OPERATION 1.0 0.9 PULSE SKIPPING MODE 1.814 MAIN SWITCH 0.8 RDS(0N) (Ω) OUTPUT VOLTAGE (V) 1.8 100 3405 G07 Output Voltage vs Load Current 1.6 50 25 75 0 TEMPERATURE (°C) 3405 G06 3405 G05 OSCILLATOR FREQUENCY (MHz) 1.50 1.35 0.784 –50 –25 1000 1.55 1.40 0.789 50 1000 1 100 10 OUTPUT CURRENT (mA) 3405 G03 Efficiency vs Output Current 90 VOUT = 1.8V 40 0.1 1000 1 100 10 OUTPUT CURRENT (mA) VIN = 5.5V 60 PULSE SKIPPING MODE Burst Mode OPERATION 20 Burst Mode OPERATION VOUT = 1.8V 55 VIN = 3.6V 80 30 IOUT = 0.1mA 60 90 70 EFFICIENCY (%) 85 VIN = 2.7V 90 V = 3.6V IN EFFICIENCY (%) 90 EFFICIENCY (%) Efficiency vs Output Current Efficiency vs Output Current 95 1.804 1.794 0.7 0.6 SYNCHRONOUS SWITCH 0.5 0.4 0.3 1.784 1.3 0.2 VIN = 3.6V 1.2 1.774 2 3 4 5 SUPPLY VOLTAGE (V) 6 3405 G08 0 100 200 300 400 LOAD CURRENT (mA) 0.1 500 600 3405 G09 0 0 1 3 2 5 4 INPUT VOLTAGE (V) 6 7 3405 G10 3405fa 3 LTC3405 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) RDS(ON) vs Temperature 600 1600 1.0 V = 2.7V IN DYNAMIC SUPPLY CURRENT (µA) VIN = 4.2V VIN = 3.6V 0.8 0.6 0.4 0.2 1400 1200 1000 800 600 PULSE SKIPPING MODE 400 200 SYNCHRONOUS SWITCH MAIN SWITCH 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 VOUT = 1.8V ILOAD = 0A 400 PULSE SKIPPING MODE 300 200 100 Burst Mode OPERATION 0 125 2 3 4 5 SUPPLY VOLTAGE (V) 6 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 3405 G12 Switch Leakage vs Temperature 100 125 3405 G13 Switch Leakage vs Input Voltage Burst Mode Operation 60 160 VIN = 5.5V 140 RUN = 0V RUN = 0V 50 120 SWITCH LEAKAGE (pA) SWITCH LEAKAGE (nA) 500 VIN = 3.6V VOUT = 1.8V ILOAD = 0A Burst Mode OPERATION 3405 G11 100 80 60 SYNCHRONOUS SWITCH 40 40 VOUT 50mV/DIV AC COUPLED 30 20 IL 100mA/DIV MAIN SWITCH 0 50 25 75 0 TEMPERATURE (°C) SW 5V/DIV SYNCHRONOUS SWITCH 10 MAIN SWITCH 20 0 –50 –25 DYNAMIC SUPPLY CURRENT (µA) 1.2 RDS(ON) (Ω) Dynamic Supply Current vs Temperature Dynamic Supply Current 100 125 0 1 2 3 4 INPUT VOLTAGE (V) 5 6 VIN = 3.6V VOUT = 1.8V ILOAD = 20mA 3405 G15 3405 G14 Pulse Skipping Mode Operation Start-Up from Shutdown RUN 2V/DIV VOUT 100mV/DIV AC COUPLED VOUT 20mV/DIV AC COUPLED VOUT 1V/DIV IL 200mA/DIV IL 200mA/DIV ILOAD 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 20mA 500ns/DIV 3405 G17 VIN = 3.6V VOUT = 1.8V ILOAD = 250mA 3405 G16 Load Step SW 5V/DIV IL 100mA/DIV 5µs/DIV 100µs/DIV 3405 G18 VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 0mA TO 250mA PULSE SKIPPING MODE 3405 G19 3405fa 4 LTC3405 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) Load Step Load Step VOUT 100mV/DIV AC COUPLED Load Step VOUT 100mV/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED IL 200mA/DIV IL 200mA/DIV ILOAD 200mA/DIV ILOAD 200mA/DIV VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 20mA TO 250mA PULSE SKIPPING MODE 3405 G20 IL 200mA/DIV ILOAD 200mA/DIV VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 20mA TO 250mA Burst Mode OPERATION 3405 G21 VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 0mA TO 250mA Burst Mode OPERATION 3405 G22 U U U PI FU CTIO S RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1µA supply current. Do not leave RUN floating. GND (Pin 2): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2µF or greater ceramic capacitor. VFB (Pin 5): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. MODE (Pin 6): Mode Select Input. To select pulse skipping mode, tie to VIN. Grounding this pin selects Burst Mode operation. Do not leave this pin floating. 3405fa 5 LTC3405 W FU CTIO AL DIAGRA U U MODE 6 SLOPE COMP 0.65V OSC OSC 4 VIN FREQ SHIFT – VFB + 5 – + 0.8V 0.4V – EA SLEEP – + S Q R Q RS LATCH RUN – OVDET 0.85V SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW OV + SHUTDOWN + 0.8V REF 5Ω + ICOMP BURST VIN 1 EN IRCMP 2 GND – 3405 BD U OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC3405 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. 3405fa 6 LTC3405 U OPERATIO (Refer to Functional Diagram) Comparator OVDET guards against transient overshoots > 6.25% by turning the main switch off and keeping it off until the fault is removed. until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. Burst Mode Operation Another important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3405 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 100mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 20µA. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier’s output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. Low Supply Operation The LTC3405 will operate with input supply voltages as low as 2.5V, but the maximum allowable output current is reduced at this low voltage. Figure 2 shows the reduction in the maximum output current as a function of input voltage for various output voltages. 600 VOUT = 1.8V MAXIMUM OUTPUT CURRENT (mA) The LTC3405 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. To enable Burst Mode operation, simply connect the MODE pin to GND. To disable Burst Mode operation and enable PWM pulse skipping mode, connect the MODE pin to VIN or drive it with a logic high (VMODE > 1.5V). In this mode, the efficiency is lower at light loads, but becomes comparable to Burst Mode operation when the output load exceeds 25mA. The advantage of pulse skipping mode is lower output ripple and less interference to audio circuitry. 500 VOUT = 1.3V 400 VOUT = 2.5V 300 200 100 0 2.5 3.0 3.5 4.0 4.5 SUPPLY VOLTAGE (V) 5.0 5.5 3405 G23 Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 210kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 1.5MHz when VFB rises above 0V. Dropout Operation As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle Figure 2. Maximum Output Current vs Input Voltage Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3405 uses a patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 3405fa 7 LTC3405 U W U U APPLICATIO S I FOR ATIO The basic LTC3405 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Table 1. Representative Surface Mount Inductors MANUFACTURER PART NUMBER MAX DC VALUE CURRENT DCR HEIGHT Taiyo Yuden LB2016T3R3M 3.3µH 280mA 0.2Ω 1.6mm Panasonic ELT5KT4R7M 4.7µH 950mA 0.2Ω 1.2mm Inductor Selection Murata LQH3C4R7M34 4.7µH 450mA 0.2Ω For most applications, the value of the inductor will fall in the range of 3.3µH to 10µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 120mA (40% of 300mA). Taiyo Yuden LB2016T4R7M 4.7µH 210mA 0.25Ω 1.6mm Panasonic ELT5KT6R8M 6.8µH 760mA 0.3Ω 1.2mm Panasonic ELT5KT100M 10µH 680mA 0.36Ω 1.2mm Sumida CMD4D116R8MC 6.8µH 620mA 0.23Ω 1.2mm ∆IL = ⎛ V ⎞ 1 VOUT ⎜ 1 − OUT ⎟ ( f)(L) ⎝ VIN ⎠ (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3405 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3405 applications. 8 2mm CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: 1/ 2 VOUT (VIN − VOUT )] [ CIN required IRMS ≅ IOMAX VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). An ESR in the range of 100mΩ to 200mΩ is necessary to provide a stable loop. For the LTC3405, the general rule for proper operation is: 0.1Ω ≤ COUT required ESR ≤ 0.6Ω ESR is a direct function of the volume of the capacitor; that is, physically larger capacitors have lower ESR. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ 3405fa LTC3405 U W U U APPLICATIO S I FOR ATIO where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Another solution is to connect the feedback resistor to the SW pin as shown in Figure 4. Taking the feedback information at the SW pin removes the phase lag due to the output capacitor resulting in a very stable loop. This configuration lowers the load regulation by the DC resistance of the inductor multiplied by the load current. This slight shift in load regulation actually helps reduce the overshoot and undershoot of the output voltage during a load transient. VIN 2.7V TO 4.2V 4 CIN 2.2µF CER VIN SW LTC3405 1 6 4.7µH 3 887k 22pF RUN VFB MODE 5 GND VOUT 1.5V COUT 4.7µF CER 1M 2 3405 F04 Using Ceramic Input and Output Capacitors Figure 4. Using All Ceramic Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When ceramic capacitors are used at the output, their low ESR cannot provide sufficient phase lag cancellation to stabilize the loop. One solution is to use a tantalum capacitor, with its higher ESR, to provide the bulk capacitance and parallel it with a small ceramic capacitor to reduce the ripple voltage as shown in Figure 3. VIN 2.7V TO 4.2V 4 CIN 2.2µF CER VIN SW 3 22pF LTC3405 1 6 4.7µH RUN VFB MODE GND 2 5 COUT1 + 1µF CER VOUT 1.5V COUT2 22µF TANT 887k 1M A third solution is to use a high value resistor to inject a feedforward signal at VFB mimicking the ripple voltage of a high ESR output capacitor. The circuit in Figure 5 shows how this technique can be easily realized. The feedforward resistor, R2B, is connected to SW as in the previous example. However, in this case, the feedback information is taken from the resistive divider, R2A and R1, at the output. This eliminates most of the load regulation degradation due to the DC resistance of the inductor while providing a stable operation similar to that obtained from a high ESR tantalum type capacitor. Using this technique, the extra feedforward resistor, R2B, must be accounted for when calculating the resistive divider as follows: R2A • R2B R2A + R2B ⎛ R2⎞ = 0.8V ⎜ 1 + ⎟ ⎝ R1⎠ R2 = R2A || R2B = VOUT VIN 2.7V TO 4.2V 4 CIN 2.2µF CER VIN SW 3 R2B 22pF 1M LTC3405 1 6 4.7µH RUN VFB MODE GND 2 5 R2A R1 215k 200k 3405 F03 Figure 3. Paralleling a Ceramic with a Tantalum Capacitor VOUT 1.5V COUT1 4.7µF CER 3405 F05 Figure 5. Feedforward Injection in an All Ceramic Capacitor Application 3405fa 9 LTC3405 U W U U APPLICATIO S I FOR ATIO In pulse skipping mode, the LTC3405 is stable with a 4.7µF ceramic output capacitor with VIN ≤ 4.2V. For single Li-Ion applications operating in pulse skipping mode, the circuit shown in Figure 6 can be used When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. 4 CIN 2.2µF CER VIN SW 3 6 VOUT 1.5V 22pF LTC3405 1 4.7µH RUN VFB MODE GND 2 1 COUT1 4.7µF CER 5 VIN = 3.6V 0.1 887k 1M 3405 F06 Figure 6. Using All Ceramic Capacitors in Pulse Skipping Mode POWER LOST (W) VIN 2.7V TO 4.2V Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3405 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 8. VOUT = 1.8V 0.01 0.001 Output Voltage Programming VOUT = 1.3V The output voltage is set by a resistive divider according to the following formula: VOUT 0.0001 0.1 1 100 10 LOAD CURRENT (mA) 1000 3405 F08 ⎛ R2⎞ = 0.8V ⎜ 1 + ⎟ ⎝ R1⎠ (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 7. 0.8V ≤ VOUT ≤ 5.5V R2 VFB LTC3405 VOUT = 3.3V VOUT = 2.5V R1 GND 3405 F07 Figure 7. Setting the LTC3405 Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Figure 8. Power Lost vs Load Current 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both 3405fa 10 LTC3405 U W U U APPLICATIO S I FOR ATIO top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.94Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 84.6mW For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.0846)(250) = 91.15°C Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. which is well below the maximum junction temperature of 125°C. Thermal Considerations Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). In most applications the LTC3405 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3405 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3405 from exceeding the maximum junction temperature, the user will need to do a thermal analysis. The goal of the thermal analysis is to determine whether the operating conditions exceed the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: T J = TA + TR where TA is the ambient temperature. As an example, consider the LTC3405 in dropout at an input voltage of 2.7V, a load current of 300mA and an Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. 3405fa 11 LTC3405 U W U U APPLICATIO S I FOR ATIO PC Board Layout Checklist 4. Keep the switching node, SW, away from the sensitive VFB node. When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3405. These items are also illustrated graphically in Figures 9 and 10. Check the following in your layout: Design Example As a design example, assume the LTC3405 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.25A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 1 L= RUN MODE ⎛ V ⎞ 1 VOUT ⎜ 1 − OUT ⎟ ( f)(∆IL ) ⎝ VIN ⎠ (3) 6 LTC3405 2 GND – VFB 5 COUT VOUT + R2 3 L1 SW VIN R1 4 CFWD CIN R3* + VIN – 3405 F09 BOLD LINES INDICATE HIGH CURRENT PATHS *ADD R3 FOR APPLICATIONS USING A CERAMIC COUT Figure 9. LTC3405 Layout Diagram VIA TO SW NODE R3* VOUT VIA TO GND VFB R1 VIN VIA TO VIN VIA TO VOUT R2 PIN 1 L1 CFWD LTC3405 SW COUT CIN GND *ADD R3 WHEN USING CERAMIC COUT 3405 F10 Figure 10. LTC3405 Suggested Layout 3405fa 12 LTC3405 U W U U APPLICATIO S I FOR ATIO Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 100mA and f = 1.5MHz in equation (3) gives: L= For the feedback resistors, choose R1 = 412k. R2 can then be calculated from equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1 = 875.5k; use 887k ⎝ 0.8 ⎠ 2.5V ⎛ 2.5V ⎞ ⎜1 − ⎟ ≅ 6.8µH 1.5MHz(100mA) ⎝ 4.2V ⎠ Figure 11 shows the complete circuit along with its efficiency curve. For best efficiency choose a 300mA or greater inductor with less than 0.3Ω series resistance. 100 CIN will require an RMS current rating of at least 0.125A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.6Ω and greater than 0.1Ω. In most cases, a tantalum capacitor will satisfy this requirement. 4 † CIN 2.2µF CER VIN SW 6.8µH* 3 6 VOUT 2.5V 22pF LTC3405 1 + VFB GND 2 5 3405 F11a 60 40 887k 412k VIN = 4.2V 70 50 COUT** 33µF TANT RUN MODE VIN = 3.6V 80 EFFICIENCY (%) VIN 2.7V TO 4.2V VIN = 2.7V 90 30 0.1 *SUMIDA CMD4D11-6R8MC ** AVX TPSB336K006R0600 † TAIYO YUDEN LMK212BJ225MG 1 100 10 OUTPUT CURRENT (mA) 1000 3405 F11b Figure 11a Figure 11b U TYPICAL APPLICATIO S Single Li-Ion to 1.8V/300mA Regulator Optimized for Small Footprint and High Efficiency VIN 2.7V TO 4.2V 4 CIN** 1µF CER VIN SW LTC3405 1 6 4.7µH* 3 1M RUN VFB MODE GND 2 5 22pF VOUT 1.8V COUT† 4.7µF CER 332k *MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC JMK107BJ105MA † 3405 TA01a TAIYO YUDEN CERAMIC JMK212BJ475MG 200k 100 VIN = 2.7V 90 VIN = 3.6V 80 EFFICIENCY (%) VOUT 100mV/DIV AC COUPLED 70 60 IL 200mA/DIV VIN = 4.2V 50 ILOAD 200mA/DIV 40 30 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3405 TA01b VIN = 3.6V 40µs/DIV ILOAD = 100mA TO 250mA 3405 TA01c 3405fa 13 LTC3405 U TYPICAL APPLICATIO S Single Li-Ion to 1.8V/300mA Regulator Using Ceramic and Tantalum Output Capacitors VIN 2.7V TO 4.2V 4 CIN** 2.2µF CER VIN SW 4.7µH* 3 22pF LTC3405 1 RUN 6 MODE VFB 5 COUT1*** + 1µF CER VOUT 1.8V COUT2† 22µF TANT 887k *MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC LMK212BJ225MG ***TAIYO YUDEN CERAMIC JMK107BJ105MA † AVX TAJA226M006R 3405 TA02a GND 698k 2 100 VIN = 2.7V VOUT 100mV/DIV AC COUPLED 90 VIN = 4.2V EFFICIENCY (%) 80 VIN = 3.6V 70 IL 200mA/DIV 60 50 ILOAD 200mA/DIV 40 30 0.1 VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 100mA TO 250mA 1000 1 100 10 OUTPUT CURRENT (mA) 3405 TA02b 3405 TA02c Single Li-Ion to 1.8V/200mA Regulator Using All Ceramic Capacitors Optimized for Smallest Footprint VIN 2.7V TO 4.2V 4 CIN** 1µF CER VIN SW LTC3405 1 6 1M RUN VFB MODE GND 2 3.3µH* 3 5 22pF VOUT 1.8V COUT† 4.7µF CER 332k 200k *TAIYO YUDEN LB2016T3R3M **TAIYO YUDEN CERAMIC JMK107BJ105MA † 3405 TA03a TAIYO YUDEN CERAMIC JMK212BJ475MG 100 VOUT 100mV/DIV AC COUPLED VIN = 2.7V 90 EFFICIENCY (%) 80 VIN = 3.6V 70 60 IL 200mA/DIV VIN = 4.2V 50 ILOAD 200mA/DIV 40 30 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3405 TA03b VIN = 3.6V 40µs/DIV ILOAD = 100mA TO 250mA 3405 TA03c 3405fa 14 LTC3405 U TYPICAL APPLICATIO S Single Li-Ion to 1.8V/300mA Regulator Using All Ceramic Capacitors Optimized for Lowest Profile, ≤ 1.2mm High VIN 2.7V TO 4.2V 4 CIN** 1µF CER VIN SW LTC3405 1 6 4.7µH* 3 22pF 1M COUT** 1µF CER RUN VFB MODE 5 GND 200k 2 VOUT 1.8V COUT** 1µF CER 332k *PANASONIC ELT5KT4R7M **TAIYO YUDEN CERAMIC JMK107BJ105MA 3405 TA04a 100 VIN = 2.7V 90 VOUT 100mV/DIV AC COUPLED EFFICIENCY (%) 80 VIN = 3.6V 70 IL 200mA/DIV VIN = 4.2V 60 50 ILOAD 200mA/DIV 40 30 0.1 VIN = 3.6V 40µs/DIV ILOAD = 100mA TO 250mA 1000 1 100 10 OUTPUT CURRENT (mA) 3405 TA04c 3405 TA04b U PACKAGE DESCRIPTIO S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 0.62 MAX 2.90 BSC (NOTE 4) 0.95 REF 1.22 REF 3.85 MAX 2.62 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 6 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 3405fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3405 U TYPICAL APPLICATIO S Single Li-Ion to 1.8V/300mA Regulator All Ceramic Capacitors with Lowest Parts Count VIN 2.7V TO 4.2V 4 CIN** 2.2µF CER VIN SW LTC3405 1 6 4.7µH* 3 887k 22pF RUN VFB MODE 5 GND 2 698k 3405 TA05a VOUT 1.8V COUT† 4.7µF CER *MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC LMK212BJ225MG † TAIYO YUDEN CERAMIC JMK212BJ475MG 100 VIN = 2.7V VOUT 100mV/DIV AC COUPLED 90 VIN = 4.2V EFFICIENCY (%) 80 70 VIN = 3.6V IL 200mA/DIV 60 50 ILOAD 200mA/DIV 40 30 0.1 1000 1 100 10 OUTPUT CURRENT (mA) VIN = 3.6V 40µs/DIV ILOAD = 100mA TO 250mA 3405 TA05b 3405 TA05c RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1174/LTC1174-3.3 LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Monolithic Switching Regulators, I OUT to 450mA, Burst Mode Operation LTC1265 1.2A, High Efficiency Step-Down DC/DC Converter Constant Off-Time, Monolithic, Burst Mode Operation LTC1474/LTC1475 Low Quiescent Current Step-Down DC/DC Converters Monolithic, IOUT to 250mA, IQ = 10µA, 8-Pin MSOP LTC1504A Monolithic Synchronous Step-Down Switching Regulator Low Cost, Voltage Mode IOUT to 500mA, VIN from 4V to 10V LT1616 600mA, 1.4MHz Step-Down DC/DC Converter 6-Pin ThinSOT, VIN from 3.6V to 25V LTC1627 Monolithic Synchronous Step-Down Switching Regulator Constant Frequency, IOUT to 500mA, Secondary Winding Regulation, VIN from 2.65V to 8.5V LTC1701 Monolithic Current Mode Step-Down Switching Regulator Constant Off-Time, IOUT to 500mA, 1MHz Operation, VIN from 2.5V to 5.5V LTC1707 Monolithic Synchronous Step-Down Switching Regulator 1.19V VREF Pin, Constant Frequency, IOUT to 600mA, VIN from 2.65V to 8.5V LTC1767 1.5A, 1.25MHz Step-Down Switching Regulator 3V to 25V Input, 8-Lead MSOP Package LTC1779 Monolithic Current Mode Step-Down Switching Regulator 550kHz, 6-Lead ThinSOT, V IN from 2.5V to 9.8V LTC1877 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, VIN Up to 10V, IQ = 10µA, IOUT to 600mA at VIN = 5V LTC1878 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA at VIN = 3.3V LTC3404 1.4MHz High Efficiency Monolithic Step-Down Regulator 1.4MHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA at VIN = 3.3V LTC3405A 1.5MHz High Efficiency Monolithic Step-Down Regulator Stable with Ceramic Output Capacitor LTC3405A-1.5/ LTC3405A-1.8 1.5MHz High Efficiency Monolithic Step-Down Regulator Fixed Output Version of LTC3405A 3405fa 16 Linear Technology Corporation LT/TP 0604 1K REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2001