ETC L6562

带极低 ηD的 过渡模 式 PFC控 制器 L甾 血
,65-
●元 器 件 卡 片
THD的
带锻 低
过 渡镆 式
PFC控 制
帮
L6562
王 守志
(临 沂师范学院 工程 学院,山
继 LCsω 和 L6561之 后 ,sT公 司不久前又推出
了一种性能 改进 的低成本经 济型功率 因数校正
(PFC)控 制器 L甾 Ω。这种工作在过渡模式的电流型
PFC控 制器与
引脚相兼容 ,其 主要不同点
"561的
AC输 人电流总谐波失
是在高线性乘法器中嵌人了
(Ⅱ
m)最
真
优化电路 ,从 而能在宽范围的 AC线 路输
人电压和一个大的负载范围内提供非常低的 Ⅱm
及高次谐波成份。
L662可 用作 PFC升 压预变换器 以
及构成符合 EC⒍ OCXl-3-2谐 波电流限
制标准的 SCXlW开 关电源 (如 IV、 台式 PC I
监视器电源等 );也 可用于高端 AC-DC
适配器/充 电器、人 口电平服务器及 Web
服务器等。
东 临沂
z9COOs)
。
“562的 其它特点如下
●具有 10.3~夕y的 宽电源电压范围
●具有低于 70FtA的 启动电流和低于 0mA的 工
作电流 ,并 且含有截止功能 ,因 而特别适用于遥控
”“
开/关 控制,并且能满足 “
蓝天使 、 能源之星”和
“
’
Energy⒛ 盯 等标准
●借助于电压误差放大器和 ±1%的 内部精密
电压参考 ,可 控制 PFC的 DC输 出电压并使其高度
:
;
;
5置
王
鳅
Ⅴ
1
|
“ 5m的 主要特 点
Lss猊 采用 8引 脚
DP和 s0封 装
starter
,
P田
芯片电路组成如图 1所 示 。与 “甾1相
比 ,Lss⒍ 的内部乘法器带有 THE,最 低
化专门电路 ,能 有效控制 AC输 人电流的
交 越失真和误 差放大器输 出纹波失 真
从而提供 高功率因数和非常低的 ⅡD。
赢
鎏
笳
,
∞
0~氵
,》 o”
冫莎
》
o卜 》
莎
》
o卜 》
》
》 丬
图 1 LssΩ 芯片的组成框图
~术 >》
》
(⑶
冫 》》 艹
尽量不要选用 MCs51系 列单片机 ,因 为该
单片机在国内的普及程度最高 ,被 研究得也最透。
(ω 产品的原创者 ,一 般具有产量大的特点 ,所
以可选用比较生僻、偏冷门的单片机来加大仿冒者
采购的难度 。
(o选 择采用新工艺 、新结构 、上市时间较短 的
单片机 ,如 朋Ⅶ⒒ AVR系 列单片机等。
(s,在 设计成本许可的条件下 :应 选用具有硬
件自毁功能的智能卡芯片 ,以 有效对付物理攻击。
(ω 如果条件许可 ,可 来用两片不同型号单片
机互为备份 ,相 互验证 ,从 而增加破解成本。
(D打 磨掉芯片型号等信息或者重
上其它
诃贸
^
∞ 之 ” 父
>》
氵 ” 大
,》
氵 卜莎
》
冫 卜》
艹
Ⅱ ◇ 》》
莎 ∞
o
・
的型号 ,以 假乱真。
当然 ,要 想从根本上防止单片机被解密 ,程 序被
盗版等侵权行为发生 ,只 能依靠法律手段来保障。
参考文献
["sergei P.skombogatov:Copy Rd∞ hm in Modem
Micmconm虬 邛⒛胧 。
[2]Ross J.Andenn,Matkus C。
tacks c,n Tamper Resistant D甜
Kuhn:1ow c∞ t
漶 ,in
At
M.L。 ms⒍ d.
(∞ 。),弘阳灯 Protocols,5山 htematioml
Work-
sh。 p,P耐s,Fmnce,AⅡ I7-9,1999。
’
收稿 日期 :zOOd-O3~17
∷
咨询编号 :00Ogao
⒛α 年第 9期
《o外 电番兖子件》
-66-
⒛∝ 年 9月
Ⅴo=400Ⅴ
PO=80W
180ko 180kΩ
100Ω
o」L^f・
lN云 ∴
0nF12kΩ
FUsE
1/250V
~VaC
~265V
~
MOs
85Ⅴ
sTP8NlW50
皿
‰
喹‰芈
lOnF I 屮Π
22uF 芈
R1
F
图 2 由 LbsΩ 控制 g9BOw1M
稳定
;
●有过电压保护功能 ,能 安全处理启动和负载
断开时产生的过电压
●在电流感测脚内嵌 RC低 通滤波电路 ,可 减
少外部元件数量和 PCB面 积
●带有源电流/灌 电流为 一bO0/SOOmA的 推挽
式输出级 ,并 带有欠压锁定 (UW【 ,)下 拉和 1sV的 电
IGBr,从 币可使变
压钳位 ,可 驱动功率 MC,sFEr或 ∷
;
;
换器输出功率高达 300W。
2 L6562的 ⅡD最 优化 电路
针对导致 ⅡD恶 化的原 因 ,吓 Ω 在其内部乘
法器单元中 ,嵌 人了 △D最 优化专门电路。该电路
能处痤 AC线 路电压过零附近积聚0q能 量 ,从 而使
桥式整流器之后的高频滤波器电容得 以充分放电
以减小交越失真 ,降 低 ⅡD。
结 合 高线 性 乘 法 器 中 的 △D最 优 化 电路
,
PFC升 压变换器 电路
L甾 ⒍ 允许在误差放大器反相输人端 (ⅡW脚 )和输
出端 (CC,MP脚 )之 间连接 RC串 联补偿网络 ,以 减小
误差放大器输 出纹波和乘法器输出的高次谐波。与
““1比 较 ,Lbs⒍ 性能有较明显的墀升 ,但 成本并
不增加。
3
应用电路及其 PFC效 果
用 “562作 控制器 的 gOw TM PFc升 压变换器
电路如图 2所 示。其中 ,磁 性元件 Tl选 用 ψ饰 13×
7磁 芯 ,初 级绕组 1Os匝 (用 ⒛ ×0.l-绞 合线 ),电
感值 L选 为 0。 TmH,次 级绕组 11匝 (用 线径 为
o。 l-的 绝缘导线 )。 LCsΩ 脚 1和 脚 2之 间连接的
R50与 C3组 成的串联 网络用作误差放大器的纹波
补偿 ,C⒛ 用作斜率补偿 。
收稿 日期 :⒛ 阴 -O3-O3
咨询编号 :o佃 鸵1
,
艹
” ” 冖 ” ” 夕 ¨ ¨ 夕
莎 莎 》 莎 》 》 》 ≥ 》 沙 莎 》 》 莎 》 莎 》 莎 万
●元 器 件 快 讯
针对
≥ 莎 ≥
CDMAzO001X的 双 频 RF CMOs接 收 芯 片
——高通
码分多址 (CDMA)数 字无线技术产品制造商
公司 (QUAIcOMM Incorpomted)宣 布推出该公司第一种针对
CDMA⒛ ∞ 1X的 双频 RF CMOs接 收芯片 RFRs1ss和 用于实
现双频 漫游 功 能 的 RΠ 61sO转 换装 置 。RFRb155还 整合 了
RFR61ss是 经过成本优化的 叫蚰0ne接 收器 ,可 提供从
射频到基带的完整降频转换 ,其 中包括整合式低 噪放大器
(LNA)、 压控振荡器 (VCo)、 本地振荡器 (∞ )生 成、
锁相环
(PLLl、 直接降频混频器以及整合基带滤波器 ,从 而可使系统
GPs接 收器 。该接 收器支持 高通公 司在全球应 用最广 的 盱
鼓sted L GPs技 术 —— gpsOnem解 决 方 案 。 RFR6155和
RFrs1sO芯 片是 血M%e双 频转换芯片 ,可 提供从
能耗降低。
基带到射频的完整升频转换 ,其 经过优化的尺寸设计可进一
RFrc1sO芯 片是极具成本效 益 的 钣MⅡe(TM)零 中频
步节约空间。
(zIF)
解 决 方 案 ,该 方 案 既 支 持 日本 低 频 CDMA/Cendar(gcxl
MI】 z),也 支持韩 国高频
m(勿
PCs(1gO0M比 )/Pcs(19IX,MIIz)和
∞ M施 )频 率 。
~
R「 rc1so采 用
s-xs-32引 脚 QFN封 装 ,RFRC1s5采
用 ⒍mx⒍ m00引 脚 QFN封 装。
′
咨询编号 :00Ogm
L6562
TRANSITION-MODE PFC CONTROLLER
1
■
■
■
■
■
■
■
■
■
■
■
■
Features
Figure 1. Packages
REALISED IN BCD TECHNOLOGY
TRANSITION-MODE CONTROL OF PFC PREREGULATORS
PROPRIETARY MULTIPLIER DESIGN FOR
MINIMUM THD OF AC INPUT CURRENT
VERY PRECISE ADJUSTABLE OUTPUT
OVERVOLTAGE PROTECTION
ULTRA-LOW (≤70µA) START-UP CURRENT
LOW (≤4 mA) QUIESCENT CURRENT
EXTENDED IC SUPPLY VOLTAGE RANGE
ON-CHIP FILTER ON CURRENT SENSE
DISABLE FUNCTION
1% (@ Tj = 25 °C) INTERNAL REFERENCE
VOLTAGE
-600/+800mA TOTEM POLE GATE DRIVER WITH
UVLO PULL-DOWN AND VOLTAGE CLAMP
DIP-8/SO-8 PACKAGES ECOPACK®
1.1 APPLICATIONS
■
PFC PRE-REGULATORS FOR:
– IEC61000-3-2 COMPLIANT SMPS
(TV,
SO-8
DIP-8
Table 1. Order Codes
Part Number
Package
L6562N
DIP-8
L6562D
SO-8
L6562DTR
Tape & Reel
DESKTOP PC, MONITOR) UP TO 300W
– HI-END AC-DC ADAPTER/CHARGER
– ENTRY LEVEL SERVER & WEB SERVER
2
Description
The L6562 is a current-mode PFC controller operating in Transition Mode (TM). Pin-to-pin compatible with the predecessor L6561, it offers improved
performance.
Figure 2. Block Diagram
COMP
MULT
2
3
CS
4
1
INV
-
MULTIPLIER AND
THD OPTIMIZER
+
40K
2.5V
VOLTAGE
OVERVOLTAGE
DETECTION
REGULATOR
5pF
+
-
VCC
8
VCC
INTERNAL
R
SUPPLY 7V
25 V
15 V
Q
S
R1
7
GD
+
R2
VREF2
DRIVER
UVLO
-
Starter
stop
ZERO CURRENT
DETECTOR
+
2.1 V
1.6 V
STARTER
-
DISABLE
6
GND
November 2005
5
ZCD
Rev. 8
1/16
L6562
2 Description (continued)
The highly linear multiplier includes a special circuit, able to reduce AC input current distortion, that allows
wide-range-mains operation with an extremely low THD, even over a large load range.
The output voltage is controlled by means of a voltage-mode error amplifier and a precise (1% @Tj =
25°C) internal voltage reference.
The device features extremely low consumption (≤70 µA before start-up and <4 mA running) and includes
a disable function suitable for IC remote ON/OFF, which makes it easier to comply with energy saving
norms (Blue Angel, EnergyStar, Energy2000, etc.).
An effective two-step OVP enables to safely handle overvoltages either occurring at start-up or resulting
from load disconnection.
The totem-pole output stage, capable of 600 mA source and 800 mA sink current, is suitable for big MOSFET or IGBT drive which, combined with the other features, makes the device an excellent low-cost solution for EN61000-3-2 compliant SMPS's up to 300W.
Table 2. Absolute Maximum Ratings
Symbol
Pin
VCC
8
---
1 to 4
IZCD
5
Ptot
Tj
Tstg
Parameter
Value
Unit
self-limited
V
-0.3 to 8
V
-50 (source)
10 (sink)
mA
1
0.65
W
Junction Temperature Operating range
-40 to 150
°C
Storage Temperature
-55 to 150
°C
IC Supply voltage (Icc = 20 mA)
Analog Inputs & Outputs
Zero Current Detector Max. Current
Power Dissipation @Tamb = 50°C
(DIP-8)
(SO-8)
Figure 3. Pin Connection (Top view)
INV
1
8
Vcc
COMP
2
7
GD
MULT
3
6
GND
CS
4
5
ZCD
Table 3. Thermal Data
Symbol
Rth j-amb
2/16
Parameter
Max. Thermal Resistance, Junction-to-ambient
SO8
Minidip
Unit
150
100
°C/W
L6562
Table 4. Pin Description
N°
Pin
Function
1
INV
Inverting input of the error amplifier. The information on the output voltage of the PFC preregulator is fed into the pin through a resistor divider.
2
COMP
Output of the error amplifier. A compensation network is placed between this pin and INV (pin
#1) to achieve stability of the voltage control loop and ensure high power factor and low THD.
3
MULT
Main input to the multiplier. This pin is connected to the rectified mains voltage via a resistor
divider and provides the sinusoidal reference to the current loop.
4
CS
Input to the PWM comparator. The current flowing in the MOSFET is sensed through a resistor,
the resulting voltage is applied to this pin and compared with an internal sinusoidal-shaped
reference, generated by the multiplier, to determine MOSFET’s turn-off.
5
ZCD
Boost inductor’s demagnetization sensing input for transition-mode operation. A negative-going
edge triggers MOSFET’s turn-on.
6
GND
Ground. Current return for both the signal part of the IC and the gate driver.
7
GD
Gate driver output. The totem pole output stage is able to drive power MOSFET’s and IGBT’s
with a peak current of 600 mA source and 800 mA sink. The high-level voltage of this pin is
clamped at about 12V to avoid excessive gate voltages in case the pin is supplied with a high
Vcc.
8
Vcc
Supply Voltage of both the signal part of the IC and the gate driver. The supply voltage upper
limit is extended to 22V min. to provide more headroom for supply voltage changes.
Table 5. Electrical Characteristics
(Tj = -25 to 125°C, VCC = 12, CO = 1 nF; unless otherwise specified)
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
22
V
SUPPLY VOLTAGE
VCC
Operating range
After turn-on
VCCon
Turn-on threshold
(1)
11
12
13
V
VCCOff
Turn-off threshold
(1)
8.7
9.5
10.3
V
Hysteresis
Zener Voltage
ICC = 20 mA
2.2
22
25
2.8
28
V
V
Hys
VZ
SUPPLY CURRENT
Istart-up Start-up Current
Iq
ICC
Iq
10.3
Before turn-on, VCC =11V
40
70
µA
Quiescent Current
After turn-on
2.5
3.75
mA
Operating Supply Current
@ 70 kHz
3.5
Quiescent Current
During OVP (either static or
dynamic) or VZCD =150 mV
5
mA
2.2
mA
-1
µA
MULTIPLIER INPUT
IMULT
Input Bias Current
VMULT
Linear Operation Range
∆V CS
--------------------∆V MUL T
K
VVFF = 0 to 4 V
0 to 3
V
Output Max. Slope
VMULT = 0 to 0.5V
VCOMP = Upper clamp
1.65
1.9
V/V
Gain (2)
VMULT = 1 V, VCOMP = 4 V
0.5
0.6
0.7
1/V
2.465
2.5
2.535
V
ERROR AMPLIFIER
VINV
IINV
Voltage Feedback Input
Threshold
Tj = 25 °C
Line Regulation
10.3 V < Vcc < 22 V
Vcc = 10.3 V to 22V
Input Bias Current
VINV = 0 to 3 V
(1)
2.44
2.56
2
5
mV
-1
µA
3/16
L6562
Table 5. Electrical Characteristics (continued)
(Tj = -25 to 125°C, VCC = 12, CO = 1 nF; unless otherwise specified)
Symbol
Parameter
Gv
Voltage Gain
GB
Gain-Bandwidth Product
Test Condition
Open loop
Min.
Typ.
Max.
Unit
60
80
dB
1
MHz
ICOMP
Source Current
Sink Current
VCOMP = 4V, VINV = 2.6 V
2.5
4.5
VCOMP
Upper Clamp Voltage
ISOURCE = 0.5 mA
5.3
5.7
6
V
Lower Clamp Voltage
ISINK = 0.5 mA (1)
2.1
2.25
2.4
V
-1
µA
200
350
ns
1.7
1.8
V
VCOMP = 4V, VINV = 2.4 V
-2
-3.5
-5
mA
mA
CURRENT SENSE COMPARATOR
ICS
td(H-L)
Input Bias Current
Delay to Output
VCS clamp Current sense reference clamp
VCSoffset
VCS = 0
Current sense offset
ZERO CURRENT DETECTOR
VZCDH Upper Clamp Voltage
VCOMP = Upper clamp
1.6
VMULT = 0
30
VMULT = 2.5V
5
mV
IZCD = 2.5 mA
5.0
5.7
6.5
V
0.3
0.65
1
V
VZCDL
Lower Clamp Voltage
IZCD = -2.5 mA
VZCDA
Arming Voltage
(positive-going edge)
(3)
2.1
V
VZCDT
Triggering Voltage
(negative-going edge)
(3)
1.6
V
IZCDb
Input Bias Current
VZCD = 1 to 4.5 V
2
µA
IZCDsrc
Source Current Capability
-2.5
IZCDsnk
Sink Current Capability
2.5
VZCDdis
Disable threshold
150
VZCDen
Restart threshold
IZCDres
Restart Current after Disable
30
75
Start Timer period
75
130
300
µs
35
40
45
µA
2.1
30
2.25
2.4
µA
V
IGDsource = 20 mA
2
2.6
IGDsource = 200 mA
2.5
3
V
IGDsink = 200 mA
0.9
1.9
V
-5.5
mA
mA
200
250
mV
350
mV
µA
STARTER
tSTART
OUTPUT OVERVOLTAGE
IOVP
Dynamic OVP triggering current
Hys
Hysteresis
Static OVP threshold
(3)
(1)
GATE DRIVER
VOH
Dropout Voltage
VOL
tf
Voltage Fall Time
30
70
ns
tr
Voltage Rise Time
40
80
ns
12
15
1.1
V
V
VOclamp
(1)
(2)
(3)
4/16
Output clamp voltage
IGDsource = 5mA; Vcc = 20V
UVLO saturation
VCC = 0 to VCCon, Isink=10mA
All parameters are in tracking
The multiplier output is given by: V cs = K ⋅ V MUL T ⋅ ( V CO MP – 2.5 )
Parameters guaranteed by design, functionality tested in production.
10
L6562
3
Typical Electrical Characteristics
Figure 4. Supply current vs. Supply voltage
Figure 6. IC consumption vs. Tj
Icc 10
[mA]
5
ICC
(mA)
Operating
Quiescent
10
2
5
Disabled or
during OVP
1
1
0.5
0.5
0.1
0.2
0.05
Co = 1nF
f = 70 kHz
Tj = 25°C
0.01
0.005
0.1
5
10
15
20
Before start-up
0.05
0
0
Vcc = 12 V
Co = 1 nF
f = 70 kHz
25
0.02
-50
0
Vcc(V)
100
150
Tj (°C)
Figure 7. Vcc Zener voltage vs. Tj
Figure 5. Start-up & UVLO vs. Tj
VccZ 28
(V)
12.5
VCC-ON
(V)
50
12
27
11.5
26
11
25
10.5
24
10
23
VCC-OFF 9.5
(V)
9
-50
0
50
Tj (°C)
100
150
22
-50
0
50
100
150
Tj (°C)
5/16
L6562
Figure 8. Feedback reference vs. Tj
Figure 11. Delay-to-output vs. Tj
VREF
2.6
(V)
tD(H-L)
500
(ns)
Vcc = 12 V
Vcc = 12 V
400
2.55
300
2.5
200
2.45
100
2.4
-50
0
50
100
0
-50
150
0
100
150
Tj (°C)
Tj (°C)
Figure 9. OVP current vs. Tj
Figure 12. Multiplier characteristic
VCOMP (pin 2)
(V)
VCS (pin 4)
(V) upper voltage
41
(µA)
clamp
Vcc = 12 V
1.6
40.5
3.5
5.0
1.4
4. 5
IOVP
50
1.2
4.0
3.2
1.0
40
0.8
3.0
0.6
39.5
0.4
2.8
0.2
2.6
39
-50
0
50
100
0
150
0
Tj (°C)
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
VMULT(pin 3) (V)
Figure 13. Multiplier gain vs. Tj
Figure 10. E/A output clamp levels vs. Tj
K
Vpin2
(V) 6
1
Vcc = 12 V
VCOMP =4 V
VMULT =1V
0.8
Upper clamp
Vcc = 12 V
5
0.6
4
0.4
3
0.2
Lower clamp
2
-50
0
50
Tj (°C)
6/16
100
150
0
-50
0
50
Tj (°C)
100
150
L6562
Figure 14. Vcs clamp vs. Tj
Figure 17. ZCD source capability vs. Tj
VCSx
2
(V)
IZCDsrc
0
(mA)
Vcc = 12 V
VZCD = lower clamp
1.8
-2
1.6
-4
1.4
1.2
1
-50
Vcc = 12 V
VCOMP= Upper clamp
0
-6
50
100
-8
-50
150
0
50
Figure 15. Start-up timer vs. Tj
Tstart 150
(µs)
100
150
Tj (°C)
Tj (°C)
Figure 18. Gate-drive output low saturation
Vpin7 [V]
4
Vcc = 12 V
Tj = 25 °C
Vcc = 11 V
SINK
140
3
130
2
120
1
110
0
100
-50
0
50
100
150
0
200
800
1,000
Figure 19. Gate-drive output high saturation
Figure 16. ZCD clamp levels vs. Tj
(V)
600
IGD[mA]
Tj (°C)
V ZCD
400
Vpin7[V]
7
-1.5
Upper clamp
6
Tj = 25 °C
Vcc = 11 V
SOURCE
-2
Vcc - 2.0
Vcc = 12 V
I ZCD = ±2.5 mA
5
Vcc - -2.5
2.5
4
-3
Vcc - 3.0
3
Vcc - -3.5
3.5
2
-4
Vcc - 4.0
1
0
-50
Lower clamp
-4.5
0
50
Tj (°C)
100
150
0
100
200
300
400
500
600
700
IGD[mA]
7/16
L6562
Figure 20. Gate-drive clamp vs. Tj
Figure 21. UVLO saturation vs. Tj
Vpin7 clamp
15
(V)
Vpin7
(V) 1.1
Vcc = 0 V
Vcc = 20 V
1
14
0.9
13
0.8
12
0.7
11
10
-50
0.6
0
50
100
150
0.5
-50
50
100
150
Tj (°C)
Tj (°C)
4
0
Application Information
4.1 Overvoltage protection
Under steady-state conditions, the voltage control loop keeps the output voltage Vo of a PFC pre-regulator
close to its nominal value, set by the resistors R1 and R2 of the output divider. Neglecting ripple components, the current through R1, IR1, equals that through R2, IR2. Considering that the non-inverting input of
the error amplifier is internally referenced at 2.5V, also the voltage at pin INV will be 2.5V, then:
Vo – 2.5- .
-------- = IR1 = --------------------I R2 = 2.5
R2
R1
If the output voltage experiences an abrupt change ∆Vo > 0 due to a load drop, the voltage at pin INV will
be kept at 2.5V by the local feedback of the error amplifier, a network connected between pins INV and
COMP that introduces a long time constant to achieve high PF (this is why ∆Vo can be large). As a result,
the current through R2 will remain equal to 2.5/R2 but that through R1 will become:
Vo – 2.5 + ∆Vo
I'R1 = ---------------------------------------- .
R1
The difference current ∆IR1=I'R1-IR2=I'R1-IR1=∆Vo/R1 will flow through the compensation network and enter the error amplifier output (pin COMP). This current is monitored inside the L6562 and if it reaches about
37 µA the output voltage of the multiplier is forced to decrease, thus smoothly reducing the energy delivered to the output. As the current exceeds 40 µA, the OVP is triggered (Dynamic OVP): the gate-drive is
forced low to switch off the external power transistor and the IC put in an idle state. This condition is maintained until the current falls below approximately 10 µA, which re-enables the internal starter and allows
switching to restart. The output ∆Vo that is able to trigger the Dynamic OVP function is then:
∆Vo = R1 ⋅ 40 ⋅ 10
–6
.
An important advantage of this technique is that the OV level can be set independently of the regulated
output voltage: the latter depends on the ratio of R1 to R2, the former on the individual value of R1. Another
advantage is the precision: the tolerance of the detection current is 12%, that is 12% tolerance on ∆Vo.
Since ∆Vo << Vo, the tolerance on the absolute value will be proportionally reduced.
Example: Vo = 400 V, ∆Vo = 40 V. Then: R1=40V/40µA=1MΩ; R2=1MΩ·2.5/(400-2.5)=6.289kΩ. The tolerance on the OVP level due to the L6562 will be 40·0.12=4.8V, that is 1.2% of the regulated value.
8/16
L6562
When the load of a PFC pre-regulator is very low, the output voltage tends to stay steadily above the nominal value, which cannot be handled by the Dynamic OVP. If this occurs, however, the error amplifier output will saturate low; hence, when this is detected, the external power transistor is switched off and the IC
put in an idle state (Static OVP). Normal operation is resumed as the error amplifier goes back into its linear region. As a result, the L6562 will work in burst-mode, with a repetition rate that can be very low.
When either OVP is activated the quiescent consumption of the IC is reduced to minimize the discharge
of the Vcc capacitor and increase the hold-up capability of the IC supply system.
4.2 THD optimizer circuit
The L6562 is equipped with a special circuit that reduces the conduction dead-angle occurring to the AC
input current near the zero-crossings of the line voltage (crossover distortion). In this way the THD (Total
Harmonic Distortion) of the current is considerably reduced.
A major cause of this distortion is the inability of the system to transfer energy effectively when the instantaneous line voltage is very low. This effect is magnified by the high-frequency filter capacitor placed after
the bridge rectifier, which retains some residual voltage that causes the diodes of the bridge rectifier to be
reverse-biased and the input current flow to temporarily stop.
Figure 22. THD optimization: standard TM PFC controller (left side) and L6562 (right side)
Input current
Input current
Rectified mains voltage
Imains
Input current
Rectified mains voltage
Imains
Input
current
Vdrain
MOSFET's drain
voltage
Vdrain
MOSFET's drain
voltage
To overcome this issue the circuit embedded in the L6562 forces the PFC pre-regulator to process more
energy near the line voltage zero-crossings as compared to that commanded by the control loop. This will
result in both minimizing the time interval where energy transfer is lacking and fully discharging the highfrequency filter capacitor after the bridge. The effect of the circuit is shown in figure 23, where the key
waveforms of a standard TM PFC controller are compared to those of the L6562.
Essentially, the circuit artificially increases the ON-time of the power switch with a positive offset added to
9/16
L6562
the output of the multiplier in the proximity of the line voltage zero-crossings. This offset is reduced as the
instantaneous line voltage increases, so that it becomes negligible as the line voltage moves toward the
top of the sinusoid.
To maximally benefit from the THD optimizer circuit, the high-frequency filter capacitor after the bridge rectifier should be minimized, compatibly with EMI filtering needs. A large capacitance, in fact, introduces a
conduction dead-angle of the AC input current in itself - even with an ideal energy transfer by the PFC preregulator - thus making the action of the optimizer circuit little effective.
Figure 23. Typical application circuit (250W, Wide-range mains)
D3 1N5406
R4
R5
180 kΩ 180 kΩ
BRIDGE
FUSE
5A/250V
Vac
(85V to 265V)
STBR606
C1
1 µF
400V
C5 12 nF
D1
STTH5L06
R14
100 Ω
D2
1N5248B
R1
1.5 MΩ
+
T
D8
1N4150
NTC
2.5 Ω
R11
750 kΩ
Vo=400V
Po=250W
R50 10 kΩ
R6
68 kΩ
R12
750 kΩ
C3 2.2 µF
C23
680 nF
R2
1.5 MΩ
-
8
5
3
L6562
2
1
R7
10 Ω
7
C6
100 µF
450V
MOS
STP12NM50
7 °C/W heat sink
6
C2
10nF
R3
22 kΩ
C29
22 µF
25V
4
C4
100 nF
R9
0.33Ω
1W
R10
0.33Ω
1W
R13
9.53 kΩ
-
Boost Inductor Spec: EB0057-C (COILCRAFT)
Figure 24. Demo board (EVAL6562-80W, Wide-range mains): Electrical schematic
R4
R5
180 kΩ 180 kΩ
T
D8
1N4150
C5 12 nF
100 Ω
D2
1N5248B
R1
750 kΩ
D1
STTH1L06
R14
R50 12 kΩ
R6
68 kΩ
BRIDGE
FUSE
4A/250V
Vac
(85V to 265V)
+
DF06M
C1
0.47 µF
400V
R2
750 kΩ
Vo=400V
Po=80W
R12
750 kΩ
C3 680 nF
8
5
3
L6562
2
6
C2
10nF
C29
22 µF
25V
Boost Inductor Spec (ITACOIL E2543/E)
E25x13x7 core, 3C85 ferrite
1.5 mm gap for 0.7 mH primary inductance
Primary: 105 turns 20x0.1 mm
Secondary: 11 turns 0.1 mm
10/16
R11
750 kΩ
C23
330 nF
-
R3
10 kΩ
NTC
2.5 Ω
C4
100 nF
1
R7
33 Ω
7
C6
47 µF
450V
MOS
STP8NM50
4
R9
0.82Ω
0.6 W
R10
0.82Ω
0.6 W
R13
9.53 kΩ
-
L6562
Figure 25. EVAL6562-80W: PCB and component layout (Top view, real size: 57 x 108 mm)
Table 6. EVAL6562N: Evaluation results at full load
Vin (VAC)
Pin (W)
Vo (VDC)
∆Vo(Vpk-pk)
Po (W)
η (%)
PF
THD (%)
85
86.4
394.79
12.8
80.16
92.8
0.998
3.6
110
84.6
394.86
12.8
80.20
94.8
0.996
4.2
135
83.8
394.86
12.8
80.20
95.7
0.991
4.9
175
83.2
394.87
15.5
80.20
96.4
0.981
6.5
220
82.9
394.87
15.7
80.20
96.7
0.956
7.8
265
82.7
394.87
15.9
80.20
97.0
0.915
9.2
Note: measurements done with the line filter shown in figure 23
Table 7. EVAL6562N: Evaluation results at half load
Vin (VAC)
Pin (W)
Vo (VDC)
∆Vo(Vpk-pk)
Po (W)
η (%)
PF
THD (%)
85
42.8
394.86
6.6
40.20
93.9
0.994
5.5
110
42.5
394.90
6.6
40.20
94.6
0.985
6.2
135
42.5
394.91
6.7
40.20
94.6
0.967
7.1
175
42.5
394.93
8.0
40.19
94.6
0.939
8.3
220
42.6
394.94
8.2
40.19
94.3
0.869
9.8
265
42.6
394.94
8.3
40.19
94.3
0.776
11.4
Note: measurements done with the line filter shown in figure 23
11/16
L6562
Table 8. EVAL6562N: No-load measurements
(*)
Vin (VAC)
Pin (W)
Vo (VDC)
∆Vo(Vpk-pk)
Po (W)
85
0.4
396.77
0.45
0
110
0.3
396.82
0.55
0
135
0.3
396.83
0.60
0
175 (*)
0.4
396.90
1.00
0
220 (*)
0.4
396.95
1.40
0
265 (*)
0.5
396.98
1.65
0
Vcc = 12V supplied externally
Figure 26. Line filter (not tested for EMI compliance) used for EVAL6562N evaluation
to the AC
source
B81133
680 nF, X2
EPCOS
B81133
470 nF, X2
EPCOS
B82732
47 mH, 1.3A
EPCOS
12/16
to
EVAL6562N
L6562
5
Package Information
In order to meet environmental requirements, ST offers these devices in ECOPACK® packages. These
packages have a Lead-free second level interconnect. The category of second Level Interconnect is
marked on the package and on the inner box label, in compliance with JEDEC Standard JESD97. The
maximum ratings related to soldering conditions are also marked on the inner box label. ECOPACK is an
ST trademark.
ECOPACK specifications are available at: www.st.com.
Figure 27. DIP-8 Mechanical Data & Package Dimensions
mm
inch
DIM.
MIN.
A
TYP.
MAX.
MIN.
3.32
TYP.
MAX.
0.131
a1
0.51
B
1.15
1.65
0.045
0.065
b
0.356
0.55
0.014
0.022
b1
0.204
0.304
0.008
0.012
0.020
D
E
10.92
7.95
9.75
0.430
0.313
0.384
e
2.54
0.100
e3
7.62
0.300
e4
7.62
0.300
F
6.6
0.260
I
5.08
0.200
L
Z
3.18
OUTLINE AND
MECHANICAL DATA
3.81
1.52
0.125
0.150
DIP-8
0.060
13/16
L6562
Figure 28. SO-8 Mechanical Data & Package Dimensions
mm
inch
DIM.
MIN.
TYP.
MAX.
MIN.
TYP.
MAX.
A
1.35
1.75
0.053
0.069
A1
0.10
0.25
0.004
0.010
A2
1.10
1.65
0.043
0.065
B
0.33
0.51
0.013
0.020
C
0.19
0.25
0.007
0.010
4.80
5.00
0.189
0.197
4.00
0.15
D
(1)
E
3.80
e
1.27
0.157
0.050
H
5.80
6.20
0.228
0.244
h
0.25
0.50
0.010
0.020
L
0.40
1.27
0.016
0.050
k
ddd
OUTLINE AND
MECHANICAL DATA
0˚ (min.), 8˚ (max.)
0.10
0.004
Note: (1) Dimensions D does not include mold flash, protrusions or gate burrs.
Mold flash, potrusions or gate burrs shall not exceed
0.15mm (.006inch) in total (both side).
SO-8
0016023 C
14/16
L6562
6
Revision History
Table 9. Revision History
Date
Revision
Description of Changes
January 2004
5
First Issue
June 2004
6
Modified the Style-look in compliance with the “Corporate Technical
Publications Design Guide”.
Changed input of the power amplifier connected to Multiplier (Fig. 2).
May 2005
7
Modified Table 2: Absolute Maximim Ratings.
November 2005
8
Added in Section 5 the ECOPACK® certicate of conformity.
15/16
L6562
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
The ST logo is a registered trademark of STMicroelectronics.
All other names are the property of their respective owners
© 2005 STMicroelectronics - All rights reserved
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16/16