10 MHz, 20 V/μs, G = 1, 2, 5, 10 iCMOS Programmable Gain Instrumentation Amplifier AD8250 Data Sheet FEATURES FUNCTIONAL BLOCK DIAGRAM Small package: 10-lead MSOP Programmable gains: 1, 2, 5, 10 Digital or pin-programmable gain setting Wide supply: ±5 V to ±15 V Excellent dc performance High CMRR 98 dB (minimum), G = 10 Low gain drift: 10 ppm/°C (maximum) Low offset drift: 1.7 μV/°C (maximum), G = 10 Excellent ac performance Fast settling time: 615 ns to 0.001% (maximum) High slew rate: 20 V/µs (minimum) Low distortion: −110 dB THD at 1 kHz High CMRR over frequency: 80 dB to 50 kHz (minimum) Low noise: 18 nV/√Hz, G = 10 (maximum) Low power: 4.1 mA DGND WR 2 A0 5 4 LOGIC –IN 1 7 OUT +IN 10 8 3 9 +VS –VS REF 06288-001 AD8250 Figure 1. 25 APPLICATIONS G = 10 20 Data acquisition Biomedical analysis Test and measurement G=5 GAIN (dB) 15 GENERAL DESCRIPTION The AD8250 user interface consists of a parallel port that allows users to set the gain in one of two ways (see Figure 1). A 2-bit word sent via a bus can be latched using the WR input. An alternative is to use the transparent gain mode where the state of the logic levels at the gain port determines the gain. 10 G=2 5 G=1 0 –5 –10 1k 10k 100k 1M 10M 100M FREQUENCY (Hz) 06288-023 The AD8250 is an instrumentation amplifier with digitally programmable gains that has GΩ input impedance, low output noise, and low distortion making it suitable for interfacing with sensors and driving high sample rate analog-to-digital converters (ADCs). It has a high bandwidth of 10 MHz, low THD of −110 dB and fast settling time of 615 ns (maximum) to 0.001%. Offset drift and gain drift are guaranteed to 1.7 μV/°C and 10 ppm/°C, respectively, for G = 10. In addition to its wide input common voltage range, it boasts a high common-mode rejection of 80 dB at G = 1 from dc to 50 kHz. The combination of precision dc performance coupled with high speed capabilities makes the AD8250 an excellent candidate for data acquisition. Furthermore, this monolithic solution simplifies design and manufacturing and boosts performance of instrumentation by maintaining a tight match of internal resistors and amplifiers. Rev. C A1 6 Figure 2. Gain vs. Frequency Table 1. Instrumentation Amplifiers by Category General Purpose AD82201 AD8221 AD8222 AD82241 AD8228 1 Zero Drift AD82311 AD85531 AD85551 AD85561 AD85571 Mil Grade AD620 AD621 AD524 AD526 AD624 Low Power AD6271 AD6231 AD82231 High Speed PGA AD8250 AD8251 AD8253 Rail-to-rail output. The AD8250 is available in a 10-lead MSOP package and is specified over the −40°C to +85°C temperature range, making it an excellent solution for applications where size and packing density are important considerations. Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©2007–2013 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD8250 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1 Input Bias Current Return Path ............................................... 17 Applications ....................................................................................... 1 Input Protection ......................................................................... 17 General Description ......................................................................... 1 Reference Terminal .................................................................... 18 Functional Block Diagram .............................................................. 1 Common-Mode Input Voltage Range ..................................... 18 Revision History ............................................................................... 2 Layout .......................................................................................... 18 Specifications..................................................................................... 3 RF Interference ........................................................................... 19 Timing Diagram ........................................................................... 5 Driving an ADC ......................................................................... 19 Absolute Maximum Ratings ............................................................ 6 Applications..................................................................................... 20 Maximum Power Dissipation ..................................................... 6 Differential Output .................................................................... 20 ESD Caution .................................................................................. 6 Setting Gains with a Microcontroller ...................................... 20 Pin Configuration and Function Descriptions ............................. 7 Data Acquisition ......................................................................... 21 Typical Performance Characteristics ............................................. 8 Outline Dimensions ....................................................................... 22 Theory of Operation ...................................................................... 15 Ordering Guide .......................................................................... 22 Gain Selection ............................................................................. 15 Power Supply Regulation and Bypassing ................................ 17 REVISION HISTORY 5/13—Rev. B to Rev. C Changed 49.9 Ω to 100 Ω in Driving an ADC Section and Figure 55 .......................................................................................... 19 11/10—Rev. A to Rev. B Changes to Voltage Offset, Offset RTI VOS, Average Temperature Coefficient Parameter in Table 2 ............................. 3 Updated Outline Dimensions ....................................................... 22 5/08—Rev. 0 to Rev. A Changes to Table 1 ............................................................................ 1 Changes to Table 2 ............................................................................ 3 Changes to Table 3.............................................................................6 Added Figure 17; Renumbered Sequentially .................................9 Changes to Figure 23...................................................................... 10 Changes to Figure 24 to Figure 26................................................ 11 Added Figure 29 ............................................................................. 11 Changes to Figure 31...................................................................... 12 Deleted Figure 43 to Figure 46; Renumbered Sequentially ...... 14 Inserted Figure 45 and Figure 46.................................................. 14 Changes to Timing for Latched Gain Mode Section ................. 16 Changes to Layout Section and Coupling Noise Section .......... 18 Changes to Figure 59...................................................................... 21 1/07—Revision 0: Initial Version Rev. C | Page 2 of 24 Data Sheet AD8250 SPECIFICATIONS +VS = 15 V, −VS = −15 V, VREF = 0 V @ TA = 25°C, G = 1, RL = 2 kΩ, unless otherwise noted. Table 2. Parameter COMMON-MODE REJECTION RATIO (CMRR) CMRR to 60 Hz with 1 kΩ Source Imbalance G=1 G=2 G=5 G = 10 CMRR to 50 kHz G=1 G=2 G=5 G = 10 NOISE Voltage Noise, 1 kHz, RTI G=1 G=2 G=5 G = 10 0.1 Hz to 10 Hz, RTI G=1 G=2 G=5 G = 10 Current Noise, 1 kHz Current Noise, 0.1 Hz to 10 Hz VOLTAGE OFFSET Offset RTI VOS Over Temperature Average Temperature Coefficient Offset Referred to the Input vs. Supply (PSR) INPUT CURRENT Input Bias Current Over Temperature Average Temperature Coefficient Input Offset Current Over Temperature Average Temperature Coefficient DYNAMIC RESPONSE Small Signal −3 dB Bandwidth G=1 G=2 G=5 G = 10 Settling Time 0.01% G=1 G=2 G=5 G = 10 Conditions Min Typ 80 86 94 98 98 104 110 110 Max Unit +IN = −IN = −10 V to +10 V dB dB dB dB +IN = −IN = −10 V to +10 V 80 86 90 90 dB dB dB dB 40 27 21 18 nV/√Hz nV/√Hz nV/√Hz nV/√Hz 2.5 2.5 1.5 1.0 μV p-p μV p-p μV p-p μV p-p pA/√Hz pA p-p ±(70 + 200/G) ±(90 + 300/G) ±(0.6 + 1.5/G) ±(2 + 7/G) ±(200 + 600/G) ±(260 + 900/G) ±(1.2 + 5/G) ±(6 + 20/G) μV μV μV/°C μV/V 5 30 40 400 30 30 160 nA nA pA/°C nA nA pA/°C 5 60 G = 1, 2, 5, 10 T = −40°C to +85°C T = −40°C to +85°C VS = ±5 V to ±15 V T = −40°C to +85°C T = −40°C to +85°C 5 T = −40°C to +85°C T = −40°C to +85°C 10 10 10 3 MHz MHz MHz MHz ΔOUT = 10 V step 585 605 605 648 Rev. C | Page 3 of 24 ns ns ns ns AD8250 Parameter Settling Time 0.001% G=1 G=2 G=5 G = 10 Slew Rate G=1 G=2 G=5 G = 10 Total Harmonic Distortion GAIN Gain Range Gain Error G=1 G = 2, 5, 10 Gain Nonlinearity G=1 G=2 G=5 G = 10 Gain vs. Temperature INPUT Input Impedance Differential Common Mode Input Operating Voltage Range Over Temperature OUTPUT Output Swing Over Temperature Short-Circuit Current REFERENCE INPUT RIN IIN Voltage Range Gain to Output DIGITAL LOGIC Digital Ground Voltage, DGND Digital Input Voltage Low Digital Input Voltage High Digital Input Current Gain Switching Time 1 tSU tHD t WR -LOW t WR -HIGH Data Sheet Conditions ΔOUT = 10 V step Min Typ Max Unit 615 635 635 685 ns ns ns ns 20 25 25 25 f = 1 kHz, RL = 10 kΩ, ±10 V, G = 1, 10 Hz to 22 kHz band-pass filter G = 1, 2, 5, 10 OUT = ±10 V V/μs V/μs V/μs V/μs dB −110 1 OUT = −10 V to +10 V RL = 10 kΩ, 2 kΩ, 600 Ω RL = 10 kΩ, 2 kΩ, 600 Ω RL = 10 kΩ, 2 kΩ, 600 Ω RL = 10 kΩ, 2 kΩ, 600 Ω All gains 10 V/V 0.03 0.04 % % 6 8 8 10 10 ppm ppm ppm ppm ppm/°C GΩ||pF GΩ||pF V V 5.3||0.5 1.25||2 VS = ±5 V to ±15 V T = −40°C to +85°C −VS + 1.5 −VS + 1.6 +VS − 1.5 +VS − 1.7 T = −40°C to +85°C −13.5 −13.5 +13.5 +13.5 37 20 +IN, −IN, REF = 0 1 +VS −VS 1 ± 0.0001 Referred to GND Referred to GND Referred to GND −VS + 4.25 DGND 2.8 0 +VS − 2.7 2.1 +VS 1 325 See Figure 3 timing diagram See Figure 3 timing diagram See Figure 3 timing diagram See Figure 3 timing diagram Rev. C | Page 4 of 24 20 10 20 40 V V mA kΩ μA V V/V V V V μA ns ns ns ns ns Data Sheet AD8250 Parameter POWER SUPPLY Operating Range Quiescent Current, +IS Quiescent Current, −IS Over Temperature TEMPERATURE RANGE Specified Performance 1 Conditions Min Typ Max Unit 4.1 3.7 ±15 4.5 4.5 4.5 V mA mA mA +85 °C ±5 T = −40°C to +85°C −40 Add time for the output to slew and settle to calculate the total time for a gain change. TIMING DIAGRAM tWR-HIGH tWR-LOW WR tHD 06288-057 tSU A0, A1 Figure 3. Timing Diagram for Latched Gain Mode (See the Timing for Latched Gain Mode Section) Rev. C | Page 5 of 24 AD8250 Data Sheet ABSOLUTE MAXIMUM RATINGS Parameter Supply Voltage Power Dissipation Output Short-Circuit Current Common-Mode Input Voltage Differential Input Voltage Digital Logic Inputs Storage Temperature Range Operating Temperature Range3 Lead Temperature (Soldering, 10 sec) Junction Temperature θJA (Four-Layer JEDEC Standard Board) Package Glass Transition Temperature Rating ±17 V See Figure 4 Indefinite1 +VS + 13 V, −VS − 13 V +VS + 13 V, −VS − 13 V2 ±VS −65°C to +125°C −40°C to +85°C 300°C 140°C 112°C/W 140°C Assumes that the load is referenced to midsupply. Current must be kept to less than 6 mA. 3 Temperature for specified performance is −40°C to +85°C. For performance to 125°C, see the Typical Performance Characteristics section. 1 2 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). Assuming that the load (RL) is referenced to midsupply, the total drive power is VS/2 × IOUT, some of which is dissipated in the package and some in the load (VOUT × IOUT). The difference between the total drive power and the load power is the drive power dissipated in the package. PD = Quiescent Power + (Total Drive Power − Load Power) V V PD = (VS × I S ) + S × OUT RL 2 In single-supply operation with RL referenced to −VS, the worst case is VOUT = VS/2. Airflow increases heat dissipation, effectively reducing θJA. In addition, more metal directly in contact with the package leads from metal traces, through holes, ground, and power planes reduces the θJA. Figure 4 shows the maximum safe power dissipation in the package vs. the ambient temperature on a four-layer JEDEC standard board. 2.00 The maximum safe power dissipation in the AD8250 package is limited by the associated rise in junction temperature (TJ) on the die. The plastic encapsulating the die locally reaches the junction temperature. At approximately 140°C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8250. Exceeding a junction temperature of 140°C for an extended period can result in changes in silicon devices, potentially causing failure. 1.75 MAXIMUM POWER DISSIPATION (W) MAXIMUM POWER DISSIPATION The still-air thermal properties of the package and PCB (θJA), the ambient temperature (TA), and the total power dissipated in the package (PD) determine the junction temperature of the die. The junction temperature is calculated as VOUT 2 – RL 1.50 1.25 1.00 0.75 0.50 0.25 0 –40 –20 0 20 40 60 80 100 120 AMBIENT TEMPERATURE (°C) Figure 4. Maximum Power Dissipation vs. Ambient Temperature ESD CAUTION TJ = TA + (PD × θJA) Rev. C | Page 6 of 24 06288-004 Table 3. Data Sheet AD8250 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS –IN 1 DGND 2 AD8250 10 +IN 9 REF 8 +VS TOP VIEW A0 4 (Not to Scale) 7 OUT A1 5 6 WR 06288-005 –VS 3 Figure 5. Pin Configuration Table 4. Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 9 10 Mnemonic −IN DGND −VS A0 A1 WR OUT +VS REF +IN Description Inverting Input Terminal. True differential input. Digital Ground. Negative Supply Terminal. Gain Setting Pin (LSB). Gain Setting Pin (MSB). Write Enable. Output Terminal. Positive Supply Terminal. Reference Voltage Terminal. Noninverting Input Terminal. True differential input. Rev. C | Page 7 of 24 AD8250 Data Sheet TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, +VS = +15 V, −VS = −15 V, RL = 10 kΩ, unless otherwise noted. 500 1400 1200 400 NUMBER OF UNITS NUMBER OF UNITS 1000 800 600 300 200 400 100 –120 –90 –60 –30 0 30 60 90 120 CMRR (µV/V) 0 06288-006 0 –30 –20 –10 0 10 20 30 INPUT OFFSET CURRENT (nA) 06288-009 200 Figure 9. Typical Distribution of Input Offset Current Figure 6. Typical Distribution of CMRR, G = 1 350 90 300 80 70 NOISE RTI (nV/ Hz) NUMBER OF UNITS 250 200 150 60 50 G=1 40 G=2 30 100 G=5 20 06288-010 G = 10 50 10 –200 –150 –100 –50 0 50 100 150 200 OFFSET VOLTAGE RTI (µV) 06288-007 0 0 1 10 100 1k 10k 100k FREQUENCY (Hz) Figure 7. Typical Distribution of Offset Voltage, VOSI Figure 10. Voltage Spectral Density Noise vs. Frequency 600 400 300 200 2µV/DIV 0 –30 –20 –10 0 10 20 INPUT BIAS CURRENT (nA) 30 1s/DIV Figure 8. Typical Distribution of Input Bias Current Figure 11. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 1 Rev. C | Page 8 of 24 06288-011 100 06288-008 NUMBER OF UNITS 500 Data Sheet AD8250 150 G = 10 130 G=5 PSRR (dB) 110 G=1 G=2 90 70 50 10 1 10 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 12. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 10 Figure 15. Positive PSRR vs. Frequency, RTI 150 18 16 130 14 G = 10 110 12 PSRR (dB) G=5 10 8 90 G=2 70 G=1 6 50 4 1 10 100 1k 10k 10 100k 1 FREQUENCY (Hz) 10 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 13. Current Noise Spectral Density vs. Frequency 06288-017 0 30 06288-013 2 Figure 16. Negative PSRR vs. Frequency, RTI 1s/DIV 9 8 7 6 5 4 3 2 1 0 0.01 1 0.1 WARMUP TIME (Minutes) 10 Figure 17. Change in Offset Voltage, RTI vs. Warmup Time Figure 14. 0.1 Hz to 10 Hz Current Noise Rev. C | Page 9 of 24 06288-117 140pA/DIV CHANGE IN OFFSET VOLTAGE, RTI (µV) 10 06288-014 CURRENT NOISE (pA/ Hz) 06288-016 1s/DIV 06288-012 30 1µV/DIV Data Sheet 10 15 8 10 6 4 ∆CMRR (µV/V) IB– 0 IB+ 2 0 –2 –5 –4 IOS –6 –10 06288-049 5 –8 –15 –40 –25 –10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) –10 –50 06288-019 INPUT BIAS CURRENT AND OFFSET CURRENT (nA) AD8250 –30 –10 10 30 50 70 90 110 130 TEMPERATURE (°C) Figure 21. CMRR vs. Temperature, G = 1 Figure 18. Input Bias Current and Offset Current vs. Temperature 25 140 G = 10 G=5 G = 10 20 120 G=5 15 G=2 G=1 GAIN (dB) 80 10 G=2 5 60 G=1 0 40 –5 1 10 100 1k 10k 100k 1M FREQUENCY (Hz) –10 1k 06288-020 20 10k 40 GAIN NONLINEARITY (10ppm/DIV) G=5 80 G=1 60 40 100M f = 1kHz 30 20 10 0 –10 –20 20 1 10 100 1k 10k 100k FREQUENCY (Hz) Figure 20. CMRR vs. Frequency, 1 kΩ Source Imbalance 1M –40 –10 06288-024 –30 06288-021 CMRR (dB) 100 G=2 10M Figure 22. Gain vs. Frequency 140 120 1M FREQUENCY (Hz) Figure 19. CMRR vs. Frequency G = 10 100k 06288-023 CMRR (dB) 100 –8 –6 –4 –2 0 2 4 6 8 10 OUTPUT VOLTAGE (V) Figure 23. Gain Nonlinearity vs. Output Voltage, G = 1, RL = 10 kΩ, 2 kΩ, 600 Ω Rev. C | Page 10 of 24 Data Sheet AD8250 16 40 0V, +13.8V 10 0 –10 –20 –40 –10 06288-025 –30 –8 –6 –4 –2 0 2 4 6 8 –3.8V, +1.9V –3.8V, –1.9V –4 –8 –13.8V, –6.9V +13.8V, –6.9V –12 0V, –14V –12 –8 16 30 INPUT COMMON-MODE VOLTAGE (V) GAIN NONLINEARITY (10ppm/DIV) f = 1kHz 20 10 0 –10 –20 06288-026 –30 –4 –2 0 2 0 4 8 4 6 8 12 16 +13.6V, +13.1V VS = ±15V 8 +0V, +3.5V 4 +4.3V, +2.1V –4.2V, +2.2V VS = ±5V 0 –4.2V, –2.0V –4 +4.3V, –2.1V 0V, –4.1V –8 –12 –14.1V, –13.6V –16 –16 –12 –8 10 0V, +13.8V –14.1V, +13.6V 12 OUTPUT VOLTAGE (V) +13.6V, –13.1V 0V, –14V –4 0 8 4 12 16 OUTPUT VOLTAGE (V) Figure 25. Gain Nonlinearity vs. Output Voltage, G = 5, RL = 10 kΩ, 2 kΩ, 600 Ω Figure 28. Input Common-Mode Voltage Range vs. Output Voltage, G = 10 40 35 f = 1kHz IB+ IB– IOS 30 30 25 20 INPUT BIAS CURRENT AND OFFSET CURRENT (nA) 10 0 –10 –20 20 15 10 5 0 –5 –40 –10 06288-027 –30 –8 –6 –4 –2 0 2 4 6 8 06288-129 GAIN NONLINEARITY (10ppm/DIV) –4 Figure 27. Input Common-Mode Voltage Range vs. Output Voltage, G = 1 40 –6 +3.8V, –2.1V 0V, –4.0V OUTPUT VOLTAGE (V) Figure 24. Gain Nonlinearity vs. Output Voltage, G = 2, RL = 10 kΩ, 2 kΩ, 600 Ω –8 +3.9V, +1.9V VS = ±5V 0 OUTPUT VOLTAGE (V) –40 –10 +13.8V, +6.9V 0V, +3.7V 4 –16 –16 10 VS = ±15V –13.8V, +6.9V 8 06288-028 20 12 06288-029 30 INPUT COMMON-MODE VOLTAGE (V) GAIN NONLINEARITY (10ppm/DIV) f = 1kHz –10 –15 –15 10 OUTPUT VOLTAGE (V) –10 –5 0 5 10 15 COMMON-MODE VOLTAGE (V) Figure 26. Gain Nonlinearity vs. Output Voltage, G = 10, RL = 10 kΩ, 2 kΩ, 600 Ω Figure 29. Input Bias Current and Offset Current vs. Common-Mode Voltage Rev. C | Page 11 of 24 AD8250 Data Sheet +VS OUTPUT VOLTAGE SWING REFERRED TO SUPPLY VOLTAGE (V) –1 +85°C +125°C –2 +25°C –40°C +2 +85°C –40°C +25°C +1 –VS 4 6 8 10 12 14 16 SUPPLY VOLTAGE (±VS) Figure 30. Input Voltage Limit vs. Supply Voltage, G = 1, VREF = 0 V, RL = 10 kΩ +125°C –0.4 –0.6 –0.8 –1.0 +85°C +1.0 +25°C –40°C +25°C –40°C +0.8 +0.6 +0.4 +0.2 –VS +125°C 06288-030 INPUT VOLTAGE REFERRED TO SUPPLY VOLTAGE (V) –0.2 +85°C +125°C 4 6 8 10 12 14 16 SUPPLY VOLTAGE (±VS) 06288-033 +VS Figure 33. Output Voltage Swing vs. Supply Voltage, G = 10, RL = 10 kΩ 15 15 +25°C +VS 10 5 FAULT CONDITION (OVER DRIVEN INPUT) G = 10 +IN 0 –IN –5 5 +85°C +125°C 0 +85°C –5 +125°C –10 –10 –40°C –40°C –VS –12 –8 –4 0 4 8 12 +25°C 16 DIFFERENTIAL INPUT VOLTAGE (V) –15 100 06288-031 –15 –16 Figure 31. Fault Current Draw vs. Input Voltage, G = 10, RL = 10 kΩ +VS +VS +125°C –0.6 –0.8 –1.0 +25°C +85°C –40°C +25°C –40°C +85°C +1.0 +0.8 +0.6 +125°C +0.4 –0.4 +85°C +125°C –0.8 –1.2 +25°C –1.6 –2.0 +2.0 +1.6 6 8 10 12 SUPPLY VOLTAGE (±VS) 14 16 Figure 32. Output Voltage Swing vs. Supply Voltage, G = 10, RL = 2 kΩ Rev. C | Page 12 of 24 +25°C –40°C +1.2 +0.8 +0.4 4 –40°C –VS +125°C +85°C 0 2 4 06288-035 OUTPUT VOLTAGE SWING REFERRED TO SUPPLY VOLTAGE (V) –0.4 06288-032 OUTPUT VOLTAGE SWING REFERRED TO SUPPLY VOLTAGE (V) 10k Figure 34. Output Voltage Swing vs. Load Resistance –0.2 +0.2 –VS 1k LOAD RESISTANCE (Ω) 06288-034 FAULT CONDITION (OVER DRIVEN INPUT) G = 10 OUTPUT VOLTAGE SWING (V) CURRENT (mA) 10 6 8 10 12 14 OUTPUT CURRENT (mA) Figure 35. Output Voltage Swing vs. Output Current 16 Data Sheet NO LOAD 47pF AD8250 100pF VOUT (V) 5V/DIV 605ns TO 0.01% 635ns TO 0.001% 2µs/DIV 2µs/DIV 06288-036 20mV/DIV TIME (µs) 06288-039 0.002%/DIV TIME (µs) Figure 36. Small Signal Pulse Response for Various Capacitive Loads Figure 39. Large Signal Pulse Response and Settling Time G = 5, RL = 10 kΩ 5V/DIV 5V/DIV 648ns TO 0.01% 685ns TO 0.001% 585ns TO 0.01% 615ns TO 0.001% 06288-037 2µs/DIV 2µs/DIV 06288-040 0.002%/DIV 0.002%/DIV TIME (µs) TIME (µs) Figure 37. Large Signal Pulse Response and Settling Time, G = 1, RL = 10 kΩ Figure 40. Large Signal Pulse Response and Settling Time G = 10, RL = 10 kΩ VOUT (V) 5V/DIV 605ns TO 0.01% 635ns TO 0.001% 20mV/DIV 2µs/DIV TIME (µs) TIME (µs) Figure 38. Large Signal Pulse Response and Settling Time G = 2, RL = 10 kΩ Figure 41. Small Signal Response G = 1, RL = 2 kΩ, CL = 100 pF Rev. C | Page 13 of 24 06288-042 2µs/DIV 06288-038 0.002%/DIV AD8250 Data Sheet –50 G G G G –55 –60 –65 =1 =2 =5 = 10 VOUT (V) THD + N (dB) –70 –75 –80 –85 –90 –95 –100 –105 2µs/DIV TIME (µs) 06288-149 20mV/DIV 06288-043 –110 –115 –120 10 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 42. Small Signal Response G = 2, RL = 2 kΩ, CL = 100 pF Figure 45. Total Harmonic Distortion + Noise vs. Frequency, 10 Hz to 22 kHz Band-Pass Filter, RL = 2 kΩ –50 G G G G –60 =1 =2 =5 = 10 VOUT (V) THD + N (dB) –70 –80 –90 TIME (µs) –110 10 06288-150 2µs/DIV 06288-044 –100 20mV/DIV 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 43. Small Signal Response G = 5, RL = 2 kΩ, CL = 100 pF 20mV/DIV 2µs/DIV TIME (µs) 06288-045 VOUT (V) Figure 46. Total Harmonic Distortion + Noise vs. Frequency, 10 Hz to 500 kHz Band-Pass Filter, RL = 2 kΩ Figure 44. Small Signal Response, G = 10, RL = 2 kΩ, CL = 100 pF Rev. C | Page 14 of 24 Data Sheet AD8250 THEORY OF OPERATION +VS +VS A0 A1 2.2kΩ +VS –IN –VS –VS 2.2kΩ 10kΩ A1 10kΩ –VS +VS DIGITAL GAIN CONTROL OUT A3 –VS +VS +VS 10kΩ A2 REF 2.2kΩ +VS –VS –VS +VS 2.2kΩ DGND WR –VS 06288-054 +IN 10kΩ –VS Figure 47. Simplified Schematic Transparent Gain Mode The easiest way to set the gain is to program it directly via a logic high or logic low voltage applied to A0 and A1. Figure 48 shows an example of this gain setting method, referred to throughout the data sheet as transparent gain mode. Tie WR to the negative supply to engage transparent gain mode. In this mode, any change in voltage applied to A0 and A1 from logic low to logic high, or vice versa, immediately results in a gain change. Table 5 is the truth table for transparent gain mode, and Figure 48 shows the AD8250 configured in transparent gain mode. All internal amplifiers employ distortion cancellation circuitry and achieve high linearity and ultralow THD. Laser trimmed resistors allow for a maximum gain error of less than 0.03% for G = 1 and minimum CMRR of 98 dB for G = 10. A pinout optimized for high CMRR over frequency enables the AD8250 to offer a guaranteed minimum CMRR over frequency of 80 dB at 50 kHz (G = 1). The balanced input reduces the parasitics that, in the past, adversely affected CMRR performance. +15V 10μF 0.1µF WR A1 A0 +IN REF –IN 10μF Logic low and logic high voltage limits are listed in the Specifications section. Typically, logic low is 0 V and logic high is 5 V; both voltages are measured with respect to DGND. See Table 2 for the permissible voltage range of DGND. The gain of the AD8250 can be set using two methods. +5V G = 10 AD8250 DGND GAIN SELECTION –15V +5V DGND 0.1µF –15V NOTE: 1. IN TRANSPARENT GAIN MODE, WR IS TIED TO −VS. THE VOLTAGE LEVELS ON A0 AND A1 DETERMINE THE GAIN. IN THIS EXAMPLE, BOTH A0 AND A1 ARE SET TO LOGIC HIGH, RESULTING IN A GAIN OF 10. 06288-055 The AD8250 is a monolithic instrumentation amplifier based on the classic, 3-op-amp topology as shown in Figure 47. It is fabricated on the Analog Devices, Inc., proprietary iCMOS® process that provides precision, linear performance, and a robust digital interface. A parallel interface allows users to digitally program gains of 1, 2, 5, and 10. Gain control is achieved by switching resistors in an internal, precision resistor array (as shown in Figure 47). Although the AD8250 has a voltage feedback topology, the gain bandwidth product increases for gains of 1, 2, and 5 because each gain has its own frequency compensation. This results in maximum bandwidth at higher gains. Figure 48. Transparent Gain Mode, A0 and A1 = High, G = 10 Rev. C | Page 15 of 24 AD8250 Data Sheet Table 5. Truth Table Logic Levels for Transparent Gain Mode Table 6. Truth Table Logic Levels for Latched Gain Mode WR A1 A0 Gain WR A1 A0 Gain −VS −VS −VS −VS Low Low High High Low High Low High 1 2 5 10 High to low High to low High to low High to low Low to low Low to high High to high Low Low High High X1 X1 X1 Low High Low High X1 X1 X1 Change to 1 Change to 2 Change to 5 Change to 10 No change No change No change Latched Gain Mode Some applications have multiple programmable devices such as multiplexers or other programmable gain instrumentation amplifiers on the same PCB. In such cases, devices can share a data bus. The gain of the AD8250 can be set using WR as a latch, allowing other devices to share A0 and A1. Figure 49 shows a schematic using this method, known as latched gain mode. The AD8250 is in this mode when WR is held at logic high or logic low, typically 5 V and 0 V, respectively. The voltages on A0 and A1 are read on the downward edge of the WR signal as it transitions from logic high to logic low. This latches in the logic levels on A0 and A1, resulting in a gain change. See the truth table in Table 6 for more information on these gain changes. +15V WR 10μF 0.1µF A1 A1 A0 +IN +5V 0V +5V 0V WR + A0 G = PREVIOUS STATE +5V 0V G = 10 AD8250 REF – –IN DGND –15V NOTE: 1. ON THE DOWNWARD EDGE OF WR, AS IT TRANSITIONS FROM LOGIC HIGH TO LOGIC LOW, THE VOLTAGES ON A0 AND A1 ARE READ AND LATCHED IN, RESULTING IN A GAIN CHANGE. IN THIS EXAMPLE, THE GAIN SWITCHES TO G = 10. X = don’t care. On power-up, the AD8250 defaults to a gain of 1 when in latched gain mode. In contrast, if the AD8250 is configured in transparent gain mode, it starts at the gain indicated by the voltage levels on A0 and A1 at power-up. Timing for Latched Gain Mode In latched gain mode, logic levels at A0 and A1 have to be held for a minimum setup time, tSU, before the downward edge of WR latches in the gain. Similarly, they must be held for a minimum hold time of tHD after the downward edge of WR to ensure that the gain is latched in correctly. After tHD, A0 and A1 can change logic levels, but the gain does not change (until the next downward edge of WR). The minimum duration that WR can be held high is t WR -HIGH, and the minimum duration that WR can be held low is t WR -LOW. Digital timing specifications are listed in Table 2. The time required for a gain change is dominated by the settling time of the amplifier. A timing diagram is shown in Figure 50. When sharing a data bus with other devices, logic levels applied to those devices can potentially feed through to the output of the AD8250. Feedthrough can be minimized by decreasing the edge rate of the logic signals. Furthermore, careful layout of the PCB also reduces coupling between the digital and analog portions of the board. Pull-up or pull-down resistors should be used to provide a well-defined voltage at the A0 and A1 pins. DGND 0.1µF 06288-056 10μF 1 Figure 49. Latched Gain Mode, G = 10 tWR-HIGH tWR-LOW WR tSU tHD 06288-057 A0, A1 Figure 50. Timing Diagram for Latched Gain Mode Rev. C | Page 16 of 24 Data Sheet AD8250 INCORRECT POWER SUPPLY REGULATION AND BYPASSING CORRECT +VS The AD8250 has high PSRR. However, for optimal performance, a stable dc voltage should be used to power the instrumentation amplifier. Noise on the supply pins can adversely affect performance. As in all linear circuits, bypass capacitors must be used to decouple the amplifier. +VS AD8250 AD8250 REF Place a 0.1 μF capacitor close to each supply pin. A 10 μF tantalum capacitor can be used farther away from the part (see Figure 51) and, in most cases, it can be shared by other precision integrated circuits. REF –VS –VS TRANSFORMER TRANSFORMER +VS +VS +VS 0.1µF WR A1 +IN 10µF AD8250 AD8250 REF A0 REF 10MΩ OUT –VS LOAD –IN –VS THERMOCOUPLE REF THERMOCOUPLE +VS DGND +VS DGND 10µF –VS 06288-058 C 0.1µF C 1 fHIGH-PASS = 2πRC AD8250 C Figure 51. Supply Decoupling, REF, and Output Referred to Ground REF R AD8250 C REF R INPUT BIAS CURRENT RETURN PATH The AD8250 input bias current must have a return path to its local analog ground. When the source, such as a thermocouple, cannot provide a return current path, one should be created (see Figure 52). –VS –VS CAPACITIVELY COUPLED CAPACITIVELY COUPLED 06288-059 AD8250 Figure 52. Creating an IBIAS Return Path INPUT PROTECTION All terminals of the AD8250 are protected against ESD. Note that 2.2 kΩ series resistors precede the ESD diodes as shown in Figure 47. The resistors limit current into the diodes and allow for dc overload conditions 13 V above the positive supply and 13 V below the negative supply. An external resistor should be used in series with each input to limit current for voltages greater than 13 V beyond either supply rail. In either scenario, the AD8250 safely handles a continuous 6 mA current at room temperature. For applications where the AD8250 encounters extreme overload voltages, external series resistors and low leakage diode clamps, such as BAV199Ls, FJH1100s, or SP720s, should be used. Rev. C | Page 17 of 24 AD8250 Data Sheet REFERENCE TERMINAL The reference terminal, REF, is at one end of a 10 kΩ resistor (see Figure 47). The instrumentation amplifier output is referenced to the voltage on the REF terminal; this is useful when the output signal needs to be offset to voltages other than its local analog ground. For example, a voltage source can be tied to the REF pin to level shift the output so that the AD8250 can interface with a single-supply ADC. The allowable reference voltage range is a function of the gain, common-mode input, and supply voltages. The REF pin should not exceed either +VS or −VS by more than 0.5 V. For best performance, especially in cases where the output is not measured with respect to the REF terminal, source impedance to the REF terminal should be kept low because parasitic resistance can adversely affect CMRR and gain accuracy. INCORRECT CORRECT AD8250 AD8250 VREF The output voltage of the AD8250 develops with respect to the potential on the reference terminal. Take care to tie REF to the appropriate local analog ground or to connect it to a voltage that is referenced to the local analog ground. Coupling Noise To prevent coupling noise onto the AD8250, do the following guidelines: • Do not run digital lines under the device. • Run the analog ground plane under the AD8250. • Shield fast switching signals with digital ground to avoid radiating noise to other sections of the board, and never run them near analog signal paths. • Avoid crossover of digital and analog signals. • Connect digital and analog ground at one point only (typically under the ADC). • Use the large traces on power supply lines to ensure a low impedance path. Decoupling is necessary; follow the guidelines listed in the Power Supply Regulation and Bypassing section. VREF + – 06288-060 OP1177 Common-Mode Rejection Figure 53. Driving the Reference Pin COMMON-MODE INPUT VOLTAGE RANGE The 3-op-amp architecture of the AD8250 applies gain and then removes the common-mode voltage. Therefore, internal nodes in the AD8250 experience a combination of both the gained signal and the common-mode signal. This combined signal can be limited by the voltage supplies even when the individual input and output signals are not. Figure 27 and Figure 28 show the allowable common-mode input voltage ranges for various output voltages, supply voltages, and gains. LAYOUT Grounding In mixed-signal circuits, low level analog signals need to be isolated from the noisy digital environment. Designing with the AD8250 is no exception. Its supply voltages are referenced to an analog ground. Its digital circuit is referenced to a digital ground. Although it is convenient to tie both grounds to a single ground plane, the current traveling through the ground wires and PCB can cause errors. Therefore, use separate analog and digital ground planes. Analog and digital ground should meet at only one point: star ground. The AD8250 has high CMRR over frequency, giving it greater immunity to disturbances, such as line noise and its associated harmonics, in contrast to typical instrumentation amplifiers whose CMRR falls off around 200 Hz. Typical instrumentation amplifiers often need common-mode filters at their inputs to compensate for this shortcoming. The AD8250 is able to reject CMRR over a greater frequency range, reducing the need for input common-mode filtering. Careful board layout maximizes system performance. To maintain high CMRR over frequency, lay out the input traces symmetrically. Ensure that the traces maintain resistive and capacitive balance; this holds for additional PCB metal layers under the input pins and traces. Source resistance and capacitance should be placed as close to the inputs as possible. Should a trace cross the inputs (from another layer), route it perpendicular to the input traces. Rev. C | Page 18 of 24 Data Sheet AD8250 RF INTERFERENCE DRIVING AN ADC RF rectification is often a problem when amplifiers are used in applications where there are strong RF signals. The disturbance can appear as a small dc offset voltage. High frequency signals can be filtered with a low-pass RC network placed at the input of the instrumentation amplifier, as shown in Figure 54. The filter limits the input signal bandwidth according to the following relationship: An instrumentation amplifier is often used in front of an ADC to provide CMRR. Usually, instrumentation amplifiers require a buffer to drive an ADC. However, the low output noise, low distortion, and low settle time of the AD8250 make it an excellent ADC driver. FilterFreq DIFF FilterFreqCM 1 2 R(2C D CC ) 1 2 RCC where CD ≥ 10 CC. +15V 0.1µF 10µF In this example, a 1 nF capacitor and a 100 Ω resistor create an antialiasing filter for the AD7612. The 1 nF capacitor stores and delivers the necessary charge to the switched capacitor input of the ADC. The 100 Ω series resistor reduces the burden of the 1 nF load from the amplifier and isolates it from the kickback current injected from the switched capacitor input of the AD7612. Selecting too small a resistor improves the correlation between the voltage at the output of the AD8250 and the voltage at the input of the AD7612 but may destabilize the AD8250. A tradeoff must be made between selecting a resistor small enough to maintain accuracy and large enough to maintain stability. +15V CC R +IN 10μF AD8250 CD R 0.1µF WR OUT +12V A1 A0 +IN REF –IN REF 10µF +5V –IN 06288-061 –15V AD7612 1nF ADR435 DGND 10μF Figure 54. RFI Suppression Values of R and CC should be chosen to minimize RFI. A mismatch between the R × CC at the positive input and the R × CC at the negative input degrades the CMRR of the AD8250. By using a value of CD that is 10 times larger than the value of CC, the effect of the mismatch is reduced and performance is improved. Rev. C | Page 19 of 24 DGND 0.1µF 06288-062 0.1µF 0.1μF 0.1μF 100Ω AD8250 CC –12V –15V Figure 55. Driving an ADC AD8250 Data Sheet APPLICATIONS DIFFERENTIAL OUTPUT SETTING GAINS WITH A MICROCONTROLLER +15V In certain applications, it is necessary to create a differential signal. High resolution ADCs often require a differential input. In other cases, transmission over a long distance can require differential signals for better immunity to interference. 10μF 0.1µF WR A1 A0 +IN Figure 57 shows how to configure the AD8250 to output a differential signal. An op amp, the AD817, is used in an inverting topology to create a differential voltage. VREF sets the output midpoint according to the equation shown in the figure. Errors from the op amp are common to both outputs and are thus common mode. Likewise, errors from using mismatched resistors cause a common-mode dc offset error. Such errors are rejected in differential signal processing by differential input ADCs or instrumentation amplifiers. MICROCONTROLLER + AD8250 REF – –IN DGND DGND 0.1µF 06288-063 10μF –15V Figure 56. Programming Gain Using a Microcontroller When using this circuit to drive a differential ADC, VREF can be set using a resistor divider from the ADC reference to make the output ratiometric with the ADC. +12V 0.1μF AMPLITUDE WR +5V A1 A0 +IN –5V AMPLITUDE + VOUTA = VIN + VREF 2 AD8250 VIN G=1 – 0.1μF +2.5V 0V –2.5V REF TIME 4.99kΩ DGND – –12V –12V 4.99kΩ 10pF + AD817 +12V VREF 0V AMPLITUDE 10μF 0.1µF –12V 0.1µF 10μF DGND VOUTB = –VIN + VREF 2 Figure 57. Differential Output with Level Shift Rev. C | Page 20 of 24 +2.5V 0V –2.5V TIME 06288-064 +12V Data Sheet AD8250 0 DATA ACQUISITION –10 The AD8250 makes an excellent instrumentation amplifier for use in data acquisition systems. Its wide bandwidth, low distortion, low settling time, and low noise enable it to condition signals in front of a variety of 16-bit ADCs. –20 –30 AMPLITUDE (dB) –40 Figure 59 shows a schematic of the AD825x data acquisition demonstration board. The quick slew rate of the AD8250 allows it to condition rapidly changing signals from the multiplexed inputs. An FPGA controls the AD7612, AD8250, and ADG1209. In addition, mechanical switches and jumpers allow users to pin strap the gains when in transparent gain mode. –50 –60 –70 –80 –90 –100 –110 –120 –140 0 This system achieved −111 dB of THD at 1 kHz and a signal-tonoise ratio of 91 dB during testing, as shown in Figure 58. 5 10 15 20 25 30 35 40 45 Figure 58. FFT of the AD825x DAQ Demo Board Using the AD8250, 1 kHz Signal JMP 0.1µF +CH3 +CH4 –CH4 –CH3 –CH2 –CH1 806Ω 806Ω 806Ω 806Ω 806Ω 806Ω 806Ω VDD 4 S1A +5V GND DGND EN DGND JMP +5V DGND 2kΩ 2 S3A 7 S4A 0Ω 0Ω DA 8 10 S4B DB 9 11 S3B 0Ω 0Ω CC +IN CD GND 15 10 + WR 5 A1 4 A0 AD8250 REF –IN 9 1 – –VS CC +VS 3 DGND 7 +IN OUT 0Ω 49.9Ω AD7612 1nF ADR435 8 A0 13 S1B A1 ALTERA EPF6010ATC144-3 DGND 6 ADG1209 12 S2B –VS 2kΩ 10µF 2 5 S2A 6 –12V 1 C4 0.1µF C3 0.1µF VSS 16 3 +12V –12V JMP 0.1µF –12V +5V 2kΩ DGND JMP +5V R8 2kΩ 06288-065 +CH2 806Ω + 10µF 14 +CH1 JMP +12V + +12V 50 FREQUENCY (kHz) DGND Figure 59. Schematic of ADG1209, AD8250, and AD7612 in the AD825x DAQ Demo Board Rev. C | Page 21 of 24 06288-066 –130 AD8250 Data Sheet OUTLINE DIMENSIONS 3.10 3.00 2.90 10 3.10 3.00 2.90 5.15 4.90 4.65 6 1 5 PIN 1 IDENTIFIER 0.50 BSC 0.95 0.85 0.75 15° MAX 1.10 MAX 0.30 0.15 6° 0° 0.23 0.13 COMPLIANT TO JEDEC STANDARDS MO-187-BA 0.70 0.55 0.40 091709-A 0.15 0.05 COPLANARITY 0.10 Figure 60. 10-Lead Mini Small Outline Package [MSOP] (RM-10) Dimensions shown in millimeters ORDERING GUIDE Model1 AD8250ARMZ AD8250ARMZ-RL AD8250ARMZ-R7 AD8250-EVALZ 1 Temperature Range –40°C to +85°C –40°C to +85°C –40°C to +85°C Package Description 10-Lead Mini Small Outline Package [MSOP] 10-Lead Mini Small Outline Package [MSOP] 10-Lead Mini Small Outline Package [MSOP] Evaluation Board Z = RoHS Compliant Part. Rev. C | Page 22 of 24 Package Option RM-10 RM-10 RM-10 Branding H00 H00 H00 Data Sheet AD8250 NOTES Rev. C | Page 23 of 24 AD8250 Data Sheet NOTES ©2007–2013 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D06288-0-5/13(C) Rev. C | Page 24 of 24