NCP1251 - ON Semiconductor

NCP1251
Current-Mode PWM
Controller for Off-line
Power Supplies
The NCP1251 is a highly integrated PWM controller capable of
delivering a rugged and high performance offline power supply in a
tiny TSOP−6 package. With a supply range up to 28 V, the controller
hosts a jittered 65 kHz or 100 kHz switching circuitry operated in peak
current mode control. When the power on the secondary side starts to
decrease, the controller automatically folds back its switching
frequency down to a minimum level of 26 kHz. As the power further
goes down, the part enters skip cycle while limiting the peak current.
Over Power Protection (OPP) is a difficult exercise especially when
no−load standby requirements drive the converter specifications. The
ON proprietary integrated OPP lets you harness the maximum
delivered power without affecting your standby performance simply
via two external resistors. A latched Over Voltage Protection (OVP) is
combined on the same pin. For ease of implementation, a latched OVP
also monitors the VCC line. They offer an efficient protection in case
of optocoupler destruction or adverse open loop operation.
Finally, a timer−based short−circuit protection offers the best
protection scheme, letting you precisely select the protection trip point
irrespective of a loose coupling between the auxiliary and the power
windings.
Features
• Fixed−Frequency 65 or 100 kHz Current−Mode Control Operation
• Internal and Adjustable Over Power Protection (OPP) Circuit
• Frequency Foldback Down to 26 kHz and Skip−Cycle in Light Load
•
•
•
•
•
•
•
•
•
•
•
•
Conditions
Internal Ramp Compensation
Internal Fixed 4 ms Soft−Start
100 ms Timer−Based Auto−Recovery Short−Circuit Protection
Frequency Jittering in Normal and Frequency Foldback Modes
Option for Auto−Recovery or Latched Short−Circuit Protection
OVP Input for Improved Robustness
Up to 28 V VCC Operation
Latched or Auto−Recovery OVP Protection on VCC
+300 mA / −500 mA Source/Sink Drive Capability
Less than 100 mW Standby Power at High Line
EPS 2.0 Compliant
These are Pb−Free Devices
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MARKING
DIAGRAM
TSOP−6
(SOT23−6)
SN SUFFIX
CASE 318G
STYLE 13
1
5xy
x
y
A
Y
W
G
5xyAYWG
G
1
= Specific Device Code
= A or U
= A, 2, C, D, or F
= Assembly Location
= Year
= Work Week
= Pb−Free Package
(Note: Microdot may be in either location)
PIN CONNECTIONS
GND
1
6
DRV
FB
2
5
VCC
OPP/Latch
3
4
CS
(Top View)
ORDERING INFORMATION
See detailed ordering, marking and shipping information on
page 2 of this data sheet.
Typical Applications
• ac−dc Converters for TVs, Set−top Boxes and Printers
• Offline Adapters for Notebooks and Netbooks
© Semiconductor Components Industries, LLC, 2015
June, 2015 − Rev. 14
1
Publication Order Number:
NCP1251/D
NCP1251
Vbulk
Vo u t
.
.
OVP
OPP
.
NCP1251
1
6
2
5
3
4
ramp
comp.
Figure 1. Typical Application Example
Pin N5
Pin Name
Function
1
GND
−
Pin Description
The controller ground.
2
FB
Feedback pin
3
OPP/OVP
Adjust the Over Power Protection
Latches off the part
Hooking an optocoupler collector to this pin will allow regulation.
4
CS
Current sense + ramp
compensation
This pin monitors the primary peak current but also offers a means to
introduce ramp compensation.
5
VCC
Supplies the controller
This pin is connected to an external auxiliary voltage and supplies the
controller. When the VCC exceeds a certain level, the part permanently
latches off.
6
DRV
Driver output
A resistive divider from the auxiliary winding to this pin sets the OPP
compensation level. When brought above 3 V, the part is fully latched off.
The driver’s output to an external MOSFET gate.
OPTIONS
Controller
Frequency
OCP
VCC OVP
OVP/OTP
NCP1251ASN65T1G
65 kHz
Latched
Latched
Latched
NCP1251BSN65T1G
65 kHz
Autorecovery
Latched
Latched
NCP1251CSN65T1G
65 kHz
Autorecovery
Autorecovery
Latched
NCP1251FSN65T1G
65 kHz
Autorecovery
Latched
Latched
NCP1251ASN100T1G
100 kHz
Latched
Latched
Latched
NCP1251BSN100T1G
100 kHz
Autorecovery
Latched
Latched
NOTE: F version has a different foldback scheme for improved efficiency.
ORDERING INFORMATION
Package
Marking
OCP Protection
VCC OVP
Protection
Switching
Frequency
5AA
Latch
Latch
65 kHz
NCP1251BSN65T1G
5A2
Autorecovery
Latch
65 kHz
NCP1251CSN65T1G
5AE
Autorecovery
Autorecovery
65 kHz
NCP1251FSN65T1G
5AF
Autorecovery
Latch
65 kHz
NCP1251ASN100T1G
5AC
Latch
Latch
100 kHz
NCP1251BSN100T1G
5AD
Autorecovery
Latch
100 kHz
Device
NCP1251ASN65T1G
Package
Shipping†
TSOP−6
(Pb−Free)
3000 /
Tape & Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
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2
NCP1251
IpFlag
BO
OPP
600−ns time
constant
Vcc logic
management
and fault timer
UVLO
Up counter
Vlatch
vdd
OVP
gone?
RST
4
20us time
constant
VOVP
power
on reset
S
Q
Rlimit
Q
vdd
R
Power on
reset
Frequency
modulation
Vcc
Iscr
1−us
blanking
65 kHz
100 kHz
clock
Clamp
S
Q
Q
R
Frequency
foldback
Drv
Vfold
Vskip
Rramp
4 ms
SS
vdd
The soft−start is activated during:
RFB
IpFlag
− the startup sequence
− the auto−recovery burst mode
/ 4.2
VFB < 1.05 V ? setpoint = 250 mV
FB
VOPP
CS
LEB
250 mV
peak current
freeze
+
Vlimit + VOPP
GND
Vlimit
Figure 2. Internal Circuit Architecture
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3
NCP1251
MAXIMUM RATINGS TABLE
Symbol
VCC
VDRVtran
Rating
Value
Unit
28
V
Maximum DRV pin voltage when DRV in H state, transient voltage (Note 1)
VCC + 0.3
V
Maximum voltage on low power pins CS, FB and OPP
−0.3 to 10
V
Power Supply voltage, VCC pin, continuous voltage
IOPP
Maximum injected negative current into the OPP pin (pin 3)
−2
mA
ISCR
Maximum continuous current into the VCC pin while in latch mode
3
mA
RqJA
Thermal Resistance Junction−to−Air
360
°C/W
Maximum Junction Temperature
150
°C
−60 to +150
°C
2
kV
200
V
1
kV
TJ,max
Storage Temperature Range
ESD Capability, Human Body Model (HBM), all pins
ESD Capability, Machine Model (MM)
ESD Capability, Charged Device Model (CDM)
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
1. The transient voltage is a voltage spike injected to DRV pin being in high state. Maximum transient duration is 100 ns.
2. This device series contains ESD protection and exceeds the following tests: Human Body Model 2000 V per JESD22, Method A114E.
Machine Model Method 200 V per JESD22, Method A115A. Charged Device Model per JEDEC Standard JESD22−C101D
3. This device contains latch−up protection and exceeds 100 mA per JEDEC Standard JESD78.
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4
NCP1251
ELECTRICAL CHARACTERISTICS
(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, Max TJ = 150°C, VCC = 12 V unless otherwise noted)
Symbol
Rating
Pin
Min
Typ
Max
Unit
VCC increasing level at which driving pulses are authorized
5
16
18
20
V
VCC(min)
VCC decreasing level at which driving pulses are stopped
5
8.2
8.8
9.4
V
VCCHYST
Hysteresis VCCON − VCC(min)
5
6.0
Clamped VCC when latched off / burst mode activation @ ICC = 500 mA
5
ICC1
Start−up current
5
ICC2
Internal IC consumption with IFB = 50 mA, FSW = 65 kHz and CL = 0 nF
5
ICC3
Internal IC consumption with IFB = 50 mA, FSW = 65 kHz and CL = 1 nF
ICC2
ICC3
SUPPLY SECTION
VCCON
VZENER
V
7.0
V
15
mA
1.4
2.2
mA
5
2.1
3.0
mA
Internal IC consumption with IFB = 50 mA, FSW = 100 kHz and CL = 0 nF
5
1.7
2.5
mA
4.0
mA
Internal IC consumption with IFB = 50 mA, FSW = 100 kHz and CL = 1 nF
5
3.1
ICCstby
Internal IC consumption while in skip cycle (VCC = 12 V, driving a typical 6 A/600 V
MOSFET)
5
550
ICCLATCH
Current flowing into VCC pin that keeps the controller latched (Note 4)
TJ = −40°C to +125°C
TJ = 0°C to +125°C
5
Current−limit resistor in series with the latch SCR
5
4.0
kW
Rlim
mA
mA
40
32
DRIVE OUTPUT
Tr
Output voltage rise−time @ CL = 1 nF, 10−90% of output signal
6
40
ns
Tf
Output voltage fall−time @ CL = 1 nF, 10−90% of output signal
6
30
ns
ROH
Source resistance
6
13
W
ROL
Sink resistance
6
6.0
W
Peak source current, VGS = 0 V – (Note 5)
6
300
mA
Peak sink current, VGS = 12 V – (Note 5)
6
500
mA
VDRVlow
DRV pin level at VCC close to VCC(min) with a 33 kW resistor to GND
6
8.0
VDRVhigh
DRV pin level at VCC = 28 V – DRV unloaded
6
10
Isource
Isink
V
12
14
V
CURRENT COMPARATOR
mA
Input Bias Current @ 0.8 V input level on pin 4
4
VLimit1
Maximum internal current setpoint – TJ = 25°C – pin 3 grounded
4
0.744
0.8
0.856
V
VLimit2
Maximum internal current setpoint – TJ = −40°C to 125°C – pin 3 grounded
4
0.72
0.8
0.88
V
Default internal voltage set point for frequency foldback trip point – 45% of Vlimit
3
357
IIB
Vfold
Vfreeze
0.02
mV
Internal peak current setpoint freeze ([31% of Vlimit)
3
250
TDEL
Propagation delay from current detection to gate off−state
4
100
TLEB
Leading Edge Blanking Duration
4
300
ns
TSS
Internal soft−start duration activated upon startup, auto−recovery
−
4
ms
IOPPo
Setpoint decrease for pin 3 biased to –250 mV – (Note 6)
3
31.3
%
IOOPv
Voltage setpoint for pin 3 biased to −250 mV – (Note 6), TJ = 25°C
3
0.51
0.55
0.60
IOOPv
Voltage setpoint for pin 3 biased to −250 mV – (Note 6), TJ = −40°C to 125°C
3
0.50
0.55
0.62
IOPPs
Setpoint decrease for pin 3 grounded
3
0
mV
150
ns
V
V
%
Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product
performance may not be indicated by the Electrical Characteristics if operated under different conditions.
4. For design robustness, we recommend to inject 60 mA as a minimum at the lowest input line voltage.
5. Guaranteed by design
6. See characterization table for linearity over negative bias voltage
7. A 1 MW resistor is connected from pin 3 to the ground for the measurement.
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5
NCP1251
ELECTRICAL CHARACTERISTICS
(For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, Max TJ = 150°C, VCC = 12 V unless otherwise noted)
Symbol
Rating
Pin
Min
Typ
Max
Unit
INTERNAL OSCILLATOR
fOSC
Oscillation frequency (65 kHz version)
−
61
65
71
kHz
fOSC
Oscillation frequency (100 kHz version)
−
92
100
108
kHz
Dmax
Maximum duty−cycle
−
76
80
84
%
fjitter
Frequency jittering in percentage of fOSC
−
±5
%
fswing
Swing frequency
−
240
Hz
FEEDBACK SECTION
Rup
Internal pull−up resistor
2
20
kW
Req
Equivalent ac resistor from FB to GND
2
16
kW
Iratio
Pin 2 to current setpoint division ratio
−
4.2
Feedback voltage below which the peak current is frozen
2
1.05
V
Vfreeze
FREQUENCY FOLDBACK
Vfold
Frequency foldback level on the feedback pin – [45% of maximum peak current
−
1.5
V
VfoldF
Frequency foldback level on the feedback pin – [59% of maximum peak current
(F version only)
−
1.9
V
Ftrans
Transition frequency below which skip−cycle occurs
−
Vfold,end
End of frequency foldback feedback level, Fsw = Fmin
350
mV
VfoldF,end
End of frequency foldback feedback level, Fsw = Fmin
(F version only)
1.5
V
22
26
30
kHz
Vskip
Skip−cycle level voltage on the feedback pin
−
300
mV
Skip
hysteresis
Hysteresis on the skip comparator – (Note 5)
−
30
mV
INTERNAL SLOPE COMPENSATION
Vramp
Internal ramp level @ 25°C – (Note 7)
4
2.5
V
Rramp
Internal ramp resistance to CS pin
4
20
kW
PROTECTIONS
Latching level input
3
Tlatch−blank
Blanking time after drive turn off
1
1.0
Tlatch−count
Number of clock cycles before latch confirmation
−
4.0
OVP detection time constant
1
600
Timer
Internal auto−recovery fault timer duration
−
100
130
160
VOVP
Latched Over voltage protection on the VCC rail
5
24
25.5
27
Delay before OVP on VCC confirmation
5
Vlatch
Tlatch−del
TOVPdel
2.7
3
20
3.3
V
ms
ns
ms
V
ms
Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product
performance may not be indicated by the Electrical Characteristics if operated under different conditions.
4. For design robustness, we recommend to inject 60 mA as a minimum at the lowest input line voltage.
5. Guaranteed by design
6. See characterization table for linearity over negative bias voltage
7. A 1 MW resistor is connected from pin 3 to the ground for the measurement.
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6
NCP1251
TYPICAL CHARACTERISTICS
72
85
84
70
83
68
81
FSW (kHz)
Dmax (%)
82
80
79
66
64
78
77
62
76
75
−50
−25
0
25
50
75
100
60
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 3.
Figure 4.
100
125
100
125
100
125
31
440
30
29
390
F_swing (Hz)
Ftrans (kHz)
28
27
26
25
340
290
240
24
23
190
22
21
−50
−25
0
25
50
75
100
140
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 5.
Figure 6.
0.89
490
0.87
FSW = 65 kHz
440
0.85
390
VLskip (mV)
Vlimit (mV)
0.83
0.81
0.79
0.77
290
240
0.75
190
0.73
0.71
−50
340
−25
0
25
50
75
100
140
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 7.
Figure 8.
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7
NCP1251
TYPICAL CHARACTERISTICS
44
0.6
39
34
IOOPV (V)
IOOPO (%)
0.58
29
24
19
−50
−25
0
25
50
75
100
0.5
−50
125
25
50
75
Figure 9.
Figure 10.
100
125
100
125
100
125
9.5
9.3
18.9
9.1
18.4
17.9
17.4
8.9
8.7
8.5
16.9
8.3
16.4
−25
0
25
50
75
100
8.1
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 11.
Figure 12.
14
16
13
14
12
12
ICC1 (mA)
11
10
9
8
10
8
6
7
4
6
2
5
−50
0
TEMPERATURE (°C)
19.4
15.9
−50
−25
TEMPERATURE (°C)
VCC(min) (V)
VCC(ON) (V)
0.54
0.52
19.9
VCC(Hyst) (V)
0.56
−25
0
25
50
75
100
0
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 13.
Figure 14.
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8
NCP1251
TYPICAL CHARACTERISTICS
2.5
2
FSW = 65 kHz
FSW = 65 kHz
2
ICC3 (mA)
ICC2 (mA)
1.5
1
0.5
0
−50
1.5
1
0.5
−25
0
25
50
75
100
0
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 15.
Figure 16.
100
125
100
125
100
125
30
10
25
ICCLatch (mA)
Vzener (V)
8
6
4
2
0
−50
20
15
10
5
−25
0
25
50
75
100
0
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 17.
Figure 18.
160
390
140
340
120
Req (kW)
Tleb (V)
290
240
100
80
60
190
40
140
90
−50
20
−25
0
25
50
75
100
0
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 19.
Figure 20.
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NCP1251
TYPICAL CHARACTERISTICS
4.8
3.4
3.3
4.6
3.2
Vlatch (V)
Iratio (−)
4.4
4.2
4
3.1
3
2.9
2.8
3.8
−25
0
25
50
75
100
2.6
−50
125
50
75
Figure 22.
80
80
60
40
20
20
25
50
75
100
0
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 23.
Figure 24.
11
100
125
100
125
100
125
60
40
0
25
TEMPERATURE (°C)
100
−25
0
Figure 21.
100
0
−50
−25
TEMPERATURE (°C)
tfall (ns)
trise (ns)
3.6
−50
2.7
35
10
30
9
25
Roh (W)
Rol (W)
8
7
6
5
20
15
4
10
3
2
−50
−25
0
25
50
75
100
5
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 25.
Figure 26.
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NCP1251
TYPICAL CHARACTERISTICS
14
100
13
12
Vdrv_low (V)
Vovp_del (ms)
80
60
40
11
10
9
20
0
−50
8
−25
0
25
50
75
100
7
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 27.
Figure 28.
12.9
100
125
100
125
100
125
4.9
12.4
4.4
11.4
TSS (ms)
Vdrv_high (V)
11.9
10.9
10.4
9.9
3.9
3.4
9.4
8.9
−50
−25
0
25
50
75
100
2.9
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 29.
Figure 30.
1.9
360
1.8
358
Vfold(CS) (mV)
Vfold(FB) (V)
1.7
1.6
1.5
1.4
356
354
1.3
352
1.2
1.1
−50
−25
0
25
50
75
100
350
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 31.
Figure 32.
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11
NCP1251
TYPICAL CHARACTERISTICS
0.41
390
340
0.37
Vskip (mV)
Vfold_end (V)
0.39
0.35
290
0.33
240
0.31
0.29
−50
−25
0
25
50
75
100
190
−50
125
50
75
TEMPERATURE (°C)
Figure 34.
100
125
100
125
1.7
1.5
Vfreeze(FB) (V)
Vfreeze (mV)
25
Figure 33.
340
290
240
1.3
1.1
0.9
−25
0
25
50
75
100
0.7
−50
125
−25
0
25
50
75
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 35.
Figure 36.
160
3.5
150
3
140
2.5
ICC (mA)
TIMER (ms)
0
TEMPERATURE (°C)
390
190
−50
−25
130
120
2
1.5
110
1
100
0.5
FSW = 65 kHz
90
−50
−25
0
25
50
75
100
0
125
0
0.5
1
1.5
2
2.5
3
TEMPERATURE (°C)
ADAPTER OUTPUT CURRENT (A)
Figure 37.
Figure 38. Controller Consumption vs.
Adapter Output Current
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3.5
NCP1251
TYPICAL CHARACTERISTICS
26.9
26.4
VOVP (V)
25.9
25.4
24.9
24.4
23.9
−50
−25
0
25
50
75
TEMPERATURE (°C)
Figure 39.
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13
100
125
NCP1251
APPLICATION INFORMATION
Introduction
The NCP1251 implements a standard current mode
architecture where the switch−off event is dictated by the
peak current setpoint. This component represents the ideal
candidate where low part−count and cost effectiveness are
the key parameters, particularly in low−cost ac−dc adapters,
open−frame power supplies etc. Capitalizing on the
NCP120X series success, the NCP1251 packs all the
necessary components normally needed in today modern
power supply designs, bringing several enhancements such
as a non−dissipative OPP.
• Current−mode operation with internal ramp
compensation: Implementing peak current mode
control at a fixed 65 kHz or 100 kHz, the NCP1251
offers an internal ramp compensation signal that can
easily by summed with the sensed current. Sub
harmonic oscillations are eliminated via the inclusion of
a single resistor in series with the current−sense
information.
• Internal OPP: By routing a portion of the negative
voltage present during the on−time on the auxiliary
winding to the dedicated OPP pin (pin 3), the user has a
simple and non−dissipative means to alter the
maximum peak current setpoint as the bulk voltage
increases. If the pin is grounded, no OPP compensation
occurs. If the pin receives a negative voltage down to
–250 mV, then a peak current reduction down to 31.3%
typical can be achieved. For an improved performance,
the maximum voltage excursion on the sense resistor is
limited to 0.8 V.
• Low startup current: Achieving a low no−load
standby power always represents a difficult exercise
when the controller draws a significant amount of
current during start−up. Due to its proprietary
architecture, the NCP1251 is guaranteed to draw less
than 15 mA typical, easing the design of low standby
power adapters.
• EMI jittering: An internal low−frequency modulation
signal varies the pace at which the oscillator frequency
is modulated. This helps by spreading out energy in
conducted noise analysis. To improve the EMI
signature at low power levels, the jittering remains
active in frequency foldback mode.
• Frequency foldback capability: A continuous flow of
pulses is not compatible with no−load/light−load
standby power requirements. To excel in this domain,
the controller observes the feedback pin and when it
reaches a level of 1.5 V, the oscillator then starts to
reduce its switching frequency as the feedback level
continues to decrease. When the feedback pin reaches
1.05 V, the peak current setpoint is internally frozen and
the frequency continues to decrease. It can go down to
•
•
•
•
26 kHz (typical) reached for a feedback level of
roughly 350 mV. At this point, if the power continues to
drop, the controller enters classical skip−cycle mode.
Internal soft−start: A soft−start precludes the main
power switch from being stressed upon start−up. In this
controller, the soft−start is internally fixed to 4 ms. The
soft−start is activated when a new startup sequence
occurs or during an auto−recovery hiccup.
OVP input: The NCP1251 includes a latch input
(pin 3) that can be used to sense an overvoltage
condition on the adapter. If this pin is brought higher
than the internal reference voltage Vlatch, then the
circuit permanently latches off. The VCC pin is pulled
down to a fixed level, keeping the controller latched.
The latch reset occurs when the user disconnects the
adapter from the mains and lets the VCC falls below the
VCC reset.
Latched OVP on VCC: It is sometimes interesting to
implement a circuit protection by sensing the VCC
level. This is what the NCP1251 does by monitoring its
VCC pin. When the voltage on this pin exceeds 25 V
typical, the pulses are immediately stopped and the part
latches off. The Vcc is maintained to 7 V typical and
remains in this state until the user unplugs the power
supply.
Short−circuit protection: Short−circuit and especially
over−load protections are difficult to implement for
transformers with high leakage inductance between
auxiliary and power windings (the aux winding level
does not properly collapse in presence of an output
short). Here, every time the internal 0.8 V maximum
peak current limit is activated (or less when OPP is
used), an error flag is asserted and a time period starts,
thanks to an internal timer. If the timer reaches
completion while the error flag is still present, the
controller stops the pulses and goes into a latch−off
phase, operating in a low−frequency burst−mode. When
the fault is cleared, the SMPS resumes operation.
Please note that some versions offer an auto−recovery
mode as described and some latch off in case of a short
circuit.
Start−up Sequence
The NCP1251 start−up voltage is made purposely high to
permit a large energy storage in a small VCC capacitor value.
This helps to operate with a small start−up current which,
together with a small VCC capacitor, will not hamper the
start−up time. To further reduce the standby power, the
start−up current of the controller is extremely low, below
15 mA maximum. The start−up resistor can therefore be
connected to the bulk capacitor or directly to the mains input
voltage to further reduce the power dissipation.
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14
NCP1251
R3
200k
3
10
R2
200k
5
D2
1N4007
D1
1N4007
11
12
R1
200k
Cbulk
22uF
input
mains
D6
1N4148
1
D5
1N4935
VCC
D4
1N4007
D3
1N4007
2
C1
4.7uF
4
aux.
C3
47uF
Figure 40. The Startup Resistor Can Be Connected to the Input Mains for Further Power Dissipation Reduction
The first step starts with the calculation of the VCC
capacitor which will supply the controller when it operates
until the auxiliary winding takes over. Experience shows
that this time t1 can be between 5 ms and 20 ms. If we
consider we need at least an energy reservoir for a t1 time of
10 ms, the VCC capacitor must be larger than:
CV CC w
I CCt 1
VCC on * VCC min
w
3m
10m
9
This calculation is purely theoretical, and assumes a
constant charging current. In reality, the take over time can
be shorter (or longer!) and it can lead to a reduction of the
VCC capacitor. Hence, a decrease in charging current and an
increase of the start−up resistor, thus reducing the standby
power. Laboratory experiments on the prototype are thus
mandatory to fine tune the converter. If we chose the 413 kW
resistor as suggested by Equation 4, the dissipated power at
high line amounts to:
w 3.3 mF
(eq. 1)
Let us select a 4.7 mF capacitor at first and experiments in
the laboratory will let us know if we were too optimistic for
the time t1. The VCC capacitor being known, we can now
evaluate the charging current we need to bring the VCC
voltage from 0 to the VCCon of the IC, 18 V typical. This
current has to be selected to ensure a start−up at the lowest
mains (85 V rms) to be less than 3 s (2.5 s for design margin):
I charge w
VCC onC VCC
2.5
w
18
4.7m
2.5
P Rstart*up +
+
I CVCC,min +
p
w 34 mA
(eq. 2)
* VCC on
(eq. 3)
R start*up
To make sure this current is always greater than 49 mA,
then the minimum value for Rstart−up can be extracted:
V ac,rmsǸ2
R start*up v
p
* VCC on
I CVCC,min
85
v
1.414
p
* 18
49m
2
4R start*up
230 2
0.827Meg
+
ǒ230
4
Ǹ2Ǔ
413k
2
(eq. 5)
+ 64 mW
Now that the first VCC capacitor has been selected, we
must ensure that the self−supply does not disappear when in
no−load conditions. In this mode, the skip−cycle can be so
deep that refreshing pulses are likely to be widely spaced,
inducing a large ripple on the VCC capacitor. If this ripple is
too large, chances exist to touch the VCCmin and reset the
controller into a new start−up sequence. A solution is to
grow this capacitor but it will obviously be detrimental to the
start−up time. The option offered in Figure 40 elegantly
solves this potential issue by adding an extra capacitor on the
auxiliary winding. However, this component is separated
from the VCC pin via a simple diode. You therefore have the
ability to grow this capacitor as you need to ensure the
self−supply of the controller without jeopardizing the
start−up time and standby power. A capacitor ranging from
22 to 47 mF is the typical value for this device.
One note on the start-up current. If reducing it helps to
improve the standby power, its value cannot fall below a
certain level at the minimum input voltage. Failure to inject
If we account for the 15 mA that will flow inside the
controller, then the total charging current delivered by the
start−up resistor must be 49 mA. If we connect the start−up
network to the mains (half−wave connection then), we know
that the average current flowing into this start−up resistor
will be the smallest when VCC reaches the VCCon of the
controller:
V ac,rmsǸ2
V ac,peak
v 413.5 kW
(eq. 4)
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15
NCP1251
enough current (30 mA) at low line will turn a converter in
fault into an auto-recovery mode since the SCR won’t
remain latched. To build a sufficient design margin, we
recommend to keep at least 60 mA flowing at the lowest input
line (80 V rms for 85 V minimum for instance). An excellent
solution is to actually combine X2 discharge and start-up
networks as proposed in Figure 13 of application note
AND8488/D.
the current−sense offset. A way to reduce the power
capability at high line is to capitalize on the negative voltage
swing present on the auxiliary diode anode. During the
power switch on−time, this point dips to −NVin, N being the
turns ratio between the primary winding and the auxiliary
winding. The negative plateau observed on Figure 42 will
have an amplitude dependant on the input voltage. The idea
implemented in this chip is to sum a portion of this negative
swing with the 0.8 V internal reference level. For instance,
if the voltage swings down to −150 mV during the on time,
then the internal peak current set point will be fixed to 0.8 −
0.150 = 650 mV. The adopted principle appears in Figure 42
and shows how the final peak current set point is
constructed.
Internal Over Power Protection
There are several known ways to implement Over Power
Protection (OPP), all suffering from particular problems.
These problems range from the added consumption burden
on the converter or the skip−cycle disturbance brought by
1 v(24)
40.0
off−time
20.0
N1(Vout +Vf)
v(24) (V)
1
0
−20.0
−N2Vbulk
on−time
−40.0
464u
472u
480u
time (s)
488u
496u
Figure 41. The Signal Obtained on the Auxiliary Winding Swings Negative During the On−time
Let’s assume we need to reduce the peak current from
2.5 A at low line, to 2 A at high line. This corresponds to a
20% reduction or a set point voltage of 640 mV. To reach this
level, then the negative voltage developed on the OPP pin
must reach:
V OPP + 640m * 800m + −160 mV
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16
(eq. 6)
NCP1251
RoppU
swings to:
Vout during toff
−N V in during ton
VCC
aux
This p oin t will
be adjusted to
reduce the ref
at hi line to the
desired level.
from FB
reset
VDD
SUM2
−
K1
OPP
ref
CS
K2
+
Io p p
0.8 V
$5%
R oppL
ref = 0.8 V + VOPP
(V O P P is negativ e)
Figure 42. The OPP Circuitry Affects the Maximum Peak Current Set Point by Summing a Negative Voltage to the
Internal Voltage Reference
Let us assume that we have the following converter
characteristics:
Vout = 19 V
Vin = 85 to 265 Vrms
N1 = Np:Ns = 1:0.25
N2 = Np:Naux = 1:0.18
Given the turns ratio between the primary and the auxiliary
windings, the on−time voltage at high line (265 Vac) on the
auxiliary winding swings down to:
V aux + −N 2V in,max + −0.18
375 + −67.5 V
Div +
0.16
[ 2.4m
67.5
(eq. 8)
If we arbitrarily fix the pull−down resistor ROPPL to 1 kW,
then the upper resistor can be obtained by:
R OPPU +
67.5 * 0.16
0.16ń1k
[ 421 kW
(eq. 9)
If we now plot the peak current set point obtained by
implementing the recommended resistor values, we obtain
the following curve (Figure 43):
(eq. 7)
To obtain a level as imposed by Equation 6, we need to
install a divider featuring the following ratio:
Peak current
setpoint
100%
80%
375
Vbulk
Figure 43. The Peak Current Regularly Reduces Down to 20% at 375 Vdc
clamped slightly below –300 mV which means that if more
current is injected before reaching the ESD forward drop,
then the maximum peak reduction is kept to 40%. If the
voltage finally forward biases the internal zener diode, then
care must be taken to avoid injecting a current beyond
–2 mA. Given the value of ROPPU, there is no risk in the
present example.
The OPP pin is surrounded by Zener diodes stacked to
protect the pin against ESD pulses. These diodes accept
some peak current in the avalanche mode and are designed
to sustain a certain amount of energy. On the other side,
negative injection into these diodes (or forward bias) can
cause substrate injection which can lead to an erratic circuit
behavior. To avoid this problem, the pin is internally
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17
NCP1251
Finally, please note that another comparator internally
fixes the maximum peak current set point to 0.8 V even if the
OPP pin is inadvertently biased above 0 V.
to reduce the operating frequency down to 26 kHz. This
value is reached at a voltage feedback level of 350 mV
typically. Below this point, if the output power continues to
decrease, the part enters skip cycle for the best noise−free
performance in no−load conditions. Figure 44 depicts the
adopted scheme for the part.
The NCP1251F version offers a means to improve
light−load efficiency by folding the switching frequency
sooner compared to the other versions. With the 1251 A, B
and C versions, the minimum frequency is reached for VFB
equals 350 mV. With the 1251F, this minimum frequency
will be obtained at a feedback voltage equal to 1.5 V,
naturally offering a better efficiency for lighter load
conditions. Figure 45 portrays the specific foldback scheme
implemented in the NCP1251F.
Frequency Foldback
The reduction of no−load standby power associated with
the need for improving the efficiency, requires a change to
the traditional fixed−frequency type of operation. This
controller implements a switching frequency foldback when
the feedback voltage passes below a certain level, Vfold, set
around 1.5 V. At this point, the oscillator enters frequency
foldback and reduces its switching frequency. The peak
current setpoint follows the feedback pin until its level
reaches 1.05 V. Below this value, the peak current freezes to
Vfold/4.2 (250 mV or 31% of the maximum 0.8 V setpoint)
and the only way to further reduce the transmitted power is
Frequency
Peak current setpoint
Fsw
FB
VCS
max
65 kHz
26 kHz
max
0.8 V
[ 0.36 V
min
[ 0.25 V
min
VFB
350 mV 1.5 V
Vfold
Vfold,end
Vfreeze Vfold
3.4 V
1.05 V
VFB
3.4 V
1.5 V
Figure 44. By Observing the Voltage on the Feedback Pin, the Controller Reduces its Switching Frequency for an
Improved Performance at Light Load
Figure 45. with NCP1251F, the frequency foldback occurs sooner as the load gets lighter.
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18
NCP1251
Auto−Recovery Short−Circuit Protection
due to the resistive starting network. When VCC reaches
VCCON, the controller attempts to re−start, checking for the
absence of the fault. If the fault is still there, the supply enters
another cycle of so−called hiccup mode. If the fault has
cleared, the power supply resumes normal operation. Please
note that the soft−start is activated during each of the re−start
sequence.
In case of output short−circuit or if the power supply
experiences a severe overloading situation, an internal error
flag is raised and starts a countdown timer. If the flag is
asserted longer than 100 ms, the driving pulses are stopped
and the VCC pin slowly goes down to around 7 V. At this
point, the controller wakes−up and the VCC builds up again
1 vcc
2 vdrv
3 ilprim
4.32
14.8
9.90
3.35
6.05
vcc in volts
15.9
3.89
ilprim in amperes
Plot1
vdrv in volts
1
23.6
Vcc (t)
VDRV (t)
2.38
2
−2.72
−2.12
1.41
ILp (t)
SS
−11.5
−8.13
445m
3
500u
1.50m
2.50m
time in seconds
3.50m
4.50m
Figure 46. An Auto−Recovery Hiccup Mode is Activated for Faults Longer than 100 ms
Slope Compensation
converters. These oscillations take place at half the
switching frequency and occur only during CCM with a
duty−cycle greater than 50%. To lower the current loop gain,
one usually injects between 50% and 100% of the inductor
downslope. Figure 47 depicts how internally the ramp is
generated. Please note that the ramp signal will be
disconnected from the CS pin, during the off time.
The NCP1251 includes an internal ramp compensation
signal. This is the buffered oscillator clock delivered only
during the on time. Its amplitude is around 2.5 V at the
maximum duty−cycle. Ramp compensation is a known
means used to cure sub harmonic oscillations in Continuous
Conduction Mode (CCM) operated current−mode
2.5 V
0V
ON
latch
reset
20k
+
Rcomp
LEB
CS
−
Rsense
from FB
setpoint
Figure 47. Inserting a Resistor in Series with the Current Sense Information Brings Ramp Compensation and
Stabilizes the Converter in CCM Operation.
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19
NCP1251
pin to the controller ground for an improved immunity to the
noise. Please make sure both components are located very
close to the controller.
In the NCP1251 controller, the oscillator ramp features a
2.5 V swing reached at a 80% duty−ratio. If the clock
operates at a 65 kHz frequency, then the available oscillator
slope corresponds to:
S ramp +
V ramp,peak
2.5
+
0.8 15m
D maxT SW
Latching Off the Controller
The OPP pin not only allows a reduction of the peak
current set point in relationship to the line voltage, it also
offers a means to permanently latch−off the part. When the
part is latched−off, the VCC pin is internally pulled down to
around 7 V and the part stays in this state until the user cycles
the VCC down and up again, e.g. by un−plugging the
converter from the mains outlet. It is important to note that
the SCR maintains its latched state as long as the injected
current stays above the minimum value of 30 mA. As the
SCR delatches for an injected current below this value, it is
the designer duty to make sure the injected current is high
enough at the lowest input voltage. Failure to maintain a
sufficiently high current would make the device auto
recover. A good design practice is to ensure at least 60 mA
at the lowest input voltage. The latch detection is made by
observing the OPP pin by a comparator featuring a 3 V
reference voltage. However, for noise reasons and in
particular to avoid the leakage inductance contribution at
turn off, a 1 ms blanking delay is introduced before the
output of the OVP comparator is checked. Then, the OVP
comparator output is validated only if its high−state duration
lasts a minimum of 600 ns. Below this value, the event is
ignored. Then, a counter ensures that 4 successive OVP
events have occurred before actually latching the part. There
are several possible implementations, depending on the
needed precision and the parameters you want to control.
The first and easiest solution is the additional resistive
divider on top of the OPP one. This solution is simple and
inexpensive but requires the insertion of a diode to prevent
disturbing the OPP divider during the on time.
(eq. 10)
+ 208 kVńs or 208 mVńms
In our flyback design, let’s assume that our primary
inductance Lp is 770 mH, and the SMPS delivers 19 V with
a Np :Ns ratio of 1:0.25. The off−time primary current slope
Sp is thus given by:
ǒVout ) VfǓ Nps
N
Sp +
Lp
+
(19 ) 0.8)
4
770m
+ 103 kAńs
(eq. 11)
Given a sense resistor of 330 mW, the above current ramp
turns into a voltage ramp of the following amplitude:
S sense + S pR sense + 103k
0.33
(eq. 12)
+ 34 kVńs or 34 mVńms
If we select 50% of the downslope as the required amount
of ramp compensation, then we shall inject a ramp whose
slope is 17 mV/ms. Our internal compensation being of
208 mV/ms, the divider ratio (divratio) between Rcomp and
the internal 20 kW resistor is:
divratio +
17m
+ 0.082
208m
(eq. 13)
The series compensation resistor value is thus:
R comp + R ramp @ divratio + 20k
0.082 [ 1.6 kW
(eq. 14)
A resistor of the above value will then be inserted from the
sense resistor to the current sense pin. We recommend
adding a small capacitor of 100 pF, from the current sense
D2
1N4148
R3
5k
11
RoppU
421k
VCC
OP P
8
aux.
winding
4
10
C1
100p
9
1
ROPPL
1k
5
OVP
OPP
Vlatch
Figure 48. A Simple Resistive Divider Brings the OPP Pin Above 3 V in Case of a VCC Voltage Runaway above
18 V
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20
NCP1251
First, calculate the OPP network with the above equations.
Then, suppose we want to latch off our controller when Vout
exceeds 25 V. On the auxiliary winding, the plateau reflects
the output voltage by the turns ratio between the power and
the auxiliary winding. In case of voltage runaway for our
19 V adapter, the plateau will go up to:
V aux,OVP + 25
0.18
+ 18 V
0.25
R OVP +
V OVPńR OPPL
+
18 * 3
3ń1k
+
15
+ 5 kW (eq. 16)
3m
In nominal conditions, the plateau establishes to around
14 V. Given the divide−by−6 ratio, the OPP pin will swing
to 14/6 = 2.3 V during normal conditions, leaving 700 mV
margin. A 100 pF capacitor can be added between the OPP
pin and GND to improve noise immunity and avoid erratic
trips in presence of external surges. Do not increase this
capacitor too much otherwise the OPP signal will be affected
by the integrating time constant.
A second solution for the OVP detection alone, is to use
a Zener diode wired as recommended by.
(eq. 15)
Since our OVP comparator trips at a 3 V level, across the
1 kW selected OPP pulldown resistor, it implies a 3 mA
current. From 3 V to go up to 18 V, we need an additional
15 V. Under 3 mA and neglecting the series diode forward
drop, it requires a series resistor of:
D3
15V
V latch * V VOP
D2
1N4148
11
ROPPU
421k
VCC
9
OPP
4
10
C1
22pF
8
aux.
winding
1
ROPPL
1k
5
Vlatch
OPP
OVP
Figure 49. A Zener Diode in Series with a Diode Helps to Improve the Noise Immunity of the System
probe connections!) and check that enough margin exists to
that respect.
For this configuration to maintain an 18 V level, we have
selected a 15 V Zener diode. In nominal conditions, the
voltage on the OPP pin is almost 0 V during the off time as
the Zener is fully blocked. This technique clearly improves
the noise immunity of the system compared to that obtained
from a resistive string as in Figure 48. Please note the
reduction of the capacitor on the OPP pin to 10 pF − 22 pF.
This capacitor is necessary because of the potential spike
coupling through the Zener parasitic capacitance from the
bias winding due to the leakage inductance. Despite the 1 ms
blanking delay at turn off. This spike is energetic enough to
charge the added capacitor C1 and given the time constant,
could make it discharge slower, potentially disturbing the
blanking circuit. When implementing the Zener option, it is
important to carefully observe the OPP pin voltage (short
Over Temperature Protection
In a lot of designs, the adapter must be protected against
thermal runaways, e.g. when the temperature inside the
adapter box increases above a certain value. Figure 50
shows how to implement a simple OTP using an external
NTC and a series diode. The principle remains the same:
make sure the OPP network is not affected by the additional
NTC hence the presence of this isolation diode. When the
NTC resistance decreases as the temperature increases, the
voltage on the OPP pin during the off time will slowly
increase and, once it passes 3 V for 4 consecutive clock
cycles, the controller will permanently latch off.
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21
NCP1251
NT C
D2
1N4148
ROPPU
841k
VCC
au x.
winding
OP P
ROPPL
2.5k
full−latch
OPP
Vlatch
Figure 50. The Internal Circuitry Hooked to Pin 3 Can Be Used to Implement Over Temperature Protection (OTP)
Back to our 19 V adapter, we have found that the plateau
voltage on the auxiliary diode was 13 V in nominal
conditions. We have selected an NTC which offers a
resistance of 470 kW at 25°C and drops to 8.8 kW at 110°C.
If our auxiliary winding plateau is 14 V and we consider a
0.6 V forward drop for the diode, then the voltage across the
NTC in fault mode must be:
V NTC + 14 * 3 * 0.6 + 10.4 V
limit at the chosen output power level. Suppose we need a
200 mV decrease from the 0.8 V set point and the on−time
swing on the auxiliary anode is −67.5 V, then we need to drop
over ROPPU a voltage of:
V ROPPU + 67.5 * 0.2 + 67.3 V
The current flowing in the pulldown resistor ROPPL in this
condition will be:
(eq. 17)
Based on the 8.8 kW NTC resistor at 110 °C, the current
through the device must be:
I NTC +
10.4
8.8k
[ 1.2 mA
I ROPPU +
3
+ 2.5 kW
1.2m
200m
2.5k
+ 80 mA
(eq. 21)
The ROPPU value is therefore easily derived:
(eq. 18)
R OPPU +
As such, the bottom resistor ROPPL, can easily be
calculated:
R OPPL +
(eq. 20)
67.3
+ 841 kW
80m
(eq. 22)
Combining OVP and OTP
The OTP and Zener−based OVP can be combined
together as illustrated by Figure 51.
(eq. 19)
Now that the pulldown OPP resistor is known, we can
calculate the upper resistor value ROPPU to adjust the power
www.onsemi.com
22
NCP1251
D3
15V
D2
1N4148
NT C
11
ROPPU
841k
VCC
8
9
au x.
winding
OPP
10
4
1
ROPPL
2.5k
5
OVP
OPP
Vlatch
Figure 51. With the NTC Back in Place, the Circuit Nicely Combines OVP, OTP and OPP on the Same Pin
network. Typical values are those given in Figure 52 and
must be selected to provide the adequate filtering function
without degrading the stand−by power by an excessive
current circulation.
In nominal VCC / output conditions, when the Zener is not
activated, the NTC can drive the OPP pin and trigger the
adapter in case of an over temperature. During nominal
temperature if the loop is broken, the voltage runaway will
be detected and the controller will shut down the converter.
In case the OPP pin is not used for either OPP or OVP, it
can simply be grounded.
Latched OVP on VCC
The VCC pin is permanently monitored by a comparator.
When the VCC exceeds 25.5 V (typical), all pulses are
immediately stopped and the VCC falls to the SCR
latched-level around 7 V typical. The controller remains in
this state as long as a sufficient current flows in the SCR, at
least 30 mA. We recommend to put a design margin there,
with a minimum current around 60 mA at the lowest input
line. With the C version, the OVP on VCC is autorecovery.
Filtering the Spikes
The auxiliary winding is the seat of spikes that can couple
to the OPP pin via the parasitic capacitances exhibited by the
Zener diode and the series diode. To prevent an adverse
triggering of the Over Voltage Protection circuitry, it is
possible to install a small RC filter before the detection
D3
15V
ad d ition al fil ter
D2
1N4148
NT C
2
11
C1
330pF
ROPPU
841k
VCC
R3
220
9
3
aux.
winding
OP P
10
4
1
ROPPL
2.5k
5
Vlatch
OVP
OPP
Figure 52. A Small RC Filter Avoids the Fast Rising Spikes from Reaching the Protection Pin of the NCP1251 in
Presence of Energetic Perturbations Superimposed on the Input Line
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23
NCP1251
PACKAGE DIMENSIONS
TSOP−6
CASE 318G−02
ISSUE V
D
ÉÉ
ÉÉ
6
E1
1
5
2
H
4
L2
GAUGE
PLANE
E
3
NOTE 5
L
M
b
C
NOTES:
1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. MAXIMUM LEAD THICKNESS INCLUDES LEAD FINISH. MINIMUM
LEAD THICKNESS IS THE MINIMUM THICKNESS OF BASE MATERIAL.
4. DIMENSIONS D AND E1 DO NOT INCLUDE MOLD FLASH,
PROTRUSIONS, OR GATE BURRS. MOLD FLASH, PROTRUSIONS, OR
GATE BURRS SHALL NOT EXCEED 0.15 PER SIDE. DIMENSIONS D
AND E1 ARE DETERMINED AT DATUM H.
5. PIN ONE INDICATOR MUST BE LOCATED IN THE INDICATED ZONE.
SEATING
PLANE
DETAIL Z
e
A
0.05
c
A1
DETAIL Z
DIM
A
A1
b
c
D
E
E1
e
L
L2
M
MIN
0.90
0.01
0.25
0.10
2.90
2.50
1.30
0.85
0.20
0°
MILLIMETERS
NOM
MAX
1.00
1.10
0.06
0.10
0.38
0.50
0.18
0.26
3.00
3.10
2.75
3.00
1.50
1.70
0.95
1.05
0.40
0.60
0.25 BSC
10°
−
STYLE 13:
PIN 1. GATE 1
2. SOURCE 2
3. GATE 2
4. DRAIN 2
5. SOURCE 1
6. DRAIN 1
RECOMMENDED
SOLDERING FOOTPRINT*
6X
0.60
6X
3.20
0.95
0.95
PITCH
DIMENSIONS: MILLIMETERS
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and the
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NCP1251/D