On the Loading of Power Modules in a Three Phase Voltage Source

IEEE Industry Applications Society
Annual Meeting
New Orleans, Louisiana, October 5-9, 1997
On the Loading of Power Modules in a Three Phase Voltage Source Converter
Vladimir Blasko and Richard Lukaszewski
Standard Drives Development
Rockwell Automation - Allen Bradley
6400 W. Enterprise Drive
Mequon, WI 53092, USA
Phone (414) 242 8294, e-mail: [email protected]
Phone (414) 242 7155, e-mail: [email protected]
Fax (414) 242 8300 for both authors
ABSTRACT -- The loading of free wheel diodes (FWD) and
IGBTs in the power modules of a Voltage Source Converter
(VSC) are investigated. In converter duty, the FWD is
utilized more then the IGBT. Presently, most FWD’s in
power modules are optimized for inverter duty. This
practice tends to undersize the FWD from a converter
viewpoint. This design philosophy is examined in the paper.
For this purpose, the original dynamic thermal model of
IGBT and FWD were developed. Model parameters were
identified from the manufacturers catalogue data. A
mathematical model of the entire VSC was developed to
compute current, voltage, power and thermal conditions for
the power module and other components in converter
operation.
I.
INTRODUCTION
The power module of Voltage Source Inverter (VSI) in an
electric drive consists an IGBT with an anti-parallel free
wheel diode (FWD). The module is highly optimized for
inverter duty applications with an assumption that drive
operates from moderately low to higher speed ranges of the
motor and thus with a modulation index greater than 0.5. In
such a duty, dominant loading is on the IGBT. The loading of
FWD is not critical. To optimize the price/performance ratio,
power module manufactures increase the forward voltage drop
on the FWD and it is typically in the range of 2.0 to 2.5 volts.
Furthermore, the thermal resistance, junction - case, of the
FWD is usually double in comparison to the thermal resistance
of the IGBT. This occurs since the ratio of the FWD to the
IGBT die size varies from about 0.4 to 0.6 depending on power
module manufacturer.
Such an “optimization” created
problems and failures of FWD’s at stalled rotor or at very low
motor speeds. These operating conditions represent a small
segment of inverter applications and hence a small segment the
power module market. This provides sound economical
justification to optimize the power module in this manner.
On the other hand, the situation is totally opposite with three
phase Voltage Source Converters (VSC), often referred in
literature as a three phase synchronous rectifiers. VSC’s are
used as the "front end" of the drives or as a part of an
integrated drive (consisting of a VSC and a VSI in the same
package sharing common DC capacitor bank). The VSC in Fig.
1, is able to maintain constant DC bus voltage, sinusoidal input
current and unity power factor [4]. Its control is similar to the
vector control of the induction motor with two main
distinctions: (a) speed feedback is replaced with output dc
voltage feedback udc and (b) instead of position of rotor flux,
the position of utility voltage space vector θ is used as the
reference angle in all the blocks for reference frames
transformations. Output of a voltage regulator U_reg creates
active q current component reference iq _ c for the synchronous
reference current regulator Iq_reg.
The another current
reference id _c is used for the control of reactive current. It is
set to zero to maintain unity power factor. The outputs of digital
synchronous reference frame current regulators Iq_reg and
Load
u dc_c
u dc
U_reg
{
i
iq _ c
Iq_reg
id _ c
iq
Power
unit
Compensation
Transformations
PWM
Id_reg
id
ia
ib
ic
L
L
L
{
3s / 2e
Transform.
sinθ
DSP Implementation
dc
cosθ
Reference
Angle
Generator
{
ua
uc
ua
ub
uc
Figure 1 VSC Block Diagram
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Id_reg are further processed in Transformation - Compensation
- PWM block to create gating pulses for the IGBT bridge in the
power unit.
In VSC applications, the direction of energy flow through
power devices is opposite to the direction of energy flow in the
VSI. Since the VSC operates mostly in motoring mode, the
majority of current loading in the power module is on the
FWD’s which supply energy to the inverter. The IGBT devices
are fully loaded only for short time during regenerating thus
having a very small duty cycle.
This paper examines the current “optimization” practice of
power module designs and points out the differences in
loading of
power modules in converter and inverter
applications. It provides the original dynamic thermal model
of IGBT and FWD and a method to identify the model
parameters from manufacture’s data. The mathematical model
of the entire VSC was developed which enabled the
computation of instantaneous values of current, voltages and
power losses during every PWM switching period. The
temperature variations, within the resolution of PWM carrier
period, of the FWD and IGBT junctions were computed as a
function of load. The influence of thermal variations on life
expectancy of the devices was analyzed. In the addition to the
power module, the loading on the other power circuit
components namely capacitor bank in DC link and fuse are
analyzed.
II. DYNAMIC THERMAL MODEL OF POWER DEVICE
The thermal impedance,
junction - case, model of
semiconductor devices was developed in [1] to [3] and is
shown in Fig. 2(a). Using partial fraction expansion, (the
ratio of two polynomials) and the impedance equation from the
schematic diagram in Fig. 2(a), the equivalent diagram in Fig.
2(b) was obtained :
R'1
R' 2
•
R' n
•
C' n
C' 2
C' 1
Z th(jc)
•
C1
•
•
C2
•
where τ i = Ri C i , i = 1,... n .
The total steady state thermal resistance, junction -case
Rth ( j −c ) equals to the sum of the resistors R1 to Rn i.e.:
n
Rth ( j −c ) = ∑ Ri
•
Cn
(2)
i =1
The manufacturer’s usually define transient thermal impedance
as a function of time. Their definition corresponds to the
temperature change on the junction when step of "unity" power
dissipation is applied. In our case, it would correspond to the
multiplication of (1), with Heviside’s unity step function and
then transforming it in the time domain by Laplace
transformation. Thus the time domain thermal impedance,
normalized to the thermal resistance Rth ( j −c ) is obtained:
z th ( j −c ) (t ) = r1 (1 + e
t /τ1
) + r2 (1 + e
t /τ 2
)+...+rn (1 + e
t /τ n
)
(3)
where
ri = Ri / Rth ( j −c ) ; i = 1,... n
(4)
The parameters ri , τ i ; i = 1,... n in (3) were identified using
LMS algorithm with the target function
m
F = F p + ∑ [z th ( j −c ) ( t k ) − z th ( j − c ) (t k )]
md
2
(5)
k =1
where Fp is a quadratic penalty function which keeps the
parameters within specified limits, m is a number of points of
approximation, zth ( j −c ) ( tk ) is the thermal impedance calculated
md
zth ( j −c ) ( tk ) is the thermal impedance from
1
•
Zth(j-c) [pu]
•
Rn
R2
(1)
manufacturer data, both at the instant tk . It was found
(a)
R1
R2
Rn
+
+...+
1 + sτ 1 1 + sτ 2
1 + sτ n
from (3) and
•
R1
Z th ( j −c ) ( s ) =
Z th(jc)
0.5
(b)
Manufacturers Data
0
Figure 2 Equivalent diagram of thermal impedance, j-c, of a
semiconductor device "transmission line" (a) and series
connection of RC elements (b)
0
0.2
0.4
t [s] 0 . 6
0.8
1
Figure 3 Manufacturers thermal impedance data, (+), with the
approximation from eq (3)
that the approximation of the thermal impedance with equation
(3), by using three terms, n=3, gives an acceptable fit. The
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Table 1 Parameters of eq (4) for the thermal impedance of the
FWD and IGBT of a 600 A, 1200 V power module, time
constants are in seconds
r3
τ1
τ2
τ3
Diode
0.2491 0.5315
0.2194
0.0024
0.0797
0.9901
IGBT
0.2629 0.3892
0.3479
0.0044
0.0736
1.1873
r1
r2
*
methods add zero sequence component uzs to the
*
abc
modulation waveforms u
original
∈{u , u , u } , [6]:
*
a
*
b
*
c
uabc = uabc + uzs ,
**
*
*
(7)
uzs = −[(1 − 2 k 0 ) + umax + (1 − k 0 )umin ]
*
quality of approximation is illustrated in Fig. 3 for a FWD in a
600 Amp, 1200 Volt IGBT power module. Table 1 lists the
parameters for the FWD and IGBT device from the same
1200V, 600A module. Note that the shortest time constant τ1
in Table 1 is smaller then 5 ms and the corresponding thermal
resistance is about 25% of the total steady state resistance.
This indicates that the junction temperature will follow
variations in power dissipation at a 50/60 Hz cycle rate.
III. LOADING OF A DIODE VS. LOADING OF IGBT
where
*
*
(8)
uabc ∈{ua , ub , uc } is a new set of modulation
**
**
**
**
waveforms at the input of PWM, umax = max{ua , ub , uc } and
*
*
*
*
umin = min{ua , ub , uc } . Factor 0 ≤ k 0 ≤ 1 is a ratio of duration
of application of zero state vector V7 and combined duration of
applications of zero state vectors V7 and V0 within a carrier
period, [6].
*
The addition of uzs to the set of modulation reference
*
*
*
*
*
Figure 4 shows the schematic diagram of a VSC power
circuit. The VSC maintains constant bus voltage during
motoring and regenerating and regulates Udc to 10 to 15%
higher [4] in value then the peak of line to line voltage (i.e.
the output value of the FWD bridge U db ) i.e.
U dc = (11
. to 15
. )U db ; U db ≈ 2 U ll ,
idc
ua
0u
Da
uc
F
C
L , R ib
ub
udc /2
0c
L, R ic
Tra
1 c
u zs dt
Tc ∫0
(9)
for the non-saturated PWM modulator, (9) is proportional to
*
uzs :
The analysis of duty cycle and power conditions in power
circuit of a VSC was done in [5] with the assumption that the
neutral Ou of utility and center point of the capacitor bank Oc
are at the same potential. This assumption is true only on a
locally average basis over carrier period for symmetrical
sinusoidal triangle comparison PWM. However, frequently
used space vector, third harmonic and discontinuous PWM
Tra
T
uzs =
(6)
where U ll is line to line voltage.
L, R i
a
voltages uabc results in a zero sequence voltage uzs between
points Ou and Oc . Locally averaged value of zero sequence
voltage uzs over carrier period Tc
C
Da
udc /2
uzs = K PWM u zs
*
where K PWM is the gain of the PWM modulator.
Using (7) to (9) the analysis of current and voltage
conditions in power circuit of a VSC from [5] can be expanded
from sinusoidal symmetrical PWM to the other PWM methods
which introduce zero sequence in order to increase linearity
and utilization of dc bus voltage.
The method to determine the duty cycle and loading of the
IGBT and the FWD with zero sequence voltage present will be
illustrated with the phase a VSC input shown in Fig. 4.
During motoring operation of the VSC, the energy will flow
from the utility towards the dc bank, and during positive half
cycle of ua , the IGBT device Tra , conducts applying the
voltage to the inductance L:
uL (Tra ) = 05
. u dc + ua + uzs
u zs
(10)
(11)
where a bar over a later u indicates locally average value.
Figure 4 VSC Power Circuit Diagram
After Tra is turned off, current commutes to the Da and energy
is transferred from the line reactors to the dc bus capacitors,
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i.e. fly - back operation. During conduction of Da the voltage
on inductance L is substantially smaller then during conduction
of the transistor:
The quasi instantaneous loses on IGBT p IGBT ,and FWD
p D , were determined as
uL ( Da ) = −05
. u dc + ua + uzs
p IGBT = iIGBT uce ( sat ) (iIGBT ) +
(12)
The duty cycles for the diode d D and IGBT d IGBT follow from
the required equilibrium of volt seconds when voltages (11)
and (12) are applied on L:
dD =
05
. u dc − ( ua + uzs )
udc
FWD during conduction. ESW are switching energy losses of
the IGBT and Err are energy recovery loses of the FWD, they
Note that of the output of averager is obtained with a hold
delay of Th . The averager was used for evaluating the highly
discontinuous variables like current through power devices
over a carrier period or fundamental period.
1 /T h
g a in 1
1 /s
In te gra to r
1
o u tpu t
Zoh, Th
Figure 5 Block Diagram of Signal Averager
and uce(sat) are current and voltage of the IGBT
(14)
The mathematical model of the complete VSC (power and
control section [7]) was built, for more precise evaluation of
loading of power devices and variations of temperature within
a period of fundamental cycle (50/60 Hz). Operation with
constant PWM frequency was assumed. Figure 5 shows a
Matlab model of the ‘local averager’ which enables
calculation of average value of signal over “hold” time Th.
in p u t
(15)
during conduction, iD and uak are current and voltage of the
IV. MATHEMATICAL MODEL OF A VSC
+
Sum
(14)
(13)
Note that during motoring operation of VSC, d D is much
higher then d IGBT and consequently, the period of conduction
of the FWD’s is substantially longer then the period of
conduction of the IGBT devices. Therefore, FWD’s take more
current during motoring then the IGBT. The situation is
opposite during regeneration when the bulk of the load current
is taken by the IGBT devices.
However, motoring is
predominant mode of operation of the drive and current loading
of diodes during motoring becomes critical.
1
p D = iD uak ( iD ) + E rr ( iD ) f PWM
where iIGBT
05
. udc + ua + u zs
u dc
d IGBT =
+ [ ESW ( on ) ( iIGBT ) + ESW ( off ) (iIGBT )] f PWM
were averaged over a carrier period,
Tc=1/fPWM , within
which the switching happened. Where, fPWM , is the PWM
switching frequency. Note that in (14) and (15) voltage and
energy losses are similar to those developed in [8] except that
they are expressed as functions of instantaneous value of
current. Their dependence on current was implemented by
look - up tables.
V. CONDITIONS IN A VSC POWER STRUCTURE
Due to the tight packaging and requirements on small stray
inductance’s, it is hard and often impossible to measure
currents through component in power structures of converters
or inverters. Very often, simulation is the only tool for the
analysis of current and power conditions in these mechanically
tightly packaged power structures. Figure 6 shows the results
of simulation; bus voltage, phase voltage current and locally
averaged currents (over PWM period) of the IGBT and FWD
in a 125 HP VSC during motoring (0 to 0.05 seconds) and
regenerating (0.05 to 0.10 seconds). The FWD and IGBT
currents in Fig. 6 were obtained at the output of averager with
Th equal to a PWM period. Note that almost all the line current
goes through FWD during motoring and loading of the IGBT is
about 10 to 15%, as expected from by (13) and (14). During
regenerating, the situation is opposite, almost all the current
goes through IGBT with small loading on the FWD.
Evidently, a power module optimized for VSC applications
should have increased current carrying capabilities for the
FWD and smaller current capabilities for the IGBT.
Therefore, the optimization practice of power module for an
inverter, when used for VSC type of applications, should be
reexamined. However, if the VSC power structure is employed
only as a regenerative brake to return energy to the utility, then
the power module designed for inverter or drive type
applications is optimal since the regenerative brake is typically
sized for only a percentage of the drive rating.
The primary function of fuse F in Fig. 4 is to protect rupture
of the power module in the case of failure by limiting the
energy from the capacitor bank.
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1
udc
Motoring
Regenerating
0.5
u dc , ua , ia
[pu]
ua
ia
0
-0.5
0
0.02
0.04
0.06
t [s]
0.08
0.1
(a)
1
Regenerating
0.5
iD
iD
, iI G B T ,
[pu]
icap(rms)
Motoring
ic a p ( r m s )
0
iIGBT
-0.5
0
0.02
0.04
0.06
t [s]
0.08
0.1
(b)
Figure 6 Loading of devices in power circuit of a VSC during
motoring and regenerating; (a) dc bus voltage udc, phase a voltage
ua and current ia and (b) currents locally averaged over carrier
period of: IGBT iIGBT , diode iD and RMS capacitor current
icap ( rms) . Base value of voltage and current are 903V and 660A
respectively.
current path is opened by the failed fuse. The nature of the
VSC topology prohibits the traditional free wheel diode
connected anti-parallel to the fuse as in the normal inverter
configuration. The fuse selection is particularly troublesome
due to the fatigue in dynamic applications with frequent
overloads which are typical for VSC applications.
To reduce current through fuse, the fuse should be connected
in series with capacitor as shown with solid line in Fig. 4. In
this case, the fuse is loaded only with capacitor current. The
position of the fuse shown with doted line in Fig. 4 is inferior
because the fuse is burdened with load and capacitor current
which makes selection of fuse with small I2t difficult. The
wave form of the RMS capacitor current calculated over the
carrier period is shown in Fig. 6(b). The RMS capacitor
current is relatively small, only about 25% of dc load current.
The proposed dc bus fuse location implies that the upstream ac
mains fuses to the VSC be appropriately sized and specified.
They should be fast acting power semiconductor type fuses.
Figure 7(a) shows the dissipated power of the FWD and
IGBT devices in the VSC power circuit during motoring
operation. Values averaged over periods of carrier frequency
and period of the fundamental are shown. Note that values
averaged over period of fundamental (marked with a bar above
symbol) are shifted right - delayed for sample and hold period,
Th which is equal to the fundamental period. However, this
delay is not noticeable on waveforms obtained when averaging
over short carrier period, Th = Tc , because of the relatively
high 5KHz PWM carrier frequency. The discontinuous PWM
method from [4] which stops switching of IGBT for
approximately 60 degrees was used. The discontinuity when
switching is stopped is visible in losses on IGBT device pIGBT.
Figure 7(b) shows FWD and IGBT junction temperatures.
Note that pulsation of FWD junction temperature Tjc_D are about
8 oC. The frequency of the pulsation’s is equal to the frequency
of utility system. According to the manufacturing data,
temperature variations smaller then 30 oC do not fatigue
devices within the power module. Temperatures T jc _ D and
T jc _ IGBT were computed using steady state thermal resistances
junction - case and locally averaged dissipation over
fundamental period. They are approximate steady state junction
temperatures. The temperature increase, junction – case, on the
FWD at rated load, is only 19 oC. This confirms that in this
particular case, the inverter duty FWD will operate reliably in
the converter duty application.
There are two main but opposing requirements governing
selecting of this fuse. The fuse should have current carrying
capability large enough to handle the rated load and any
overload current requirements and on the other hand, small
enough I2t to avoid rupture of the failed IGBT module.
“Premature” failure of the undersized fuse causes voltage break
down and failure of power module. This occurs if the fuse
opens prematurely and trapped inductive energy in the load and
parasitics manifests itself as a high transient voltage since the
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800
700
600
pD
IGBT and FWD losses [W]
500
p IGBT
400
300
200
pD
100
0
-100
p IGBT
0
0.02
0.06
0.04
t [s]
(a)
0.08
0.1
applications and de-rating should be done to achieve
reliable operation. For proper sizing of the devices, a
precise thermal impedance model was developed. A
third order thermal impedance model provided a good
match with manufacturer’s data. The model of the
complete converter, together with a model of transient
thermal impedance was developed and used for analysis
of current, voltage, power and thermal condition in a
VSC power structure. The variations of junction
temperature on the IGBT and FWD vary from 2 to 8 oC
and repeat at the utility frequency of 50/60 Hz at rated
load conditions. In the particular case analyzed, a FWD
from a 600A, 1200V IGBT power module optimized for
inverter - drive duty application, proved to be able to
operate satisfactory in VSC duty type of applications.
Finally, a preferred location of the dc bus fuse has been
defined. It is in series with the dc bus capacitors and not
in the traditional inverter location, i.e., in series with
the dc bus load.
REFERENCES
20
T jc_D , T jc_IGBT
[deg. C]
[1] F. W. Gutzwiller and T. D. Sylivan, 'Power
Semiconductor Ratings Under Transient and Intermittent
18
Loads', General Electric, Application Note 200.9, 6/61.
[2] S. K. Ghandhi,’Semiconductor Power Devices
16
Physics of Operation and Fabrication Technology’, John
T jc_D
Wiley & Sons, pp. 301 - 306, New York, London,
14
T jc_D
T jc_IGBT
Sidney, Toronto, 1977.
12
[3] G. L. Skibinski and W. A. Sethares, 'Thermal
Parameter Estimation Using Recursive Identification',
10
IEEE IAS Annual Conference Record 1990, pp. 15811588.
8
[4] V. Kaura and V. Blasko, “Operation of a Voltage
Source Converter at Increased Utility Voltage,” IEEE
6
Trans. On Power Electronics., vol. 12, No. 1, pp. 1324
137, January 1997.
T jc_IGBT
[5] B. T. Ooi, J. C. Salmon, J. W. Dixon and A. B.
2
Kulkarni, ‘A 3-Phase Controlled Current PWM
Converter with Leading Power Factor’, IEEE Ind. App.
0
0
0.02
0.04
0.06
0.08
0.1 Soc. Conf. Record, 1985, pp. 1008 - 1014.
[6] V. Blasko, ’A Hybrid PWM Strategy Combining
t [s]
(b)
Modified Space Vector and Triangle Comparison
Methods’, in IEEE PESC96 Conference Record, Volume
Figure 7 (a) dissipation on diode p D and IGBT p IGBT averaged over II, pp. 1872 -1878, June 23-27, 1996, Baveno, Italy.
[7] V. Blasko and V. Kaura, ‘A New Mathematical
carrier period and dissipation on the same components p D and Model and Control of a Three-Phase AC-DC Voltage
p IGBT averaged over fundamental period and (b) temperatures junction - Source Converter’, IEEE Transactions of Power
Electronics, Vol. 12, No. 1, pp. 116-123, January1997.
case on diode and IGBT
[8] IGBTMOD and Intellimod -Intelligent Power
Modules, Applications and Technical Data Book, First
VI. CONCLUSION
Edition, October 1994, Powerex.
Power modules in converters applications have opposite
modes of operation than modules in inverter applications, i.e.,
the FWD’s are more heavily loaded than the IGBT devices.
Since the standard power modules are optimized for inverter
applications, the FWD’s may be overloaded for converter
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