MIC26400

MIC26400
5A Hyper Speed ControlTM
Synchronous DC/DC Buck Regulator
SuperSwitcher IITM
General Description
Features
The Micrel MIC26400 is a constant-frequency, synchronous
buck regulator featuring a unique digitally modified adaptive
ON-time control architecture. The MIC26400 operates over
an input supply range of 4.5V to 26V and provides a
regulated output at up to 5A of output current. The output
voltage is adjustable down to 0.8V with a typical accuracy of
±1%, and the device operates at a switching frequency of
300kHz.
Micrel’s Hyper Speed ControlTM architecture allows for ultrafast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This digitally modified adaptive tON ripple control architecture
combines the advantages of fixed frequency operation and
fast transient response in a single device.
The MIC26400 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup” mode shortcircuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
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Hyper Speed ControlTM architecture enables
- High delta V operation (VIN = 26V and VOUT = 0.8V)
- Small output capacitance
4.5V to 26V input voltage
Adjustable output from 0.8V to 5.5V (±1% accuracy)
Any CapacitorTM Stable
- Zero ESR to high-ESR output capacitance
5A output current capability
300kHz switching frequency
Internal compensation
Up to 95% efficiency
6ms Internal soft-start
Foldback current limit and “hiccup” mode short-circuit
protection
Thermal shutdown
Supports safe start-up into a pre-biased load
–40°C to +125°C junction temperature range
28-pin 5mm × 6mm MLF® package
Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
____________________________________________________________________________________________________________
Typical Application
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
July 2011
M9999-070111-C
Micrel, Inc.
MIC26400
Ordering Information
Part Number
Voltage
Switching Frequency
Junction Temperature
Range
Package
Lead Finish
MIC26400YJL
Adjustable
300kHz
−40°C to +125°C
28-Pin 5mm × 6mm MLF®
Pb-Free
Pin Configuration
28-Pin 5mm X 6mm MLF® (YJL)
Pin Description
Pin Number
13, 14, 15,
16, 17, 18, 19
Pin Name
PVIN
Pin Function
High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 26V. Input capacitors between the PVIN pins and the power ground (PGND) are required.
Note that the connection must be kept short.
24
EN
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA).
25
FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
26
SGND
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin
to the PGND Pad on the top layer. (see PCB Layout Guidelines for details.)
27
VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC26400. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin-to-PGND
must be placed next to the IC.
PGND
Power Ground. PGND is the ground path for the MIC26400 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of
input capacitors, and the negative terminals of output capacitors. The loop for the power ground
should be as small as possible and separate from the Signal ground (SGND) loop.
2, 5, 6, 7, 8, 21
22
CS
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high speed switching on this pin, the CS pin should be routed away from sensitive
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.
20
BST
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected
between the BST pin and the SW pin.
4, 9, 10, 11, 12
SW
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain.
23
VIN
Power Supply Voltage (Input): Requires bypass capacitor to SGND.
1, 3, 28
NC
No Connect.
July 2011
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MIC26400
Absolute Maximum Ratings(1, 2)
Operating Ratings(3)
PVIN to PGND................................................ −0.3V to +28V
VIN to PGND ....................................................−0.3V to PVIN
VDD to PGND ................................................... −0.3V to +6V
VSW, VCS to PGND .............................. −0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 34V
VEN to PGND ...................................... −0.3V to (VDD + 0.3V)
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Supply Voltage (PVIN, VIN)................................. 4.5V to 26V
Output Voltage (VOUT) ...................................... 0.8V to 5.5V
Bias Voltage (VDD)............................................ 4.5V to 5.5V
Enable Input (VEN) ................................................. 0V to VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF® (θJA) ....................................36°C/W
Electrical Characteristics(5)
PVIN = VIN =12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
26
V
V
Power Supply Input
Input Voltage Range (VIN, PVIN)
4.5
VDD Bias Voltage
Operating Bias Voltage (VDD)
Under-Voltage Lockout Trip Level
VDD Rising
4.5
5
5.5
2.4
2.7
3.2
UVLO Hysteresis
50
Quiescent Supply Current
Shutdown Supply Current
VFB = 1.5V
VDD = VBST = 5.5V, VIN = 26V
SW = unconnected, VEN = 0V
V
mV
1.4
3
mA
0.7
2
mA
Reference
Feedback Reference Voltage
0°C ≤ TJ ≤ 85°C (±1.0%)
0.792
0.8
0.808
−40°C ≤ TJ ≤ 125°C (±1.5%)
0.788
0.8
0.812
V
Load Regulation
IOUT = 0A to 5A
0.2
%
Line Regulation
VIN = (VOUT + 3.0V) to 26V
0.1
%
FB Bias Current
VFB = 0.8V
5
nA
0.85
V
Enable Control
EN Logic Level High
4.5V < VDD < 5.5V
EN Logic Level Low
4.5V < VDD < 5.5V
EN Bias Current
VEN = 0V
1.2
0.78
0.4
50
V
µA
Oscillator
Switching Frequency
Maximum duty cycle
Minimum duty cycle
(6)
(7)
225
VFB = 0V
300
87
VFB > 0.8V
Minimum Off-time
375
kHz
%
0
%
360
ns
6
ms
Soft-Start
Soft-Start time
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MIC26400
Electrical Characteristics(5) (Continued)
PVIN = VIN =12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Current-Limit Threshold
VFB = 0.8V
6
13
A
Short Circuit Current
VFB = 0V
6
A
Short Circuit Protection
Internal FETs
Top-MOSFET RDS (ON)
ISW = 1A
43
mΩ
Bottom-MOSFET RDS (ON)
ISW = 1A
12.5
mΩ
SW Leakage Current
VIN = 26V, VSW = 26V, VEN = 0V, VBST = 31.5 V
60
µA
VIN Leakage Current
VIN = 26V, VSW = 0V, VEN = 0V, VBST = 31.5V
25
µA
Thermal Protection
Over-temperature Shutdown
TJ Rising
Over-temperature Shutdown
Hysteresis
155
°C
10
°C
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
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MIC26400
Typical Characteristics
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
8.0
6.0
4.0
VOUT = 1.2V
IOUT = 0A
2.0
VDD = 5V
SWITCHING
0.0
10
16
12
8
VDD = 5V
4
VEN = 0V
0
4
10
16
22
28
10
6
4
VOUT = 1.2V
2
VDD = 5V
SWITCHING
16
22
28
4
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
10
VOUT = 1.2V
VDD = 5V
20
0.8%
VDD = 5V
IOUT = 0A to 5A
0.6%
0.4%
0.2%
15
10
5
VOUT = 1.2V
VDD = 5V
IOUT = 0A
0.792
0.0%
4
10
16
22
28
VOUT = 1.2V
CURRENT LIMIT (A)
TOTAL REGULATION (%)
0.800
22
Current Limit
vs. Input Voltage
1.0%
0.804
16
INPUT VOLTAGE (V)
Total Regulation
vs. Input Voltage
0.808
0.796
8
0
4
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
SUPPLY CURRENT (mA)
20
SHUTDOWN CURRENT (µA)
SUPPLY CURRENT (mA)
10.0
VDD Operating Supply Current
vs. Input Voltage
28
0
4
10
16
22
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
28
4
10
16
22
28
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
SWITCHING FREQUENCY (kHz)
390
VOUT = 1.2V
VDD = 5V
345
IOUT = 0A
300
255
210
4
10
16
22
28
INPUT VOLTAGE (V)
July 2011
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M9999-070111-C
Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
VDD Operating Supply Current
vs. Temperature
VDD Shutdown Current
vs. Temperature
VDD = 5V
IOUT = 0A
SWITCHING
6.0
4.0
2.0
0.8
2.7
0.6
VIN = 12V
0.4
IOUT = 0A
VDD = 5V
0.2
VEN = 0V
-20
10
40
70
100
130
-50
VIN = 12V
2.4
-20
10
40
70
100
130
-50
-20
10
40
70
VIN Operating Supply Current
vs. Temperature
VIN Shutdown Current
vs. Temperature
Current Limit
vs. Temperature
IOUT = 0A
SWITCHING
6.0
4.0
2.0
10.0
25
8.0
20
CURRENT LIMIT (A)
SUPPLY CURRENT (µA)
VOUT = 1.2V
VDD = 5V
6.0
4.0
VIN = 12V
VDD = 5V
2.0
-20
10
40
70
100
130
-20
10
40
70
TEMPERATURE (°C)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
Load Regulation
vs. Temperature
100
130
IOUT = 0A
0.800
0.796
0.792
10
40
70
100
10
40
70
0.5%
VIN = 12V
0.8%
VOUT = 1.2V
VDD = 5V
IOUT = 0A to 5A
0.6%
0.4%
0.2%
VIN = 6V to 26V
0.4%
VOUT = 1.2V
VDD = 5V
0.3%
0.2%
0.1%
0.0%
-50
130
-20
Line Regulation
vs. Temperature
0.0%
-20
130
VIN = 12V
VOUT = 1.2V
VDD = 5V
TEMPERATURE (°C)
LINE REGULATION (%)
LOAD REGULATION (%)
VIN = 12V
VOUT = 1.2V
VDD = 5V
100
10
-50
1.0%
0.804
130
0
-50
0.808
100
15
5
IOUT = 0A
0.0
0.0
-20
10
40
70
100
130
TEMPERATURE (°C)
TEMPERATURE (°C)
-50
-20
10
40
70
100
130
TEMPERATURE (°C)
EN Bias Current
vs. Temperature
Switching Frequency
vs. Temperature
100
345
V IN = 12V
330
EN BIAS CURRENT (µA)
SWITCHING FREQUENCY (kHz)
2.5
TEMPERATURE (°C)
8.0
V OUT = 1.2V
V DD = 5V
315
IOUT = 0A
300
285
270
255
80
60
40
VIN = 12V
VOUT = 1.2V
20
VDD = 5V
0
-50
-20
10
40
70
TEMPERATURE (°C)
July 2011
Falling
TEMPERATURE (°C)
VIN = 12V
-50
2.6
TEMPERATURE (°C)
10.0
-50
Rising
2.3
0
-50
SUPPLY CURRENT (mA)
2.8
VDD THRESHOLD (V)
VOUT = 1.2V
8.0
0.0
FEEDBACK VOLTAGE (V)
1
VIN = 12V
SUPPLY CURRENT (mA)
SUPPLY CURRENT (mA)
10.0
VDD UVLO Threshold
vs. Temperature
100
130
-50
-20
10
40
70
100
130
TEMPERATURE (°C)
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Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
Efficiency
vs. Output Current
100
12VIN
85
24VIN
80
75
70
65
VOUT = 1.2V
60
VDD = 5V
55
OUTPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
0.804
0.800
VIN = 12V
0.796
VOUT = 1.2V
50
1
2
3
4
5
0
1
2
3
4
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current
Switching Frequency
vs. Output Current
0.5%
VDD = 5V
0.3%
0.2%
0.1%
0.0%
1
2
3
4
330
300
270
VIN = 12V
240
VDD = 5V
VOUT = 1.2V
V OUT = 1.2V
1
2
3
4
July 2011
4
4.4
4.2
TA
25ºC
85ºC
125ºC
4
3.8
0
5
1
2
3
4
6
7
Die Temperature* (VIN = 24V)
vs. Output Current
40
20
VIN = 12V
VOUT = 1.2V
40
20
V IN = 24V
V OUT = 1.2V
VDD = 5V
5
5
OUTPUT CURRENT (A)
V DD = 5V
0
3
VFB < 0.8V
VDD = 5V
60
V DD = 5V
0
5
3.4
0
DIE TEMPERATURE (°C)
V IN = 5V
4
VIN = 5V
4.6
Die Temperature* (VIN = 12V)
vs. Output Current
20
3
Output Voltage (VIN = 5V)
vs. Output Current
OUTPUT CURRENT (A)
40
2
3.6
210
60
2
1
OUTPUT CURRENT (A)
360
5
60
OUTPUT CURRENT (A)
VDD = 5V
5
Die Temperature* (VIN = 5V)
vs. Output Current
1
VIN = 12V
VOUT = 1.2V
1.192
4.8
OUTPUT CURRENT (A)
0
1.196
0
OUTPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
VOUT = 1.2V
0
1.2
5
390
VIN = 6V to 26V
0.4%
1.204
1.188
0.792
0
1.208
VDD = 5V
DIE TEMPERATURE (°C)
EFFICIENCY (%)
90
LINE REGULATION (%)
1.212
0.808
95
DIE TEMPERATURE (°C)
Output Voltage
vs. Output Current
Feedback Voltage
vs. Output Current
0
0
1
2
3
OUTPUT CURRENT (A)
7
4
5
0
1
2
3
4
5
OUTPUT CURRENT (A)
M9999-070111-C
Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
Efficiency (VIN = 5V)
vs. Output Current
100
95
EFFICIENCY (%)
3.3V
2.5V
1.8V
1.5V
90
1.2V
1.0V
0.9V
0.8V
85
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
90
85
80
70
0
1
2
3
4
5
6
7
1
4
5
6
0.8V
10
8
1.5V
6
4
VIN = 5V
VOUT = 0.8, 1.2, 1.5V
0
-25
0
25
50
75
100
6
3.3V
4
2
VIN = 5V
VOUT = 1.8, 2.5, 3.3V
-50
OUTPUT CURRENT (A)
8
5V
6
4
VIN = 12V
2
5
6
7
100
125
0.8V
10
8
1.8V
6
4
2
VIN = 12V
VOUT = 0.8, 1.2, 1.8V
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
AMBIENT TEMPERATURE (°C)
Thermal Derating*
vs. Ambient Temperature
12
2.5V
4
0
AMBIENT TEMPERATURE (°C)
12
3
12
8
125
Thermal Derating*
vs. Ambient Temperature
10
2
Thermal Derating*
vs. Ambient Temperature
14
1.8V
10
AMBIENT TEMPERATURE (°C)
14
1
OUTPUT CURRENT (A)
0
-50
0
7
12
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
3
Thermal Derating*
vs. Ambient Temperature
14
12
OUTPUT CURRENT (A)
2
OUTPUT CURRENT (A)
Thermal Derating*
vs. Ambient Temperature
2
1.0V
0.9V
0.8V
80
70
0
OUTPUT CURRENT (A)
14
85
75
75
80
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
90
OUTPUT CURRENT (A)
EFFICIENCY (%)
95
95
EFFICIENCY (%)
100
Efficiency (VIN = 24V)
vs. Output Current
Efficiency (VIN = 12V)
vs. Output Current
10
0.8V
8
2.5V
6
4
2
VIN = 24V
VOUT = 0.8, 1.2, 2.5V
VOUT = 2.5, 3.3, 5V
0
0
-50
-25
0
25
50
75
100
AMBIENT TEMPERATURE (°C)
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26400 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
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Micrel, Inc.
MIC26400
Functional Characteristics
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MIC26400
Functional Characteristics (Continued)
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Micrel, Inc.
MIC26400
Functional Characteristics (Continued)
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MIC26400
Functional Diagram
Figure 1. MIC26400 Block Diagram
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MIC26400
It is not recommended to use MIC26400 with a OFF-time
close to tOFF(min) during steady-state operation. Also, as
VOUT increases, the internal ripple injection will increase
and reduce the line regulation performance. Therefore,
the maximum output voltage of the MIC26400 should be
limited to 5.5V. Please refer to “Setting Output Voltage”
subsection in “Application Information” for more details.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 26V to 1.0V. The minimum tON
measured on the MIC26400 evaluation board is about
184ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC26400 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Functional Description
The MIC26400 is an adaptive ON-time synchronous
step-down DC/DC regulator. It is designed to operate
over a wide input voltage range from 4.5V to 26V and
provides a regulated output voltage at up to 5A of output
current. A digitally modified adaptive ON-time control
scheme is employed in to obtain a constant switching
frequency and to simplify the control compensation.
Over-current protection is implemented without the use
of an external sense resistor. The device includes an
internal soft-start function which reduces the power
supply input surge current at start-up by controlling the
output voltage rise time.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC26400. The output voltage is sensed by the
MIC26400 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
t ON(estimated) =
VOUT
VIN × 300kHz
(1)
Where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, the MIC26400 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET. The maximum duty cycle is obtained from the
360ns tOFF(min):
D max =
t S − t OFF(min)
tS
= 1−
360ns
tS
(2)
Figure 2. MIC26400 Control Loop Timing
where tS = 1/300kHz = 3.33μs.
July 2011
13
M9999-070111-C
Micrel, Inc.
MIC26400
Figure 3 shows the operation of the MIC26400 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26400 converter.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC26400 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC26400 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC26400 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the inductor current is
greater than 13A, then the MIC26400 turns off the highside MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The current-limit threshold has a foldback
characteristic related to the feedback voltage, as shown
in Figure 4.
Figure 3. MIC26400 Load Transient Response
Unlike true current-mode control, the MIC26400 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26400 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC26400 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in “Application Information” for more details
about the ripple injection technique.
July 2011
Current-Limit Thresold
vs. Feedback Voltage
CURRENT-LIMIT THRESHOLD (A)
20.0
16.0
12.0
8.0
4.0
0.0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC26400 Current Limit Foldback Characteristic
14
M9999-070111-C
Micrel, Inc.
MIC26400
MOSFET Gate Drive
The Block Diagram of Figure 1 shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reversed biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA x 3.33μs/0.1μF = 333mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
July 2011
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
15
M9999-070111-C
Micrel, Inc.
MIC26400
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by the equation below:
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below:
L=
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × 20% × IOUT(max)
2
PINDUCTOR(Cu) = IL(RMS) × RWINDING
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
(4)
PWINDING(Ht) =
RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Where:
fSW = switching frequency, 300kHz
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
ΔIL(pp) =
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × L
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
(5)
(6)
The RMS inductor current is used to calculate the I2R
losses in the inductor.
ESR COUT ≤
2
IL(RMS) = IOUT(max) +
ΔIL(PP)
12
2
ΔVOUT(pp)
(10)
ΔIL(PP)
(7)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC26400 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
July 2011
(9)
Where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
(8)
16
M9999-070111-C
Micrel, Inc.
MIC26400
The peak input current is equal to the peak inductor
current, so:
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated below:
2
ΔVOUT(pp)
ΔVIN = IL(pk) × ESRCIN
ΔIL(PP)
⎛
⎞
2
⎟ + ΔIL(PP) × ESR C
= ⎜⎜
OUT
⎟
×
×
C
f
8
⎝ OUT SW
⎠
(11)
(
)
(14)
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
Where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
ICIN(RMS) ≈ IOUT(max) × D × (1 − D)
(15)
The power dissipated in the input capacitor is:
As described in the “Theory of Operation” subsection in
“Functional Description”, the MIC26400 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated below:
ICOUT (RMS) =
ΔIL(PP)
PDISS(CIN) = ICIN(RMS)2 × ESRCIN
Ripple Injection
The VFB ripple required for proper operation of the
MIC26400 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
the MIC26400 will lose control and the output voltage is
not regulated. In order to have some amount of VFB
ripple, a ripple injection method is applied for low output
voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
As shown in Figure 5a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
(12)
12
The power dissipated in the output capacitor is:
2
PDISS(COUT ) = ICOUT (RMS) × ESR COUT
(13)
ΔVFB(pp) =
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR.
July 2011
(16)
R2
× ESR COUT × ΔIL (pp)
R1 + R2
(17)
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 5b. The typical Cff value is between 1nF and
100nF.
17
M9999-070111-C
Micrel, Inc.
MIC26400
With the feedforward capacitor, the feedback voltage
ripple is very close to the output voltage ripple:
ΔVFB(pp) ≈ ESR × ΔIL (pp)
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 5c. The injected ripple
is:
(18)
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
ΔVFB(pp) = VIN × K div × D × (1 - D) ×
K div =
R1//R2
R inj + R1//R2
1
fSW × τ
(19)
(20)
Where:
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
Figure 5a. Enough Ripple at FB
In equations (19) and (20), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
= << 1
fSW × τ τ
(21)
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using equation (22),
Figure 5b. Inadequate Ripple at FB
Figure 5c. Invisible Ripple at FB
K div =
July 2011
18
ΔVFB(pp)
VIN
×
fSW × τ
D × (1 − D)
(22)
M9999-070111-C
Micrel, Inc.
MIC26400
Then the value of Rinj is obtained as:
R inj = (R1//R2) × (
1
− 1)
K div
(23)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26400 requires two resistors to set the output
voltage as shown in Figure 6.
Figure 7. Internal Ripple Injection
Thermal Measurements
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation:
VOUT = VFB × (1 +
R1
)
R2
(24)
Where VFB = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
R2 =
VFB × R1
VOUT − VFB
(25)
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC26400, as shown
in Figure 7. The inverting input voltage VINJ is clamped to
1.2V. As VOUT is increased, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected back as a DC error on the FB
terminal. Therefore, the maximum output voltage of the
MIC26400 should be limited to 5.5V to avoid this line
regulation problem.
July 2011
19
M9999-070111-C
Micrel, Inc.
MIC26400
Inductor
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC26400 converter.
IC
•
The signal ground pin (SGND) must be connected
directly to the ground planes. Do not route the
SGND pin to the PGND Pad on the top layer.
•
Place the IC close to the point of load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
Place the input capacitor next.
•
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
Keep both the PVIN and PGND connections short.
•
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
•
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
July 2011
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the lowside MOSFET.
To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
RC Snubber
Input Capacitor
•
Keep the inductor connection to the switch node
(SW) short.
•
The 2.2µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
•
•
•
20
Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
M9999-070111-C
Micrel, Inc.
MIC26400
Evaluation Board Schematic
Figure 8. Schematic of MIC26400 Evaluation Board
(J13, R13, R15 are for testing purposes)
July 2011
21
M9999-070111-C
Micrel, Inc.
MIC26400
Bill of Materials
Item
C1
C2
C3
C4, C5, C13
Part Number
B41125A7227M
C8, C9
C11
C12
GRM32ER71H475KA88L
12106D107MAT2A
GRM32ER60J107ME20L
GRM188R71H104KA93D
GRM21BR71A225KA01L
R2, R16
R3, R4
R5
R6
R7
R8
July 2011
Murata(3)
Qty
220µF Aluminum Capacitor, SMD, 35V
1
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
1
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
3
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
3
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V
2
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
22nF Ceramic Capacitor, X7R, Size 0603, 50V
1
Small Signal Schottky Diode
1
5.6V Zener Diode
1
4.0µH Inductor, 12A Saturation Current
1
AVX(2)
Murata(3)
(4)
Murata(3)
C2012X7R1A225K
TDK
06035C102KAT2A
AVX(2)
GRM188R71H102KA01D
Murata(3)
C1608X7R1H102K
TDK
06035C223KAZ2A
AVX(2)
GRM188R71H223K
Murata(3)
Open
R1
AVX(2)
AVX(2)
C15
Q1
Murata(3)
0805ZC225MAT2A
Open
L1
AVX(2)
TDK
C14
D2
EPCOS
C1608X7R1H104K
C1608X7R1H223K
D1
Description
(1)
Open
12105C475KAZ2A
06035C104KAT2A
C6, C7, C10
Manufacturer
SD103AWS-7
SD103AWS
CMDZ5L6
HCF1305-4R0-R
FCX619
CRCW06034R75FKEA
CRCW08051R21FKEA
CRCW060310K0FKEA
CRCW060380K6FKEA
CRCW060340K2FKEA
CRCW060320K0FKEA
CRCW060311K5FKEA
(4)
TDK
Diodes Inc(6)
Vishay(7)
Central Semi(8)
(9)
Cooper Bussmann
(6)
ZETEX
50V NPN Transistor
1
(7)
4.75Ω Resistor, Size 0603, 1%
1
(7)
1.21Ω Resistor, Size 0805, 1%
2
(7)
10kΩ Resistor, Size 0603, 1%
2
(7)
80.6kΩ Resistor, Size 0603, 1%
1
(7)
40.2kΩ Resistor, Size 0603, 1%
1
(7)
20kΩ Resistor, Size 0603, 1%
1
(7)
11.5kΩ Resistor, Size 0603, 1%
1
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
22
M9999-070111-C
Micrel, Inc.
MIC26400
Bill of Materials (Continued)
Item
R9
Part Number
CRCW06038K06FKEA
Manufacturer
Description
Qty
(7)
8.06kΩ Resistor, Size 0603, 1%
1
(7)
Vishay Dale
R10
CRCW06034K75FKEA
Vishay Dale
4.75kΩ Resistor, Size 0603, 1%
1
R11
CRCW06033K24FKEA
Vishay Dale(7)
3.24kΩ Resistor, Size 0603, 1%
1
CRCW06031K91FKEA
(7)
1.91kΩ Resistor, Size 0603, 1%
1
(7)
0Ω Resistor, Size 0603, 5%
1
(7)
5.23kΩ Resistor, Size 0603, 1%
1
(7)
49.9Ω Resistor, Size 0603, 1%
1
26V/5A Synchronous Buck DC/DC Regulator
1
R12
R13
R14
R15
U1
CRCW06030000FKEA
CRCW06035K23FKEA
CRCW060349R9FKEA
MIC26400YJL
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Micrel. Inc.
(10)
Notes:
1. EPCOS: www.epcos.com.
2. AVX: www.avx.com.
3. Murata: www.murata.com.
4. TDK: www.tdk.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Central Semi: www.centralsemi.com.
9. Cooper Bussmann: www.cooperbussmann.com.
10. Micrel, Inc.: www.micrel.com.
July 2011
23
M9999-070111-C
Micrel, Inc.
MIC26400
PCB Layout
Figure 9. MIC26400 Evaluation Board Top Layer
Figure 10. MIC26400 Evaluation Board Mid-Layer 1 (Ground Plane)
July 2011
24
M9999-070111-C
Micrel, Inc.
MIC26400
PCB Layout (Continued)
Figure 11. MIC26400 Evaluation Board Mid-Layer 2
Figure 12. MIC26400 Evaluation Board Bottom Layer
July 2011
25
M9999-070111-C
Micrel, Inc.
MIC26400
Recommended Land Pattern
July 2011
26
M9999-070111-C
Micrel, Inc.
MIC26400
Package Information
28-Lead 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
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relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2010 Micrel, Incorporated.
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