MIC24053 12V, 9A High-Efficiency Buck Regulator SuperSwitcher II General Description Features The Micrel MIC24053 is a constant-frequency, synchronous buck regulator featuring a unique adaptive on-time control architecture. The MIC24053 operates over an input supply range of 4.5V to 19V and provides a regulated output of up to 9A of output current. The output voltage is adjustable down to 0.8V with a guaranteed accuracy of ±1%, and the device operates at a switching frequency of 600kHz. • Hyper Speed Control architecture enables - High Delta V operation (VIN = 19V and VOUT = 0.8V) - Small output capacitance • 4.5V to 19V voltage input • 9A output current capability, up to 95% efficiency • Adjustable output from 0.8V to 5.5V • ±1% feedback accuracy • Any Capacitor stable-zero-to-high ESR • 600kHz switching frequency • No external compensation • Power Good (PG) output • Foldback current-limit and “hiccup mode” short-circuit protection • Supports safe startup into a pre-biased load • –40°C to +125°C junction temperature range • Available in 28-pin 5mm × 6mm QFN package Micrel’s Hyper Speed Control architecture allows for ultrafast transient response while reducing the output capacitance and also makes (High VIN)/(Low VOUT) operation possible. This adaptive tON ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. The MIC24053 offers a full suite of features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, foldback current limit, “hiccup mode” short-circuit protection, and thermal shutdown. An open-drain Power Good (PG) pin is provided. ® The 9A Hyper Light Load part, MIC24054, is also available on Micrel’s web site. All support documentation is available on Micrel’s web site at: www.micrel.com. Applications • Servers, workstations • Routers, switches, and telecom equipment • Base stations ___________________________________________________________________________________________________________ Typical Application Efficiency (VIN = 12V) vs. Output Current 100 95 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V EFFICIENCY (%) 90 85 80 75 70 65 60 55 VIN = 12V 50 0 2 4 6 8 10 12 OUTPUT CURRENT (A) Hyper Speed Control, SuperSwitcher II, and Any Capacitor are trademarks of Micrel, Inc. Hyper Light Load is a registered trademark of Micrel, Inc. Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com November 2012 M9999-110712-A Micrel, Inc. MIC24053 Ordering Information Part Number Switching Frequency Voltage Package Junction Temperature Range Lead Finish MIC24053YJL 600kHz Adjustable 28-Pin 5mm × 6mm QFN −40°C to +125°C Pb-Free Pin Configuration 28-Pin 5mm × 6mm QFN (JL) (Top View) Pin Description Pin Number Pin Name 1 PVDD 5V Internal Linear Regulator output. PVDD supply is the power MOSFET gate drive supply voltage created by internal LDO from VIN. When VIN < +5.5V, PVDD should be tied to PVIN pins. A 2.2µF ceramic capacitor from the PVDD pin to PGND (pin 2) must be placed next to the IC. 2, 5, 6, 7, 8, 21 PGND Power Ground. PGND is the ground path for the MIC24053 buck converter power stage. The PGND pins connect to the low-side N-Channel internal MOSFET gate drive supply ground, the sources of the MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (SGND) loop. 3 NC No Connect. 4, 9, 10, 11, 12 SW Switch Node output. Internal connection for the high-side MOSFET source and low-side MOSFET drain. Because of the high-speed switching on this pin, the SW pin should be routed away from sensitive nodes. 13, 14, 15, 16, 17, 18, 19 PVIN High-Side N-internal MOSFET Drain Connection input. The PVIN operating voltage range is from 4.5V to 19V. Input capacitors between the PVIN pins and the power ground (PGND) are required; keep the connection short. BST Boost output. Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is connected between the PVDD pin and the BST pin. A boost capacitor of 0.1μF is connected between the BST pin and the SW pin. Adding a small resistor at the BST pin can slow down the turn-on time of high-side N-Channel MOSFETs. 20 November 2012 Pin Function 2 M9999-110712-A Micrel, Inc. MIC24053 Pin Description (Continued) Pin Number Pin Name Pin Function 22 CS Current Sense input. The CS pin senses current by monitoring the voltage across the low-side MOSFET during the OFF-time. Current sensing is necessary for short circuit protection. To sense the current accurately, connect the low-side MOSFET drain to SW using a Kelvin connection. The CS pin is also the high-side MOSFET’s output driver return. 23 SGND Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer; see PCB Layout Recommendations for details. 24 FB Feedback input. Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 25 PG Power Good output. Open-drain output. The PG pin is externally tied with a resistor to VDD. A high output is asserted when VOUT > 92% of nominal. 26 EN Enable input. A logic level control of the output. The EN pin is CMOS-compatible. Logic high = enable, logic low = shutdown. In the off state, the device’s supply current is greatly reduced (typically 5µA). Do not leave the EN pin floating. 27 VIN Power Supply Voltage input. Requires bypass capacitor to SGND. 28 VDD 5V Internal Linear Regulator output. VDD supply is the power MOSFET gate drive supply voltage and the supply bus for the IC. VDD is created by internal LDO from VIN. When VIN < +5.5V, tie VDD to PVIN pins. A 1µF ceramic capacitor from the VDD pin to SGND pins must be placed next to the IC. November 2012 3 M9999-110712-A Micrel, Inc. MIC24053 Absolute Maximum Ratings(1) Operating Ratings(3) PVIN to PGND............................................... −0.3V to +29V VIN to PGND ................................................. −0.3V to PVIN PVDD, VDD to PGND ..................................... −0.3V to +6V VSW , VCS to PGND ............................. −0.3V to (PVIN +0.3V) VBST to VSW ........................................................ −0.3V to 6V VBST to PGND .................................................. −0.3V to 35V VFB, VPG to PGND ............................. −0.3V to (VDD + 0.3V) VEN to PGND ....................................... −0.3V to (VIN +0.3V) PGND to SGND............................................ −0.3V to +0.3V Junction Temperature .............................................. +150°C Storage Temperature (TS) ......................... −65°C to +150°C Lead Temperature (soldering, 10s) ............................ 260°C (2). ESD Rating ................................................ ESD Sensitive Supply Voltage (PVIN, VIN) .............................. 4.5V to 19V PVDD, VDD Supply Voltage (PVDD, VDD) ..... 4.5V to 5.5V Enable Input (VEN) .................................................. 0V to VIN Junction Temperature (TJ) ........................ −40°C to +125°C Maximum Power Dissipation ...................................... Note 4 (4) Package Thermal Resistance 5mm x 6mm QFN-28 (θJA) ................................ 28°C/W Electrical Characteristics(5) PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units 19 V Power Supply Input 4.5 Input Voltage Range (VIN, PVIN) Quiescent Supply Current VFB = 1.5V (non-switching) Shutdown Supply Current VEN = 0V 730 1500 µA 5 10 µA 5 5.4 V 4.2 4.5 VDD Supply Voltage VDD Output Voltage VIN = 7V to 19V, IDD = 40mA 4.8 VDD UVLO Threshold VDD Rising 3.7 VDD UVLO Hysteresis Dropout Voltage (VIN – VDD) 400 IDD = 25mA 380 V mV 600 mV 5.5 V DC-DC Controller Output-Voltage Adjust Range (VOUT) 0.8 Reference Feedback Reference Voltage 0°C ≤ TJ ≤ 85°C (±1.0%) 0.792 0.8 0.808 −40°C ≤ TJ ≤ 125°C (±1.5%) 0.788 0.8 0.812 V Load Regulation IOUT = 0A to 9A (Continuous Mode) 0.25 % Line Regulation VIN = 4.5V to 19V 0.25 % FB Bias Current VFB = 0.8V 50 nA Enable Control 1.8 EN Logic Level High V EN Logic Level Low 0.6 V EN Bias Current 6 30 µA 600 750 kHz VEN = 12V Oscillator (6) VOUT = 2.5V (7) VFB = 0V 82 % VFB = 1.0V 0 % 300 ns Switching Frequency Maximum Duty Cycle Minimum Duty Cycle 450 Minimum Off-Time November 2012 4 M9999-110712-A Micrel, Inc. MIC24053 Electrical Characteristics(5) (Continued) PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate -40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units Soft-Start Soft-Start Time 3 ms Short-Circuit Protection Peak Inductor Current-Limit Threshold Short-Circuit Current VFB = 0.8V, TJ = 25°C 12.5 14 20 A VFB = 0.8V, TJ = 125°C 11.25 14 20 A VFB = 0V 8 A Top-MOSFET RDS (ON) ISW = 3A 27 mΩ Bottom-MOSFET RDS (ON) ISW = 3A 10.5 mΩ SW Leakage Current VEN = 0V 60 µA VIN Leakage Current VEN = 0V 25 µA 95 %VOUT Internal FETs Power Good (PG) PG Threshold Voltage Sweep VFB from Low to High 85 92 PG Hysteresis Sweep VFB from High to Low 5.5 %VOUT PG Delay Time Sweep VFB from Low to High 100 µs PG Low Voltage Sweep VFB < 0.9 × VNOM, IPG = 1mA 70 TJ Rising 160 °C 15 °C 200 mV Thermal Protection Overtemperature Shutdown Overtemperature Shutdown Hysteresis Notes: 1. Exceeding the absolute maximum rating can damage the device. 2. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF. 3. The device is not guaranteed to function outside its operating range. 4. PD(MAX) = (TJ(MAX) – TA)/θJA, where θJA depends on the printed circuit layout. A 5in2, 4-layer, 0.62”, FR-4 PCB with 2oz finish copper weight per layer is used for the θJA. 5. Specification for packaged product only. 6. Measured in test mode. 7. The maximum duty-cycle is limited by the fixed mandatory off-time (tOFF) of typically 300ns. November 2012 5 M9999-110712-A Micrel, Inc. MIC24053 Typical Characteristics VIN Operating Supply Current vs. Input Voltage 60 20 15 10 VOUT = 1.8V 5 IOUT = 0A SWITCHING REN = OPEN 8 45 30 15 4 7 10 13 16 VFB = 0.9V 0 4 19 7 10 13 16 7 4 19 10 16 13 INPUT VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V) Feedback Voltage vs. Input Voltage Total Regulation vs. Input Voltage Output Current Limit vs. Input Voltage 0.800 0.796 VOUT = 1.8V 0.1% 0.0% -0.1% 10 13 16 7 VOUT = 1.8V 10 13 16 19 600 550 500 450 VOUT = 1.8V 12 8 4 IOUT = 0A November 2012 95% 90% 85% VFB = 0.8V 80% 0 INPUT VOLTAGE (V) 19 VEN = VIN 350 16 16 100% VPG THRESHOLD/V REF (%) EN INPUT CURRENT (µA) 650 13 PG/VREF Ratio vs. Input Voltage 16 13 10 INPUT VOLTAGE (V) Enable Input Current vs. Input Voltage 700 10 7 4 INPUT VOLTAGE (V) Switching Frequency vs. Input Voltage 7 5 0 4 19 INPUT VOLTAGE (V) 400 10 IOUT = 0A to 9A -0.2% 7 15 VOUT = 1.8V IOUT = 0A 0.792 19 20 CURRENT LIMIT (A) TOTAL REGULATION (%) 0.2% 0.804 4 4 IDD = 10mA 0.808 4 6 2 0 0 FEEDBACK VOLTAGE (V) 10 VEN = 0V VDD VOLTAGE (V) SHUTDOWN CURRENT (µA) SUPPLY CURRENT (mA) 25 FREQUENCY (kHz) VDD Output Voltage vs. Input Voltage VIN Shutdown Current vs. Input Voltage 19 4 7 10 13 INPUT VOLTAGE (V) 6 16 19 4 7 10 13 16 19 INPUT VOLTAGE (V) M9999-110712-A Micrel, Inc. MIC24053 Typical Characteristics (Continued) VIN Shutdown Current vs. Temperature VIN Operating Supply Current vs. Temperature VDD UVLO Threshold vs. Temperature 5 14 30.0 20.0 15.0 10.0 VIN = 12V VOUT = 1.8V 5.0 IOUT = 0A SWITCHING 0.0 FEEBACK VOLTAGE (V) 0.808 0 25 50 75 100 10 8 6 4 VIN = 12V IOUT = 0A VEN = 0V 2 Falling 3 2 1 Hyst -50 -25 0 25 50 75 100 -50 125 -25 0 25 50 75 100 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) Feedback Voltage vs. Temperature Load Regulation vs. Temperature Line Regulation vs. Temperature 1.0% 0.804 0.800 0.796 VIN = 12V VOUT = 1.8V 0.792 0.2% 0.5% 0.0% -0.5% VIN = 12V VOUT = 1.8V -25 0 25 50 75 100 IOUT =0A to 9A 0.0% -0.1% -0.2% -0.3% -0.4% VIN = 4.5V to 19V VOUT = 1.8V IOUT = 0A -0.6% -50 125 0.1% -0.5% -1.0% 0.788 -25 0 25 50 75 100 125 -50 -25 TEMPERATURE (°C) TEMPERATURE (°C) 0 25 50 75 100 125 TEMPERATURE (°C) VDD vs. Temperature Switching Frequency vs. Temperature Output Current Limit vs. Temperature 20 6 700 125 0.3% IOUT = 0A -50 4 0 0 125 LINE REGULATION (%) -25 LOAD REGULATION (%) -50 12 VDD THRESHOLD (V) SHUTDOWN CURRENT (µA) SUPPLY CURRENT (mA) Rising 25.0 CURRENT LIMIT (A) 5 600 VDD (V) FREQUENCY (kHz) 650 550 500 450 VIN = 12V VOUT = 1.8V 3 VIN = 12V VOUT = 1.8V 400 4 15 10 5 VIN = 12V VOUT = 1.8V IOUT = 0A IOUT = 0A 0 2 350 -50 -25 0 25 50 75 TEMPERATURE (°C) November 2012 100 125 -50 -25 0 25 50 75 TEMPERATURE (°C) 7 100 125 -50 -25 0 25 50 75 100 125 TEMPERATURE (°C) M9999-110712-A Micrel, Inc. MIC24053 Typical Characteristics (Continued) 0.808 FEEDBACK VOLTAGE (V) 700 600 550 500 450 400 350 VIN = 12V IOUT = 0A 300 250 1.814 0.804 0.800 0.796 VIN = 12V VOUT = 1.8V 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 0 1.5 OUTPUT VOLTAGE (V) 4.5 6 7.5 0.0% -0.5% VIN = 4.5V to 19V VOUT = 1.8V -1.0% 6 7.5 550 VIN = 12V VOUT = 1.8V POWER DISSIPATION (W) 3.5 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 90 85 80 75 70 65 60 VIN = 5V 50 2 4 6 8 4 6 8 OUTPUT CURRENT (A) 3.8 TA 25ºC 85ºC 125ºC 3.4 0 10 10 2 12 4 6 8 10 12 OUTPUT CURRENT (A) Die Temperature* (VIN = 5V) vs. Output Current 100 VIN = 5V 3.0 2.5 VOUT = 3.3V 2.0 1.5 1.0 VOUT = 0.8V 80 60 40 20 VIN = 5V VOUT = 1.8V 0.5 0 0.0 2 9 4.2 IC Power Dissipation (VIN = 5V) vs. Output Current 4.0 7.5 VIN = 5V VFB < 0.8V 4.6 OUTPUT CURRENT (A) 95 6 3.0 0 100 4.5 Output Voltage (VIN = 5V) vs. Output Current 600 9 3 OUTPUT CURRENT (A) 650 Efficiency (VIN = 5V) vs. Output Current 0 1.5 5.0 OUTPUT CURRENT (A) 55 VIN = 12V VOUT = 1.8V 0 500 4.5 1.791 9 OUTPUT VOLTAGE (V) 0.5% FREQUENCY (kHz) LINE REGULATION (%) 3 700 3 1.796 Switching Frequency vs. Output Current 1.0% 1.5 1.800 OUTPUT CURRENT (A) Line Regulation vs. Output Current 0 1.805 1.782 0.792 0 1.810 1.787 DIE TEMPERATURE (°C) FREQUENCY (kHz) 650 1.819 OUTPUT VOLTAGE (V) 750 EFFICIENCY (%) Output Voltage vs. Output Current Feedback Voltage vs. Output Current Switching Frequency vs. Output Voltage 0 1.5 3 4.5 6 OUTPUT CURRENT (A) 7.5 9 0 1.5 3 4.5 6 7.5 9 OUTPUT CURRENT (A) Die Temperature* : The temperature measurement was taken at the hottest point on the MIC24053 case mounted on a 5in2, 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight per layer; see the Thermal Measurements section. Actual results depend on the size of the PCB, ambient temperature, and proximity to other heat-emitting components. November 2012 8 M9999-110712-A Micrel, Inc. MIC24053 Typical Characteristics (Continued) Efficiency (VIN = 12V) vs. Output Current IC Power Dissipation (VIN = 12V) vs. Output Current 100 4.0 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 90 EFFICIENCY (%) POWER DISSIPATION (W) 95 85 80 75 70 65 60 VIN = 12V 3.5 3.0 2.5 VOUT = 5V 2.0 1.5 1.0 VOUT = 0.8V 0.5 VIN = 12V 55 0.0 50 0 2 4 6 8 OUTPUT CURRENT (A) 10 12 0 1.5 3 4.5 6 7.5 9 OUTPUT CURRENT (A) Die Temperature* : The temperature measurement was taken at the hottest point on the MIC24053 case mounted on a 5in2, 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight per layer; see the Thermal Measurements section. Actual results depend on the size of the PCB, ambient temperature, and proximity to other heat-emitting components. November 2012 9 M9999-110712-A Micrel, Inc. MIC24053 Functional Characteristics November 2012 10 M9999-110712-A Micrel, Inc. MIC24053 Functional Characteristics (Continued) November 2012 11 M9999-110712-A Micrel, Inc. MIC24053 Functional Characteristics (Continued) November 2012 12 M9999-110712-A Micrel, Inc. MIC24053 Functional Diagram Figure 1. MIC24053 Block Diagram November 2012 13 M9999-110712-A Micrel, Inc. MIC24053 The maximum duty cycle is obtained from the 300ns tOFF(min): Functional Description The MIC24053 is an adaptive ON-time synchronous step-down DC/DC regulator with an internal 5V linear regulator and a Power Good (PG) output. It is designed to operate over a wide input-voltage range, from 4.5V to 19V, and provides a regulated output voltage at up to 9A of output current. It uses an adaptive ON-time control scheme to get a constant switching frequency and to simplify the control compensation. Overcurrent protection is implemented without using an external sense resistor. The device includes an internal soft-start function, which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Dmax = Continuous Mode In continuous mode, the MIC24053 feedback pin (FB) senses the output voltage through the voltage divider (R1 and R2), and compares it to a 0.8V reference voltage (VREF) at the error comparator through a low-gain transconductance (gm) amplifier. If the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the error comparator triggers the control logic and generates an ON-time period. The ON-time period length is predetermined by the “FIXED tON ESTIMATION” circuitry: VOUT VIN × 600kHz Eq. 1 where VOUT is the output voltage and VIN is the power stage input voltage. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. In most cases, the OFF-time period length depends on the feedback voltage. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time (tOFF(min)), which is about 300ns, the MIC24053 control logic applies the tOFF(min) instead. tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the highside MOSFET. November 2012 tS = 1− 300ns tS Eq. 2 where tS = 1/600kHz = 1.66µs. Micrel does not recommend using the MIC24053 with an OFF-time close to tOFF(min) during steady-state operation. Also, as VOUT increases, the internal ripple injection increases and reduces the line regulation performance. Therefore, the maximum output voltage of the MIC24053 should be limited to 5.5V and the maximum external ripple injection should be limited to 200mV. Please refer to the Setting Output Voltage subsection in Application Information for more details. The actual ON-time and resulting switching frequency vary with the part-to-part variation in the rise and fall times of the internal MOSFETs, the output load current, and variations in the VDD voltage. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications, such as 18V to 1.0V. The minimum tON measured on the MIC24053 evaluation board is about 100ns. During load transients, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop operation, we will analyze both the steady-state and load transient scenarios. Figure 2 shows the MIC24053 control-loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple to trigger the ON-time period. The feedback voltage ripple is proportional to the output voltage ripple and the inductor current ripple. The ON-time is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. Theory of Operation The MIC24053 operates in a continuous mode, as shown in Figure 1. t ON(estimated) = t S − t OFF(min) 14 M9999-110712-A Micrel, Inc. MIC24053 Unlike true current-mode control, the MIC24053 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. The MIC24053 control loop has the advantage of eliminating the need for slope compensation. To meet the stability requirements, the MIC24053 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV. If a low-ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to the Ripple Injection subsection in Application Information for more details about the ripple injection technique. Figure 2. MIC24053 Control Loop Timing Figure 3 shows the operation of the MIC24053 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This causes the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time (tOFF(min)) is generated to charge CBST because the feedback voltage is still below VREF. Then, the next ON-time period is triggered because of the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency after the output has stabilized at the new load current level. Because of the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in the MIC24053 converter. VDD Regulator The MIC24053 provides a 5V regulated output for input voltage VIN ranging from 5.5V to 19V. When VIN < 5.5V, tie VDD to the PVIN pins to bypass the internal linear regulator. Soft-Start Soft-start reduces the power supply input surge current at start-up by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time draws a lower input surge current. The MIC24053 implements an internal digital soft-start by making the 0.8V reference voltage (VREF) ramp from 0 to 100% in about 3ms with 9.7mV steps. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. After the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function correctly. Current Limit The MIC24053 uses the RDS(ON) of the internal low-side power MOSFET to sense overcurrent conditions. This method reduces cost, board space, and power losses taken by a discrete current sense resistor. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. Figure 3. MIC24053 Load Transient Response November 2012 15 M9999-110712-A Micrel, Inc. MIC24053 MOSFET Gate Drive The Block Diagram (Figure 1) shows a bootstrap circuit consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reverse biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA, so a 0.1μF to 1μF capacitor is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle; that is, ΔBST = 10mA x 1.67μs/0.1μF = 167mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor (RG), which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the VDD supply voltage. The nominal low-side gate drive voltage is VDD and the nominal high-side gate drive voltage is approximately VDD – VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. In each switching cycle of the MIC24053 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. If the peak inductor current is greater than 14A, the MIC24053 turns off the high-side MOSFET and a soft-start sequence is triggered. This mode of operation is called “hiccup mode.” Its purpose is to protect the downstream load in case of a hard short. The load current-limit threshold has a foldback characteristic related to the feedback voltage as shown in Figure 4. Current Limit Threshold vs. Feedback Voltage CURRENT LIMIT THRESHOLD (A) 20 16 12 8 4 0 0.0 0.2 0.4 0.6 0.8 1.0 FEEDBACK VOLTAGE (V) Figure 4. MIC24053 Current-Limit Foldback Characteristic Power Good (PG) The Power Good (PG) pin is an open-drain output that indicates logic high when the output is nominally 92% of its steady-state voltage. A pull-up resistor of more than 10kΩ should be connected from PG to VDD. November 2012 16 M9999-110712-A Micrel, Inc. MIC24053 The proper selection of core material and minimizing the winding resistance is required to maximize efficiency. The high-frequency operation of the MIC24053 requires the use of ferrite materials for all but the most costsensitive applications. Lower-cost iron powder cores may be used, but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by Equation 7: Application Information Inductor Selection Selecting the output inductor requires values for inductance, peak, and RMS currents. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents also require more output capacitance to smooth out the larger ripple current. Smaller peak-topeak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss, and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by Equation 3: 2 L= PINDUCTOR(Cu) = IL(RMS) × RWINDING VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × 20% × IOUT(max) Eq. 3 The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. where: fSW = switching frequency, 600kHz 20% = ratio of AC ripple current to DC output current VIN(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is: ∆IL(pp) = VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × L PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C)) Eq. 8 where: TH = temperature of wire under full load T20°C = ambient temperature RWINDING(20°C) = room temperature winding resistance (usually specified by the manufacturer) Eq. 4 The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) =IOUT(max) + 0.5 × ΔIL(pp) Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are ceramic, low-ESR aluminum electrolytic, OSCON, and POSCAP. The output capacitor’s ESR is usually the main cause of the output ripple. It also affects the stability of the control loop. Eq. 5 2 The RMS inductor current is used to calculate the I R losses in the inductor. 2 IL(RMS) = IOUT(max) + November 2012 ΔIL(PP) 12 Eq. 7 2 Eq.6 17 M9999-110712-A Micrel, Inc. MIC24053 Input Capacitor Selection The input capacitor for the power stage input (VIN) should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning on the input supply. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple primarily depends on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: The maximum value of ESR is calculated by Equation 9: ESR COUT ≤ ΔVOUT(pp) Eq. 9 ΔIL(PP) where: ΔVOUT(pp) = peak-to-peak output voltage ripple ΔIL(PP) = peak-to-peak inductor current ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated in Equation 10: 2 ΔVOUT(pp) ΔVIN = IL(pk) × ESRCIN ΔIL(PP) 2 + ΔIL(PP) × ESR C = OUT C × f × 8 OUT SW Eq. 10 ( ) The input capacitor must be rated for the input current ripple. The RMS value of the input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: where: D = duty cycle COUT = output capacitance value fSW = switching frequency As described in the Theory of Operation subsection in the Functional Description section, the MIC24053 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors’ value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are used for the output capacitors, a ripple injection method should be applied to provide enough feedback voltage ripple. Please refer to the Ripple Injection subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 11: ICOUT (RMS) = ΔIL(PP) Eq. 13 ICIN(RMS) ≈ IOUT(max) × D × (1 − D) Eq. 14 The power dissipated in the input capacitor is: 2 PDISS(CIN) = ICIN(RMS) × ESRCIN Eq. 15 Ripple Injection The VFB ripple required for proper operation of the MIC24053 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator cannot sense it, then the MIC24053 will lose control and the output voltage is not regulated. To have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. Eq. 11 12 The power dissipated in the output capacitor is: 2 PDISS(COUT ) = ICOUT (RMS) × ESR COUT November 2012 Eq. 12 18 M9999-110712-A Micrel, Inc. MIC24053 The applications are divided into three situations according to the amount of the feedback voltage ripple: 1. Enough ripple at the feedback voltage due to the large ESR of the output capacitors. As shown in Figure 5, the converter is stable without any ripple injection. The feedback voltage ripple is: ΔVFB(pp) R2 = × ESR COUT × ΔIL (pp) R1 + R2 Figure 7. Invisible Ripple at FB Eq. 16 In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node (SW) using a resistor (Rinj) and a capacitor (Cinj), as shown in Figure 7. The injected ripple is: where ΔIL(pp) is the peak-to-peak value of the inductor current ripple. 2. Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. ΔVFB(pp) = VIN × K div × D × (1 - D) × The output voltage ripple is fed into the FB pin through a feedforward capacitor (Cff) in this situation, as shown in Figure 6. The typical Cff value is between 1nF and 100nF. With the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: ΔVFB(pp) ≈ ESR × ΔIL (pp) K div = R1//R2 R inj + R1//R2 1 fSW × τ Eq. 18 Eq. 19 where: VIN = power stage input voltage D = duty cycle fSW = switching frequency τ = (R1//R2//Rinj) × Cff Eq. 17 3. Virtually no ripple at the FB pin voltage due to the very-low ESR of the output capacitors. In Equations 18 and 19, it is assumed that the time constant associated with Cff must be much greater than the switching period: 1 T = << 1 fSW × τ τ Eq. 20 If the voltage divider resistors (R1 and R2) are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirement. Also, a 100nF injection capacitor (Cinj) is used in order to be considered as short for a wide range of the frequencies. Figure 5. Enough Ripple at FB Figure 6. Inadequate Ripple at FB November 2012 19 M9999-110712-A Micrel, Inc. MIC24053 The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in the kΩ range. Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 19: K div = ΔVFB(pp) VIN × fSW × τ D × (1 − D) Eq. 21 Figure 8. Voltage-Divider Configuration Then the value of Rinj is calculated as: R inj = (R1//R2) × ( 1 K div − 1) In addition to the external ripple injection added at the FB pin, internal ripple injection is added at the inverting input of the comparator inside the MIC24053, as shown in Figure 9. The inverting input voltage (VINJ) is clamped to 1.2V. As VOUT increases, the swing of VINJ is clamped. The clamped VINJ reduces the line regulation because it is reflected as a DC error on the FB terminal. Therefore, the maximum output voltage of the MIC24053 should be limited to 5.5V to avoid this problem. Eq. 22 Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. Setting Output Voltage The MIC24053 requires two resistors to set the output voltage as shown in Figure 8. The output voltage is determined by Equation 23: VOUT = VFB × (1 + R1 ) R2 Eq. 23 where VFB = 0.8V. A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 = VFB × R1 VOUT − VFB November 2012 Figure 9. Internal Ripple Injection Eq. 24 20 M9999-110712-A Micrel, Inc. MIC24053 Thermal Measurements It is a good idea to measure the IC’s case temperature to make sure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get false results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are to use a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire, or higher (smaller wire size), to minimize the wire heatsinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on small form factor ICs. However, an IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. November 2012 21 M9999-110712-A Micrel, Inc. MIC24053 PCB Layout Guidelines NOTE: Inductor To minimize EMI and output noise, follow these layout recommendations. PCB layout is critical to achieve reliable, stable, and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal, and return paths. Follow these guidelines to ensure proper MIC24053 regulator operation: IC • • A 2.2µF ceramic capacitor, which is connected to the PVDD pin, must be located right at the IC. The PVDD pin is very noise sensitive and placement of the capacitor is critical. Use wide traces to connect to the PVDD and PGND pins. • Keep the inductor connection to the switch node (SW) short. • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (SW) away from the feedback (FB) pin. • Connect the CS pin directly to the SW pin to accurately sense the voltage across the low-side MOSFET. • To minimize noise, place a ground plane underneath the inductor. • The inductor can be placed on the opposite side of the PCB with respect to the IC. It does not matter whether the IC or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. The input and output capacitors must be placed on the same side of the board as the IC. A 1µF ceramic capacitor must be placed right between VDD and the signal ground (SGND). SGND must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. Output Capacitor • Place the IC close to the point-of-load (POL). • • Use fat traces to route the input and output power lines. Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. • Keep signal and power grounds separate and connected at only one location. • Phase margin changes as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. • The feedback trace should be separate from the power trace and connected as near as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. Input Capacitor • Place the input capacitor next. • Place the input capacitors on the same side of the board and as close to the IC as possible. • Keep both the PVIN pin and PGND connections short. • Place several vias to the ground plane close to the input capacitor ground terminal. • Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. • Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. • If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. • In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply when power is suddenly applied. November 2012 Optional RC Snubber • 22 Place the RC snubber on either side of the board and as close to the SW pin as possible. M9999-110712-A Micrel, Inc. MIC24053 Evaluation Board Schematic Figure 10. Schematic of MIC24053 Evaluation Board (J11, R13, R15 are for testing purposes) November 2012 23 M9999-110712-A Micrel, Inc. MIC24053 Evaluation Board Schematic (Continued) Figure 11. Schematic of MIC24053 Evaluation Board (Optimized for Smallest Footprint) November 2012 24 M9999-110712-A Micrel, Inc. MIC24053 Bill of Materials Item Part Number C1 Open 12103C475KAT2A C2, C3 GRM32DR71E475KA61K C3225X7R1E475K C13, C15 C6, C7, C10 GRM32ER60J107ME20L C12 Murata (2) 4.7µF Ceramic Capacitor, X7R, Size 1210, 25V 2 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 2 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 3 1.0µF Ceramic Capacitor, X7R, Size 0603, 10V 1 2.2µF Ceramic Capacitor, X5R, Size 0603, 10V 1 4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1 220µF Aluminum Capacitor, 35V 1 40V, 350mA Schottky Diode. SOD323 1 2.2µH Inductor, 15A Saturation Current 1 (3) TDK AVX Murata 06035C104KAT2A AVX GRM188R71H104KA93D GRM188R71A105KA61D Murata TDK AVX Murata C1608X7R1A105K TDK 0603ZD225KAT2A AVX GRM188R61A225KE34D Murata C1608X5R1A225K TDK 06035C472KAZ2A AVX GRM188R71H472K Murata C1608X7R1H472K TDK C14 B41851F7227M C11, C16 Open SD103AWS D1 (1) TDK 0603ZC105KAT2A C9 Qty. AVX C3225X5R0J107M C1608X7R1H104K C8 Description Open 12106D107MAT2A C4, C5 Manufacturer SD103AWS-7 SD103AWS (4) EPCOS (5) MCC Diodes Inc (6) (7) Vishay HCF1305-2R2-R R1 CRCW06032R21FKEA Vishay Dale 2.21Ω Resistor, Size 0603, 1% 1 R2 CRCW06032R00FKEA Vishay Dale 2.00Ω Resistor, Size 0603, 1% 1 R3 CRCW060319K6FKEA Vishay Dale 19.6kΩ Resistor, Size 0603, 1% 1 R4 CRCW06032K49FKEA Vishay Dale 2.49kΩ Resistor, Size 0603, 1% 1 R5 CRCW060320K0FKEA Vishay Dale 20.0kΩ Resistor, Size 0603, 1% 1 R6, R14, R17 CRCW060310K0FKEA Vishay Dale 10.0kΩ Resistor, Size 0603, 1% 3 R7 CRCW06034K99FKEA Vishay Dale 4.99kΩ Resistor, Size 0603, 1% 1 R8 CRCW06032K87FKEA Vishay Dale 2.87kΩ Resistor, Size 0603, 1% 1 R9 CRCW06032K006FKEA Vishay Dale 2.00kΩ Resistor, Size 0603, 1% 1 R10 CRCW06031K18FKEA Vishay Dale 1.18kΩ Resistor, Size 0603, 1% 1 R11 CRCW0603806RFKEA Vishay Dale 806Ω Resistor, Size 0603, 1% 1 R12 CRCW0603475RFKEA Vishay Dale 475Ω Resistor, Size 0603, 1% 1 November 2012 Cooper Bussmann (8) L1 25 M9999-110712-A Micrel, Inc. MIC24053 Bill of Materials (Continued) Item Part Number Manufacturer R13 CRCW06030000FKEA Vishay Dale 0Ω Resistor, Size 0603, 5% 1 R15 CRCW060349R9FKEA Vishay Dale 49.9Ω Resistor, Size 0603, 1% 1 R16, R18 CRCW06031R21FKEA Vishay Dale 1.21Ω Resistor, Size 0603, 1% 2 R20 Open All Reference designators ending with “A” Open U1 MIC24053YJL 12V, 9A High-Efficiency Buck Regulator 1 (9) Micrel. Inc. Description Qty. Notes: 1. AVX: www.avx.com. 2. Murata: www.murata.com. 3. TDK: www.tdk.com. 4. EPCOS: www.epcos.com. 5. MCC: www.mccsemi.com. 6. Diode Inc.: www.diodes.com. 7. Vishay: www.vishay.com. 8. Cooper Bussmann: www.cooperbussmann.com. 9. Micrel, Inc.: www.micrel.com. November 2012 26 M9999-110712-A Micrel, Inc. MIC24053 PCB Layout Recommendations Figure 12. MIC24053 Evaluation Board Top Layer Figure 13. MIC24053 Evaluation Board Mid-Layer 1 (Ground Plane) November 2012 27 M9999-110712-A Micrel, Inc. MIC24053 PCB Layout Recommendations (Continued) Figure 14. MIC24053 Evaluation Board Mid-Layer 2 Figure 15. MIC24053 Evaluation Board Bottom Layer November 2012 28 M9999-110712-A Micrel, Inc. MIC24053 Package Information(1) 28-Pin 5mm × 6mm QFN (JL) Note: 1. Package information is correct as of the publication date. For updates and most current information, go to www.micrel.com. November 2012 29 M9999-110712-A Micrel, Inc. MIC24053 MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2012 Micrel, Incorporated. November 2012 30 M9999-110712-A