MICREL MIC26603YJL

MIC26603
28V, 6A Hyper Light Load™
Synchronous DC/DC Buck Regulator
SuperSwitcher IIG™
General Description
Features
The Micrel MIC26603 is a constant-frequency,
synchronous DC/DC buck regulator featuring adaptive ontime control architecture. The MIC26603 operates over a
supply range of 4.5V to 28V. It has an internal linear
regulator which provides a regulated 5V to power the
internal control circuitry. MIC26603 operates at a constant
600kHz switching frequency in continuous-conduction
mode and can be used to provide up to 6A of output
current. The output voltage is adjustable down to 0.8V.
•
•
Micrel’s Hyper Light Load™ architecture provides the same
high-efficiency and ultra-fast transient response as the
Hyper Speed Control™ architecture under medium to heavy
loads, but also maintains high efficiency under light load
conditions
by
transitioning
to
variable-frequency,
discontinuous-mode operation.
The MIC26603 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, thermal shutdown, internal
soft-start to reduce the inrush current, foldback current
limit and “hiccup mode” short-circuit protection. The
MIC26603 includes a Power Good (PG) output to allow
simple sequencing.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
•
•
•
•
•
•
•
•
•
•
•
•
Hyper Light Load™ efficiency – up to 80% at 10mA
Hyper Speed Control™ architecture enables
− High Delta V operation (VIN = 28V and VOUT = 0.8V)
− Small output capacitance
Input voltage range: 4.5V to 28V
Output current up to 6A
Up to 95% Efficiency
Adjustable output voltage from 0.8V to 5.5V
±1% FB accuracy
Any CapacitorTM stable − zero-to-high ESR
600kHz switching frequency
Power Good (PG) output
Foldback current-limit and “hiccup mode” short-circuit
protection
Safe start-up into pre-biased loads
5mm x 6mm MLF® package
–40°C to +125°C junction temperature range
Applications
• Distributed power systems
• Telecom/networking infrastructure
• Printers, scanners, graphic cards, and video cards
___________________________________________________________________________________________________________
Typical Application
Efficiency (VIN = 12V)
vs. Output Current
100
95
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
EFFICIENCY (%)
90
85
80
75
70
65
60
55
50
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
Hyper Speed Control, Hyper Light Load, SuperSwitcher II, and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
July 2011
M9999-071311-A
Micrel, Inc.
MIC26603
Ordering Information
Part Number
Voltage
Switching Frequency
Junction Temperature
Range
Package
Lead Finish
MIC26603YJL
Adjustable
600kHz
–40°C to +125°C
28-Pin 5mm × 6mm MLF®
Pb-Free
Pin Configuration
28-Pin 5mm x 6mm MLF® (YJL)
Pin Description
Pin Number
Pin Name
1
PVDD
3
NC
No Connect.
4, 9, 10, 11, 12
SW
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain. Due to the high speed switching on this pin, the SW pin should be routed away from sensitive
nodes.
PGND
Power Ground. PGND is the ground path for the MIC26603 buck converter power stage. The PGND
pins connect to the low-side N-Channel internal MOSFET gate drive supply ground, the sources of
the MOSFETs, the negative terminals of input capacitors, and the negative terminals of output
capacitors. The loop for the power ground should be as small as possible and separate from the
Signal ground (SGND) loop.
PVIN
High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 26V. Input capacitors between the PVIN pins and the power ground (PGND) are required
and keep the connection short.
BST
Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the PVDD pin and the BST pin. A boost capacitor of 0.1μF is connected between
the BST pin and the SW pin. Adding a small resistor at the BST pin can slow down the turn-on time
of high-side N-Channel MOSFETs.
2, 5, 6, 7, 8, 21
13,14,15,
16,17,18,19
20
July 2011
Pin Function
5V Internal Linear Regulator (Output): PVDD supply is the power MOSFET gate drive supply voltage
and created by internal LDO from VIN. When VIN < +5.5V, PVDD should be tied to PVIN pins.
A 2.2µF ceramic capacitor from the PVDD pin to PGND (pin 2) must be place next to the IC.
2
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Micrel, Inc.
MIC26603
Pin Description (Continued)
Pin Number
Pin Name
Pin Function
22
CS
Current Sense (input): The CS pin senses current by monitoring the voltage across the low-side
MOSFET during the OFF-time. The current sensing is necessary for short circuit protection and zero
current cross comparator. In order to sense the current accurately, connect the low-side MOSFET
drain to SW using a Kelvin connection. The CS pin is also the high-side MOSFET’s output driver
return.
23
SGND
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin
to the PGND Pad on the top layer, see PCB layout guidelines for details.
24
FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is
regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the
desired output voltage.
25
PG
Power Good (Output): Open Drain Output. The PG pin is externally tied with a resistor to VDD. A
high output is asserted when VOUT > 92% of nominal.
26
EN
Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high =
enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced
(typically 5µA). The EN pin should not be left open.
27
VIN
Power Supply Voltage (Input): Requires bypass capacitor to SGND.
28
VDD
5V Internal Linear Regulator (Output): VDD supply is the supply bus for the IC control circuit. VDD is
created by internal LDO from VIN. When VIN < +5.5V, VDD should be tied to PVIN pins. A 1.0µF
ceramic capacitor from the VDD pin to SGND pins must be place next to the IC.
July 2011
3
M9999-071311-A
Micrel, Inc.
MIC26603
Absolute Maximum Ratings(1,2)
Operating Ratings(3)
PVIN to PGND................................................ −0.3V to +29V
VIN to PGND ....................................................−0.3V to PVIN
PVDD, VDD to PGND ......................................... −0.3V to +6V
VSW, VCS to PGND .............................. −0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 35V
VFB, VPG to PGND............................... −0.3V to (VDD + 0.3V)
VEN to PGND ........................................ −0.3V to (VIN +0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Supply Voltage (PVIN, VIN)................................. 4.5V to 28V
PVDD, VDD Supply Voltage (PVDD, VDD)......... 4.5V to 5.5V
Enable Input (VEN) .................................................. 0V to VIN
Junction Temperature (TJ) ........................ −40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF®-24L (θJA) .............................28°C/W
Electrical Characteristics(5)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate -40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
28
V
450
750
µA
5
10
µA
Power Supply Input
Input Voltage Range (VIN, PVIN)
4.5
Quiescent Supply Current
VFB = 1.5V (non-switching)
Shutdown Supply Current
VEN = 0V
VDD Supply Voltage
VDD Output Voltage
VIN = 7V to 28V, IDD = 40mA
4.8
5
5.4
V
VDD UVLO Threshold
VDD Rising
3.7
4.2
4.5
V
VDD UVLO Hysteresis
400
Dropout Voltage (VIN – VDD)
IDD = 25mA
380
mV
600
mV
5.5
V
DC/DC Controller
Output-Voltage Adjust Range
(VOUT)
0.8
Reference
0°C ≤ TJ ≤ 85°C (±1.0%)
0.792
0.8
0.808
−40°C ≤ TJ ≤ 125°C (±1.5%)
0.788
0.8
0.812
V
Load Regulation
IOUT = 1A to 6A (Continuous Mode)
0.25
%
Line Regulation
VIN = 4.5V to 28V
0.25
%
FB Bias Current
VFB = 0.8V
50
500
nA
Enable Control
EN Logic Level High
V
1.8
EN Logic Level Low
EN Bias Current
VEN = 12V
0.6
V
6
30
µA
600
750
kHz
Oscillator
Switching Frequency (6)
Maximum Duty Cycle
Minimum Duty Cycle
(7)
450
VFB = 0V
82
VFB = 1.0V
Minimum Off-Time
July 2011
4
%
0
%
300
ns
M9999-071311-A
Micrel, Inc.
MIC26603
Electrical Characteristics(5) (Continued)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Soft-Start
Soft-Start time
5
ms
Short-Circuit Protection
Current-Limit Threshold
VFB = 0.8V, TJ = 25°C
7.5
13
17
A
Current-Limit Threshold
VFB = 0.8V, TJ = 125°C
6.6
13
17
A
Short-Circuit Current
VFB = 0V
2.7
A
Top-MOSFET RDS (ON)
ISW = 3A
42
mΩ
Bottom-MOSFET RDS (ON)
ISW = 3A
12.5
mΩ
SW Leakage Current
VEN = 0V
60
µA
VIN Leakage Current
VEN = 0V
25
µA
95
%VOUT
Internal FETs
Power Good
Power Good Threshold Voltage
Sweep VFB from Low to High
Power Good Hysteresis
Sweep VFB from High to Low
5.5
%VOUT
Power Good Delay Time
Sweep VFB from Low to High
100
µs
Power Good Low Voltage
Sweep VFB < 0.9 × VNOM, IPG = 1mA
70
TJ Rising
160
°C
15
°C
85
92
200
mV
Thermal Protection
Over-Temperature Shutdown
Over-Temperature Shutdown
Hysteresis
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. A 5 square inch 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight
per layer is used for the θJA.
5. Specification for packaged product only.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 300ns.
July 2011
5
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Micrel, Inc.
MIC26603
Typical Characteristics
IOUT = 0A
SWITCHING
0.6
0.4
0.2
4
10
16
22
VEN = 0V
REN = Open
40
20
4
10
16
22
4
28
0.800
V OUT = 1.8V
IOUT = 1A
0.792
16
22
VOUT = 1.8V
IOUT = 1A to 6A
0.5%
0.0%
-0.5%
10
INPUT VOLTAGE (V)
16
22
VOUT = 1.8V
IOUT = 1A
500
22
INPUT VOLTAGE (V)
July 2011
28
VEN = VIN
8
4
22
28
PG Threshold/VREF Ratio
vs. Input Voltage
100%
12
16
INPUT VOLTAGE (V)
VPG THRESHOLD/VREF (%)
EN INPUT CURRENT (µA)
600
16
10
Enable Input Current
vs. Input Voltage
650
10
5
4
28
16
4
10
INPUT VOLTAGE (V)
700
550
15
0
4
Switching Frequency
vs. Input Voltage
28
VOUT = 1.8V
-1.0%
28
22
20
CURRENT LIMIT (A)
TOTAL REGULATION (%)
0.804
16
Current Limit
vs. Input Voltage
1.0%
10
10
INPUT VOLTAGE (V)
Total Regulation
vs. Input Voltage
0.808
4
V FB = 0.9V
IDD = 10mA
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
0.796
4
0
0
28
6
2
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
8
VDD VOLTAGE (V)
VOUT = 1.8V
0.8
0.0
FREQUENCY (kHz)
10
60
SHUTDOWN CURRENT (µA)
SUPPLY CURRENT (mA)
1.0
VDD Output Voltage
vs. Input Voltage
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
95%
90%
85%
VREF = 0.7V
80%
0
4
10
16
22
INPUT VOLTAGE (V)
6
28
4.0
10.0
16.0
22.0
28.0
INPUT VOLTAGE (V)
M9999-071311-A
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
VIN Operating Supply Current
vs. Temperature
1.0
VIN Shutdown Current
vs. Temperature
8
VDD UVLO Threshold
vs. Temperature
5
0.6
0.4
VIN = 12V
VOUT = 1.8V
IOUT = 0A
SWITCHING
0.2
VDD THRESHOLD (V)
SUPPLY CURRENT (uA)
SUPPLY CURRENT (mA)
Rising
0.8
6
4
VIN = 12V
2
IOUT = 0A
4
Falling
3
2
1
Hyst
VEN = 0V
0.0
-50
-25
0
25
50
75
100
0
0
125
-50
-25
0
TEMPERATURE (°C)
75
100
-50
125
0
VOUT = 1.8V
IOUT = 1A
0.800
0.796
50
75
100
125
Line Regulation
vs. Temperature
0.2%
LINE REGULATION (%)
VIN = 12V
25
TEMPERATURE (°C)
0.4%
0.804
-25
Load Regulation
vs. Temperature
LOAD REGULATION (%)
FEEBACK VOLTAGE (V)
50
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
0.808
25
0.2%
0.0%
VIN = 12V
-0.2%
VOUT = 1.8V
0.1%
0.0%
VIN = 4.5V to 28V
VOUT = 1.8V
IOUT = 1A
-0.1%
IOUT =1A to 6A
0.792
-50
-25
0
25
50
75
100
-0.2%
-0.4%
125
-50
TEMPERATURE (°C)
-25
0
25
50
75
100
-50
125
-25
TEMPERATURE (°C)
Switching Frequency
vs. Temperature
700
25
50
75
100
125
100
125
Current Limit
vs. Temperature
VDD
vs. Temperature
6
0
TEMPERATURE (°C)
25
V IN = 12V
5
IOUT = 1A
VDD (V)
FREQUENCY (kHz)
CURRENT LIMIT (A)
V OUT = 1.8V
650
600
550
4
3
VIN = 12V
20
15
10
V IN = 12V
V OUT = 1.8V
5
IOUT = 0A
500
-50
-25
0
25
50
75
TEMPERATURE (°C)
July 2011
100
125
0
2
-50
-25
0
25
50
75
TEMPERATURE (°C)
7
100
125
-50
-25
0
25
50
75
TEMPERATURE (°C)
M9999-071311-A
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Feedback Voltage
vs. Output Current
Efficiency
vs. Output Current
100
80
24V IN
70
60
VOUT = 1.8V
50
0
1
2
3
4
5
1.814
OUTPUT VOLTAGE
FEEDBACK VOLTAGE (V)
12V IN
0.804
0.800
0.796
VIN = 12V
0
OUTPUT CURRENT (A)
3
4
5
1.791
0
6
1
2
3
4
5
6
OUTPUT CURRENT (A)
Output Voltage (VIN = 5V)
vs. Output Current
5
OUTPUT VOLTAGE (V)
0.0%
-0.5%
600
VIN = 12V
550
VOUT = 1.8V
0
1
2
3
4
5
1
6
2
Efficiency (VIN = 5V)
vs. Output Current
IC POWER DISSIPATION (W)
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
85
80
75
70
VIN = 5V
55
50
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
July 2011
3
4
5
3.8
TA
25ºC
85ºC
125ºC
3.4
0
6
1
2
7
8
4
5
6
7
8
Die Temperature* (VIN = 5V)
vs. Output Current
60
V IN = 5V
V OUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V
2.0
1.5
1.0
3.3V
0.8V
0.5
3
OUTPUT CURRENT (A)
IC Power Dissipation
vs. Output Current
2.5
95
90
4.2
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
100
VFB < 0.8V
4.6
3
500
-1.0%
EFFICIENCY (%)
2
650
VOUT = 1.8V
FREQUENCY (kHz)
LINE REGULATION (%)
1
Switching Frequency
vs. Output Current
700
VIN = 4.6V to 28V
60
1.796
VIN = 5V
0.5%
65
1.800
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current
1.0%
1.805
1.782
0.792
6
VIN = 12V
VOUT = 1.8V
1.810
1.787
VOUT = 1.8V
DIE TEMPERATURE (°C)
EFFICIENCY (%)
1.819
0.808
90
Output Voltage
vs. Output Current
50
40
30
20
VIN = 5V
VOUT = 1.8V
10
0
0.0
0
1
2
3
4
OUTPUT CURRENT (A)
8
5
6
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
M9999-071311-A
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Efficiency (VIN = 12V)
vs. Output Current
2.5
80
75
70
65
60
55
50
0
1
2
3
4
5
6
7
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
2.0
1.5
1.0
5.0V
0.5
0
80
75
70
IC POWER DISSIPATION (W)
EFFICIENCY (%)
85
65
60
55
50
2
3
4
5
2
3
4
5
6
7
1.5
1.0
5.0V
0.8V
0.5
OUTPUT CURRENT (A)
8
1.5V
VIN = 5V
V OUT = 0.8, 1.2, 1.5V
0
-25
0
25
50
75
100
AMBIENT TEMPERATURE (°C)
July 2011
125
3
4
5
6
50
40
30
20
VIN = 24V
VOUT = 1.8V
10
0
0
1
2
3
4
5
0
6
1
10
3.3V
VIN = 5V
VOUT = 1.8, 2.5, 3.3V
2
4
5
6
12
8
4
3
Thermal Derating*
vs. Ambient Temperature
1.8V
6
2
OUTPUT CURRENT (A)
Thermal Derating*
vs. Ambient Temperature
0.8V
2
60
OUTPUT CURRENT (A)
10
-50
1
OUTPUT CURRENT (A)
12
2
VIN = 12V
VOUT = 1.8V
10
0
0.0
8
12
4
20
Die Temperature* (VIN = 24V)
vs. Output Current
VIN = 24V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
Thermal Derating*
vs. Ambient Temperature
6
30
6
2.0
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
1
IC Power Dissipation (VIN = 24V)
vs. Output Current
2.5
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
90
1
40
OUTPUT CURRENT (A)
Efficiency (VIN = 24V)
vs. Output Current
0
50
0
0.0
8
OUTPUT CURRENT (A)
95
0.8V
OUTPUT CURRENT (A)
EFFICIENCY (%)
85
IC POWER DISSIPATION (W)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
90
60
VIN = 12V
DIE TEMPERATURE (°C)
95
DIE TEMPERATURE (°C)
100
Die Temperature* (VIN = 12V)
vs. Output Current
IC Power Dissipation (VIN = 12V)
vs. Output Current
10
0.8V
8
1.8V
6
4
VIN = 12V
VOUT = 0.8, 1.2, 1.8V
2
0
0
-50
-25
0
25
50
75
100
AMBIENT TEMPERATURE (°C)
9
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
M9999-071311-A
Micrel, Inc.
MIC26603
Typical Characteristics (Continued)
Thermal Derating*
vs. Ambient Temperature
Thermal Derating*
vs. Ambient Temperature
12
10
2.5V
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
12
8
5V
6
4
VIN = 12V
VOUT = 2.5, 3.3, 5V
2
0
-50
-25
0
25
50
75
100
AMBIENT TEMPERATURE (°C)
125
10
0.8V
8
6
2.5V
4
VIN = 24V
VOUT = 0.8, 1.2, 2.5V
2
0
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26603 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz finish copper weight per layer; see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
July 2011
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Micrel, Inc.
MIC26603
Functional Characteristics
July 2011
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Micrel, Inc.
MIC26603
Functional Characteristics (Continued)
July 2011
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Micrel, Inc.
MIC26603
Functional Characteristics (Continued)
July 2011
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Micrel, Inc.
MIC26603
Functional Diagram
Figure 1. MIC26603 Block Diagram
July 2011
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M9999-071311-A
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MIC26603
The maximum duty cycle is obtained from the 300ns
tOFF(min):
Functional Description
The MIC26603 is an adaptive, ON-time, synchronous
step-down, DC/DC regulator with an internal 5V linear
regulator and a Power Good (PG) output. It is designed
to operate over a wide input voltage range from 4.5V to
28V and provides a regulated output voltage at up to 7A
of output current. An adaptive ON-time control scheme is
employed in to obtain a constant switching frequency
and to simplify the control compensation. Over-current
protection is implemented without the use of an external
sense resistor. The device includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time.
Dmax =
Continuous Mode
In continuous mode, the output voltage is sensed by the
MIC26603 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
VOUT
VIN × 600kHz
Eq. 1
where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
300ns, the MIC26603 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
July 2011
tS
= 1-
300ns
tS
Eq. 2
where tS = 1/600kHz = 1.66μs.
It is not recommended to use MIC26603 with a OFF-time
close to tOFF(min) during steady-state operation. Also, as
VOUT increases, the internal ripple injection will increase
and reduce the line regulation performance. Therefore,
the maximum output voltage of the MIC26603 should be
limited to 5.5V and the maximum external ripple injection
should be limited to 200mV.Please refer to “Setting
Output Voltage” subsection in Application Information for
more details.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 24V to 1.0V. The minimum tON
measured on the MIC26603 evaluation board is about
100ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios.
Figure 2 shows the MIC26603 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Theory of Operation
The MIC26603 is able to operate in either continuous
mode or discontinuous mode. The operating mode is
determined by the output of the Zero Cross comparator
(ZC) as shown in Figure 1.
t ON(estimated) =
t S - t OFF(min)
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MIC26603
Unlike true current-mode control, the MIC26603 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26603 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC26603 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
Figure 2. MIC26603 Control Loop Timing
Figure 3 shows the operation of the MIC26603 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26603 converter.
Discontinuous Mode
In continuous mode, the inductor current is always
greater than zero; however, at light loads the MIC26603
is able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 4. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current. The MIC26603 wakes
up and turns on the high-side MOSFET when the
feedback voltage VFB drops below 0.8V.
The MIC26603 has a zero crossing comparator that
monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time. If
the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC26603 automatically powers down
most of the IC circuitry and goes into a low-power mode.
Once the MIC26603 goes into discontinuous mode, both
LSD and HSD are low, which turns off the high-side and
low-side MOSFETs. The load current is supplied by the
output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 4 shows the control loop timing in
discontinuous mode.
Figure 3. MIC26603 Load Transient Response
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Micrel, Inc.
MIC26603
Current Limit
The MIC26603 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC26603 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the inductor current is
greater than 13A, then the MIC26603 turns off the highside MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The load current-limit threshold has a fold
back characteristic related to the feedback voltage as
shown in Figure 5.
Figure 4. MIC26603 Control Loop Timing
(Discontinuous Mode)
Current Limit Threshold
vs. Feedback Voltage
During discontinuous mode, the zero crossing
comparator and the current-limit comparator are turned
off. The bias current of most circuits are reduced. As a
result, the total power supply current during
discontinuous mode is only about 450μA, allowing the
MIC26603 to achieve high efficiency in light load
applications.
CURRENT LIMIT THRESHOLD (A)
20
16
12
VDD Regulator
The MIC26603 provides a 5V regulated output for input
voltage VIN ranging from 5.5V to 28V. When VIN < 5.5V,
VDD should be tied to PVIN pins to bypass the internal
linear regulator.
4
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC26603 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 5ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
July 2011
8
Figure 5. MIC26603 Current Limit
Foldback Characteristic
Power Good (PG)
The Power Good (PG) pin is an open drain output which
indicates logic high when the output is nominally 92% of
its steady state voltage. A pull-up resistor of more than
10kΩ should be connected from PG to VDD.
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MIC26603
MOSFET Gate Drive
The Block Diagram (Figure 1) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA × 1.67μs/0.1μF = 167mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
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MIC26603
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high-frequency operation of the MIC26603 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 7:
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated in Equation 3:
L=
PINDUCTOR(Cu) = IL(RMS)2 × RWINDING
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × 20% × IOUT(max)
Eq. 3
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
where:
fSW = switching frequency, 600kHz
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
ΔIL(pp) =
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × L
PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 8
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Eq. 4
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view.
Eq. 5
The RMS inductor current is used to calculate the I2R
losses in the inductor.
2
IL(RMS) = IOUT(max) +
July 2011
ΔIL(PP)
12
Eq. 7
2
Eq. 6
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M9999-071311-A
Micrel, Inc.
MIC26603
The maximum value of ESR is calculated:
ESR COUT ≤
The power dissipated in the output capacitor is:
2
ΔVOUT(pp)
PDISS(COUT ) = ICOUT (RMS) × ESR COUT
Eq. 9
ΔIL(PP)
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
2
ΔVOUT(pp)
ΔIL(PP)
⎞
⎛
2
⎟ + ΔIL(PP) × ESR C
= ⎜⎜
OUT
⎟
C
f
8
×
×
OUT
SW
⎠
⎝
Eq. 10
(
)
ΔVIN = IL(pk) × ESRCIN
where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
July 2011
ΔIL(PP)
12
Eq. 13
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
As described in the “Theory of Operation” subsection in
Functional Description, the MIC26603 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 11:
ICOUT (RMS) =
Eq. 12
ICIN(RMS) ≈ IOUT(max) × D × (1 − D)
Eq. 14
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)2 × ESRCIN
Eq. 15
Ripple Injection
The VFB ripple required for proper operation of the
MIC26603 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC26603 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
Eq. 11
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M9999-071311-A
Micrel, Inc.
MIC26603
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 6a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
ΔVFB(pp) =
R2
× ESR COUT × ΔIL (pp)
R1 + R2
Figure 6c. Invisible Ripple at FB
Eq. 16
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 6c. The injected ripple
is:
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
ΔVFB(pp) = VIN × K div × D × (1 - D) ×
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 6b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
ΔVFB(pp) ≈ ESR × ΔIL (pp)
K div =
R1//R2
R inj + R1//R2
1
fSW × τ
Eq. 18
Eq. 19
where
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
Eq. 17
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors.
In Equations 21 and 22, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
= << 1
fSW × τ τ
Eq. 20
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Figure 6a. Enough Ripple at FB
Figure 6b. Inadequate Ripple at FB
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Micrel, Inc.
MIC26603
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 24:
K div =
ΔVFB(pp)
VIN
×
fSW × τ
D × (1- D)
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC26603, as shown
in Figure 8. The inverting input voltage VINJ is clamped to
1.2V. As VOUT is increased, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected as a DC error on the FB terminal.
Therefore, the maximum output voltage of the MIC26603
should be limited to 5.5V to avoid this problem.
Eq. 21
Then the value of Rinj is obtained as:
R inj = (R1//R2) × (
1
K div
− 1)
Eq. 22
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26603 requires two resistors to set the output
voltage as shown in Figure 7.
The output voltage is determined by the equation:
VOUT = VFB × (1 +
R1
)
R2
Eq. 23
Figure 8. Internal Ripple Injection
where, VFB = 0.8V.
A typical value of R1 can be between 3kΩ and 10kΩ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, it will
decrease the efficiency of the power supply, especially
at light loads. Once R1 is selected, R2 can be calculated
using:
R2 =
VFB × R1
VOUT − VFB
Thermal Measurements
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Eq. 24
Figure 7. Voltage-Divider Configuration
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MIC26603
Inductor
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC26603 regulator.
IC
•
A 2.2µF ceramic capacitor, which is connected to
the PVDD pin, must be located right at the IC. The
PVDD pin is very noise sensitive and placement of
the capacitor is very critical. Use wide traces to
connect to the PVDD and PGND pins.
•
A 1.0uF ceramic capacitor must be placed right
between VDD and the signal ground SGND. The
SGND must be connected directly to the ground
planes. Do not route the SGND pin to the PGND
Pad on the top layer.
•
Place the IC close to the point-of-load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
Place the input capacitor next.
•
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
Keep both the PVIN pin and PGND connections
short.
•
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
•
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
July 2011
Keep the inductor connection to the switch node
(SW) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the lowside MOSFET.
•
To minimize noise, place a ground plane underneath
the inductor.
•
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
Output Capacitor
Input Capacitor
•
•
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
Optional RC Snubber
•
23
Place the RC snubber on either side of the board
and as close to the SW pin as possible.
M9999-071311-A
Micrel, Inc.
MIC26603
Evaluation Board Schematic
Figure 9. Schematic of MIC26603 Evaluation Board
(J11, R13, R15 are for testing purposes)
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Micrel, Inc.
MIC26603
Bill of Materials
Item
Part Number
C1
Open
12105C475KAZ2A
C2, C3
GRM32ER71H475KA88L
C3225X7R1H475K
C4, C13, C15
C6, C7, C10
C8
C9
C12
GRM32ER60J107ME20L
C11, C16
R1
R2
R3
R4
R5
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
2
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
3
1.0µF Ceramic Capacitor, X7R, Size 0603, 10V
1
2.2µF Ceramic Capacitor, X5R, Size 0603, 10V
1
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
1
220µF Aluminum Capacitor, 35V
1
40V, 350mA, Schottky Diode, SOD323
1
Cooper Bussmann(8) 2.2µH Inductor, 15A Saturation Current
1
(3)
TDK
AVX(1)
Murata(2)
(3)
06035C104KAT2A
AVX(1)
GRM188R71H104KA93D
Murata(2)
(3)
C1608X7R1H104K
TDK
0603ZC105KAT2A
AVX(1)
GRM188R71A105KA61D
Murata(2)
(3)
C1608X7R1A105K
TDK
0603ZD225KAT2A
AVX(1)
GRM188R61A225KE34D
Murata(2)
(3)
C1608X5R1A225K
TDK
06035C472KAZ2A
AVX(1)
GRM188R71H472K
Murata(2)
B41851F7227M
(3)
TDK
EPCOS(4)
Open
SD103AWS-7
SD103AWS
L1
Murata(2)
TDK
SD103AWS
D1
Qty.
AVX(1)
C3225X5R0J107M
C1608X7R1H472K
C14
Description
Open
12106D107MAT2A
C5
Manufacturer
HCF1305-2R2-R
CRCW06032R21FKEA
CRCW06032R00FKEA
CRCW060319K6FKEA
CRCW06032K49FKEA
CRCW060320K0FKEA
MCC(5)
Diodes Inc(6)
(7)
Vishay
(7)
2.21Ω Resistor, Size 0603, 1%
1
(7)
2.00Ω Resistor, Size 0603, 1%
1
(7)
19.6kΩ Resistor, Size 0603, 1%
1
(7)
2.49kΩ Resistor, Size 0603, 1%
1
(7)
20.0kΩ Resistor, Size 0603, 1%
1
(7)
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
R6, R14, R17
CRCW060310K0FKEA
Vishay Dale
10.0kΩ Resistor, Size 0603, 1%
3
R7
CRCW06034K99FKEA
Vishay Dale(7)
4.99kΩ Resistor, Size 0603, 1%
1
CRCW06032K87FKEA
(7)
2.87kΩ Resistor, Size 0603, 1%
1
(7)
2.00kΩ Resistor, Size 0603, 1%
1
(7)
1.18kΩ Resistor, Size 0603, 1%
1
(7)
806Ω Resistor, Size 0603, 1%
1
(7)
475Ω Resistor, Size 0603, 1%
1
R8
R9
R10
R11
R12
July 2011
CRCW06032K006FKEA
CRCW06031K18FKEA
CRCW0603806RFKEA
CRCW0603475RFKEA
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
25
M9999-071311-A
Micrel, Inc.
MIC26603
Bill of Materials (Continued)
Item
R13
Part Number
CRCW06030000FKEA
Manufacturer
Description
Qty.
(7)
0Ω Resistor, Size 0603, 5%
1
(7)
Vishay Dale
R15
CRCW060349R9FKEA
Vishay Dale
49.9Ω Resistor, Size 0603, 1%
1
R16, R18
CRCW06031R21FKEA
Vishay Dale(7)
1.21Ω Resistor, Size 0603, 1%
2
R20
Open
U1
MIC26603YJL
Micrel. Inc.(9)
28V, 6A Hyper Light Load™ Synchronous
DC/DC Buck Regulator
1
Notes:
1.
AVX: www.avx.com.
2.
Murata: www.murata.com.
3.
TDK: www.tdk.com.
4.
EPCOS: www.epcos.com.
5.
SANYO: www.sanyo.com.
6.
Diode Inc.: www.diodes.com.
7.
Vishay: www.vishay.com.
8.
Cooper Bussmann: www.cooperbussmann.com.
9.
Micrel, Inc.: www.micrel.com.
July 2011
26
M9999-071311-A
Micrel, Inc.
MIC26603
PCB Layout Recommendations
Figure 10. MIC26603 Evaluation Board Top Layer
Figure 11. MIC26603 Evaluation Board Mid-Layer 1 (Ground Plane)
July 2011
27
M9999-071311-A
Micrel, Inc.
MIC26603
PCB Layout Recommendations (Continued)
Figure 12. MIC26603 Evaluation Board Mid-Layer 2
Figure 13. MIC26603 Evaluation Board Bottom Layer
July 2011
28
M9999-071311-A
Micrel, Inc.
MIC26603
Recommended Land and Solder Stencil Pattern
July 2011
29
M9999-071311-A
Micrel, Inc.
MIC26603
Package Information
28-Pin 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2011 Micrel, Incorporated.
July 2011
30
M9999-071311-A