Measurement and Control Circuit Collection

Application Note 45
June 1991
Measurement and Control Circuit Collection
Diapers and Designs on the Night Shift
Jim Williams
Introduction
During my wife’s pregnancy I wondered what it would
really be like when the baby was finally born. Before that
time, there just wasn’t much mothering and fathering to
do. As a consolation, we busied ourselves watching the
baby’s heartbeat (Figure 1) on a thrown-together fetal heart
monitor (see References).
feedings. As such, the circuits are annotated with the
number of feedings required for their completion; e.g., a
“3-bottle circuit” took three feedings. The circuit’s degree
of difficulty, and Michael’s degree of cooperation, combined
to determine the bottle rating, which is duly recorded in
each figure.
Low Noise and Drift Chopped Bipolar Amplifier
Figure 2’s circuit combines the low noise of an LT®1028
with a chopper based carrier modulation scheme to achieve
an extraordinarily low noise, low drift DC amplifier. DC
drift and noise performance exceed any currently available
monolithic amplifier. Offset is inside 1μV, with drift less
than 0.05μV/°C. Noise in a 10Hz bandwidth is less than
40nV, far below monolithic chopper-stabilized amplifiers.
A = 500μV/DIV
HORIZ = 500ms/DIV
(0.1Hz TO 30Hz BANDPASS)
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Figure 1. Michael’s Fetal Heartbeat 4 1/2 Months into Pregnancy
When Michael was born things got noticeably busier in
a hurry. My wife and I split up the evening duties. I got
the night shift, 2 am to 7 am. After a few weeks, Michael
and I got the hang of it and things began to go (relatively)
smoothly. The two of us had mastered feedings, naps, crying jags, bottles, diapers and such and we began looking
around for something to do. I decided to introduce Michael
to the glories of late night circuit hacking. I first learned
about wee hours circuit design at MIT in the 1970s. There
was a subculture there that loaded up on pizza, soft drinks,
and junk food, took it all into the lab, and closed the door
until long after daylight. I was an enthusiastic convert.
Michael and I changed the rules just a bit. We loaded up
on formula, diapers, and bottles and went into the lab.
The circuits in this collection represent our efforts, which
stopped when he (more or less) began sleeping through
the night. Most of the breadboarding occurred between
feedings, with design reviews and discussions during
Bias current, set by the bipolar LT1028 input, is about
25nA. These specifications suit demanding transducer
signal conditioning situations such as high resolution
scales and magnetic search coils.
The 74C04 inverters form a simple 2-phase square wave
clock running at about 350Hz. The oscillator provides
complementary drive to S1 and S2, causing A1 to see a
chopped version of the input voltage. A1 amplifies this
AC signal. A1’s square wave output is synchronously
demodulated by S3 and S4. Because these switches are
synchronously driven with the input chopper, proper
amplitude and polarity information is presented to A2,
the DC output amplifier. This stage integrates the square
wave into a DC voltage, providing the output. The output
is divided down (R2 and R1) and fed back to the input
chopper where it serves as a zero signal reference. Gain,
in this case 1000, is set by the R1-R2 ratio. Because A1
is AC-coupled, its DC offset and drift do not affect overall
circuit offset, resulting in the extremely low offset and
drift noted.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
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Figure 2. The Chopped Bipolar Amplifier. Noise Is Inside 40nV with 0.05μV/ °C Drift
Figure 3, a noise plot of the amplifier in a 0.1Hz to 10Hz
bandwidth, shows less than 40nV of peak-to-peak noise. A1
and the 60Ω resistance of S1-S2 contribute about equally
to form this noise. When using this amplifier it is important
to realize that A1’s bias current flowing through the input
source impedance causes additional noise. In general, to
maintain low noise performance, source resistance should
be kept below 500Ω. Fortunately, transducers such as
strain gauge bridges, RTDs, and magnetic detectors are
well below this figure.
50nV
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10 SECONDS
Figure 3. Noise in a 0.1Hz to 10Hz Bandwidth in Less Than 40nV
with 0.05μV/°C Drift
Low Noise and Drift-Chopped FET Amplifier
Figure 4’s circuit combines the low drift of a chopperstabilized amplifier with a pair of low noise FETs. The
result is an amplifier with 0.05μV/°C drift, offset within
5μV, 50pA bias current, and 200nV noise in a 0.1Hz to
10Hz bandwidth. The noise performance is especially
noteworthy; it is almost eight times better than monolithic
chopper-stabilized amplifiers.
FET pair Q1 differentially feeds A2 to form a simple low
noise op amp. Feedback, provided by R1 and R2, sets
closed-loop gain (in this case 1000) in the usual fashion.
Although Q1 has extraordinarily low noise characteristics,
its 15mV offset and 25μV/°C drift are poor. A1, a chopperstabilized amplifier, corrects these deficiencies. It does this
by measuring the difference between the amplifier’s inputs
and adjusting Q1A’s channel current to minimize the difference. Q1’s skewed drain values ensure that A1 will be
able to capture the offset. A1 supplies whatever current
is required into Q1A’s channel to force offset within 5μV.
Additionally, A1’s low bias current does not appreciably
add to the overall 50pA amplifier bias current. As shown,
the amplifier is set up for a noninverting gain of 1000,
although other gains and inverting operation are possible.
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Figure 4. Chopper-Stabilized FET Pair Combines Low Bias, Offset and Drift with 200nV Noise
Figure 5 is a plot of noise measured in a 0.1Hz to 10Hz
bandwidth. The performance obtained is almost an order
of magnitude better than any monolithic chopper-stabilized
amplifier, while retaining low offset and drift.
0.25μV
"/t'
10 SECONDS
Figure 5. Noise Performance for Figure 4. A1’s Low Offset and
Drift are Retained, but Noise Is Almost Ten Times Better
A2’s optional overcompensation can be used (capacitor to
ground) to optimize damping for low closed-loop gains.
Stabilized, Wideband Cable Driving Amplifier with
Low Input Capacitance
Figure 6’s amplifier has over 20MHz of small-signal
bandwidth driving 100mA loads, capacitance or cable.
Input capacitance is below 1.5pF and bias current about
100pA. The output is fully protected. These features make
this amplifier ideal as an ATE pin amplifier, video A/D
input buffer, or cable driver. The amplifier also permits
wideband probing when oscilloscope probe loading is not
tolerable. The overall amplifier is composed of a low input
capacitance FET, two LT1010 buffers, and a discrete gain
stage. A3 acts as a DC restoration loop. The 33Ω resistors
sense A1’s operating current, biasing Q3 and Q4. These
devices furnish complementary voltage gain to A2, which
provides the circuit’s output. Feedback is from A2’s output to A1’s output, which is a low impedance point. This
“current mode” feedback permits fixed bandwidth over
a wide range of closed-loop gains. This contrasts with
normal feedback schemes where bandwidth degrades as
closed-loop gain increases.
A3’s stabilizing loop compensates large offsets in the
signal path, which are dominated by mismatch in Q3 and
Q4. A3 measures the DC difference between the amplifier’s
input and output and biases the signal path to correct
for offset. Correction is implemented by controlling Q1’s
channel current via Q2. The channel current sets Q1’s VGS,
allowing A3 to control overall circuit offset. The 9k to 1k
feedback divider feeding A3 is selected to equal the gain
ratio of the circuit, in this case 10.
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Figure 6. Stabilized, Wideband Cable Driving Amplifier with Low Input Capacitance
The feedback scheme makes A1’s output look like the
negative input of the amplifier, with closed-loop gain set by
the ratio of the 470Ω and 51Ω resistors. The outstanding
feature of this connection is that the bandwidth becomes
relatively independent of closed-loop gain over a reasonable range. For this circuit, small-signal bandwidth exceeds
20MHz over gains of 1 to 20. The loop is quite stable, and
the 10pF value at A2’s input provides good damping over
a wide range of gains.
Figure 7 shows large-signal performance at a gain of 10
driving 10 feet of cable. A fast input pulse (trace A) produces the output shown (trace B). Response is quick and
clean with no slew residue or poor dynamics.
Voltage Programmable, Ground Referred Current
Source
Precise, voltage programmable, ground referred current
sources are usually complex and require trimming. Figure 8’s simple, powerful configuration produces output
current in strict accordance with the sign and magnitude of
the control voltage. Dynamic response is well controlled,
A = 0.5V/DIV
B = 5V/DIV
HORIZ = 100ns/DIV
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Figure 7. Wideband Amplifier’s Response Driving A 10 Foot Cable
and no trimming is required. The circuit’s accuracy and
stability are almost entirely dependent upon resistor R.
A1, biased by VIN, drives current through R (in this case
10Ω) and the load. Instrumentation amplifier A2, operating
at a gain of 100, senses across R. A2’s output closes a
loop back to A1. Because A1’s loop forces a fixed voltage
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Figure 8. Voltage Programmable Current Source
Is Simple and Precise
Figure 9. Current Source Dynamics Are Clean, with No Slew
Residue or Aberrations
across R, the current through the load is constant. The
10k-0.05μF combination sets A1’s roll-off, and the circuit
is stable.
filtered, and applied to the load. A3 senses load current
across the 16Ω shunt and drives T2’s center tap. Q9 and
Q10, receiving complementary drive picked off from T1’s
secondary, modulate T2’s DC center-tap voltage. T2’s
secondary receives this information, with flip-flop driven
Q6-Q7 demodulating it back to DC at T2’s center tap. T2’s
center tap voltage is fed A2, completing an isolated control
loop. Changes in the circuit’s input voltage cause this loop
to adjust the load current accordingly. Conversely, load
resistance changes have no effect, because the loop forces
whatever voltage is necessary to maintain a constant 16Ω
shunt voltage. Because T1 can supply up to 50V, load current remains fixed over load resistance swings from 0Ω
to 2500Ω. Power supply shifts are similarly rejected by
the loop, and the transformer modulation-demodulation
scheme permits 0.05% accuracy and stability over temperature and a 250V common mode range. Greater common
mode voltages are possible with increased transformer
breakdown ratings.
Assuming an errorless component for R, the circuit’s initial
error is dominated by A2’s 0.05% gain specification and
its 5ppm/°C temperature coefficient. High grade film or
wirewound resistors will maintain this level of performance.
Figure 9 shows dynamic response for a full-scale input
step. Trace A is the voltage control input while trace B
shows the output current. Response is clean, with no slew
residue or aberrations.
5V Powered, Fully Floating 4mA to 20mA Current
Loop Transmitter
4mA to 20mA current-loop transmitters are frequently
required in industrial process control. Often, because
of uncertain or dangerous common mode voltages, it
is desirable that the generated 4mA to 20mA current be
completely galvanically isolated from the transmitter’s
input. Figure 10’s circuit does this while operating from
a single 5V supply.
A2’s positive input assumes a bias dependent upon the
input and the 4mA trim setting. Under these conditions
A2’s output heads positive, turning on Q1 and Q2. Q2’s
collector drives T1’s primary, which is chopped by Q3 and
Q4. Complementary chopper drive comes from the 74C74
flip-flop outputs, with oscillator I1 setting a 25kHz clock
rate. T1’s output, producing voltage step-up, is rectified,
Several subtleties aid circuit performance. I2-I3 and I4-I5
provide drive delays to Q6 and Q7. These delays approximate the delay through T1 to modulator pair Q9/Q10. This
helps the four transistors switch simultaneously, aiding
modulator-demodulator accuracy. Zener connected Q5
ensures that T1 produces enough voltage to power A3 and
Q9/Q10, even when the load is 0Ω. Q8, similarly Zener
connected, clamps gate drive to Q9 and Q10, improving
modulator linearity by preventing excessive gate drive variations over operating conditions. The diodes in A3’s output
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Figure 10. 5V Powered, Fully Floating 4mA to 20mA Current Loop Transmitter
ensure proper loop start-up. They prevent T2’s center tap
from receiving any bias until A3 has enough power supply
voltage to function normally. To calibrate this circuit apply 0V input and adjust the 4mA trim for 4.00mA output
(0.064V across the 16Ω shunt). Next, apply 2.56V input
and set the 20mA trimmer for 20.00mA output (0.3200V
across the 16Ω shunt). Repeat this procedure until both
points are fixed. Note that the 2.56V input range is directly
compatible with D/A converter outputs, permitting digital
control.
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AN45-6
Application Note 45
Transistor ∆VBE Based Thermometer
The LTC®1043 contains switches whose state is controlled
by an on-chip oscillator. The 0.01μF capacitor at Pin 16
sets oscillator frequency at about 500Hz. Q1 operates as a
switched-value current source, alternating between about
10μA and 100μA (trace A, Figure 12) as the LTC1043 commutates switch Pin 12 and Pin 14. The two currents’ exact
value is unimportant, so long as their ratio remains constant. Because of this, Q1 requires no reference, although
its emitter resistor’s ratio is precise. The alternating 10μA
to 100μA stepped current to the sensor transistor (Q2)
Low cost makes transistors potentially attractive as temperature sensors. Almost all transistor-sensed thermometer circuits utilize the base-emitter diode voltage shift with
temperature as the sensing mechanism. Unfortunately,
the absolute diode voltage is unpredictable, necessitating
circuit calibration. Additionally, if the transistor sensor ever
requires replacement, the calibration must be repeated.
This constraint often negates the transistor sensor’s cost
and convenience advantages.
Figure 11’s transistor sensor thermometer overcomes
this difficulty. The circuit provides a 0V to 10V output
corresponding to a 0°C to 100°C temperature excursion
at the sensor transistor. Accuracy is ±1°C. No calibration
is required, and any common small-signal NPN transistor can serve as the sensor. The circuit is based on the
predictable relationship between current and voltage in
a transistor VBE junction.1 At room temperature, the VBE
junction diode shifts 59.16mV per decade of current. The
temperature dependence of this constant is 0.33%/°C, or
198μV/°C. This ΔVBE versus current relationship holds,
regardless of the VBE diode’s absolute value.
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Figure 12. Waveforms for the ∆VBE Based Thermometer
Note 1: See References 1 through 4.
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Application Note 45
Cold junction compensation is included, and accuracy is
within 1°C with stable 0.1°C resolution. Additionally, the
circuit functions from a single supply, which may range from
4.75V to 10V. Maximum current consumption is 360μA.
causes the theoretical 59.16mV (25°C) excursion (trace B)
to appear across the VBE junction. This signal is coupled to
a switched demodulator via C1, which strips off Q2’s DC
bias. LTC1043 switch Pin 2 (trace C) sees only the 59mV
waveform, which is referenced to ground via demodulator
action at Pin 5 and Pin 6. Pin 5, connected to capacitor C2,
sits at Pin 2’s DC peak value. A1 amplifies this DC signal,
with the LT1004 providing offset so 0°C equals 0V. The
optional 10k resistor protects against ESD events, which
may occur if Q2 is located at the end of a cable.
The LT1025 provides an appropriately scaled cold junction
compensation voltage to the type K thermocouple. As a
result, the voltage at schematic point “A” varies from 0mV
to 4.06mV over a sensed 0°C to 100°C range (type K slope
is 40.6μV/°C). The remaining components form a voltageto-frequency converter that directly converts this millivolt
level signal without the usual DC gain stage. A1’s negative
input is biased by the thermocouple. A1’s output drives a
crude V-F converter, comprised of Q2, the 74C14 inverters,
and associated components. Each V-F output pulse causes
a fixed quantity of charge to be dispensed into C3 from
C2 via the LTC201 based charge pump. C3 integrates the
charge packets, producing a voltage at A1’s positive input.
A1’s output forces the V-F converter to run at whatever
frequency is required to balance the amplifier’s inputs. This
feedback action eliminates drift and nonlinearities in the
V-F converter as an error term and the output frequency
Using the components shown, the circuit achieves ±1°C
accuracy over a sensed 0°C to 100°C range. Substituting
randomly selected 2N3904s and 2N2222s for Q2 showed
less than 0.4°C spread over 25 devices from various
manufacturers.
Micropower, Cold Junction Compensated
Thermocouple-to-Frequency Converter
Figure 13 is a complete, digital output, thermocouple signal
conditioner. The circuit produces a 0kHz to 1kHz output in
response to a sensed 0°C to 100°C temperature excursion.
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Application Note 45
is solely a function of the DC conditions at A1’s inputs.
The 0.02μF capacitor forms a dominant response pole at
A1, stabilizing the loop. Chopper stabilized A1’s low VOS
offset and drift eliminate offset error in the circuit, despite
an output LSB value of only 4.06μV (0.1°C).
Figure 14 details circuit operation. A1’s output biases current source Q2, producing a ramp (trace A, Figure 14) across
C1. When the ramp crosses I1’s threshold, the cascaded
inverter chain switches, producing complementary outputs
at I1 (trace B) and I2 (trace C). I3’s RC delayed response
(trace D) turns on diode connected Q1, discharging C1 and
resetting the ramp. The ramp aberrations before the reset
are due to transient I1 input currents during switching (near
top of ramp). Q1’s VBE diode rounding and reverse charge
transfer (bottom of ramp) account for the discontinuities
during the ramp’s low point.
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The complementary I1-I2 outputs clock the LTC201 switch
based charge pump. C2 is alternately charged to the
LT1004’s reference voltage via S1 and S4 and discharged
into C3 through S2 and S3. Each time this cycle occurs,
C3’s voltage is forced up (trace E). C3’s average voltage
is set by the 6.81k to 1.5k trimmer resistance across it.
A1 servo controls the repetition rate of the V-F to bring
its inputs to the same value, closing a control loop. The
0.02μF capacitor smooths A1’s response to DC.
To calibrate this circuit, disconnect the thermocouple and
drive point “A” with 4.06mV. Next, set the 1.5k trimmer
for exactly 1000Hz output. Connect the thermocouple and
the circuit is ready for use. Recalibration is not required
if the thermocouple is replaced.
It is worth noting that this circuit can directly digitize any
millivolt level signal by deleting the LT1025 thermocouple
pair and directly driving point “A.”
Relative Humidity Signal Conditioner
Relative humidity is a difficult physical parameter to
transduce, and most transducers require fairly complex
signal conditioning circuitry. Figure 15 combines simple
circuitry with a capacitively based transducer to achieve
good results. This circuit, which runs from a 9V battery,
is accurate within 2% in the 5% to 90% relative humidity
range.
The sensor specified has a nominal 500pF capacitance at
RH = 76%, with a slope of 1.7pF/% RH. The average voltage
across the device must be zero. This prevents deleterious
electrochemical migration in the sensor. LTC1043 section
“A,” driven by an internal oscillator, alternately charges
the sensor from a resistively scaled portion of the LT1004
reference and discharges it into A1’s summing point. Note
that the switching is arranged so that sensor related current
flows out of A1’s summing point. The 0.1μF series capacitor
ensures the sensor sees the required zero average voltage,
with the 22MΩ resistor preventing charge accumulation,
which would stop current flow. The average current out
of A1’s summing point is balanced by packets of charge
delivered by the LTC1043 switched capacitor section “C”
in A1’s feedback loop. The 0.1μF feedback capacitor gives
A1 an integrator-like response, and its output is DC. As
such, changes in sensor capacitance are seen as DC shifts
in A1’s output. A1 responds by raising its output positive
to whatever DC potential is required to maintain its summing point at zero.
To allow 0% RH to equal 0V, offsetting is required. The
signal and feedback terms biasing A1’s summing point
are expressed in charge form. Because of this, the offset
must also be delivered to the summing point as charge,
instead of a simple DC current. If this is not done, the circuit
will be affected by drift in the LTC1043’s internal oscillator. LTC1043 section “B” serves this function, delivering
LT1004 referenced offsetting charge to A1.
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Drift terms in this circuit include the LT1004 and the ratio
stability of the sensor and the polystyrene capacitors.
These terms are well within the sensor’s 2% accuracy
specification, and temperature compensation is not required. To calibrate this circuit, place the sensor in a 5% RH
environment and set the “5% RH trim” for 50mV output.
Next, place the sensor in a 90% RH environment and set
the “90% trim” for 900mV output. Repeat this procedure
until both points are fixed. If known RH environments are
unavailable, the capacitance versus RH table in Figure 15
may be utilized, although it applies for an ideal sensor. The
capacitor values may be built-up or directly dialed out on
a precision variable air capacitor (General Radio #722D).
Inexpensive Precision Electronic Barometer
Until recently, precision electronically based pressure measurements required expensive transducers. Capacitive and
bonded strain gauge based approaches provide unmatched
results, but costs are often prohibitive. Additionally, if low
power operation is desired, signal conditioning for these
devices can become complex.
Semiconductor based pressure transducers becoming
available offer significant improvement over earlier devices. Figure 16’s circuit utilizes such a device to form a
low cost barometer. The LT1027 reference and A1 form a
current source to put precisely 1.5mA through transducer
T1, in accordance with the manufacturer’s specifications.
Instrumentation amplifier A3 takes a differential gain of
10 from T1’s bridge output. A2 provides additional gain
to yield a calibrated output directly in inches of mercury.
T1’s manufacturer specifies a nominal 115mV at full scale,
although each device is supplied with precise calibration
data. This information considerably simplifies calibration.
To calibrate the circuit, simply adjust the potentiometer at
A1 until the output corresponds to the scale factor supplied with the unit.
This circuit, compared to a long column mercury barometer,
tracked ambient pressure variations from 29.75" to 30.32"
over three months with only two counts of uncertainty.
Additionally, over 50 turn-on/turn-off cycles had no measurable effect. Changes in pressure, particularly rapid ones,
correlated quite nicely to changing weather conditions.
1.5V Powered Radiation Detector
Figure 17’s circuit provides an audible “tick” signal each
time radiation or a cosmic ray passes through the detector.
The LT1073 switching regulator pulses T1. T1 takes gain via
its turns ratio and drives a voltage tripler, providing 500V
bias to the detector. R1 and R2 provide scaled feedback to
an45f
AN45-10
Application Note 45
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Figure 16. A Simple, Inexpensive Precision Barometer
C2
T1
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D1
Q1
2N3906
1N4148
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212mV
REF.
–
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VIN
+
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500V
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+
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–
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COILTRONICS CTX10052-1
68pF
PROJECTS UNLIMITED AT11K
600V
MUR1100
0.1μF, 200V
0.1μF, 400V
0.1μF, 600V
VICTOREEN SLIM-MOX-108
LND-712 LND CORP., OCEANSIDE, N.Y.
10M
DETECTOR
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Figure 17. 1.5V Powered Radiation Detector
an45f
AN45-11
Application Note 45
the LT1073, closing a control loop. The 0.01μF lag adds AC
hysteresis and the Schottky diode clamps negative going
T1 excursions. When radiation or a cosmic ray strikes
the detector, impedance drops briefly, transferring a quick
negative going spike through the 68pF capacitor. This spike
triggers the LT1073’s auxiliary gain block, configured here
as a comparator. Q1 and Q2 provide additional gain to
drive the audible beeper. About 10 to 15 cosmic rays per
minute are recorded in a normal environment.
This quartz stabilized 4kHz oscillator has less than 9ppm
(0.0009%) distortion in its 10VP-P output.
To understand circuit operation, temporarily assume A2’s
output is grounded. With the crystal removed, A1 and
the A3 power buffer form a noninverting amplifier with
a grounded input. The gain is set by the ratio of the 47k
resistor to the 50k potentiometer—opto-isolator pair.
Inserting the crystal closes a positive feedback path at
the crystal’s resonant frequency, and oscillations occur.
A4 compares A3’s positive peaks with the LT1004 2.5V
negative reference. The diode in series with the LT1004
provides temperature compensation for A3’s rectifier diode.
A4 biases the LED portion of the opto-isolator, controlling
9ppm Distortion, Quartz Stabilized Oscillator
A spectrally pure sine wave oscillator is required for data
converter, filter and audio testing. Figure 18 provides a
stable frequency output with extremely low distortion.
OUTPUT
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Figure 18. Quartz Stabilized 4kHz Oscillator with 9ppm Distortion
an45f
AN45-12
Application Note 45
the photoresistor’s resistance. This sets loop gain to a
value permitting stable amplitude oscillations. The 10μF
capacitor stabilizes this amplitude control loop.
A2’s function is to eliminate the common mode swing seen
by A1. This dramatically reduces distortion due to A1’s
common mode rejection limitations. A2 does this by servo
controlling the 560kΩ-photocell junction to maintain its
negative input at 0V. This action eliminates common mode
swing at A1, leaving only the desired differential signal.
Q1 and the LTC201 switch form a start-up loop. When power
is first applied oscillations may build very slowly. Under
these conditions A4’s output saturates positive, turning
on Q1. The LTC201 switch turns on, shunting the 2kΩ
resistor across the 50kΩ potentiometer. This raises A1’s
loop gain, forcing a rapid build-up of oscillations. When
oscillations rise high enough A4 comes out of saturation,
Q1 and the switch go off and the loop functions normally.
The circuit is adjusted for minimum distortion by adjusting the 50kΩ potentiometer while monitoring A3’s output
with a distortion analyzer. This trim sets the voltage across
the photocell to the optimum value for lowest distortion.
The circuit’s power supply should be well regulated and
bypassed to ensure the distortion figures quoted.
After trimming, A3’s output (trace A, Figure 19) contains
less than 9ppm (0.0009%) distortion. Residual distortion
components (trace B) include noise and second harmonic
residue. Oscillation frequency, set by crystal tolerance,
is typically within 50ppm with less than 2.5ppm/°C drift.
A = 5V/DIV
#QQN
DISTORTION
HORIZ = 100μs/DIV
"/t'
Figure 19. Oscillator Output and its 9ppm Distortion Residue
1.5V Powered Temperature Compensated Crystal
Oscillator
Many single cell systems require a stable clock source.
Crystal oscillators which run from 1.5V are relatively easy
to construct. However, if good stability over temperature
is required, things become more difficult. Ovenizing
the crystal is one approach, but power consumption is
excessive. An alternate method provides open loop, frequency correcting bias to the oscillator. The bias value is
determined by absolute temperature. In this fashion, the
oscillator’s thermal drift, which is repeatable, is corrected.
The simplest way to do this is by slightly varying the
crystal’s resonance point with a variable shunt or series
impedance. Varactor diodes, the capacitance of which
varies with reverse voltage, are commonly employed for
this purpose. Unfortunately, these diodes require volts
of reverse bias to generate significant capacitance shift,
making direct 1.5V powered operation impossible.
Figure 20 improves the temperature stability of a 1.5V
powered crystal oscillator by a factor of 20. It does this
by slightly tuning the crystal’s resonance as ambient
temperature varies. Q1 and associated components
form a 1MHz Colpitts oscillator which normally has a
temperature coefficient of about 1ppm/°C. The remainder
of the circuit implements the temperature correction.
The LM134 senses ambient temperature, converting it
to a current which flows through the 30.1k resistor. This
resistor’s voltage is subtracted from a reference potential
by A1. The stable subtraction voltage is derived from the
LT1073’s 212mV reference via Q2 and the 73.2k to 27.4k
resistors. Feedback from Q2’s collector to the LT1073’s
auxiliary amplifier closes the reference loop, which also
powers the Colpitts oscillator. The 47μF capacitor frequency
compensates the loop.
A1’s output controls the remaining portion of the LT1073,
which is configured as a voltage step-up switching regulator. L1’s high voltage inductive events are rectified and
stored in the 47μF output capacitor, resulting in a steppedup DC potential. This potential is fed back to A1, closing
a control loop. Because A1 is biased by the temperature
sensitive LM134, the loop’s output varies with ambient
temperature in a controlled manner. Q3’s drop forces the
step-up converter to always run, regardless of the loop’s
required output voltage. This permits smooth and continuous varactor bias from 0V to 3.9V over a 0°C to 70°C
an45f
AN45-13
Application Note 45
1N4148
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Figure 20. 1.5V Powered Temperature Compensated Crystal Oscillator
ambient operating environment. This output is applied to
the varactor diode in the oscillator circuit. The varactor’s
capacitance, a function of its DC bias, thus varies with
ambient temperature. This change in capacitance shifts
the crystal’s resonant frequency, opposing temperature
induced crystal drift. For the values given in the circuit
and the crystal cut specified, residual oscillator drift is
only 0.05ppm/°C. This compares favorably with 1ppm/°C
drift with no compensation used. The circuit functions
from 1.7V down to 1.1V with no specification degradation. Current drain is only 230μA. Applications include
portable high accuracy clocks, survival radios, and secure
communications.
90μA Precision Voltage-to-Frequency Converter
Figure 21 is a micropower voltage-to-frequency converter.
A 0V to 5V input produces a 0kHz to 10kHz output with a
linearity of 0.05%. Gain drift is 80ppm/°C. Maximum current consumption is only 90μA, almost 30 times lower than
currently available V-F converters. To understand circuit
operation, assume C1’s positive input is slightly below
its negative input (C2’s output is low). The input voltage
causes a positive going ramp at C1’s positive input (trace
A, Figure 22). C1’s output is low, biasing the CMOS inverter
output high. This allows current to flow from Q1’s emitter,
through the inverter supply pin to the 100pF capacitor. The
an45f
AN45-14
Application Note 45
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0V TO 5V
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Figure 21. V-to-F Converter Achieves 0.05% Linearity While Requiring Only 90μA Supply Current
2.2μF capacitor provides high frequency bypass, maintaining low impedance at Q1’s emitter. Diode connected
Q6 provides a path to ground. The 100pF unit charges to
a voltage that is a function of Q1’s emitter potential and
Q6’s drop. When the ramp at C1’s positive input goes high
enough, C1’s output goes high (trace B) and the inverter
switches low (trace C). The Schottky clamp prevents
CMOS inverter input overdrive. This action pulls current
from C1’s positive input capacitor via the Q5-100pF route
(trace D). This current removal resets C1’s positive input
ramp to a potential slightly below ground, forcing C1’s
output to go low. The 50pF capacitor furnishes AC positive
feedback, ensuring that C1’s output remains positive long
enough for a complete discharge of the 100pF capacitor.
The Schottky diode prevents C1’s input from being driven
outside its negative common mode limit. When the 50pF
unit’s feedback decays, C1 again switches low and the
entire cycle repeats. The oscillation frequency depends
directly on the input voltage derived current.
A = 50mV/DIV
B = 5V/DIV
C = 5V/DIV
D = 1mA/DIV
"/t'
HORIZ = 20μs/DIV
Figure 22. Micropower V-to-F Converter's Waveforms
an45f
AN45-15
Application Note 45
Q1’s emitter voltage must be carefully controlled to get
low drive. Q3 and Q4 temperature compensate Q5 and Q6
while Q2 compensates Q1’s VBE. The two LT1034s are the
actual voltage reference and the LM334 current source
provides 35μA bias to the stack. The current drive provides
excellent supply immunity (better than 40ppm/V) and also
aids circuit temperature coefficient. It does this by utilizing
the LM334’s 0.3%/°C temperature coefficient to slightly
temperature modulate the voltage drop in the Q2-Q4 trio.
This correction’s sign and magnitude directly oppose that
of the –120ppm/°C, 100pF polystyrene capacitor, aiding
overall circuit stability.
The Q1 emitter-follower efficiently delivers charge to the
100pF capacitor. Both base and collector current end up in
the capacitor. The CMOS inverter provides low loss SPDT
reference switching without significant drive losses. The
100pF capacitor draws only small transient currents during
its charge and discharge cycles. The 50pF-47k positive
feedback combination draws insignificantly small switching currents. Figure 23, a plot of supply current versus
operating frequency, reflects the low power design. At zero
frequency, the LT1017’s quiescent current and the 35μA
reference stack bias account for all current drain. There
are no other paths for loss. As frequency scales up, the
charge/discharge cycle of the 100pF capacitor introduces
the 1.5μA/kHz increase shown.
100
CURRENT CONSUMPTION (μA)
90
80
SLOPE = 1.5μA/kHz
70
60
50
40
30
20
10
0
0 1
2
3
4
5
6
7
8
FREQUENCY (kHz)
9 10 11 12
"/t'
Figure 23. Current Consumption vs Frequency for the V-to-F
Converter
Circuit start-up or overdrive can cause the circuit’s ACcoupled feedback to latch. If this occurs, C1’s output
goes high. C2, detecting this via the inverter and the
2.7M-0.1μF lag, also goes high. This lifts C1’s negative
input and grounds the positive input with Q7, initiating
normal circuit action.
Because the charge pump is directly coupled to C1’s output,
response is fast. The output settles within one cycle for a
fast input step. To calibrate this circuit, apply 50mV and
select the value at C1’s input for a 100Hz output. Then, apply
5V and trim the input potentiometer for a 10kHz output.
Bipolar (AC) Input V-F Converter
No currently available V-F converter will accept bipolar
(AC) inputs. This feature is desirable in power line monitoring and other applications. Figure 24’s V-F converter
accepts ±10V inputs, producing a 0kHz to 10kHz output.
Linearity is 0.04%, and temperature coefficient measures
about 50ppm/°C. To understand circuit operation, assume
a bipolar square wave (trace A, Figure 25) is applied to
the input. During the input’s positive phase, A1’s output
(trace B) swings negative, driving current through C1 via
the full wave diode bridge. A1’s current causes C1 to ramp
linearly. Instrumentation amplifier A2, operating at a gain
of 10, looks differentially across C1. A2’s output (trace C)
biases comparator A3’s negative input. When A2’s output
crosses zero, A3 fires (trace D). AC positive feedback to
A3’s positive input (trace E) “hangs up” A3’s output for
about 20μs. The Q1 level shifter drives ground referred
inverters I1 and I2 to deliver biphase drive (traces G and
H) to the LTC201 switch. The LTC201, set up as a charge
pump, places C2 across C1 each time the inverters switch,
resetting C1 to a lower voltage. The LT1004 reference, along
with C2’s value, determines how much charge is removed
from C1 each time the charge pump cycles. Thus, each
time A2’s output tries to cross zero, C2 is switched across
C1, resetting it to a small negative voltage and forcing A1
to begin recharging it. The frequency of this oscillatory
behavior is directly proportional to the input derived current
into A1. During the time C1 is ramping toward zero the
LTC201 switches C2 across the LT1004, preparing it for the
next discharge cycle. The action is the same for negative
input excursions (see Figure 25), except that A1’s output
phasing is reversed. A2, looking differentially across A1’s
diode bridge, sees the same signal as for positive inputs
and circuit action is identical. A4, detecting A1’s output
polarity, provides a sign bit output (trace F).
an45f
AN45-16
Application Note 45
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Figure 24. Bipolar (AC) Input V-to-F Converter
Figure 26, an amplitude expanded version of A1 and A2’s
outputs, shows detail. Trace A is the input, while trace B
and trace C are A1 and A2’s outputs, respectively. Complementary bias points and ramping action are clearly visible
in A1’s output, while A2 responds identically for both input
phases. A1’s output bias points are established by the two
conducting bridge diodes. When the input switches polarity,
A1 responds immediately and oscillation frequency settles
within 1 to 2 cycles of final value.
"7%*7
B = 1V/DIV
C = 0.5V/DIV
D = 100V/DIV
E = 50V/DIV
'7%*7
G = 50V/DIV
H = 50V/DIV
HORIZ = 500μs/DIV
"/t'
Figure 25. Waveforms for the Bipolar Input V-to-F Converter
Start-up or overdrive conditions could cause this loop to
latch. A start-up mechanism, adapted from oscilloscope
trigger circuitry, precludes latch-up.2 If C1 charges past
the point where C2 can reset it, loop closure ceases. A2’s
Note 2: See References 5 and 6.
an45f
AN45-17
Application Note 45
output saturates positive, causing A3 to go negative. A3’s
prolonged negative state, detected by the R1-C3 filter, pulls
its negative input toward –15V. When A3’s negative input
crosses zero, its output changes state and charges R1-C3
positively. A3’s input rises above zero, causing output
reversal and free-running oscillation commences. As in
normal mode, the 100pF-33k RC aids transitions. A3’s
oscillations are transmitted to the LTC201 based charge
pump via A1 and the inverters. C2 pumps charge out of
C1, driving the voltage across it toward zero. A2 comes
out of positive saturation and heads negative, eliminating
A = 50V/DIV
B = 0.5V/DIV
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HORIZ = 500μs/DIV
Figure 26. Detail of Integrator and Differential Amplifier Outputs
positive bias at A3’s input. A3’s free-running oscillation
stops, and normal loop action begins.
To calibrate this circuit apply either a –10V or a +10V input
and set the 10kΩ trimmer for exactly 10kHz output. The
low offsets of A1 and A2 permit operation down to a few
hertz with no zero trim required.
1.5V Powered, 350ps Rise Time Pulse Generator
Verifying the rise time limit of wideband test equipment
setups is a difficult task. In particular, the “end-to-end”
rise time of oscilloscope-probe combinations is often
required to assure measurement integrity. Conceptually,
a pulse generator with rise times substantially faster than
the oscilloscope-probe combination can provide this information. Figure 27’s circuit does this, providing a 1ns
pulse with rise and fall times inside 350ps. Pulse amplitude
is 10V with a 50Ω source impedance. This circuit, built
into a small box and powered by a 1.5V battery, provides
a simple, convenient way to verify the rise time capability
of almost any oscilloscope-probe combination.
The LT1073 switching regulator and associated components supply the necessary high voltage. The LT1073 forms
a flyback voltage boost regulator. Further voltage step-up is
obtained from a diode-capacitor voltage doubler network.
L1 periodically receives charge, and its flyback discharge
+90V AVALANCHE BIAS
3
1M
+1.5V
L1
150μH
Ω
IL
VIN
SW1
R1
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an45f
AN45-18
Application Note 45
delivers high voltage events to the doubler network. A
portion of the doubler network’s DC output is fed back to
the LT1073 via the R1, R2 divider, closing a control loop.
The regulator’s 90V output is applied to Q1 via the R3-C1
combination. Q1, a 40V breakdown device, non-destructively avalanches when C1 charges high enough.3 The
result is a quickly rising, very fast pulse across R4. C1
discharges, Q1’s collector voltage falls and breakdown
ceases. C1 then recharges until breakdown again occurs.
This action causes free-running oscillation at about 200kHz.
Figure 28 shows the output pulse. A 1GHz sampling
oscilloscope (Tektronix 556 with 1S1 sampling plug-in)
measures the pulse at 10V high with about a 1ns base.
Rise time is 350ps, with fall time also indicating 350ps.
The figures may actually be faster, as the 1S1 is specified
with a 350ps rise time limit.4
VIN 7507
+
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Figure 29. Alternate 90V DC-DC Converter
VERT = 2V/DIV
HORIZ = 200ps/DIV
"/t'
Figure 28. Avalanche Pulse Generator Output Pulse. Waveform
Has 350ps Rise and Fall Times. Slightly Under Damped Turn-Off
Is Probably Due to Test Fixture Limitations
Q1 may require selection to get avalanche behavior. Such
behavior, while characteristic of the device specified, is not
guaranteed by the manufacturer. A sample of 50 Motorola
2N2369s, spread over a 12 year date code span, yielded
82%. All “good” devices switched in less than 600ps. C1
is selected for a 10V amplitude output. Value spread is
typically 2pF to 4pF. Ground plane type construction with
high speed layout techniques are essential for good results
from this circuit. Current drain from the 1.5V battery version is about 5mA.
Note 3: See References 7.
Note 4: I’m sorry, but 1GHz is the fastest scope in my house.
For those applications which must run from higher voltage
inputs, Figure 29 is included. This circuit, which operates
from inputs of 4V to 20V, will also power the avalanche
stage. Cascoded high voltage transistor Q1 combines
with the LT1072 switching regulator to form a high voltage switched mode control loop. The LT1072 pulse width
modulates Q1 at its 40kHz clock rate. L1’s inductive events
are rectified and stored in the 2μF output capacitor. The
1MΩ to 12kΩ divider provides feedback to the LT1072. The
diode and RC at Q1’s base damp inductor related parasitic
behavior. The circuit’s output drives the avalanche stage
in similar fashion to the LT1073 based circuit.
A Simple Ultralow Dropout Regulator
Switching regulator post regulators, battery-powered
apparatus, and other applications frequently require low
dropout linear regulators. Often, battery life is significantly
affected by the regulator’s dropout performance. Figure 30’s simple circuit offers lower dropout voltage than
any monolithic regulator. Dropout is below 50mV at 1A,
increasing to only 450mA at 5A. Line and load regulation
are within 5mV, and initial output accuracy is inside 1%.
an45f
AN45-19
Application Note 45
Circuit operation is straightforward. The 3-pin LT1123
regulator (TO-92 package) servo controls Q1’s base to
maintain its feedback pin (FB) at 5V. The 10μF output
capacitor provides frequency compensation. If the circuit
is located more than six inches from the input source,
the optional 10μF capacitor should bypass the input. The
optional 20Ω resistor limits LT1123 power dissipation
and is selected based upon the maximum expected input
voltage (see Figure 31).
Normally, configurations of this type offer unpredictable
short-circuit protection. Here, the MJE1123 transistor
Q1
MJE1123
INPUT
+
+5VOUT
+
10μF*
600Ω
10μF
20Ω*
DRIVE
LT1123 FB
GND
* = OPTIONAL (SEE TEXT)
MJE1123 = MOTOROLA
shown has been specially designed for use with the LT1123.
Because of this, beta based current limiting is practical.
Excessive output current causes the LT1123 to pull down
harder on Q1 until beta limiting occurs. Under these conditions the controlled pull-down current combines with Q1’s
beta and safe operating area characteristics to provide
reliable short-circuit limiting. Figure 32 details current
limit characteristics for 30 randomly selected transistors.
Figure 33 shows dropout characteristics. Even at 5A,
dropout is about 450mV, decreasing to only 50mV at 1A.
Monolithic regulators cannot approach these figures, primarily because monolithic power transistors do not offer
Q1’s combination of high beta and excellent saturation. For
POWER LIMITING RESISTOR VALUE IN OHMS
Additionally, the regulator is fully short-circuit protected,
and has a no load quiescent current of 600μA.
110
100
90
80
70
60
50
40
ASSUME: VBE = 0.9V
7-57
SAT
IDRIVE MAX = 150mA
20
10
5
"/t'
10
15
20
INPUT VOLTAGE (V)
"/t'
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
4.00 4.25 4.50 4.75 5.00 5.25 5.50 5.75 6.00
Figure 31. LT1123 Power Dissipation Limiting Resistor Value
vs Input Voltage
0.75
0.60
DROPOUT VOLTAGE (V)
NUMBER OF UNITS
Figure 30. The Ultralow Dropout Regulator. LT1123
Combines with Specially Designed Transistor for
Low Dropout and Short-Circuit Protection
0.45
0.15
0
0
1
2
4
5
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
"/t'
Figure 32. Short Circuit Current for 30 Randomly Selected
MJE1123 Transistors at VIN = 7V
"/t'
Figure 33. Dropout Voltage vs Output Current
an45f
AN45-20
Application Note 45
Cold Cathode Fluorescent Lamp Power Supply
3.0
Current generation portable computers utilize back-lit
LCD displays. Cold Cathode Fluorescent Lamps (CCFL)
provide the highest available efficiency for back lighting the
display. These lamps require high voltage AC to operate,
mandating an efficient, high voltage DC/AC converter. In
addition to good efficiency, the converter should deliver
the lamp drive in the form of a sine wave. This is desirable
to minimize RF emissions. Such emissions can cause interference with other devices, as well as degrading overall
operating efficiency.
DROPOUT VOLTAGE (V)
2.5
LT138
2.0
1.5
LT1084
1.0
LT1185
0.5
LT1123/2N4276
LT1123/MJE1123
0
0
1
3
2
4
5
OUTPUT CURRENT (A)
"/t'
Figure 34. Dropout Voltage vs Output Current for Various
Regulators
comparison, Figure 34 compares the circuit’s performance
against some popular monolithic regulators. Dropout is
10 times better than 138 types, and significantly better
than the other types shown. Because of Q1’s high beta,
base drive loss is only 1% to 2% of output current, even
at full 5A output. This maintains high efficiency under the
low VIN – VOUT conditions the circuit will typically operate at. As an exercise, the MJE1123 was replaced with a
2N4276, a Germanium device. This combination provided
even lower dropout performance, although current limit
characteristics cannot be guaranteed.
Figure 35 shows a simple way to add shutdown to the
regulator. A CMOS inverter or gate biases Q2 to control
LT1123 bias. When Q2’s base is driven, the loop functions
normally. With Q2 unbiased, the circuit goes into shutdown
and pulls no current.
Q1
MJE1123
INPUT
5.1k
SHUTDOWN
CMOS
INVERTER
OR GATE
9
7
D1
1N4148
L1
5
1
2
+VIN
3
4
+
10μF
D2
1N4148
C1
0.02μF
Q1
MPS650
1k
L2 562Ω*
300μH
5
6
8
Q2
MPS650
1N5818
+VIN
4.5V TO +20V
VIN
E1
VSW
E2
VFB
GND
10μF
LAMP
33pF
3kV
7
LT1072
+5VOUT
+
300Ω
Figure 36 meets these requirements. Efficiency is 78%,
with an input voltage range of 4.5V to 20V. 82% efficiency
is possible if the LT1072 is driven from a low voltage (e.g.,
3V to 5V) source. Additionally, lamp intensity is continuously and smoothly variable from zero to full intensity.
1
VC
+
10k
50kΩ
INTENSITY
ADJUST
3
2
+
2μF
1μF
Q2
MPSA12
"/t'
DRIVE
LT1123 FB
GND
C1 = MUST BE A LOW LOSS CAPACITOR.
METALIZED POLYCARB
WIMA FKP2 (GERMAN) RECOMMENDED.
L1 = SUMIDA-6345-020 OR COILTRONICS-CTX110092-1.
PIN NUMBERS SHOWN FOR COILTRONICS UNIT
L2 = COILTRONICS-CTX300-4
* = 1% FILM RESISTOR
DO NOT SUBSTITUTE COMPONENTS
"/t'
Figure 35. Shutdown for the Low Dropout Regulator
Figure 36. Cold Cathode Fluorescent
Lamp Power Supply
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AN45-21
Application Note 45
When power is applied the LT1072 switching regulator’s
feedback pin is below the devices internal 1.23V reference,
causing full duty cycle modulation at the VSW pin (trace A,
Figure 37). L2 conducts current (trace B), which flows
A = 20V/DIV
B = 0.4A/DIV
C = 20V/DIV
D = 20V/DIV
E = 1000V/DIV
F = 5V/DIV
A AND B HORIZ =10μs/DIV
C THRU F HORIZ = 20μs/DIV
TRIGGERS FULLY INDEPENDENT
"/t'
Figure 37. Waveforms for the Cold Cathode Fluorescent Lamp
Power Supply. Note Independent Triggering on Traces A and B
and C Through F.
from L1’s center tap, through the transistors, into L2. L2’s
current is deposited in switched fashion to ground by the
regulator’s action.
L1 and the transistors comprise a current driven Royer
class converter5 which oscillates at a frequency primarily set by L1’s characteristics and the 0.02μF capacitor.
LT1072 driven L2 sets the magnitude of the Q1-Q2 tail
current, and hence L1’s drive level. The 1N5818 diode
maintains L2’s current flow when the LT1072 is off. The
LT1072’s 40kHz clock rate is asynchronous from the
Royer converters (≈60kHz) rate, accounting for trace B’s
waveform thickening.
The 0.02μF capacitor combines with L1’s characteristics
to produce sine wave voltage drive at the Q1 and Q2 collectors (traces C and D, respectively). L1 furnishes voltage
step-up, and about 1400VP-P appears at its secondary
(trace E). Current flows through the 33pF capacitor into
the lamp. On negative waveform cycles the lamp’s current
is steered to ground via D1. Positive waveform cycles are
directed, via D2, to the ground referred 562Ω-50k potentiometer chain. The positive half-sine appearing across these
resistors (trace F) represents 1/2 the lamp current. This
signal is filtered by the 10k-1μF pair and presented to the
LT1072’s feedback pin. This connection closes a control
loop which regulates lamp current. The 2μF capacitor at
the LT1072’s VC pin provides stable loop compensation.
The loop forces the LT1072 to switch-mode modulate L2’s
average current to whatever value is required to maintain
a constant current in the lamp. The constant current’s
value, and hence lamp intensity, may be varied with the
potentiometer. The constant current drive allows full 0% to
100% intensity control with no lamp dead zones or “popon” at low intensities. Additionally, lamp life is enhanced
because current cannot increase as the lamp ages.
Several points should be kept in mind when observing this
circuit’s operation. L1’s high voltage secondary can only
be monitored with a wideband, high voltage probe fully
specified for this type of measurement. The vast majority
of oscilloscope probes will break down and fail if used for
this measurement.6 Tektronix probe type P-6009 (acceptable) or types P6013A and P6015 (preferred) probes must
be used to read L1’s output.
Another consideration involves observing waveforms. The
LT1072’s switching frequency is completely asynchronous
from the Q1-Q2 Royer converter’s switching. As such,
most oscilloscopes cannot simultaneously trigger and
display all the circuit’s waveforms. Figure 37 was obtained
using a dual beam oscilloscope (Tektronix 556). LT1072
related traces A and B are triggered on one beam, while
the remaining traces are triggered on the other beam.
Single beam instruments with alternate sweep and trigger
switching (e.g., Tektronix 547) can also be used, but are
less versatile and restricted to four traces.
Note 5: See References 8.
Note 6: Don’t say we didn’t warn you!
an45f
AN45-22
Application Note 45
References
1. Verster, T.C., “P-N Junction as an Ultralinear Calculable
Thermometer,” Electronic Letters, Vol. 4, pg. 175, May,
1968.
2. Verster, T.C., “The Silicon Transistor as a Temperature
Sensor,” International Symposium on Temperature,
1971, Washington, D.C.
3. Type 7D13 Plug-In Operating and Service Manual,
Tektronix, Inc., 1971.
4. Sheingold, D.H., “Nonlinear Circuits Handbook,”
Chapter 3-1, “Basic Considerations,” pgs. 165-166,
Analog Devices, Inc., 1974.
5. Oscilloscope Trigger Circuits, “Automatic Trigger,” pgs.
39-49, Tektronix Concept Series, 1969.
6. Type 547 Oscilloscope Operating and Service Manual,
“Automatic Stability Circuit,” pgs. 3-8, Tektronix, Inc.,
1964.
7. Type 111 Pretrigger Pulse Generator Operating and
Service Manual, Tektronix, Inc., 1960.
8. Bright, Pittman and Royer, “Transistors As On-Off
Switches in Saturable Core Circuits,” Electrical
Manufacturing, December, 1954.
9. Morrison, John C., MD, Editor. “Antepartal Fetal
Surveillance,” Obstetrics and Gynecology Clinics
of North America, Volume 17:1, March, 1990, W.B.
Saunders Co.
10. Atkinson, P., Woodcock, J.P., “Doppler Ultrasound,”
London, Academic Press, 1982.
11. Doppler, J.C., “Uber das farbigte Licht der
Dopplersterne und einigr anderer Gestirne des
Himmels,” Abhandl d Konigl Bomischen Gesellschaft
der Wissenschaften 2:466, 1843.
12. FitzGerald, D.E., Drumm, J.E., “Noninvasive
measurement of fetal circulation using ultrasound:
A new method,” Br Med J 2:1450, 1977.
13. Hata, T., Aoki, S., Hata, K., et al, “Intracardiac blood
flow velocity waveforms in normal fetuses in utero,”
Am J Cardiol 58:464, 1987.
14. Pourcelot, L., “Applications clinique de l’examen
Doppler transcutane,” In Pourcelot, L. (ed),
Velocimetric Ultrasonore Doppler, INSERM 34:213,
1974.
15. Shung, K.K., Sigelman, R.A., Reid, J.M., “Scattering
of ultrasound by blood,” IEEE Trans Biomed Eng
BME-23:460, 1976.
16. Stuart, B., Drumm, J., FitzGerald, D.E., et al, “Fetal
blood velocity waveforms in normal pregnancy,” Br
J Obstet Gynaecol 87:780, 1980.
17. Stabile, I., Bilardo, C., Panella, M., et al, “Doppler
measurement of uterine blood flow in the first
trimester of normal and complicated pregnancies,”
Trophoblast Res 3:301, 1988.
an45f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN45-23
Application Note 45
an45f
AN45-24
Linear Technology Corporation
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