DS6237AB 04

®
RT6237A/B
7A, 18V, 500kHz, ACOTTM Synchronous Step-Down Converter
General Description
Features
The RT6237A/B is a high-performance 500kHz, 7A stepdown regulator with internal power switches and
synchronous rectifiers. It features quick transient response
using its Advanced Constant On-Time (ACOTTM) control

architecture that provides stable operation with small
ceramic output capacitors and without complicated
external compensation, among other benefits. The input
voltage range is from 4.5V to 18V and the output is
adjustable from 0.7V to 8V. The proprietary ACOTTM control
improves upon other fast response constant on-time
architectures, achieving nearly constant switching
frequency over line, load, and output voltage ranges. Since
there is no internal clock, response to transients is nearly
instantaneous and inductor current can ramp quickly to
maintain output regulation without large bulk output
capacitance. The RT6237A/B is stable with and optimized
for ceramic output capacitors. With internal 40mΩ switches
and 16mΩ synchronous rectifiers, the RT6237A/B displays
excellent efficiency and good behavior across a range of
applications, especially for low output voltages and low
duty cycles. Cycle-by-cycle current limit provides
protection against shorted outputs, input under-voltage
lockout, externally-adjustable soft-start, output under- and
over-voltage protection, and thermal shutdown provide safe
and smooth operation in all operating conditions. The
RT6237A/B is available in the UQFN-14L 2x3 (FC)
package, with exposed thermal pad.
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Fast Transient Response
Advanced Constant On-Time (ACOTTM) Control
4.5V to 18V Input Voltage Range
Adjustable Output Voltage from 0.7V to 8V
7A Output Current
40mΩ
Ω Internal High-Side N-MOSFET and 16mΩ
Ω
Internal Low-Side N-MOSFET
Steady 500kHz Switching Frequency
Up to 95% Efficiency
Optimized for All Ceramic Capacitors
Externally-Adjustable, Pre-Biased Compatible SoftStart
Cycle-by-Cycle Current Limit
Input Under-Voltage Lockout
Output Over- and Under-Voltage Protection
Power Good Output
Thermal Shutdown
Applications
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Industrial and Commercial Low Power Systems
Computer Peripherals
LCD Monitors and TVs
Green Electronics/Appliances
Point of Load Regulation for High-Performance DSPs,
FPGAs, and ASICs
Simplified Application Circuit
VIN
EN Signal
Power Good
RT6237A/B
VIN
SW
EN
VOUT
BOOT
FB
PGOOD
PVCC
SS
GND
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
DS6237A/B-04 May 2016
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
1
RT6237A/B
Ordering Information
Marking Information
RT6237A/B
RT6237ALGQUF
09 : Product Code
Package Type
QUF : UQFN-14L 2x3 (U-Type) (FC)
09W
W : Date Code
Lead Plating System
G : Green (Halogen Free and Pb Free)
UVP Option
H : Hiccup Mode UVP
L : Latched OVP & UVP
RT6237BLGQUF
07 : Product Code
07W
W : Date Code
A : PSM
B : PWM
Note :
RT6237AHGQUF
0A : Product Code
Richtek products are :

RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.

0AW
W : Date Code
Suitable for use in SnPb or Pb-free soldering processes.
RT6237BHGQUF
08 : Product Code
Pin Configurations
SS
EN
GND
(TOP VIEW)
14
13
12
08W
GND
FB
2
10
GND
PVCC
3
9
GND
PGOOD
4
8
VIN
5
6
7
VIN
11
SW
1
BOOT
AGND
W : Date Code
UQFN-14L 2x3 (FC)
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is a registered trademark of Richtek Technology Corporation.
DS6237A/B-04 May 2016
RT6237A/B
Functional Pin Description
Pin No.
Pin Name
Pin Function
1
AGND
Analog GND.
2
FB
Feedback Voltage Input. It is used to regulate the output of the converter to a set
value via an external resistive voltage divider. The feedback reference voltage is
0.7V typically.
3
PVCC
Internal Regulator Output. Connect a 1F capacitor to GND to stabilize output
voltage.
4
PGOOD
Power Good Indicator Open-Drain Output.
5
BOOT
Bootstrap Supply for High-Side Gate Driver. This capacitor is needed to drive the
power switch's gate above the supply voltage. It is connected between the SW and
BOOT pins to form a floating supply across the power switch driver. A 0.1F
capacitor is recommended for use.
6
SW
Switch Node. Connect this pin to an external L-C filter.
7, 8
VIN
Power Input. The input voltage range is from 4.5V to 18V. Must bypass with a
suitably large (10F x 2) ceramic capacitor.
9, 10, 11, 12
GND
Ground.
13
EN
Enable Control Input. A logic-high enables the converter; a logic-low forces the IC
into shutdown mode reducing the supply current to less than 10A.
14
SS
Soft-Start Time Setting. An external capacitor should be connected between this
pin and GND.
Function Block Diagram
BOOT
PVCC
VIN
PVCC
Reg
Min.
Off
VIBIAS
PVCC
VIN
VREF
UGATE
Control
OC
Driver
SW
LGATE
UV & OV
PVCC
SW
6µA
Ripple
Gen.
SS
FB
VIN
SW
GND
GND SW
+
Comparator
On-Time
Comparator
0.9 VREF
FB
PGOOD
+
-
EN
EN
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
DS6237A/B-04 May 2016
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
3
RT6237A/B
Detailed Description
The RT6237A/B is a high-performance 500kHz 7A stepdown regulators with internal power switches and
synchronous rectifiers. It features an Advanced Constant
On-Time (ACOTTM) control architecture that provides
stable operation with ceramic output capacitors without
complicated external compensation, among other benefits.
The ACOTTM control mode also provides fast transient
response, especially for low output voltages and low duty
cycles.
The input voltage range is from 4.5V to 18V and the output
is adjustable from 0.7V to 8V. The proprietary ACOTTM
control scheme improves upon other constant on-time
architectures, achieving nearly constant switching
frequency over line, load, and output voltage ranges. The
RT6237A/B are optimized for ceramic output capacitors.
Since there is no internal clock, response to transients is
nearly instantaneous and inductor current can ramp quickly
to maintain output regulation without large bulk output
capacitance.
Constant On-Time (COT) Control
The heart of any COT architecture is the on-time one shot.
Each on-time is a pre-determined “fixed” period that is
triggered by a feedback comparator. This robust
arrangement has high noise immunity and is ideal for low
duty cycle applications. After the on-time one-shot period,
there is a minimum off-time period before any further
regulation decisions can be considered. This arrangement
avoids the need to make any decisions during the noisy
time periods just after switching events, when the
switching node (SW) rises or falls. Because there is no
fixed clock, the high-side switch can turn on almost
immediately after load transients and further switching
pulses can ramp the inductor current higher to meet load
requirements with minimal delays.
Traditional current mode or voltage mode control schemes
typically must monitor the feedback voltage, current
signals (also for current limit), and internal ramps and
compensation signals, to determine when to turn off the
high-side switch and turn on the synchronous rectifier.
Weighing these small signals in a switching environment
is difficult to do just after switching large currents, making
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4
those architectures problematic at low duty cycles and in
less than ideal board layouts.
Because no switching decisions are made during noisy
time periods, COT architectures are preferable in low duty
cycle and noisy applications. However, traditional COT
control schemes suffer from some disadvantages that
preclude their use in many cases. Many applications require
a known switching frequency range to avoid interference
with other sensitive circuitry. True constant on-time control,
where the on-time is actually fixed, exhibits variable
switching frequency. In a step-down converter, the duty
factor is proportional to the output voltage and inversely
proportional to the input voltage. Therefore, if the on-time
is fixed, the off-time (and therefore the frequency) must
change in response to changes in input or output voltage.
Modern pseudo-fixed frequency COT architectures greatly
improve COT by making the one-shot on-time proportional
to VOUT and inversely proportional to VIN. In this way, an
on-time is chosen as approximately what it would be for
an ideal fixed-frequency PWM in similar input/output
voltage conditions. The result is a big improvement but
the switching frequency still varies considerably over line
and load due to losses in the switches and inductor and
other parasitic effects.
Another problem with many COT architectures is their
dependence on adequate ESR in the output capacitor,
making it difficult to use highly-desirable, small, low-cost,
but low-ESR ceramic capacitors. Most COT architectures
use AC current information from the output capacitor,
generated by the inductor current passing through the
ESR, to function in a way like a current mode control
system. With ceramic capacitors the inductor current
information is too small to keep the control loop stable,
like a current mode system with no current information.
ACOTTM Control Architecture
Making the on-time proportional to VOUT and inversely
proportional to VIN is not sufficient to achieve good
constant-frequency behavior for several reasons. First,
voltage drops across the MOSFET switches and inductor
cause the effective input voltage to be less than the
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DS6237A/B-04 May 2016
RT6237A/B
measured input voltage and the effective output voltage to
be greater than the measured output voltage. As the load
changes, the switch voltage drops change causing a
switching frequency variation with load current. Also, at
light loads if the inductor current goes negative, the switch
dead-time between the synchronous rectifier turn-off and
the high-side switch turn-on allows the switching node to
rise to the input voltage. This increases the effective on
time and causes the switching frequency to drop
noticeably.
One way to reduce these effects is to measure the actual
switching frequency and compare it to the desired range.
This has the added benefit eliminating the need to sense
the actual output voltage, potentially saving one pin
connection. ACOTTM uses this method, measuring the
actual switching frequency and modifying the on-time with
a feedback loop to keep the average switching frequency
in the desired range.
To achieve good stability with low-ESR ceramic capacitors,
ACOTTM uses a virtual inductor current ramp generated
inside the IC. This internal ramp signal replaces the ESR
ramp normally provided by the output capacitor's ESR.
The ramp signal and other internal compensations are
optimized for low-ESR ceramic output capacitors.
ACOTTM One-Shot Operation
The RT6237A/B control algorithm is simple to understand.
The feedback voltage, with the virtual inductor current ramp
added, is compared to the reference voltage. When the
combined signal is less than the reference and the ontime one-shot is triggered, as long as the minimum offtime one-shot is clear and the measured inductor current
(through the synchronous rectifier) is below the current
limit. The on-time one-shot turns on the high-side switch
and the inductor current ramps up linearly. After the on
time, the high-side switch is turned off and the synchronous
rectifier is turned on and the inductor current ramps down
linearly. At the same time, the minimum off-time one-shot
is triggered to prevent another immediate on-time during
the noisy switching time and allow the feedback voltage
and current sense signals to settle. The minimum off-time
is kept short (230ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed.
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
DS6237A/B-04 May 2016
Discontinuous Operating Mode (RT6237A Only)
After soft-start, the RT6237A operates in fixed frequency
mode to minimize interference and noise problems. The
RT6237A uses variable-frequency discontinuous switching
at light loads to improve efficiency. During discontinuous
switching, the on-time is immediately increased to add
“hysteresis” to discourage the IC from switching back to
continuous switching unless the load increases
substantially.
The IC returns to continuous switching as soon as an ontime is generated before the inductor current reaches zero.
The on-time is reduced back to the length needed for
500kHz switching and encouraging the circuit to remain
in continuous conduction, preventing repetitive mode
transitions between continuous switching and
discontinuous switching.
Current Limit
The RT6237A/B current limit is a cycle-by-cycle “valley”
type, measuring the inductor current through the
synchronous rectifier during the off-time while the inductor
current ramps down. The current is determined by
measuring the voltage between Source and Drain of the
synchronous rectifier. If the inductor current exceeds the
current limit, the on-time one-shot is inhibited (Mask high
side signal) until the inductor current ramps down below
the current limit. Thus, only when the inductor current is
well below the current limit is another on time permitted.
This arrangement prevents the average output current from
greatly exceeding the guaranteed current limit value, as
typically occurs with other valley-type current limits. If
the output current exceeds the available inductor current
(controlled by the current limit mechanism), the output
voltage will drop. If it drops below the output under-voltage
protection level the IC will stop switching (see next section).
Output Under-Voltage Protection
Hiccup Mode
The RT6237AH/RT6237BH provide Hiccup Mode UnderVoltage Protection (UVP). When the FB voltage drops
below 60% of the feedback reference voltage, the output
voltage drops below the UVP trip threshold for longer than
270μs (typical) then IC's UVP is triggered. UVP function
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RT6237A/B
will be triggered to shut down switching operation. If the
UVP condition remains for a period, the RT6237 will retry
automatically. When the UVP condition is removed, the
converter will resume operation. The UVP is disabled
during soft-start period. During hiccup mode, the shutdown
time is determined by the capacitor at SS. A 2μA current
source discharges VSS from its starting voltage (normally
VPVCC). The IC remains shut down until VSS reaches
0.2V, about 10ms for a 3.9nF capacitor. At that point the
IC begins to charge the SS capacitor at 6μA, and a normal
start-up occurs. If the fault remains, UVP protection will
be enabled when VSS reaches 2.2V (typical). The IC will
then shut down and discharge the SS capacitor from the
2.2V level, taking about 4ms for a 3.9nF SS capacitor.
Latch Mode
For the RT6237AL/RT6237BL, it provides Latch-Off Mode
Under Voltage Protection (UVP). When the FB voltage
drops below 60% of the feedback reference voltage, the
output voltage drops below the UVP trip threshold for longer
than 270μs (typical) then IC's UVP is triggered. UVP
function will be triggered to shut down switching operation.
In shutdown condition, the RT6237 can be reset by EN
pin or power input VIN.
Output Over-Voltage Protection
If the output voltage VOUT rises above the regulation level
and lower 1.2 times regulation level, the high-side switch
naturally remains off and the synchronous rectifier turns
on. For RT6237BL, if the output voltage remains high, the
synchronous rectifier remains on until the inductor current
reaches the low side current limit. If the output voltage
still remains high, then IC's switches remain that the
synchronous rectifier turns on and high-side MOS keeps
off to operate at typical 500kHz switching protection, again
if inductor current reaches low side current limit, the
synchronous rectifier will turn off until next protection
clock. If the output voltage exceeds the OVP trip threshold
(1.2 times regulation level) for longer than 10μs (typical),
then IC's output Over-Voltage Protection (OVP) is
triggered. RT6237BL chip enters latch mode. For
RT6237AL, if the output voltage VOUT rises above the
regulation level and lower 1.2 times regulation level, the
high-side switch naturally remains off and the synchronous
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6
rectifier turns on until the inductor current reaches zero
current. If the output voltage remains high, then IC's
switches remain off. If the output voltage exceeds the OVP
trip threshold (1.2 times regulation level) for longer than
10μs (typical), the IC's OVP is triggered. RT6237AL chip
enters latch mode. For RT6237BH, if the output voltage
remains high, the synchronous rectifier remains on until
the inductor current reaches the low side current limit. If
the output voltage still remains high, the synchronous
rectifier turns on and high-side MOSFET keeps off to
operate at typical 500kHz switching protection, again if
inductor current reaches low side current limit, the
synchronous rectifier will turn off until next protection
clock. RT6237BH is without OVP latch function and
recover when OV condition release.
For RT6237AH, if the output voltage remains high, the
synchronous rectifier remains on until the inductor current
reaches zero current. If the output voltage still remains
high, then IC's switches remain off. RT6237AH is without
OVP latch function and recover when OV condition release.
Latch-Off Mode
The RT6237AL/BL uses latch-off mode OVP and UVP.
When the protection function is triggered, the IC will shut
down in Latch-Off Mode. The IC stops switching, leaving
both switches open, and is latched off. To restart operation,
toggle EN or power the IC off and then on again.
Shut-Down, Start-Up and Enable (EN)
The enable input (EN) has a logic-low level of 0.4V. When
VEN is below this level the IC enters shutdown mode and
supply current drops to less than 10μA. When VEN exceeds
its logic-high level of 1.2V the IC is fully operational.
Between these 2 levels there are 2 thresholds (1V typical
and 1.2V typical). Switching operation begins when VEN
exceeds the upper threshold, and then switching
operation stops when V EN decreases to the lower
threshold. Since EN is a low voltage input, it must be
connected to VIN (up to 18V) with a pull-up resistor for
automatic start-up.
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DS6237A/B-04 May 2016
RT6237A/B
Input Under-Voltage Lockout
PGOOD Comparator
In addition to the enable function, the RT6237A/B feature
an Under-Voltage Lockout (UVLO) function that monitors
the internal linear regulator output (VIN). To prevent
operation without fully-enhanced internal MOSFET
switches, this function inhibits switching when VIN drops
below the UVLO-falling threshold. The IC resumes
switching when VIN exceeds the UVLO-rising threshold
PGOOD is an open-drain output controlled by a comparator
connected to the feedback signal. If FB exceeds 90% of
the internal reference voltage, PGOOD will be high
impedance. Otherwise, the PGOOD output is connected
to GND.
Soft-Start (SS)
The RT6237A/B soft-start uses an external pin (SS) to
clamp the output voltage and allow it to slowly rise. After
VEN is high and VIN exceeds its UVLO threshold, the IC
begins to source 6μA from the SS pin. An external capacitor
at SS is used to adjust the soft-start timing. Following
below equation to get the minimum capacitance range in
order to avoid UV occur.
T=
COUT  VOUT  0.75  1.2
ILIM  Load Current   0.8
CSS 
T  6μA
VREF
Do not leave SS unconnected. During start-up, while the
SS capacitor charges, the RT6237A/B operates in
discontinuous switching mode with very small pulses. This
prevents negative inductor currents and keeps the circuit
from sinking current. Therefore, the output voltage may
be pre-biased to some positive level before start-up. Once
the VSS ramp charges enough to raise the internal
reference above the feedback voltage, switching will begin
and the output voltage will smoothly rise from the prebiased level to its regulated level. After VSS rises above
about 2.2V output over- and under-voltage protections are
enabled and the RT6237A/B begins continuous-switching
operation.
External Bootstrap Capacitor (CBOOT)
Connect a 0.1μF low ESR ceramic capacitor between
BOOT and SW. This bootstrap capacitor provides the gate
driver supply voltage for the high-side N-channel MOSFET
switch.
Some of case, such like duty ratio is higher than 65%
application or input voltage is lower than 5.5V which are
recommended to add an external bootstrap diode between
an external 5V and BOOT pin for efficiency improvement
The bootstrap diode can be a low cost one such as IN4148
or BAT54. The external 5V can be a 5V fixed input from
system or a 5V output of the RT6237A/B. Note that the
external boot voltage must be lower than 5.5V.
Over-Temperature Protection
The RT6237A/B includes an Over-Temperature Protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. The OTP will shut down switching
operation when the junction temperature exceeds 150°C.
Once the junction temperature cools down by
approximately 20°C the IC will resume normal operation
with a complete soft-start. For continuous operation,
provide adequate cooling so that the junction temperature
does not exceed 150°C.
Internal Regulator (PVCC)
An internal linear regulator (PVCC) produces a 5V supply
from VIN. The 5V power supplies the internal control
circuit, such as internal gate drivers, PWM logic, reference,
analog circuitry, and other blocks. 1μF ceramic capacitor
for decoupling and stability is required.
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
DS6237A/B-04 May 2016
is a registered trademark of Richtek Technology Corporation.
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7
RT6237A/B
Absolute Maximum Ratings
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(Note 1)
Supply Voltage, VIN -----------------------------------------------------------------------------------------------Switch Voltage, SW -----------------------------------------------------------------------------------------------Switch Voltage, <10ns --------------------------------------------------------------------------------------------BOOT Voltage -------------------------------------------------------------------------------------------------------EN to GND ------------------------------------------------------------------------------------------------------------Other Pins ------------------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C
UQFN-14L 2x3 (FC) ------------------------------------------------------------------------------------------------Package Thermal Resistance (Note 2)
UQFN-14L 2x3 (FC), θJA ------------------------------------------------------------------------------------------UQFN-14L 2x3 (FC), θJC ------------------------------------------------------------------------------------------Junction Temperature Range -------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------Storage Temperature Range -------------------------------------------------------------------------------------ESD Susceptibility (Note 3)
HBM (Human Body Model) ----------------------------------------------------------------------------------------
Recommended Operating Conditions

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
−0.3V to 21V
−0.3V to (VIN + 0.3V)
−3V to (VIN + 0.3V)
−0.3V to 27.3V
−0.3V to 6V
−0.3V to 6V
2.1W
47.5°C/W
4.1°C/W
150°C
260°C
−65°C to 150°C
2kV
(Note 4)
Supply Voltage, VIN ------------------------------------------------------------------------------------------------ 4.5V to 18V
Junction Temperature Range -------------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range -------------------------------------------------------------------------------------- −40°C to 85°C
Electrical Characteristics
(VIN = 12V, TA = 25°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Current
Supply Current (Shutdown)
ISHDN
VEN = 0V
--
1.5
6
A
Supply Current (Quiescent)
IQ
VEN = 2V, VFB = 0.7V
--
0.6
0.9
mA
Logic-High
1.1
1.2
1.3
Hysteresis
--
0.2
--
Logic Threshold
EN Input
Voltage
V
VFB Voltage and Discharge Resistance
Feedback Voltage
VFB
4.5V  VIN  18V
0.692
0.7
0.708
V
Feedback Current
IFB
VFB = 0.71V
0.1
--
0.1
A
VPVCC
6V  VIN  18V, 0 < IPVCC  5mA
--
5
--
V
Line Regulation
6V  VIN  18V, IPVCC = 5mA
--
--
5
mV
Load Regulation
0  IPVCC  20mA
--
--
20
mV
VIN = 6V, VPVCC = 4V, TA = 25C
--
210
--
mA
VPVCC Output
VPVCC Output Voltage
Output Current
IPVCC
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is a registered trademark of Richtek Technology Corporation.
DS6237A/B-04 May 2016
RT6237A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
--
40
--
RDS(ON)_L
--
16
--
ILIM
8
9.5
11
TSD
--
150
--
--
20
--
--
175
--
ns
RDS(ON)
Switch On-Resistance
RDS(ON)_H
VBOOT  VSW = 5V
m
Current Limit
Valley Current Limit
A
Thermal Shutdown
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis TSD
C
On-Time Timer Control
On-Time
tON
VIN = 12V, VOUT = 1.05V
Minimum On-Time
tON(MIN)
--
60
--
ns
Minimum Off-Time
tOFF(MIN)
--
200
--
ns
VSS = 0V
5
6
7
A
Wake Up VPVCC
4
4.2
4.4
Hysteresis
--
0.5
--
FB Rising
85
90
95
%
FB Falling
--
80
--
%
PGOOD = 0.1V
10
20
--
mA
115
120
125
%
--
10
--
s
UVP Detect
55
60
65
Hysteresis
--
17
--
--
270
--
s
--
tSS
x 1.7
--
--
Soft-Start
SS Charge Current
UVLO
UVLO Threshold
V
Power Good
PGOOD Threshold
PGOOD Sink Current
Output Under-Voltage and Over-Voltage Protection
OVP Trip Threshold
OVP Detect
OVP Propagation Delay
UVP Trip Threshold
UVP Propagation Delay
UVP Enable Delay
Relative to Soft-Start Time
%
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may
affect device reliability.
Note 2. θJA is measured at TA = 25°C on a highly thermal conductive four-layer test board. θJC is measured at the exposed pad
of the package.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
DS6237A/B-04 May 2016
is a registered trademark of Richtek Technology Corporation.
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9
RT6237A/B
Typical Application Circuit
VIN
7, 8
C1
10µF x 2
C2
0.1µF
RT6237A/B
VIN
SW
4 PGOOD
BOOT
13 EN
Enable
C5
10nF
FB
14 SS
PVCC
L1
1µH
6
VOUT
1V
C6
0.1µF
5
R1
20k
C3
C7
22µF x 3
2
3
AGND GND
9, 10, 11, 12
1
C4
1µF
R2
46.6k
VPVCC
Table 1. Suggested Component Values
VOUT (V)
R1 (k)
R2 (k)
C3 (pF)
L1 (H)
C7 (F)
1
20
46.4
--
1
66
1.5
53.6
46.4
10
1
66
1.8
73.2
46.4 10
1
66
2.5
120
46.4 10
2
66
5
287
46.4 10
2.8
66
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RT6237A/B
Typical Operating Characteristics
Efficiency vs. Output Current
98
94
94
90
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
86
82
78
=
=
=
=
=
=
=
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
98
1V
1.1V
1.2V
1.5V
1.8V
3.3V
5V
90
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
86
82
78
RT6237A : PSM, VIN = 12V, fSW = 500kHz
RT6237B : PWM, VIN = 12V, fSW = 500kHz
70
70
0
1
2
3
4
5
6
0
7
1
2
Efficiency vs. Output Current
98
94
94
86
82
=
=
=
=
=
=
1V
1.1V
1.2V
1.5V
1.8V
3.3V
Efficiency (%)
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
4
5
6
7
Efficiency vs. Output Current
98
90
3
Output Current (A)
Output Current (A)
Efficiency (%)
1V
1.1V
1.2V
1.5V
1.8V
3.3V
5V
74
74
78
90
VOUT
VOUT
VOUT
VOUT
VOUT
VOUT
86
82
=
=
=
=
=
=
1V
1.1V
1.2V
1.5V
1.8V
3.3V
78
74
74
RT6237A : PSM, VIN = 5V, fSW = 500kHz
70
0
1
2
3
4
5
6
RT6237B : PWM, VIN = 5V, fSW = 500kHz
70
0
7
1
2
Output Current (A)
RT6237A
1.09
1.08
1.08
Output Voltage (V)
1.09
1.07
1.06
IOUT = 0A
IOUT = 3A
IOUT = 6A
1.04
4
5
6
7
Output Voltage vs. Input Voltage
1.10
1.05
3
Output Current (A)
Output Voltage vs. Input Voltage
1.10
Output Voltage (V)
=
=
=
=
=
=
=
1.03
RT6237B
1.07
1.06
IOUT = 0A
IOUT = 3A
IOUT = 6A
1.05
1.04
1.03
VOUT = 1V
VOUT = 1V
1.02
1.02
4
6
8
10
12
14
16
Input Voltage (V)
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18
4
6
8
10
12
14
16
18
Input Voltage (V)
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RT6237A/B
Output Voltage vs. Output Current
Output Voltage vs. Output Current
1.10
RT6237A
1.09
1.09
1.08
1.08
Output Voltage (V)
Output Voltage (V)
1.10
1.07
1.06
VIN = 17V
VIN = 12V
VIN = 4.5V
1.05
1.04
1.03
RT6237B
1.07
1.06
VIN = 17V
VIN = 12V
VIN = 4.5V
1.05
1.04
1.03
VOUT = 1V
VOUT = 1V
1.02
1.02
0
1
2
3
4
5
6
7
0
1
2
Output Current (A)
3
4
5
6
7
Output Current (A)
Output Voltage vs. Temperature
Frequency vs. Input Voltage
1.03
600
580
Frequency (kHz)1
Output Voltage (V)
1.02
1.01
1.00
VIN = 17V
VIN = 12V
VIN = 4.5V
0.99
560
540
520
500
480
460
440
0.98
VOUT = 1V, IOUT = 0.5A
0.97
420
VOUT = 3.3V, IOUT = 0A
400
-50
-25
0
25
50
75
100
125
4
6
Temperature (°C)
8
10
12
14
16
18
Input Voltage (V)
Frequency vs. Temperature
Load Transient Response
550
RT6237A
Frequency (kHz)1
530
VOUT
(50mV/Div)
510
490
IOUT
(5A/Div)
470
VOUT = 1V
450
-50
-25
0
25
50
75
100
125
VIN = 12V, VOUT = 1V, IOUT = 0.1A to 7A
Time (100μs/Div)
Temperature (°C)
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Load Transient Response
Load Transient Response
RT6237B
RT6237A
VOUT
(50mV/Div)
VOUT
(50mV/Div)
IOUT
(5A/Div)
IOUT
(5A/Div)
VIN = 12V, VOUT = 1V, IOUT = 0.1A to 7A
VIN = 12V, VOUT = 1V, IOUT = 3.5A to 7A
Time (100μs/Div)
Time (100μs/Div)
Output Ripple Voltage
Output Ripple Voltage
RT6237A
RT6237B
VOUT
(10mV/Div)
VOUT
(10mV/Div)
VLX
(10V/Div)
VLX
(10V/Div)
ILX
(0.5A/Div)
ILX
(3A/Div)
VIN = 12V, VOUT = 1V, IOUT = 50mA
VIN = 12V, VOUT = 1V, IOUT = 3.5A
Time (20μs/Div)
Time (2μs/Div)
Output Ripple Voltage
Power On from EN
RT6237B
RT6237A
VOUT
(10mV/Div)
VEN
(5V/Div)
VOUT
(1V/Div)
VLX
(10V/Div)
VLX
(10V/Div)
ILX
(3A/Div)
VIN = 12V, VOUT = 1V, IOUT = 7A
Time (2μs/Div)
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DS6237A/B-04 May 2016
ILX
(10A/Div)
VIN = 12V, VOUT = 1V, IOUT = 7A
Time (5ms/Div)
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RT6237A/B
Power Off from EN
VIN
(5V/Div)
RT6237A
VEN
(5V/Div)
VIN = 12V, VOUT = 1V, IOUT = Short
VOUT
(1V/Div)
VOUT
(1V/Div)
VLX
(10V/Div)
VLX
(10V/Div)
ILX
(10A/Div)
UVP Short (Latch Mode)
VIN = 12V, VOUT = 1V, IOUT = 7A
Time (5ms/Div)
ILX
(10A/Div)
Time (2ms/Div)
UVP Short (Hiccup Mode)
VIN
(5V/Div)
VIN = 12V, VOUT = 1V, IOUT = Short
VOUT
(500mV/Div)
VLX
(10V/Div)
ILX
(10A/Div)
Time (10ms/Div)
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RT6237A/B
Application information
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor value
is generally flexible and is ultimately chosen to obtain the
best mix of cost, physical size, and circuit efficiency.
Lower inductor values benefit from reduced size and cost
and they can improve the circuit's transient response, but
they increase the inductor ripple current and output voltage
ripple and reduce the efficiency due to the resulting higher
peak currents. Conversely, higher inductor values increase
efficiency, but the inductor will either be physically larger
or have higher resistance since more turns of wire are
required and transient response will be slower since more
time is required to change current (up or down) in the
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (ΔIL) about
15% to 40% of the desired full output load current.
Calculate the approximate inductor value by selecting the
input and output voltages, the switching frequency (fSW),
the maximum output current (IOUT(MAX)) and estimating a
ΔIL as some percentage of that current.
VOUT   VIN  VOUT 
L=
VIN  fSW  IL
Once an inductor value is chosen, the ripple current (ΔIL)
is calculated to determine the required peak inductor
current.
VOUT   VIN  VOUT 
IL =
VIN  fSW  L
I
IL(PEAK) = IOUT(MAX)  L
2
I
IL(VALLEY) = IOUT(MAX)  L
2
Inductor saturation current should be chosen over IC's
current limit.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source and
to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS current
rating (and voltage rating, of course). The RMS input ripple
current (IRMS) is a function of the input voltage, output
voltage, and load current :
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DS6237A/B-04 May 2016
V
VIN
IRMS = IOUT(MAX)  OUT
1
VIN
VOUT
Ceramic capacitors are most often used because of their
low cost, small size, high RMS current ratings, and robust
surge current capabilities. However, take care when these
capacitors are used at the input of circuits supplied by a
wall adapter or other supply connected through long, thin
wires. Current surges through the inductive wires can
induce ringing at the RT6237A/B input which could
potentially cause large, damaging voltage spikes at VIN.
If this phenomenon is observed, some bulk input
capacitance may be required. Ceramic capacitors (to meet
the RMS current requirement) can be placed in parallel
with other types such as tantalum, electrolytic, or polymer
(to reduce ringing and overshoot).
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled to
meet the RMS current, size, and height requirements of
the application. The typical operating circuit uses two 10μF
and one 0.1μF low ESR ceramic capacitors on the input.
Output Capacitor Selection
The RT6237A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level and
transient response requirements for sag (undershoot on
positive load steps) and soar (overshoot on negative load
steps).
Output Ripple
Output ripple at the switching frequency is caused by the
inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are similar
in amplitude and both should be considered if ripple is
critical.
VRIPPLE = VRIPPLE(ESR)  VRIPPLE(C)
VRIPPLE(ESR) = IL  RESR
IL
VRIPPLE(C) =
8  COUT  fSW
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RT6237A/B
Feed-forward Capacitor (Cff)
Soft-Start (SS)
The RT6237A/B are optimized for ceramic output
capacitors and for low duty cycle applications. However
for high-output voltages, with high feedback attenuation,
the circuit's response becomes over-damped and transient
response can be slowed. In high-output voltage circuits
(VOUT > 3.3V) transient response is improved by adding a
small “feed-forward” capacitor (Cff) across the upper FB
divider resistor (Figure 1), to increase the circuit's Q and
reduce damping to speed up the transient response without
affecting the steady-state stability of the circuit. Choose
a suitable capacitor value that following below step.
The RT6237A/B soft-start uses an external capacitor at
SS to adjust the soft-start timing according to the following
equation :

Get the BW the quickest method to do transient
response form no load to full load. Confirm the damping
frequency. The damping frequency is BW.
t  ms  
CSS  nF   0.7
ISS μA 
Following below equation to get the minimum capacitance
range in order to avoid UV occur.
COUT  VOUT  0.6  1.2
(ILIM  Load Current)  0.8
T  6μA
CSS 
VREF
T
Do not leave SS unconnected.
Enable Operation (EN)
For automatic start-up, the low-voltage EN pin must be
connected to VIN with a 100kΩ resistor. EN can be
externally pulled to VIN by adding a resistor-capacitor
delay (REN and CEN in Figure 2). Calculate the delay time
using EN's internal threshold where switching operation
begins (1.2V, typical).
BW
VOUT
R1
Cff
FB
RT6237A/B
R2
GND
Figure 1. Cff Capacitor Setting
An external MOSFET can be added to implement digital
control of EN (Figure 3). In this case, a 100kΩ pull-up
resistor, REN, is connected between VIN and the EN pin.
MOSFET Q1 will be under logic control to pull down the
EN pin. To prevent enabling circuit when VIN is smaller
than the VOUT target value or some other desired voltage
level, a resistive voltage divider can be placed between
the input voltage and ground and connected to EN to create
an additional input under voltage lockout threshold (Figure
4).
EN
VIN

Cff can be calculated base on below equation :
Cff 
1
2  3.1412  R1 BW  0.8
REN
CEN
EN
RT6237A/B
GND
Figure 2. External Timing Control
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RT6237A/B
VIN
REN
100k
External BOOT Bootstrap Diode
EN
Q1
Enable
RT6237A/B
GND
Figure 3. Digital Enable Control Circuit
VIN
REN1
External BOOT Capacitor Series Resistance
EN
REN2
RT6237A/B
GND
Figure 4. Resistor Divider for Lockout Threshold Setting
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected to
FB. The output voltage is set according to the following
equation :
VOUT = 0.7 x (1 + R1 / R2)
VOUT
R1
FB
RT6237A/B
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode between
VIN (or VINR) and the BOOT pin to improve enhancement
of the internal MOSFET switch and improve efficiency.
The bootstrap diode can be a low cost one such as 1N4148
or BAT54.
R2
GND
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low power
loss and good efficiency, but also slow enough to reduce
EMI. Switch turn-on is when most EMI occurs since VSW
rises rapidly. During switch turn-off, SW is discharged
relatively slowly by the inductor current during the dead
time between high-side and low-side switch on-times. In
some cases it is desirable to reduce EMI further, at the
expense of some additional power dissipation. The switch
turn-on can be slowed by placing a small (<47Ω)
resistance between BOOT and the external bootstrap
capacitor. This will slow the high-side switch turn-on and
VSW's rise. To remove the resistor from the capacitor
charging path (avoiding poor enhancement due to
undercharging the BOOT capacitor), use the external diode
shown in figure 6 to charge the BOOT capacitor and place
the resistance between BOOT and the capacitor/diode
connection.
5V
Figure 5. Output Voltage Setting
BOOT
Place the FB resistors within 5mm of the FB pin. Choose
R2 between 10kΩ and 100kΩ to minimize power
consumption without excessive noise pick-up and
calculate R1 as follows :
R2  (VOUT  0.7)
R1 
0.7
For output voltage accuracy, use divider resistors with 1%
or better tolerance.
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DS6237A/B-04 May 2016
RT6237A/B
0.1µF
SW
Figure 6. External Bootstrap Diode
PVCC Capacitor Selection
Decouple PVCC to GND with a 1μF ceramic capacitor.
High grade dielectric (X7R, or X5R) ceramic capacitors
are recommended for their stable temperature and bias
voltage characteristics.
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RT6237A/B
Thermal Considerations
Layout Consideration
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :

Follow the PCB layout guidelines for optimal
performance of the device.

Keep the traces of the main current paths as short and
wide as possible.

Put the input capacitor as close as possible to VIN and
VIN pins.
PD(MAX) = (TJ(MAX) − TA) / θJA

SW node is with high frequency voltage swing and
should be kept at small area. Keep analog components
away from the SW node to prevent stray capacitive noise
pickup.

Connect feedback network behind the output capacitors.
Keep the loop area small. Place the feedback
components near the device.

Connect all analog grounds to common node and then
connect the common node to the power ground behind
the output capacitors.

An example of PCB layout guide is shown in Figure 8
and Figure 9 for reference.
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications, the
maximum junction temperature is 125°C. The junction to
ambient thermal resistance, θJA, is layout dependent. For
UQFN-14L 2x3 (FC) package, the thermal resistance, θJA,
is 47.5°C/W on a standard four-layer thermal test board.
The maximum power dissipation at TA = 25°C can be
calculated by the following formula :
PD(MAX) = (125°C − 25°C) / (47.5°C/W) = 2.1W for
UQFN-14L 2x3 (FC) package
The maximum power dissipation depends on the operating
ambient temperature for fixed T J(MAX) and thermal
resistance, θJA. The derating curve in Figure 7 allows the
designer to see the effect of rising ambient temperature
on the maximum power dissipation.
Maximum Power Dissipation (W)1
2.4
Four-Layer PCB
2.0
1.6
1.2
0.8
0.4
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
Figure 7. Derating Curve of Maximum Power Dissipation
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RT6237A/B
Connect IC Pin Trace as wide as possible for thermal consideration
AGND must be
connected clear ground.
Add via for thermal consideration
VIN
REN
Internal Regulator Output.
Connect a 1µF capacitor to GND
to stabilize output voltage.
EN
GND
12
11
GND
FB
2
10
GND
PVCC
3
9
GND
PGOOD
4
8
VIN
Power Good Indicator
Open-Drain Output.
5
6
7
VIN
5V
13
1
SW
VOUT
14
AGND
BOOT
R2
R1
SS
CSS
The feedback components
must be connected as close
to the device as possible.
VIN
CIN
GND
Keep sensitive components
away from this CBOOT.
Top Layer
SW should be connected to inductor by
wide and short trace. Keep sensitive
components away from this trace .
VOUT
Figure 8. PCB Layout Guide (Top Layer)
Add via for thermal consideration
VIN
GND
Bottom Layer
Figure 9. PCB Layout Guide (Bottom Layer)
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RT6237A/B
Suggested Inductors for Typical Application Circuit
Component
Supplier
Part No.
Inductance (H)
DCR (m)
Dimensions (mm)
WE
7443320100
1
1.85
12.1 x 11.4 x 9.5
WE
744325120
1.2
1.8
10.2 x 10.2 x 4.7
WE
7443551200
2
2.6
12.8 x 12.8 x 6.2
Recommended component selection for Typical Application.
Component Supplier
Part No.
Capacitance (F)
Case Size
MURATA
GRM31CR61E106K
10
1206
TDK
C3225X5R1E106K
10
1206
TAIYO YUDEN
TMK316BJ106ML
10
1206
MURATA
GRM31CR60J476M
47
1206
TDK
C3225X5R0J476M
47
1210
TAIYO YUDEN
EMK325BJ476MM
47
1210
MURATA
GRM32ER71C226M
22
1210
TDK
C3225X5R1C226M
22
1210
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RT6237A/B
Outline Dimension
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.500
0.600
0.020
0.024
A1
0.000
0.050
0.000
0.002
A3
0.100
0.152
0.004
0.006
b
0.200
0.300
0.008
0.012
D
1.900
2.100
0.075
0.083
E
2.900
3.100
0.114
0.122
e
0.500
0.020
K
0.325
0.013
L
0.400
0.500
0.016
0.020
L1
2.325
2.425
0.092
0.095
L2
0.825
0.925
0.032
0.036
L3
0.300
0.400
0.012
0.016
L4
1.825
1.925
0.072
0.076
L5
0.325
0.425
0.013
0.017
U-Type 14L QFN 2x3 (FC) Package
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RT6237A/B
Footprint Information
Package
Number
of Pin
UQFN2*3-14(FC)
14
Footprint Dimension (mm)
P
Ax
Ay
By
C*4 C1*3
C2
C3
C4*4
C5
D*14
K
K1
0.500 2.800 3.800 2.250 0.850 0.775 2.775 1.275 0.750 2.275 0.250 0.325 0.250
Tolerance
±0.050
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot
assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be
accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
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DS6237A/B-04 May 2016