DS6233AB 01

RT6233A/B
3A Peak, 18V, 500kHz, ACOTTM Step-Down Converter
General Description
Features
The RT6233A/B is a high-efficiency, monolithic
synchronous step-down DC/DC converter that can
deliver up to 3A peak output current from a 4.5V to 18V
input supply. The RT6233A/B adopts ACOT
architecture to allow the transient response to be
improved and keep in constant frequency.
Cycle-by-cycle current limit provides protection against
shorted outputs and soft-start eliminates input current
surge during start-up. Fault conditions also include


Monotonic Start-Up into Pre-biased Outputs
Input Under-Voltage Lockout
output under voltage protection, output over current
protection, and thermal shutdown.

Over-Current Protection and Hiccup
Ordering Information









Package Type
QW : WDFN-8L 2x3 (W-Type)


Lead Plating System
G : Green (Halogen Free and Pb Free)
Adjustable Soft-Start
Power Good Indicator
Applications

RT6233A/B
Integrated 195m/105m MOSFETs
4.5V to 18V Supply Voltage Range
500kHz Switching Frequency
ACOT Control
0.8V  1.5% Voltage Reference
Set-Top Boxes
Portable TVs
Access Point Routers
DSL Modems
LCD TVs
Marking Information
UVP Option
H : Hiccup
RT6233AHGQW
0T : Product Code
W : Date Code
A : PSM Mode
B : PWM Mode
0TW
Note :
Richtek products are :

RT6233BHGQW
0SW
requirements of IPC/JEDEC J-STD-020.

0S : Product Code
W : Date Code
RoHS compliant and compatible with the current
Suitable for use in SnPb or Pb-free soldering processes.
Simplified Application Circuit
RT6233A/B
VIN
VIN
BOOT
CIN
CBOOT
L
VOUT
LX
Enable
EN
PGOOD
PGOOD
R1
CFF
COUT
FB
R2
SS
GND
CSS
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June 2015
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1
RT6233A/B
Pin Configurations
BOOT
GND
LX
VIN
1
2
3
4
GND
(TOP VIEW)
9
8
7
6
5
SS
PGOOD
EN
FB
WDFN-8L 2x3
Functional Pin Description
Pin No.
Pin Name
Pin Function
BOOT
Bootstrap Supply for High-Side Gate Driver. Connect a 100nF or greater
capacitor from LX to BOOT to power the high-side switch.
GND
System Ground. Provides the ground return path for the control circuitry and
low-side power MOSFET.
3
LX
Switch Node. LX is the switching node that supplies power to the output and
connect the output LC filter from LX to the output load.
4
VIN
Power Input. Supplies the power switches of the device.
5
FB
Feedback Voltage Input. This pin is used to set the desired output voltage via
an external resistive divider. The feedback voltage is 0.8V typically.
6
EN
Enable Control Input. Floating this pin or connecting this pin to GND can
disable the device and connecting this pin to logic high can enable the device.
7
PGOOD
Power Good Indicator Output. This pin is an open-drain logic output that is
pulled to ground when the output voltage is lower or higher than its specified
threshold under the conditions of OVP, OTP, dropout, EN shutdown, or during
slow start.
8
SS
Soft-Start Control Input. Connect a capacitor from SS to GND to set the
soft-start period.
1
2,
9 (Exposed Pad)
Function Block Diagram
BOOT
VIN
VIN
PVCC
Reg
VCC
VIBIAS
UV &OV
VREF Protection
PVCC
UGATE
VCC
1µA
OC
Ripple
Gen.
LX
Driver
LGATE
+
- -
SS
FB
Control
Comparator
On-Time
GND
PGOOD
LX
EN
FB
0.9VREF
+
-
Comparator
EN
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DS6233A/B-01T00
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RT6233A/B
Operation
The RT6233A/B is a synchronous step-down converter
with advanced constant on-time control mode. Using
the ACOTTM control mode can reduce the output
capacitance and provide fast transient response. It can
minimize the component size without additional
external compensation network.
UVLO Protection
To protect the chip from operating at insufficient supply
voltage, the UVLO is needed. When the input voltage
of VIN is lower than the UVLO falling threshold voltage,
the device will be lockout.
Current Protection
Thermal Shutdown
The inductor current is monitored via the internal
switches cycle-by-cycle. Once the output voltage drops
under UV threshold, the RT6233A/B will enter hiccup
mode.
When the junction temperature exceeds the OTP
threshold value, the IC will shut down the switching
operation. Once the junction temperature cools down
and is lower than the OTP lower threshold, the
converter will autocratically resume switching.
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RT6233A/B
Absolute Maximum Ratings
(Note 1)

Supply Input Voltage --------------------------------------------------------------------------------- 0.3V to 20V

Switch Node Voltage, LX ---------------------------------------------------------------------------- 0.3V to (VIN + 0.3V)

BOOT Pin Voltage ------------------------------------------------------------------------------------ (VLX – 0.3V) to (VIN + 6.3V)

Other Pins ----------------------------------------------------------------------------------------------- 0.3V to 6V

Power Dissipation, PD @ TA = 25C
WDFN-8L 2x3 ------------------------------------------------------------------------------------------ 1.667W

Package Thermal Resistance
(Note 2)
WDFN-8L 2x3, JA ------------------------------------------------------------------------------------ 60C/W
WDFN-8L 2x3, JC ------------------------------------------------------------------------------------ 8C/W

Lead Temperature (Soldering, 10 sec.) ---------------------------------------------------------- 260C

Junction Temperature -------------------------------------------------------------------------------- 150C

Storage Temperature Range ----------------------------------------------------------------------- 65C to 150C

ESD Susceptibility
(Note 3)
HBM (Human Body Model) ------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
(Note 4)

Supply Input Voltage --------------------------------------------------------------------------------- 4.5V to 18V

Ambient Temperature Range----------------------------------------------------------------------- 40C to 85C

Junction Temperature Range ---------------------------------------------------------------------- 40C to 125C
Electrical Characteristics
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
4.5
--
18
Under-Voltage Lockout
Threshold
VUVLO
3.6
3.9
4.2
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
--
340
--
mV
Shutdown Current
ISHDN
VEN = 0V
--
--
5
µA
Quiescent Current
IQ
VEN = 2V, VFB = 0.85V
--
0.5
--
mA
--
1
--
µA
VEN Rising
1.3
1.5
1.7
VEN Falling
1.18
1.28
1.38
V
Soft-Start
Internal Charge Current
Enable Voltage
EN Voltage Threshold
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V
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DS6233A/B-01T00
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RT6233A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
0.788
0.8
0.812
V
--
195
--
--
105
--
Feedback Voltage
VFB_TH
4.5V ≤ VIN ≤ 18V
High-Side On-Resistance
RDS(ON)_H
VBOOT − VLX = 4.8V
Low-Side On-Resistance
RDS(ON)_L
Feedback Voltage Threshold
Internal MOSFET
mΩ
Current Limit
High-Side Switch Current Limit ILIM_H
Low-Side Switch Valley Current
ILIM_L
Limit
Switching Frequency
--
5.8
--
3.1
3.8
--
Switching Frequency
fOSC
400
500
--
kHz
Maximum Duty Cycle
DMAX
--
86
--
%
Minimum On Time
tON(MIN)
--
60
--
Minimum Off Time
tOFF(MIN)
--
240
--
FB Rising
--
90
--
FB Falling
--
85
--
--
125
--
%
--
10
--
µS
UVP Detect
45
50
55
Hysteresis
--
10
--
--
5
--
A
On-Time Timer Control
ns
Power Good
PGOOD Threshold
%
Output Under Voltage and Over Voltage Protections
OVP Trip Threshold
OVP Detect
OVP Propagation Delay
UVP Trip Threshold
UVP Propagation Delay
%
µS
Thermal Shutdown
Thermal Shutdown Threshold
TSD
--
150
--
Thermal Shutdown Hysteresis
TSD
--
20
--
°C
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the
operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect
device reliability.
Note 2. JA is measured at TA = 25C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. JC is
measured at the exposed pad of the package.
Note 3. Devices are ESD sensitive. Handling precaution recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
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June 2015
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RT6233A/B
Typical Application Circuit
RT6233A/B
VIN
4.5V to 18V
4
CIN
22μF
VIN
BOOT
LX
1
CBOOT
0.1μF
L
3
6 EN
Enable
7
PGOOD
PGOOD
R1
FB
CFF
COUT
44μF
VOUT
5
R2
8
CSS
10nF
SS
GND
2, 9 (Exposed Pad)
Table 1. Recommended Components Selection
VOUT (V)
R1 (k)
R2 (k)
CFF (pF)
L (H)
COUT (F)
5.0
126
24
10 to 22
4.7
22 x 2
3.3
75
24
10 to 22
3.6
22 x 2
2.5
51
24
10 to 22
3.6
22 x 2
1.8
30
24
10 to 22
2.2
22 x 2
1.5
21
24
--
2.2
22 x 2
1.2
12
24
--
2.2
22 x 2
1.0
6
24
--
2.2
22 x 2
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RT6233A/B
Typical Operating Characteristics
Efficiency vs. Output Current
100
90
90
80
80
70
VIN = 4.5V
60
VIN = 12V
50
VIN = 18V
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
40
30
20
70
VIN = 4.5V
60
VIN = 12V
50
VIN = 18V
40
30
20
10
10
RT6233A, VOUT = 1.2V
0
RT6233B, VOUT = 1.2V
0
0
0.5
1
1.5
2
2.5
0
0.5
1
Output Current (A)
2.5
Efficiency vs. Output Current
100
100
90
90
80
80
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.2V
40
30
VOUT = 5V
70
Efficiency (%)
Efficiency (%)
2
Output Current (A)
Efficiency vs. Output Current
VOUT = 3.3V
60
VOUT = 1.2V
50
40
30
20
20
10
10
RT6233A, VIN = 12V
0
RT6233B, VIN = 12V
0
0
0.5
1
1.5
2
2.5
0
0.5
Output Voltage vs. Output Current
1.23
1.23
Output Voltage (V)
1.24
RT6233A
1.21
1.20
RT6233B
1.19
1.18
1.17
1.5
2
2.5
Output Voltage vs. Input Voltage
1.24
1.22
1
Output Current (A)
Output Current (A)
Output Voltage (V)
1.5
1.22
1.21
1.20
1.19
IOUT = 1A
1.18
IOUT = 2A
IOUT = 2.5A
1.17
VIN = 12V, VOUT = 1.2V
VOUT = 1.2V
1.16
1.16
0
0.5
1
1.5
2
Output Current (A)
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June 2015
2.5
4
6
8
10
12
14
16
18
Input Voltage (V)
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RT6233A/B
Switching Frequency vs. Output Current
Switching Frequency vs. Temperature
650
Switching Frequency (kHz)1
Switching Frequency (kHz)1
700
600
RT6233B
500
400
300
RT6233A
200
100
600
550
500
450
400
350
IOUT = 1A
0
300
0
0.5
1
1.5
2
2.5
-50
-25
0
Output Current (A)
75
100
125
Feedback Voltage vs. Temperature
0.84
0.84
0.83
0.83
Feedback Voltage (V)
Feedback Voltage (V)
50
Temperature (°C)
Feedback Voltage vs. Input Voltage
0.82
0.81
0.80
0.79
0.78
0.82
0.81
0.80
0.79
0.78
0.77
0.77
0.76
0.76
4
6
8
10
12
14
16
-50
18
-25
0
25
50
75
100
125
Temperature (°C)
Input Voltage (V)
Inductor Valley Current Limit
vs. Input Voltage
Inductor Valley Current Limit
vs. Temperature
5.0
5.0
Inductor Valley Current Limit (A)
Inductor Valley Current Limit (A)
25
4.5
4.0
3.5
3.0
2.5
Low-Side Switch
2.0
4
6
8
10
12
14
16
Input Voltage (V)
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18
4.5
4.0
3.5
3.0
2.5
Low-Side Switch
2.0
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6233A/B
Shutdown Current vs. Temperature
18
3.5
15
Shutdown Current (μA)1
Shutdown Current (μA)1
Shutdown Current vs. Input Voltage
4.0
3.0
2.5
2.0
1.5
4
6
8
10
12
14
16
9
6
3
VEN = 0V
VEN = 0V
1.0
12
0
-50
18
-25
0
Quiescent Current vs. Input Voltage
75
100
125
Quiescent Current vs. Temperature
1200
1100
1100
Quiescent Current (μA)
1200
1000
900
800
700
1000
900
800
700
600
600
VEN = 2V, VFB = 0.85V
VEN = 2V, VFB = 0.85V
500
500
4
6
8
10
12
14
16
18
-50
-25
0
25
50
75
100
125
Temperature (°C)
Input Voltage (V)
EN Threshold vs. Temperature
Input UVLO vs. Temperature
1.7
4.1
1.6
Rising
EN Threshold (V)
4.0
Input UVLO (V)
50
Temperature (°C)
Input Voltage (V)
Quiescent Current (μA)
25
3.9
3.8
Falling
3.7
3.6
Rising
1.5
1.4
Falling
1.3
1.2
1.1
3.5
1.0
3.4
-50
-25
0
25
50
75
100
Temperature (°C)
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125
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6233A/B
Load Transient Response
VOUT
(100mV/Div)
IOUT
(1A/Div)
VOUT
(100mV/Div)
RT6233A,VIN = 12V,VOUT = 1.2V, IOUT = 0 to 2.5A
IOUT
RT6233B,VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2.5A
(1A/Div)
Time (100s/Div)
Time (100s/Div)
Load Transient Response
Load Transient Response
VOUT
(100mV/Div)
VOUT
(100mV/Div)
IOUT
(1A/Div)
Load Transient Response
RT6233A, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 3A
IOUT
(1A/Div)
Time (100s/Div)
Time (100s/Div)
Voltage Ripple
Voltage Ripple
VLX
(10V/Div)
VLX
(10V/Div)
VOUT
(20mV/Div)
VOUT
(20mV/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 1A
Time (1s/Div)
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RT6233B, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 3A
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (1s/Div)
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RT6233A/B
Power On from Input Voltage
VIN
(20V/Div)
Power Off from Input Voltage
VIN
(20V/Div)
VLX
(10V/Div)
VLX
(10V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
IOUT
(2A/Div)
Time (25ms/Div)
Time (5ms/Div)
Power On from Enable Voltage
Power Off from Enable Voltage
VEN
(5V/Div)
VEN
(5V/Div)
VPGOOD
(5V/Div)
VPGOOD
(5V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (5ms/Div)
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VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (2.5ms/Div)
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RT6233A/B
Application Information
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor
value is generally flexible and is ultimately chosen to
obtain the best mix of cost, physical size, and circuit
efficiency. Lower inductor values benefit from reduced
size and cost and they can improve the circuit's
transient response, but they increase the inductor
ripple current and output voltage ripple and reduce the
efficiency due to the resulting higher peak currents.
Conversely, higher inductor values increase efficiency,
but the inductor will either be physically larger or have
higher resistance since more turns of wire are required
and transient response will be slower since more time
is required to change current (up or down) in the
meet the desired output current. If needed, reduce the
inductor ripple current (IL) to increase the average
inductor current (and the output current) while ensuring
that IL(PEAK) does not exceed the upper current limit
level.
For best efficiency, choose an inductor with a low DC
resistance that meets the cost and size requirements.
For low inductor core losses some type of ferrite core is
usually best and a shielded core type, although
possibly larger or more expensive, will probably give
fewer EMI and other noise problems.
Considering the Typical Operating Circuit for 1.2V
output at 2.5A and an input voltage of 12V, using an
inductor ripple of 0.5A (20%), the calculated inductance
value is :
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (IL)
L
about 20% to 50% of the desired full output load
current. Calculate the approximate inductor value by
selecting the input and output voltages, the switching
frequency (f SW ), the maximum output current
(IOUT(MAX)) and estimating a IL as some percentage of
that current.
The ripple current was selected at 0.5A and, as long as
we use the calculated 4.3H inductance, that should be
the actual ripple current amount. The ripple current and
required peak current as below :
L=
VOUT   VIN  VOUT 
VIN  fSW  IL
1.2  12  1.2
 4.3μH
12  500kHz  0.5A
IL =
1.2  12  1.2 
= 0.5A
12  500kHz  4.3μH
and IL(PEAK) = 2.5A  0.5A = 3A
2
Once an inductor value is chosen, the ripple current
(IL) is calculated to determine the required peak
For the 4.3H value, the inductor's saturation and
inductor current.
thermal rating should exceed at least at least 3A. For
IL =
VOUT   VIN  VOUT 
I
and IL(PEAK) = IOUT(MAX)  L
VIN  fSW  L
2
To guarantee the required output current, the inductor
needs a saturation current rating and a thermal rating
that exceeds IL(PEAK). These are minimum requirements.
To maintain control of inductor current in overload and
short circuit conditions, some applications may desire
current ratings up to the current limit value. However,
the IC's output under-voltage shutdown feature make
this unnecessary for most applications.
IL(PEAK) should not exceed the minimum value of IC's
upper current limit level or the IC may not be able to
more conservative, the rating for inductor saturation
current must be equal to or greater than switch current
limit of the device rather than the inductor peak current.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source
and to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS
current rating (and voltage rating, of course). The RMS
input ripple current (IRMS) is a function of the input
voltage, output voltage, and load current :
IRMS = IOUT(MAX) 
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VOUT
VIN
VIN
1
VOUT
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RT6233A/B
Ceramic capacitors are most often used because of
their low cost, small size, high RMS current ratings, and
robust surge current capabilities. However, take care
For the Typical Operating Circuit for 1.2V output and an
inductor ripple of 0.5A, with 2 x 22F output
capacitance each with about 5m ESR including PCB
when these capacitors are used at the input of circuits
supplied by a wall adapter or other supply connected
through long, thin wires. Current surges through the
inductive wires can induce ringing at the RT6233A/B
input which could potentially cause large, damaging
voltage spikes at VIN. If this phenomenon is observed,
some bulk input capacitance may be required. Ceramic
capacitors (to meet the RMS current requirement) can
be placed in parallel with other types such as tantalum,
electrolytic, or polymer (to reduce ringing and
overshoot).
trace resistance, the output voltage ripple components
are :
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled
to meet the RMS current, size, and height requirements
of the application. The typical operating circuit uses two
10F and one 0.1F low ESR ceramic capacitors on
VRIPPLE(ESR) = 0.5A  5m = 2.5mV
0.5A
= 2.84mV
8  44μF  500kHz
= 2.5mV  2.84mV = 5.34mV
VRIPPLE(C) =
VRIPPLE
Output Transient Undershoot and Overshoot
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load
steps up and down abruptly. The ACOT transient
response is very quick and output transients are
usually small.
However, the combination of small ceramic output
the input.
capacitors (with little capacitance), low output voltages
(with little stored charge in the output capacitors), and
Output Capacitor Selection
low duty cycle applications (which require high
inductance to get reasonable ripple currents with high
The RT6233A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level
and transient response requirements for sag
(undershoot on positive load steps) and soar
(overshoot on negative load steps).
Output Ripple
Output ripple at the switching frequency is caused by
the inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are
similar in amplitude and both should be considered if
ripple is critical.
VRIPPLE = VRIPPLE(ESR)  VRIPPLE(C)
VRIPPLE(ESR) = IL  RESR
VRIPPLE(C) =
input voltages) increases the size of voltage variations
in response to very quick load changes. Typically, load
changes occur slowly with respect to the IC's 650kHz
switching frequency.
But some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings
in response to very fast load steps.
The output voltage transient undershoot and overshoot
each have two components : the voltage steps caused
by the output capacitor's ESR, and the voltage sag and
soar due to the finite output capacitance and the
inductor current slew rate. Use the following formulas
to check if the ESR is low enough (typically not a
problem with ceramic capacitors) and the output
capacitance is large enough to prevent excessive sag
and soar on very fast load step edges, with the chosen
inductor value.
IL
8  COUT  fSW
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RT6233A/B
The amplitude of the ESR step up or down is a function
of the load step and the ESR of the output capacitor :
VESR _STEP = IOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the
maximum duty cycle. The maximum duty cycle during a
fast transient is a function of the on-time and the
minimum off-time since the ACOTTM control scheme
will ramp the current using on-times spaced apart with
minimum off-times, which is as fast as allowed.
Calculate the approximate on-time (neglecting
parasites) and maximum duty cycle for a given input
and output voltage as :
tON =
VOUT
tON
and DMAX =
VIN  fSW
tON  tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we
can neglect both of these since the on-time increase
compensates for the voltage losses. Calculate the
output voltage sag as :
VSAG =
L  (IOUT )2
2  COUT   VIN(MIN)  DMAX  VOUT 
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR =
L  (IOUT )2
2  COUT  VOUT
For the Typical Operating Circuit for 1.2V output, the
circuit has an inductor 4.3H and 2 x 22F output
capacitance with 5m ESR each. The ESR step is
2.5A x 2.5m = 6.25mV which is small, as expected.
The output voltage sag and soar in response to full
0A-2.5A-0A instantaneous transients are :
1.2V
= 200ns
12V  500kHz
200ns
and DMAX =
= 0.455
200ns  240ns
tON =
where 240ns is the minimum off time.
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VSAG =
4.3μH  (2.5A)2
= 72mV
2  44μF  12V  0.455  1.2V 
VSOAR =
4.3μH  (2.5A)2
= 254.5mV
2  44μF  1.2V
The sag is about 6% of the output voltage and the soar
is a full 21.2% of the output voltage. The ESR step is
negligible here but it does partially add to the soar, so
keep that in mind whenever using higher-ESR output
capacitors.
The soar is typically much worse than the sag in high
input, low-output step-down converters because the
high input voltage demands a large inductor value
which stores lots of energy that is all transferred into
the output if the load stops drawing current. Also, for a
given inductor, the soar for a low output voltage is a
greater voltage change and an even greater
percentage of the output voltage.
Any sag is always short-lived, since the circuit quickly
sources current to regain regulation in only a few
switching cycles. With the RT6233B, any overshoot
transient is typically also short-lived since the converter
will sink current, reversing the inductor current sharply
until the output reaches regulation again. The
RT6233A discontinuous operation at light loads
prevents sinking current so, for that IC, the output
voltage will soar until load current or leakage brings the
voltage down to normal.
Most applications never experience instantaneous full
load steps and the RT6233A/B high switching
frequency and fast transient response can easily
control voltage regulation at all times. Also, since the
sag and soar both are proportional to the square of the
load change, if load steps were reduced to 1A (from the
3A examples preceding) the voltage changes would be
reduced by a factor of almost ten. For these reasons
sag and soar are seldom an issue except in very
low-voltage CPU core or DDR memory supply
applications, particularly for devices with high clock
frequencies and quick changes into and out of sleep
modes. In such applications, simply increasing the
amount of ceramic output capacitor (sag and soar are
directly proportional to capacitance) or adding extra
bulk capacitance can easily eliminate any excessive
voltage transients.
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RT6233A/B
In any application with large quick transients, always
calculate soar to make sure that over-voltage
protection will not be triggered. Under-voltage is not
Therefore, the required minimum capacitance for the
12V to 1.2V converter is :
11
COUT  3  5.23  10
12V  4.3μH
COUT  3.04μF
likely since the threshold is very low (50%), that
function has a long delay (5s), and the IC will quickly
return the output to regulation. Over-voltage protection
has a minimum threshold of 125% and short delay of
10s and can actually be triggered by incorrect
component choices, particularly for the RT6233A which
does not sink current.
Any ESR in the output capacitor lowers the required
minimum output capacitance, sometimes considerably.
For the rare application where that is needed and
useful, the equation including ESR is given here :
Output Capacitors Stability Criteria
COUT 
The e RT6233A/B's ACOTTM control architecture uses
an internal virtual inductor current ramp and other
As can be seen, setting RESR to zero and simplifying
compensation that ensures stability with any
reasonable output capacitor. The internal ramp allows
the IC to operate with very low ESR capacitors and the
IC is stable with very small capacitances. Therefore,
output capacitor selection is nearly always a matter of
meeting output voltage ripple and transient response
requirements, as discussed in the previous sections.
For the sake of the unusual application where ripple
voltage is unimportant and there are few transients
(perhaps battery charging or LED lighting) the stability
criteria are discussed below.
The equations giving the minimum required
capacitance for stable operation include a term that
depends on the output capacitor's ESR. The higher the
ESR, the lower the capacitance can be and still ensure
stability. The equations can be greatly simplified if the
ESR term is removed by setting ESR to zero. The
resulting equation gives the worst-case minimum
required capacitance and it is usually sufficiently small
that there is usually no need for the more exact
equation :
11
COUT  3  5.23  10
VIN  L
VOUT
2  fSW  VIN  (RESR  13647  L  VOUT )
the equation yields the previous simpler equation. To
allow for the capacitor's temperature and bias voltage
coefficients, use at least double the calculated
capacitance and use a good quality dielectric such as
X5R or X7R with an adequate voltage rating since
ceramic capacitors exhibit considerable capacitance
reduction as their bias voltage increases.
Feed-forward Capacitor (Cff)
The RT6233A/B are optimized for ceramic output
capacitors and for low duty cycle applications. However
for high-output voltages, with high feedback attenuation,
the circuit's response becomes over-damped and
transient response can be slowed. In high-output
voltage circuits (VOUT > 3.3V) transient response is
improved by adding a small “feed-forward” capacitor
(Cff) across the upper FB divider resistor (Figure 1), to
increase the circuit's Q and reduce damping to speed
up the transient response without affecting the
steady-state stability of the circuit. Choose a suitable
capacitor value that following below step.

Get the BW the quickest method to do transient
response form no load to full load. Confirm the
damping frequency. The damping frequency is BW.
The worst-case high capacitance requirement is for low
VIN and small inductance, so a 12V to 1.2V converter
is used for an example. Using the inductance equation
in a previous section to determine the required
inductance :
L=
1.2  12  1.2 
= 4.3μH
12  500kHz  0.5A
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RT6233A/B
Figure 2). Calculate the delay time using EN's internal
threshold where switching operation begins (1.5V,
typical).
BW
VOUT
R1
Cff
FB
RT6233A/B
R2
GND
An external MOSFET can be added to implement
digital control of EN when no system voltage above 2V
is available (Figure 3). In this case, a 100k pull-up
resistor, REN, is connected between VIN and the EN
pin. MOSFET Q1 will be under logic control to pull
down the EN pin. To prevent enabling circuit when VIN
is smaller than the VOUT target value or some other
desired voltage level, a resistive voltage divider can be
placed between the input voltage and ground and
connected to EN to create an additional input under
voltage lockout threshold (Figure 4).
EN
VIN
Figure 1. Cff Capacitor Setting

REN
EN
RT6233A/B
CEN
Cff can be calculated base on below equation :
Cff 
1
2  3.1412  R1 BW  0.8
Soft-Start (SS)
The RT6233A/B soft-start uses an external capacitor at
SS to adjust the soft-start timing according to the
following equation :
t ms  
CSS nF   0.8V
ISS μA 
Following below equation to get the minimum
capacitance range in order to avoid UV occur.
COUT  VOUT  0.75  1.2
(ILIM  Load Current)  0.8
T  1μA
CSS 
VREF
T
In addition, the chosen CSS must follow the ratio criteria
of CBOOT/CSS > 20 and the SS pin cannot be floating.
Enable Operation (EN)
For automatic start-up the low-voltage EN pin can be
connected to VIN through a 100k resistor. Its large
hysteresis band makes EN useful for simple delay and
timing circuits. EN can be externally pulled to VIN by
adding a resistor-capacitor delay (REN and CEN in
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GND
Figure 2. External Timing Control
VIN
REN
100k
EN
Q1
Enable
RT6233A/B
GND
Figure 3. Digital Enable Control Circuit
VIN
REN1
EN
REN2
RT6233A/B
GND
Figure 4. Resistor Divider for Lockout Threshold
Setting
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected
to FB. The output voltage is set according to the
following equation :
VOUT = 0.8V x (1 + R1 / R2)
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RT6233A/B
VOUT
5V
R1
FB
RT6233A/B
BOOT
R2
RT6233A/B
LX
GND
Figure 5. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin.
Choose R2 between 10k and 100k to minimize
power consumption without excessive noise pick-up
and calculate R1 as follows :
R1 
R2  (VOUT  VREF )
VREF
For output voltage accuracy, use divider resistors with
1% or better tolerance.
External BOOT Bootstrap Diode
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode
between VIN (or VINR) and the BOOT pin to improve
enhancement of the internal MOSFET switch and
improve efficiency. The bootstrap diode can be a low
cost one such as 1N4148 or BAT54.
External BOOT Capacitor Series Resistance
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low
power loss and good efficiency, but also slow enough
to reduce EMI. Switch turn-on is when most EMI occurs
since VLX rises rapidly. During switch turn-off, LX is
discharged relatively slowly by the inductor current
during the dead time between high-side and low-side
switch on-times. In some cases it is desirable to reduce
EMI further, at the expense of some additional power
dissipation. The switch turn-on can be slowed by
0.1μF
Figure 6. External Bootstrap Diode
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature.
The maximum power dissipation can be calculated by
the following formula :
PD(MAX) = (TJ(MAX)  TA) / JA
where TJ(MAX) is the maximum junction temperature,
TA is the ambient temperature, and JA is the junction to
ambient thermal resistance.
For recommended operating condition specifications,
the maximum junction temperature is 125C. The
junction to ambient thermal resistance, JA, is layout
dependent. For WDFN-8L 2x3 package, the thermal
resistance, JA, is 60C/W on a standard four-layer
thermal test board. The maximum power dissipation at
TA = 25C can be calculated by the following formula :
PD(MAX) = (125C  25C) / (60C/W) = 1.667W for
WDFN-8L 2x3 package
The maximum power dissipation depends on the
operating ambient temperature for fixed TJ(MAX) and
thermal resistance, JA. The derating curve in Figure 7
allows the designer to see the effect of rising ambient
temperature on the maximum power dissipation.
placing a small (<47) resistance between BOOT and
the external bootstrap capacitor. This will slow the
high-side switch turn-on and VLX's rise. To remove the
resistor from the capacitor charging path (avoiding poor
enhancement due to undercharging the BOOT
capacitor), use the external diode shown in Figure 6 to
charge the BOOT capacitor and place the resistance
between BOOT and the capacitor/diode connection.
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RT6233A/B
CSS
Four-Layer PCB
BOOT
GND
LX
VIN
CBOOT
1.5
1
2
3
4
9
L
LX should be connected
to inductor by wide and
short trace, and keep
VOUT
sensitive components
away from this trace.
1.0
VOUT
GND
GND
Maximum Power Dissipation (W)1
2.0
8
7
6
5
SS
PGOOD
EN
FB
R1
R2
Place the feedback
as close to the IC
as possible.
CIN
GND
COUT
Place the input and output
capacitors as close to the IC
as possible.
0.5
Figure 8. PCB Layout Guide
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
Figure 7. Derating Curve of Maximum Power
Dissipation
Layout Considerations
Follow the PCB layout
performance of the device.
guidelines
for
optimal

Keep the traces of the main current paths as short
and wide as possible.

Put the input capacitor as close as possible to VIN
and VIN pins.

LX node is with high frequency voltage swing and
should be kept at small area. Keep analog
components away from the LX node to prevent stray
capacitive noise pickup.

Connect feedback network behind the output
capacitors. Keep the loop area small. Place the
feedback components near the device.

Connect all analog grounds to common node and
then connect the common node to the power ground
behind the output capacitors.

An example of PCB layout guide is shown is Figure
8 for reference.
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RT6233A/B
Outline Dimension
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min
Max
Min
Max
A
0.700
0.800
0.028
0.031
A1
0.000
0.050
0.000
0.002
A3
0.175
0.250
0.007
0.010
b
0.200
0.300
0.008
0.012
D
1.900
2.100
0.075
0.083
D2
1.550
1.650
0.061
0.065
E
2.900
3.100
0.114
0.122
E2
1.650
1.750
0.065
0.069
e
L
0.500
0.350
0.020
0.450
0.014
0.018
W-Type 8L DFN 2x3 Package
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume
responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and
reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may
result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
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June 2015
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