RT6213A/B 3A Peak, 18V, 500kHz, ACOTTM Step-Down Converter General Description Features The RT6213A/B is a high-efficiency, monolithic synchronous step-down DC/DC converter that can deliver up to 3A peak output current from a 4.5V to 18V input supply. The RT6213A/B adopts ACOT architecture to allow the transient response to be improved and keep in constant frequency. Cycle-by-cycle current limit provides protection against shorted outputs and soft-start eliminates input current surge during start-up. Fault conditions also include output under voltage protection, output over current protection, and thermal shutdown. Package Type J6F : TSOT-23-6 (FC) LCD TVs Lead Plating System G : Green (Halogen Free and Pb Free) Marking Information UVP Option H : Hiccup RT6213AHGJ6F 09= : Product Code DNN : Date Code 09=DNN PSM/PWM A : PSM Mode B : PWM Mode RT6213BHGJ6F Note : Richtek products are : Set-Top Boxes Portable TVs Access Point Routers DSL Modems RT6213A/B 0.8V 1.5% Voltage Reference Internal Start-Up into Pre-biased Outputs Compact Package: TSOT-23-6 pin Input Under-Voltage Lockout Over-Current Protection and Hiccup Applications Ordering Information Integrated 150m/70m MOSFETs 4.5V to 18V Supply Voltage Range 500kHz Switching Frequency ACOT Control 08= : Product Code DNN : Date Code 08=DNN RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. Suitable for use in SnPb or Pb-free soldering processes. Simplified Application Circuit RT6213A/B BOOT VIN VIN CIN CBOOT L VOUT LX Enable EN R1 GND Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 CFF COUT FB R2 is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6213A/B Pin Configurations (TOP VIEW) BOOT EN 6 FB 5 4 2 3 GND LX VIN TSOT-23-6 (FC) Functional Pin Description Pin No. Pin Name Pin Function 1 GND System Ground. Provides the ground return path for the control circuitry and low-side power MOSFET. 2 LX Switch Node. LX is the switching node that supplies power to the output and connect the output LC filter from LX to the output load. 3 VIN Power Input. Supplies the power switches of the device. 4 FB Feedback Voltage Input. This pin is used to set the desired output voltage via an external resistive divider. The feedback voltage is 0.8V typically. 5 EN Enable Control Input. Floating this pin or connecting this pin to GND can disable the device and connecting this pin to logic high can enable the device. 6 BOOT Bootstrap Supply for High-Side Gate Driver. Connect a 100nF or greater capacitor from LX to BOOT to power the high-side switch. Function Block Diagram BOOT VIN VIN PVCC Reg Minoff PVCC VIBIAS VREF UGATE OC Control LX Driver LGATE UV &OV GND GND LX PVCC EN + -- LX Ripple Gen. Comparator EN VIN On-Time LX FB Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Operation The RT6213A/B is a synchronous step-down converter with advanced constant on-time control mode. Using the ACOTTM control mode can reduce the output capacitance and provide fast transient response. It can minimize the component size without additional external compensation network. UVLO Protection To protect the chip from operating at insufficient supply voltage, the UVLO is needed. When the input voltage of VIN is lower than the UVLO falling threshold voltage, the device will be lockout. Current Protection Thermal Shutdown The inductor current is monitored via the internal switches cycle-by-cycle. Once the output voltage drops under UV threshold, the RT6213A/B will enter hiccup mode. When the junction temperature exceeds the OTP threshold value, the IC will shut down the switching operation. Once the junction temperature cools down and is lower than the OTP lower threshold, the converter will autocratically resume switching. Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6213A/B Absolute Maximum Ratings (Note 1) Supply Input Voltage --------------------------------------------------------------------------------- 0.3V to 20V Switch Node Voltage, LX ---------------------------------------------------------------------------- 0.3V to (VIN + 0.3V) BOOT Pin Voltage ------------------------------------------------------------------------------------ (VLX – 0.3V) to (VIN + 6.3V) Other Pins ----------------------------------------------------------------------------------------------- 0.3V to 6V Power Dissipation, PD @ TA = 25C TSOT-23-6 (FC) --------------------------------------------------------------------------------------- 1.667W Package Thermal Resistance (Note 2) TSOT-23-6 (FC), JA --------------------------------------------------------------------------------- 60C/W TSOT-23-6 (FC), JC --------------------------------------------------------------------------------- 8C/W Lead Temperature (Soldering, 10 sec.) ---------------------------------------------------------- 260C Junction Temperature -------------------------------------------------------------------------------- 150C Storage Temperature Range ----------------------------------------------------------------------- 65C to 150C ESD Susceptibility (Note 3) HBM (Human Body Model) ------------------------------------------------------------------------- 2kV Recommended Operating Conditions (Note 4) Supply Input Voltage --------------------------------------------------------------------------------- 4.5V to 18V Ambient Temperature Range----------------------------------------------------------------------- 40C to 85C Junction Temperature Range ---------------------------------------------------------------------- 40C to 125C Electrical Characteristics (VIN = 12V, TA = 25C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max 4.5 -- 18 Unit Supply Voltage VIN Supply Input Operating Voltage VIN Under-Voltage Lockout Threshold VUVLO Under-Voltage Lockout Threshold Hysteresis VUVLO Shutdown Current ISHDN Quiescent Current IQ V 3.6 3.9 4.2 -- 340 -- mV VEN = 0V -- -- 5 µA VEN = 2V, VFB = 0.85V -- 0.5 -- mA -- 1000 -- µS VEN Rising 1.4 1.5 1.6 VEN Falling 1.18 1.28 1.38 Soft-Start Soft-Start Time Enable Voltage Enable Voltage Threshold Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 V is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Parameter Symbol Test Conditions Min Typ Max Unit 0.788 0.8 0.812 V -- 150 -- -- 70 -- High-Side Switch Current Limit ILIM_H -- 5.8 -- Low-Side Switch Valley Current ILIM_L Limit 3.1 3.8 -- fOSC 400 500 -- kHz Maximum Duty Cycle DMAX -- 86 -- % Minimum On Time tON(MIN) -- 60 -- Minimum Off Time tOFF(MIN) -- 240 -- -- 125 -- % -- 10 -- µS UVP Detect 45 50 55 Hysteresis -- 10 -- -- 5 -- Feedback Voltage VFB_TH 4.5V ≤ VIN ≤ 18V High-Side On-Resistance RDS(ON)_H VBOOT − VLX = 4.8V Low-Side On-Resistance RDS(ON)_L Feedback Voltage Threshold Internal MOSFET mΩ Current Limit A Switching Frequency Switching Frequency On-Time Timer Control nS Output Under Voltage and Over Voltage Protections OVP Trip Threshold OVP Detect OVP Propagation Delay UVP Trip Threshold UVP Propagation Delay % µS Thermal Shutdown Thermal Shutdown Threshold TSD -- 150 -- Thermal Shutdown Hysteresis TSD -- 20 -- °C Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. JA is measured at TA = 25C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. The first layer of copper area is filled JC is measured at the exposed pad of the package. Note 3. Devices are ESD sensitive. Handling precaution recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT6213A/B Typical Application Circuit RT6213A/B 3 VIN CIN 22μF BOOT LX 5 Enable VIN 6 2 L 2.2μH EN GND 1 FB CBOOT 0.1μF VOUT R1 12k CFF Open COUT 44μF 4 R2 24k Table 1. Recommended Components Selection VOUT (V) R1 (k) R2 (k) CFF (pF) L (H) COUT (F) 5.0 126 24 10 to 22 4.7 22*2 3.3 75 24 10 to 22 3.6 22*2 2.5 51 24 10 to 22 3.6 22*2 1.8 30 24 10 to 22 2.2 22*2 1.5 21 24 -- 2.2 22*2 1.2 12 24 -- 2.2 22*2 1.0 6 24 -- 2.2 22*2 Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Typical Operating Characteristics Efficiency vs. Output Current 100 90 90 80 80 70 VIN = 4.5V 60 VIN = 12V 50 VIN = 18V Efficiency (%) Efficiency (%) Efficiency vs. Output Current 100 40 30 20 70 VIN = 4.5V 60 VIN = 12V 50 VIN = 18V 40 30 20 10 10 RT6213A, VOUT = 1.2V 0 0 0.5 1 1.5 2 RT6213B, VOUT = 1.2V 0 2.5 0 0.5 1 Output Current (A) Efficiency vs. Output Current 100 90 90 2.5 80 70 VOUT = 5V 60 VOUT = 3.3V Efficiency (%) Efficiency (%) 2 Efficiency vs. Output Current 100 80 VOUT = 1.2V 50 40 30 20 70 VOUT = 5V 60 VOUT = 3.3V VOUT = 1.2V 50 40 30 20 10 10 RT6213A, VIN = 12V 0 RT6213B, VIN = 12V 0 0 0.5 1 1.5 2 2.5 0 0.5 1 Output Current (A) 2 2.5 Output Voltage vs. Input Voltage 1.23 1.23 1.22 1.22 1.21 Output Voltage (V) 1.24 RT6213A 1.21 1.5 Output Current (A) Output Voltage vs. Output Current Output Voltage (V) 1.5 Output Current (A) 1.20 RT6213B 1.19 1.18 IOUT = 1A 1.20 1.19 IOUT = 2A 1.18 IOUT = 2.5A 1.17 1.16 1.17 VOUT = 1.2V VIN = 12V, VOUT = 1.2V 1.15 1.16 0 0.5 1 1.5 2 Output Current (A) Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 2.5 4 6 8 10 12 14 16 18 Input Voltage (V) is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT6213A/B Switching Frequency vs. Temperature Switching Frequency vs. Output Current 650 Switching Frequency (kHz)1 Switching Frequency (kHz)1 700 600 RT6213B 500 400 RT6213A 300 200 100 600 550 500 450 400 350 IOUT = 1A 0 300 0 0.5 1 1.5 2 2.5 -50 -25 0 Output Current (A) 50 75 100 125 Feedback Voltage vs. Temperature 0.84 0.83 0.83 Feedback Voltage (V) Feedback Voltage (V) Feedback Voltage vs. Input Voltage 0.84 0.82 0.81 0.80 0.79 0.78 0.77 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.76 4 6 8 10 12 14 16 18 -50 -25 0 Input Voltage (V) 25 50 75 100 125 Temperature (°C) Inductor Valley Current Limit vs. Input Voltage Inductor Valley Current Limit vs. Temperature 5.0 4.5 4.0 3.5 3.0 2.5 Low-Side Switch 2.0 Inductor Valley Current Limit (A) 5.0 Inductor Valley Current Limit (A) 25 Temperature (°C) 4.5 4.0 3.5 3.0 2.5 Low-Side Switch 2.0 4 6 8 10 12 14 16 Input Voltage (V) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 18 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Shutdown Current vs. Temperature 18 3.5 15 Shutdown Current (μA)1 Shutdown Current (μA)1 Shutdown Current vs. Input Voltage 4.0 3.0 2.5 2.0 1.5 VEN = 0V 1.0 4 6 8 10 12 14 16 12 9 6 3 VEN = 0V 0 18 -50 -25 0 Input Voltage (V) Quiescent Current vs. Input Voltage 50 1200 1100 1100 1000 100 125 900 800 700 1000 900 800 700 600 600 VEN = 2V, VFB = 0.85V 500 VEN = 2V, VFB = 0.85V 500 4 6 8 10 12 14 16 18 -50 -25 0 Input Voltage (V) 4.0 1.6 Enable Threshold (V) 1.7 Rising 3.8 3.7 Falling 3.6 50 75 100 125 Enable Threshold vs. Temperature 4.1 3.9 25 Temperature (°C) Input UVLO vs. Temperature Input UVLO (V) 75 Quiescent Current vs. Temperature 1200 Quiescent Current (μA) Quiescent Current (μA) 25 Temperature (°C) 3.5 1.5 Rising 1.4 1.3 Falling 1.2 1.1 3.4 1.0 -50 -25 0 25 50 75 100 Temperature (°C) Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 125 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT6213A/B Load Transient Response Load Transient Response RT6213B RT6213A VOUT VOUT (100mV/Div) (100mV/Div) IOUT (1A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2.5A IOUT (1A/Div) Time (100s/Div) Time (100s/Div) Load Transient Response Load Transient Response RT6213B RT6213A VOUT VOUT (100mV/Div) (100mV/Div) IOUT (1A/Div) IOUT (1A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 0 to 3A VIN = 12V, VOUT = 1.2V, IOUT = 0 to 3A Time (100s/Div) Time (100s/Div) Voltage Ripple Voltage Ripple VLX VLX (10V/Div) (10V/Div) VOUT VOUT (20mV/Div) (20mV/Div) VIN = 12V, VOUT = 1.2V, IOUT = 1A Time (1s/Div) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2.5A VIN = 12V, VOUT = 1.2V, IOUT = 2.5A Time (1s/Div) is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Power On from Input Voltage Power Off from Input Voltage VIN VIN (20V/Div) (20V/Div) VLX (10V/Div) VLX (10V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2.5A IOUT (2A/Div) Time (2.5ms/Div) Time (25ms/Div) Power On from Enable Power Off from Enable VEN (5V/Div) VEN (5V/Div) VLX (10V/Div) VLX (10V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2.5A Time (2.5ms/Div) Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 VIN = 12V, VOUT = 1.2V, IOUT = 2.5A June 2015 VIN = 12V, VOUT = 1.2V, IOUT = 2.5A Time (2.5ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT6213A/B Application Information Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (IL) about 20% to 50% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (f SW ), the maximum output current (IOUT(MAX)) and estimating a IL as some percentage of that current. L= VOUT VIN VOUT VIN fSW IL Once an inductor value is chosen, the ripple current (IL) is calculated to determine the required peak inductor current. IL = VOUT VIN VOUT I and IL(PEAK) = IOUT(MAX) L VIN fSW L 2 To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and short circuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. IL(PEAK) should not exceed the minimum value of IC's upper current limit level or the IC may not be able to meet the desired output current. If needed, reduce the inductor ripple current (IL) to increase the average inductor current (and the output current) while ensuring that IL(PEAK) does not exceed the upper current limit level. For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Considering the Typical Operating Circuit for 1.2V output at 2.5A and an input voltage of 12V, using an inductor ripple of 0.5A (20%), the calculated inductance value is : L 1.2 12 1.2 4.3μH 12 500kHz 0.5A The ripple current was selected at 0.5A and, as long as we use the calculated 4.3H inductance, that should be the actual ripple current amount. The ripple current and required peak current as below : IL = 1.2 12 1.2 = 0.5A 12 500kHz 4.3μH and IL(PEAK) = 2.5A 0.5A = 2.75A 2 For the 4.3H value, the inductor's saturation and thermal rating should exceed at least 2.75A. For more conservative, the rating for inductor saturation current must be equal to or greater than switch current limit of the device rather than the inductor peak current. Input Capacitor Selection The input filter capacitors are needed to smooth out the switched current drawn from the input power source and to reduce voltage ripple on the input. The actual capacitance value is less important than the RMS current rating (and voltage rating, of course). The RMS input ripple current (IRMS) is a function of the input voltage, output voltage, and load current : IRMS = IOUT(MAX) Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 VOUT VIN VIN 1 VOUT is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Ceramic capacitors are most often used because of their low cost, small size, high RMS current ratings, and robust surge current capabilities. However, take care For the Typical Operating Circuit for 1.2V output and an inductor ripple of 0.5A, with 2 x 22F output capacitance each with about 5m ESR including PCB when these capacitors are used at the input of circuits supplied by a wall adapter or other supply connected through long, thin wires. Current surges through the inductive wires can induce ringing at the RT6213A/B input which could potentially cause large, damaging voltage spikes at VIN. If this phenomenon is observed, some bulk input capacitance may be required. Ceramic capacitors (to meet the RMS current requirement) can be placed in parallel with other types such as tantalum, electrolytic, or polymer (to reduce ringing and overshoot). trace resistance, the output voltage ripple components are : Choose capacitors rated at higher temperatures than required. Several ceramic capacitors may be paralleled to meet the RMS current, size, and height requirements of the application. The typical operating circuit uses two 10F and one 0.1F low ESR ceramic capacitors on VRIPPLE(ESR) = 0.5A 5m = 2.5mV 0.5A = 2.84mV 8 44μF 500kHz = 2.5mV 2.84mV = 5.34mV VRIPPLE(C) = VRIPPLE Output Transient Undershoot and Overshoot In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. However, the combination of small ceramic output the input. capacitors (with little capacitance), low output voltages Output Capacitor Selection (with little stored charge in the output capacitors), and The RT6213A/B are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output Ripple Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE = VRIPPLE(ESR) VRIPPLE(C) VRIPPLE(ESR) = IL RESR VRIPPLE(C) = IL 8 COUT fSW Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 650kHz switching frequency. But some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. The output voltage transient undershoot and overshoot each have two components : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value. is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6213A/B The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR _STEP = IOUT x RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOTTM control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasites) and maximum duty cycle for a given input and output voltage as : tON = VOUT tON and DMAX = VIN fSW tON tOFF(MIN) The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increase compensates for the voltage losses. Calculate the output voltage sag as : VSAG = L (IOUT )2 2 COUT VIN(MIN) DMAX VOUT The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR = L (IOUT )2 2 COUT VOUT For the Typical Operating Circuit for 1.2V output, the circuit has an inductor 4.3H and 2 x 22F output capacitance with 5m ESR each. The ESR step is 2.5A x 2.5m = 6.25mV which is small, as expected. The output voltage sag and soar in response to full 0A-2.5A-0A instantaneous transients are : 1.2V = 200ns 12V 500kHz 200ns and DMAX = = 0.455 200ns 240ns tON = where 240ns is the minimum off time. Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 VSAG = 4.3μH (2.5A)2 = 72mV 2 44μF 12V 0.455 1.2V VSOAR = 4.3μH (2.5A)2 = 254.5mV 2 44μF 1.2V The sag is about 6% of the output voltage and the soar is a full 21.2% of the output voltage. The ESR step is negligible here but it does partially add to the soar, so keep that in mind whenever using higher-ESR output capacitors. The soar is typically much worse than the sag in high input, low-output step-down converters because the high input voltage demands a large inductor value which stores lots of energy that is all transferred into the output if the load stops drawing current. Also, for a given inductor, the soar for a low output voltage is a greater voltage change and an even greater percentage of the output voltage. Any sag is always short-lived, since the circuit quickly sources current to regain regulation in only a few switching cycles. With the RT6213B, any overshoot transient is typically also short-lived since the converter will sink current, reversing the inductor current sharply until the output reaches regulation again. The RT6213A discontinuous operation at light loads prevents sinking current so, for that IC, the output voltage will soar until load current or leakage brings the voltage down to normal. Most applications never experience instantaneous full load steps and the RT6213A/B high switching frequency and fast transient response can easily control voltage regulation at all times. Also, since the sag and soar both are proportional to the square of the load change, if load steps were reduced to 1A (from the 3A examples preceding) the voltage changes would be reduced by a factor of almost ten. For these reasons sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B In any application with large quick transients, always calculate soar to make sure that over-voltage protection will not be triggered. Under-voltage is not likely since the threshold is very low (50%), that function has a long delay (10s), and the IC will quickly return the output to regulation. Over-voltage protection has a minimum threshold of 125% and short delay of 10s and can actually be triggered by incorrect component choices, particularly for the RT6213A which does not sink current. Output Capacitors Stability Criteria The e RT6213A/B's ACOTTM control architecture uses an internal virtual inductor current ramp and other compensation that ensures stability with L= 1.2 12 1.2 = 4.3μH 12 500kHz 0.5A Therefore, the required minimum capacitance for the 12V to 1.2V converter is : 11 COUT 3 5.23 10 12V 4.3μH COUT 3.04μF Any ESR in the output capacitor lowers the required minimum output capacitance, sometimes considerably. For the rare application where that is needed and useful, the equation including ESR is given here : COUT VOUT 2 fSW VIN (RESR 13647 L VOUT ) any reasonable output capacitor. The internal ramp allows the IC to operate with very low ESR capacitors and the IC is stable with very small capacitances. Therefore, output capacitor selection is nearly always a matter of meeting output voltage ripple and transient response requirements, as discussed in the previous sections. For the sake of the unusual application where ripple As can be seen, setting RESR to zero and simplifying the equation yields the previous simpler equation. To allow for the capacitor's temperature and bias voltage coefficients, use at least double the calculated capacitance and use a good quality dielectric such as X5R or X7R with an adequate voltage rating since ceramic capacitors exhibit considerable capacitance reduction as their bias voltage increases. voltage is unimportant and there are few transients Feed-forward Capacitor (Cff) (perhaps battery charging or LED lighting) the stability The RT6213A/B are optimized for ceramic output criteria are discussed below. capacitors and for low duty cycle applications. However for high-output voltages, with high feedback attenuation, The equations giving the minimum required capacitance for stable operation include a term that depends on the output capacitor's ESR. The higher the ESR, the lower the capacitance can be and still ensure stability. The equations can be greatly simplified if the ESR term is removed by setting ESR to zero. The resulting equation gives the worst-case minimum required capacitance and it is usually sufficiently small that there is usually no need for the more exact equation : 11 COUT 3 5.23 10 VIN L the circuit's response becomes over-damped and transient response can be slowed. In high-output voltage circuits (VOUT > 3.3V) transient response is improved by adding a small “feed-forward” capacitor (Cff) across the upper FB divider resistor (Figure 1), to increase the circuit's Q and reduce damping to speed up the transient response without affecting the steady-state stability of the circuit. Choose a suitable capacitor value that following below step. Get the BW the quickest method to do transient response form no load to full load. Confirm the damping frequency. The damping frequency is BW. The worst-case high capacitance requirement is for low VIN and small inductance, so a 12V to 1.2V converter is used for an example. Using the inductance equation in a previous section to determine the required inductance : Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT6213A/B EN REN VIN EN RT6213A/B CEN GND BW Figure 2. External Timing Control VIN VOUT REN 100k EN Q1 Enable R1 RT6213A/B GND Cff FB RT6213A/B Figure 3. Digital Enable Control Circuit R2 GND VIN REN1 EN REN2 RT6213A/B Figure 1. Cff Capacitor Setting GND Cff can be calculated base on below equation : 1 Cff 2 3.1412 R1 BW 0.8 Figure 4. Resistor Divider for Lockout Threshold Setting Output Voltage Setting Enable Operation (EN) For automatic start-up the low-voltage EN pin can be connected to VIN through a 100k resistor. Its large hysteresis band makes EN useful for simple delay and timing circuits. EN can be externally pulled to VIN by adding a resistor-capacitor delay (REN and CEN in Figure 2). Calculate the delay time using EN's internal Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected to FB. The output voltage is set according to the following equation : VOUT = 0.8V x (1 + R1 / R2) VOUT threshold where switching operation begins (1.5V, typical). R1 FB An external MOSFET can be added to implement RT6213A/B digital control of EN when no system voltage above 2V is available (Figure 3). In this case, a 100k pull-up resistor, REN, is connected between VIN and the EN pin. MOSFET Q1 will be under logic control to pull down the EN pin. To prevent enabling circuit when VIN is smaller than the VOUT target value or some other desired voltage level, a resistive voltage divider can be placed between the input voltage and ground and connected to EN to create an additional input under voltage lockout threshold (Figure 4). R2 GND Figure 5. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose R2 between 10k and 100k to minimize power consumption without excessive noise pick-up and calculate R1 as follows : R1 R2 (VOUT VREF ) VREF For output voltage accuracy, use divider resistors with 1% or better tolerance. Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B External BOOT Bootstrap Diode Thermal Considerations When the input voltage is lower than 5.5V it is For continuous operation, do not exceed absolute maximum junction temperature. The maximum power recommended to add an external bootstrap diode between VIN (or VINR) and the BOOT pin to improve enhancement of the internal MOSFET switch and improve efficiency. The bootstrap diode can be a low cost one such as 1N4148 or BAT54. dissipation depends on the thermal resistance of the IC package, PCB layout, rate of surrounding airflow, and difference between junction and ambient temperature. The maximum power dissipation can be calculated by the following formula : External BOOT Capacitor Series Resistance PD(MAX) = (TJ(MAX) TA) / JA The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, but also slow enough to reduce EMI. Switch turn-on is when most EMI occurs where TJ(MAX) is the maximum junction temperature, TA is the ambient temperature, and JA is the junction to ambient thermal resistance. since VLX rises rapidly. During switch turn-off, LX is discharged relatively slowly by the inductor current during the dead time between high-side and low-side switch on-times. In some cases it is desirable to reduce EMI further, at the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (<47) resistance between BOOT and the external bootstrap capacitor. This will slow the high-side switch turn-on and VLX's rise. To remove the resistor from the capacitor charging path (avoiding poor enhancement due to undercharging the BOOT capacitor), use the external diode shown in Figure 6 to charge the BOOT capacitor and place the resistance between BOOT and the capacitor/diode connection. resistance, JA, is 60C/W on a standard four-layer thermal test board. The maximum power dissipation at TA = 25C can be calculated by the following formula : PD(MAX) = (125C 25C) / (60C/W) = 1.667W for TSOT-23-6 (FC) package The maximum power dissipation depends on the operating ambient temperature for fixed TJ(MAX) and thermal resistance, JA. The derating curve in Figure 7 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation. BOOT 0.1μF LX Figure 6. External Bootstrap Diode Maximum Power Dissipation (W)1 2.0 5V RT6213A/B For recommended operating condition specifications, the maximum junction temperature is 125C. The junction to ambient thermal resistance, JA, is layout dependent. For TSOT-23-6 (FC) package, the thermal Four-Layer PCB 1.5 1.0 0.5 0.0 0 25 50 75 100 125 Ambient Temperature (°C) Figure 7. Derating Curve of Maximum Power Dissipation Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT6213A/B Layout Considerations Follow the PCB layout performance of the device. guidelines for optimal Keep the traces of the main current paths as short and wide as possible. Connect all analog grounds to common node and then connect the common node to the power ground behind the output capacitors. For better thermal performance, to design a wide and thick plane for GND pin or to add a lot of vias to GND plane. An example of PCB layout guide is shown in Figure Put the input capacitor as close as possible to VIN and VIN pins. LX node is with high frequency voltage swing and 8 for reference. should be kept at small area. Keep analog components away from the LX node to prevent stray capacitive noise pickup. Connect feedback network behind the output capacitors. Keep the loop area small. Place the feedback components near the device. Place the feedback as close to the IC as possible. VOUT R1 BOOT EN The only GND pin must be connected to a wide and thick plane for thermal consideration. 6 R2 FB 5 4 2 3 GND LX VIN GND CBOOT LX should be connected to inductor by wide and short trace, and keep sensitive components away from this trace. CIN L COUT VOUT Place the input and output capacitors as close to the IC as possible. Figure 8. PCB Layout Guide Copyright © 2015 Richtek Technology Corporation. All rights reserved. www.richtek.com 18 is a registered trademark of Richtek Technology Corporation. DS6213A/B-01 June 2015 RT6213A/B Outline Dimension Dimensions In Millimeters Symbol Dimensions In Inches Min. Max. Min. Max. A 0.700 1.000 0.028 0.039 A1 0.000 0.100 0.000 0.004 B 1.397 1.803 0.055 0.071 b 0.300 0.559 0.012 0.022 C 2.591 3.000 0.102 0.118 D 2.692 3.099 0.106 0.122 e 0.950 0.037 H 0.080 0.254 0.003 0.010 L 0.300 0.610 0.012 0.024 TSOT-23-6 (FC) Surface Mount Package Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. Copyright © 2015 Richtek Technology Corporation. All rights reserved. DS6213A/B-01 June 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 19