® RT2858B 3A, 18V, 650kHz ACOTTM Synchronous Step-Down Converter General Description Features The RT2858B is a synchronous DC/DC step-down converter with Advanced Constant On-Time (ACOTTM) z mode control. It achieves high power density to deliver up to 3A output current from a 4.5V to 18V input supply. The proprietary ACOTTM mode offers an optimal transient response over a wide range of loads and all kinds of ceramic capacitors, which allows the device to adopt very low ESR output capacitors for ensuring performance stabilization. In addition, the RT2858B keeps an excellent constant switching frequency under line and load variation and the integrated synchronous power switches with the ACOTTM mode operation provides high efficiency in whole output current load range. Cycle-by-cycle current limit provides an accurate protection by a valley detection of low-side MOSFET and external soft-start setting eliminates input current surge during startup. Protection functions also include output under voltage protection and thermal shutdown. z z z z z z z z z z ACOTTM Control for Fast Transient, fSW Stability, and Robust Loop Stability with all-MLCC COUT 4.5V to 18V Input Voltage Range 3A Output Current RDSON 120mΩ/ Ω/50mΩ Ω/ Ω for High Efficiency Across IOUT Range and Competitive Advantage for IOUT > 1.5A Advanced Constant On-Time Control Support All Ceramic Capacitors Up to 95% Efficiency 650kHz f SW ; Start-Up into Pre-Biased Load; Adjustable Soft-Start; Internal Bootstrap Adjustable Output Voltage from 0.765V to 8V Enable; UVLO; OCP (Cycle-by-Cycle); and OTP (150°°C) RoHS Compliant and Halogen Free Applications z z z z z Industrial and Commercial Low Power Systems Computer Peripherals LCD Monitors and TVs Green Electronics/Appliances Point of Load Regulation for High-Performance DSPs, FPGAs, and ASICs Simplified Application Circuit RT2858B VIN SW VIN C1 C2 L1 C6 BOOT Enable C5 EN FB SS PVCC GND Load Transient Response C4 VOUT VOUT (50mV/Div) C7 C3 R1 VPVCC R2 IOUT (1A/Div) VIN = 12V, VOUT = 1.05V, IOUT = 0A to 3A Time (100μs/Div) Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT2858B Ordering Information Marking Information RT2858B RT2858BHGSP Package Type SP : SOP-8 (Exposed Pad-Option 2) Lead Plating System G : Green (Halogen Free and Pb Free) H : Hiccup Mode OVP and UVP N : OVP and UVP disable Note : Richtek products are : ` RT2858BHGSP : Product Number RT2858BH GSPYMDNN YMDNN : Date Code RT2858BNGSP RT2858BNGSP : Product Number RT2858BN GSPYMDNN YMDNN : Date Code RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. ` Suitable for use in SnPb or Pb-free soldering processes. Pin Configurations (TOP VIEW) EN 8 FB 2 PVCC SS 3 4 GND 9 VIN 7 BOOT 6 SW 5 GND SOP-8 (Exposed Pad) Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Absolute Maximum Ratings z z z z z z z z z z z (Note 1) Supply Voltage, VIN ----------------------------------------------------------------------------------------------Switch Voltage, SW ----------------------------------------------------------------------------------------------<10ns ----------------------------------------------------------------------------------------------------------------BOOT to SW -------------------------------------------------------------------------------------------------------PVCC to VIN --------------------------------------------------------------------------------------------------------Other Pins -----------------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C SOP-8 (Exposed Pad) -------------------------------------------------------------------------------------------Package Thermal Resistance (Note 2) SOP-8 (Exposed Pad), θJA --------------------------------------------------------------------------------------SOP-8 (Exposed Pad), θJC -------------------------------------------------------------------------------------Junction Temperature Range ------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) -----------------------------------------------------------------------Storage Temperature Range ------------------------------------------------------------------------------------ESD Susceptibility (Note 3) HBM (Human Body Model) --------------------------------------------------------------------------------------- Recommended Operating Conditions z z z −0.3V to 21V −0.8V to (VIN + 0.3V) −5V to 25V −0.3V to 6V −18V to 0.3V −0.3V to 21V 2.041W 49°C/W 8°C/W 150°C 260°C −65°C to 150°C 2kV (Note 4) Supply Voltage, VIN ----------------------------------------------------------------------------------------------- 4.5V to 18V Junction Temperature Range ------------------------------------------------------------------------------------- −40°C to 125°C Ambient Temperature Range ------------------------------------------------------------------------------------- −40°C to 85°C Electrical Characteristics (VIN = 12V, TA = −40°C to 85°C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit Supply Current Shutdown Current ISHDN EN = 0V, TA = 25°C -- 1 10 μA Quiescent Current IQ EN = 5V, VFB = 0.8V, T A = 25°C -- 1 1.3 mA Logic-High 2 -- 18 Logic-Low -- -- 0.4 Logic Threshold EN Input Voltage V VFB Voltage and Discharge Resistance Feedback Threshold Voltage VFB T A = 25°C 0.757 0.765 0.773 V Feedback Input Current IFB VFB = 0.8V, T A = 25°C −0.1 0 0.1 μA VPVCC 6V ≤ VIN ≤ 18V, 0 < IPVCC < 5mA, TA = 25°C 4.8 5.1 5.4 V VPVCC Output VPVCC Output Voltage Line Regulation 6V ≤ VIN ≤ 18V, IPVCC = 5mA -- -- 20 mV Load Regulation 0 < IPVCC < 5mA -- -- 100 mV VIN = 6V, VPVCC = 4V, TA = 25°C -- 70 -- mA Output Current IPVCC Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT2858B Parameter Symbol Test Conditions Min Typ Max Unit RDS(ON) Switch-On Resistance High-Side RDS(ON)_H VBOOT – SW = 5V, TA = 25°C -- 120 -- Low-Side RDS(ON)_L TA = 25°C -- 50 -- I LIM 4 5 6 A TSD -- 150 -- °C -- 20 -- °C mΩ Current Limit Current Limit Thermal Shutdown Thermal Shutdown Threshold Thermal Shutdown Hysteresis ΔTSD On-Time Timer Control On-Time tON VIN = 12V, VOUT = 1.05V -- 135 -- ns Minimum Off-Time tOFF(MIN) VFB = 0.7V, TA = 25°C -- 260 310 ns Soft-Start SS Charge Current VSS = 0V 1.4 2 2.6 μA SS Discharge Current VSS = 0.5V 0.1 0.2 -- mA VIN Rising to Wake up VPVCC 3.6 3.85 4.1 V 130 350 400 mV UVLO UVLO Threshold Hysteresis Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC is measured at the exposed pad of the package. The PCB copper area with exposed pad is 70mm2 (please see PCB Layout section for recommended shape & board physical design guidance). Note 3. Devices are ESD sensitive. Handling precaution is recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Typical Operating Characteristics Efficiency vs. Load Current Output Voltage vs. Input Voltage 100 1.10 VIN = 4.5V 90 1.09 1.08 Output Voltage (V) Efficiency (%) 80 70 60 VIN = 12V 50 40 VIN = 18V 30 20 1.07 1.06 1.05 1.04 1.03 1.02 10 1.01 VOUT = 1.05V 0 0.001 0.01 0.1 VIN = 4.5V to 18V, VOUT = 1.05V, IOUT = 0A 1.00 1 10 4 6 8 Load Current (A) 12 14 18 Output Voltage vs. Output Current 1.065 5.15 1.060 5.10 1.055 Output Voltage (V) 5.20 5.05 5.00 4.95 4.90 4.85 VIN = 18V 1.050 1.045 VIN = 12V 1.040 1.035 1.030 VOUT = 1.05V VIN = 12V, VOUT = 5V, IOUT = 0A 4.80 1.025 -50 -25 0 25 50 75 100 125 0 0.5 1 Temperature (°C) 680 0.78 Reference Voltage (V) 0.80 660 640 620 VOUT = 1.05V, IOUT = 0.3A 4 6 8 10 12 14 16 Input Voltage (V) Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 2 2.5 3 Reference Voltage vs. Temperature 700 600 1.5 Output Current (A) Frequency vs. Input Voltage Frequency (kHz)1 16 Input Voltage (V) Output Voltage vs. Temperature Output Voltage (V) 10 18 0.76 0.74 0.72 VIN = 12V, VOUT = 0.765V 0.70 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT2858B Current Limit vs. Temperature Current Limit vs. Input Voltage 6.0 7.0 6.5 Current Limit (A) Current Limit (A) 5.5 5.0 4.5 4.0 Peak Current 6.0 5.5 5.0 4.5 Valley Current 4.0 3.5 VIN = 12V, VOUT = 1.05V 3.5 -50 -25 0 25 50 75 100 VOUT = 1.05V 3.0 125 4 6 8 10 12 14 Temperature (°C) Input Voltage (V) Load Transient Response VOUT Ripple 16 18 VOUT (10mV/Div) VOUT (50mV/Div) VSW (5V/Div) IOUT (1A/Div) VIN = 12V, VOUT = 1.05V, IOUT = 3A VIN = 12V, VOUT = 1.05V, IOUT = 0A to 3A Time (100μs/Div) Time (500ns/Div) Power On from VIN Power Off from VIN VIN (5V/Div) VIN (5V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) I SW (2A/Div) VIN = 12V, VOUT = 1.05V, IOUT = 3A Time (1ms/Div) Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 VIN = 12V, VOUT = 1.05V, IOUT = 3A Time (5ms/Div) is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Power On from EN Power Off from EN EN (2V/Div) EN (2V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.05V, IOUT = 3A Time (1ms/Div) Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 I SW (2A/Div) VIN = 12V, VOUT = 1.05V, IOUT = 3A Time (5ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT2858B Functional Pin Description Pin No. Pin Name Pin Function 1 EN Enable Control Input. A logic-high enables the converter; a logic-low forces the IC into shutdown mode reducing the supply current to less than 10μA. 2 FB Feedback Voltage Input. It is used to regulate the output of the converter to a set value via an external resistive voltage divider. The feedback threshold voltage is 0.765V typically. 3 PVCC Regulator Output for Internal Circuit. Connect a 1μF capacitor to GND to stabilize output voltage. 4 SS Soft-Start Time Setting. SS controls the soft-start period. Connect a capacitor from SS to GND to set the soft-start period. A 3.9nF capacitor sets the soft-start period of VOUT to 2.6ms. GND Ground. The Exposed pad should be soldered to a large PCB and connected to GND for maximum thermal dissipation. 6 SW Switch Node. Connect this pin to an external L-C filter. 7 BOOT Bootstrap Supply for High-Side Gate Driver. Connect a 0.1μF or greater ceramic capacitor between the BOOT to SW pins. 8 VIN Power Input. The input voltage range is from 4.5V to 18V. Must bypass with a suitably large ( ≥10μF x 2) ceramic capacitor. 5, 9 (Exposed Pad) Function Block Diagram BOOT PVCC Internal Regulator VIN PVCC VIBIAS Over Current PVCC Protection VREF UGATE Switch Controller PVCC SW Driver Ripple Gen. LGATE GND SW 2µA + - - SS FB FB Comparator On-Time EN EN Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Operation In normal operation, the high-side N-MOSFET is turned on when the FB Comparator sets the Switch Controller, and it is turned off when On-Time Controller resets the Switch Controller. While the high-side N-MOSFET is turned off, the low-side N-MOSFET is turned on and waits for the FB Comparator to set the beginning of next cycle. The FB Comparator sets the Switch Controller by comparing the feedback signal (FB) from output voltage with the internal 0.765V reference. When load transient induces VOUT drop, the FB voltage will be less than its threshold voltage. This means that the high-side NMOSFET will turn on again immediately after minimum off-time expired. The switching frequency will vary during the transient period thus can provide a very fast transient response. After the load transient finished, the RT2858B will be back to steady state with a constant switching frequency. Enable Internal Regulator Provide internal power for logic control and switch gate drivers. On-Time Controller Control on-time according to VIN and SW to obtain constant switching frequency. OVP/UVP Protection The RT2858B detects over and under voltage conditions by monitoring the feedback voltage on FB pin. The two functions are enabled after approximately 1.7 times the soft-start time. When the feedback voltage becomes higher than 120% of the target voltage, the OVP comparator will go high to turn off both internal high side and low side MOSFETs. When the feedback voltage is lower than 70% of the target voltage for 250μs, the UVP comparator will go high to turn off both internal high side and low side MOSFETs. Activate internal regulator once EN input level is higher than the target level. Force IC to enter shutdown mode when the EN input level is lower than 0.4V Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT2858B Typical Application Circuit VIN C1 10µF x 2 C2 0.1µF RT2858B 6 8 SW VIN 1 Enable 5, 9 (Exposed Pad) C5 3.9nF BOOT EN FB GND 4 SS PVCC 7 L1 1.4µH C6 0.1µF C3 C7 22µF x 2 R1 8.25k VOUT 1.05V/3A 2 3 VPVCC R2 22.1k C4 1µF Table 1. Suggested Component Values (VIN = 12V) VOUT (V) R1 (kΩ) R2 (kΩ) C3 (pF) L1 (μH) C7 (μF) 1 6.81 22.1 -- 1.4 22 to 68 1.05 8.25 22.1 -- 1.4 22 to 68 1.2 12.7 22.1 -- 1.4 22 to 68 1.8 30.1 22.1 5 to 22 2 22 to 68 2.5 49.9 22.1 5 to 22 2 22 to 68 3.3 73.2 22.1 5 to 22 2 22 to 68 5 124 22.1 5 to 22 3.3 22 to 68 7 180 22.1 5 to 22 3.3 22 to 68 Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Design Procedure Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (ΔIL) about 20-50% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (f SW), the maximum output current (IOUT(MAX)) and estimating a ΔIL as some percentage of that current. L= VOUT × ( VIN − VOUT ) VIN × fSW × ΔIL Once an inductor value is chosen, the ripple current (ΔIL) is calculated to determine the required peak inductor current. VOUT × ( VIN − VOUT ) ΔI ΔIL = and IL(PEAK) = IOUT(MAX) + L VIN × fSW × L 2 To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and shortcircuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. IL(PEAK) should not exceed the minimum value of IC's upper current limit level or the IC may not be able to meet the desired output current. If needed, reduce the inductor ripple current (ΔIL) to increase the average inductor current (and the output current) while ensuring that IL(PEAK) does not exceed the upper current limit level. Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Considering the Typical Operating Circuit for 1.05V output at 3A and an input voltage of 12V, using an inductor ripple of 1A (33%), the calculated inductance value is : L= 1.05V × (12V − 1.05V ) = 1.47μH 12V × 650kHz × 1A The ripple current was selected at 1A and, as long as we use the calculated 1.47μH inductance, that should be the actual ripple current amount. Typically the exact calculated inductance is not readily available and a nearby value is chosen. In this case 1.4μH was available and actually used in the typical circuit. To illustrate the next calculation, assume that for some reason is was necessary to select a 1.8μH inductor (for example). We would then calculate the ripple current and required peak current as below : 1.05V × (12V − 1.05V ) ΔIL = = 0.82A 12V × 650kHz × 1.8μH and IL(PEAK) = 3A + 0.82 = 3.41A 2 For the 1.8μH value, the inductor's saturation and thermal rating should exceed 3.41A. Since the actual value used was 1.4μH and the ripple current exactly 1A, the required peak current is 3.53A. Input Capacitor Selection The input filter capacitors are needed to smooth out the switched current drawn from the input power source and to reduce voltage ripple on the input. The actual capacitance value is less important than the RMS current rating (and voltage rating, of course). The RMS input ripple current (IRMS) is a function of the input voltage, output voltage, and load current : IRMS = IOUT × VOUT × ( VVIN − VOUT ) VVIN Ceramic capacitors are most often used because of their low cost, small size, high RMS current ratings, and robust surge current capabilities. However, take care when these is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT2858B capacitors are used at the input of circuits supplied by a wall adapter or other supply connected through long, thin wires. Current surges through the inductive wires can induce ringing at the RT2858B's input which could potentially cause large, damaging voltage spikes VIN. If this phenomenon is observed, some bulk input capacitance may be required. Ceramic capacitors (to meet the RMS current requirement) can be placed in parallel with other types such as tantalum, electrolytic, or polymer (to reduce ringing and overshoot). Choose capacitors rated at higher temperatures than required. Several ceramic capacitors may be paralleled to meet the RMS current, size, and height requirements of the application. The typical operating circuit uses two 10μF and one 0.1μF low ESR ceramic capacitors on the input. Output Capacitor Selection The RT2858B are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output Ripple Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) VRIPPLE(ESR) = ΔIL × RESR ΔIL VRIPPLE(C) = 8 × COUT × fSW For the Typical Operating Circuit for 1.05V output and an inductor ripple of 1A, with 2 x 22μF output capacitance each with about 10mΩ ESR including PCB trace resistance, the output voltage ripple components are : VRIPPLE(C) = 1A = 4.4mV 8 × 44μF × 0.65MHz VRIPPLE = 5mV + 4.4mV = 9.4mV Output Transient Undershoot and Overshoot In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. However, the combination of small ceramic output capacitors (with little capacitance), low output voltages (with little stored charge in the output capacitors), and low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 650kHz switching frequency. But some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. The output voltage transient undershoot and overshoot each have two components : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value. The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor: VESR_STEP = ΔIOUT × RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOTTM control scheme will ramp the current using on-times spaced apart with minimum off-times, which is VRIPPLE(ESR) = 1A × 5mΩ = 5mV Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B as fast as allowed. Calculate the approximate on-time (neglecting parasitics) and maximum duty cycle for a given input and output voltage as : VOUT tON and DMAX = VIN × fSW tON + tOFF(MIN) tON = The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increase compensates for the voltage losses. Calculate the output voltage sag as : VSAG = L × (ΔIOUT )2 2 × COUT × ( VIN(MIN) × DMAX − VOUT ) The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR = L × (ΔIOUT )2 2 × COUT × VOUT For the Typical Operating Circuit for 1.05V output, the circuit has an inductor 1.4μH and 2 x 22μF output capacitance with 5mΩ ESR each. The ESR step is 3A x 2.5mΩ = 7.5mV which is small, as expected. The output voltage sag and soar in response to full 0A-3A-0A instantaneous transients are : tON = 1.05V = 135ns 12V × 650kHz and DMAX VSAG = 135ns = = 0.34 135ns + 260ns 1.4μH × (3A)2 = 47mV 2 × 44μF × (12V × 0.34 − 1.05V ) VSOAR = 1.4μH × (3A)2 = 136mV 2 × 44μF × 1.05V The sag is about 4% of the output voltage and the soar is a full 13% of the output voltage. The ESR step is negligible here but it does partially add to the soar, so keep that in mind whenever using higher-ESR output capacitors. The soar is typically much worse than the sag in highinput, low-output step-down converters because the high input voltage demands a large inductor value which stores lots of energy that is all transferred into the output if the load stops drawing current. Also, for a given inductor, the soar for a low output voltage is a greater voltage change Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 and an even greater percentage of the output voltage. This is illustrated by comparing the previous to the next example. The Typical Operating Circuit for 12V to 3.3V with a 2μH inductor and 2 x 22μF output capacitance can be used to illustrate the effect of a higher output voltage. The output voltage sag and soar in response to full 0A-3A-0A instantaneous transients are calculated as follows : t ON = 3.3V = 423ns 12V × 650kHz and DMAX = 423ns = 0.62 423ns + 260ns VSAG = 2μH × (3A)2 = 49.5mV 2 × 44μF × (12V × 0.62 − 3.3V ) VSOAR = 2μH × (3A)2 = 62mV 2 × 44μF × 3.3V In this case the sag is about 1.5% of the output voltage and the soar is only 2% of the output voltage. Any sag is always short-lived, since the circuit quickly sources current to regain regulation in only a few switching cycles. With the RT2858B, any overshoot transient is typically also short-lived since the converter will sink current, reversing the inductor current sharply until the output reaches regulation again. Most applications never experience instantaneous full load steps and the RT2858B's high switching frequency and fast transient response can easily control voltage regulation at all times. Also, since the sag and soar both are proportional to the square of the load change, if load steps were reduced to 1A (from the 3A examples preceding) the voltage changes would be reduced by a factor of almost ten. For these reasons sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT2858B Output Capacitors Stability Criteria The RT2858B's ACOTTM control architecture uses an internal virtual inductor current ramp and other compensation that ensures stability with any reasonable output capacitor. The internal ramp allows the IC to operate with very low ESR capacitors and the IC is stable with very small capacitances. Therefore, output capacitor selection is nearly always a matter of meeting output voltage ripple and transient response requirements, as discussed in the previous sections. For the sake of the unusual application where ripple voltage is unimportant and there are few transients (perhaps battery charging or LED lighting) the stability criteria are discussed below. The equations giving the minimum required capacitance for stable operation include a term that depends on the output capacitor's ESR. The higher the ESR, the lower the capacitance can be and still ensure stability. The equations can be greatly simplified if the ESR term is removed by setting ESR to zero. The resulting equation gives the worst-case minimum required capacitance and it is usually sufficiently small that there is usually no need for the more exact equation. Any ESR in the output capacitor lowers the required minimum output capacitance, sometimes considerably. For the rare application where that is needed and useful, the equation including ESR is given here : VOUT COUT ≥ 2 × fSW × VIN × (RESR + 13647 × L × VOUT ) As can be seen, setting RESR to zero and simplifying the equation yields the previous simpler equation. To allow for the capacitor's temperature and bias voltage coefficients, use at least double the calculated capacitance and use a good quality dielectric such as X5R or X7R with an adequate voltage rating since ceramic capacitors exhibit considerable capacitance reduction as their bias voltage increases. The required output capacitance (COUT) is a function of the inductor value (L) and the input voltage (VIN) : −11 COUT ≥ 5.64 × 10 VIN × L The worst-case high capacitance requirement is for low VIN and small inductance, so a 5V to 3.3V converter is used for an example. Using the inductance equation in a previous section to determine the required inductance : 3.3V × ( 5V − 3.3V ) L= = 1.73μH 5V × 650kHz × 1A Therefore, the required minimum capacitance for the 5V to 3.3V converter is : −11 COUT ≥ 5.64 × 10 = 6.5μF 5V × 1.73μH Using the 12V to 1.05V typical application as another example : −11 COUT ≥ 5.64 × 10 = 3.4μF 12V × 1.4μH Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Applications Information Current-Sinking Applications Soft-Start (SS) The RT2858B's is not recommended for current sinking applications even though its continuous switching operation allows the IC to sink some current. Sinking enables a fast recovery from output voltage overshoot caused by load transients and is normally useful for applications requiring negative currents, such as DDR VTT bus termination applications and changing-output voltage applications where the output voltage needs to slew quickly from one voltage to another. However, the IC's negative current limit is set low (about 1.6A) and the current limit behavior latches the synchronous rectifier off until the high-side switch's next pulse, to prevent the possibility of IC damage from large negative currents. Therefore, sinking current is not necessarily available at all times. The RT2858B soft-start uses an external capacitor at SS to adjust the soft-start timing according to the following equation : C (nF) × 1.065V tSS (ms) = SS ISS (μA) The available capacitance range is from 2.7nF to 220nF. If a 3.9nF capacitor is used, the typical soft-start will be 2ms. Do not leave SS unconnected. If implementing applications where current-sinking may occur, take care to allow for the current that is delivered to the input supply. A step-down converter in sinking operation functions like a backwards step-up converter. The current that is sunk at its output terminals is delivered up to its input terminals. If this current has no outlet, the input voltage will rise. A good arrangement for long-term sinking applications is for a sinking supply (supply A) that is sinking current sourced from supply B, to both be powered by the same input supply. That way, any current delivered back to the input by supply A is current that just left the input through supply B. In this way, the current simply makes a round trip and the input supply will not rise. In cases where this is not possible, make sure that there are sufficient other loads on the input supply to prevent that supply's voltage from rising high enough to cause damage to itself or any of its loads. In cases where the sinking is not long-term, such as output-voltage slewing applications, make sure there is sufficient input capacitance to control any input voltage rise. The worst-case voltage rise is : C × ΔVOUT ΔVIN = OUT CIN Enable Operation (EN) For automatic start-up the high-voltage EN pin can be connected to VIN, either directly or through a 100kΩ resistor. Its large hysteresis band makes EN useful for simple delay and timing circuits. EN can be externally pulled to VIN by adding a resistor-capacitor delay (REN and CEN in Figure 1). Calculate the delay time using EN's internal threshold where switching operation begins (1.2V, typical). An external MOSFET can be added to implement digital control of EN when no system voltage above 2V is available (Figure 2). In this case, a 100kΩ pull-up resistor, REN, is connected between VIN and the EN pin. MOSFET Q1 will be under logic control to pull down the EN pin. To prevent enabling circuit when VIN is smaller than the VOUT target value or some other desired voltage level, a resistive voltage divider can be placed between the input voltage and ground and connected to EN to create an additional input undervoltage lockout threshold (Figure 3). EN VIN REN EN RT2858B CEN GND Figure 1. External Timing Control VIN Enable REN 100k EN Q1 RT2858B GND Figure 2. Digital Enable Control Circuit Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT2858B VIN External BOOT Bootstrap Diode REN1 EN REN2 RT2858B GND Figure 3. Resistor Divider for Lockout Threshold Setting Output Voltage Setting Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected to FB. The output voltage is set according to the following equation : R1 VOUT = 0.765 × (1+ ) R2 When the input voltage is lower than 5.5V it is recommended to add an external bootstrap diode between VIN (or VCC) and the BOOT pin to improve enhancement of the internal MOSFET switch and improve efficiency. The bootstrap diode can be a low cost one such as 1N4148 or BAT54. 5V BOOT RT2858B 0.1µF SW VOUT Figure 5. External Bootstrap Diode R1 External BOOT Capacitor Series Resistance FB RT2858B R2 GND Figure 4. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose R2 between 10kΩ and 100kΩ to minimize power consumption without excessive noise pick-up and calculate R1 as follows : R2 × (VOUT − 0.765V) R1 = 0.765V For output voltage accuracy, use divider resistors with 1% or better tolerance. Under-Voltage Lockout Protection The RT2858B feature an Under-Voltage Lockout (UVLO) function that monitors the internal linear regulator output (VPVDD) and prevents operation if VPVDD is too low. In some multiple input voltage applications, it may be desirable to use a power input that is too low to allow VPVDD to exceed the UVLO threshold. In this case, if there is another lowpower supply available that is high enough to operate the VPVDD regulator, connecting that supply to VCC will allow the IC to operate, using the lower-voltage high-power supply for the DC/DC power path. Because of the internal linear regulator, any supply regulated or unregulated) between 4.5V and 18V will operate the IC. Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, but also slow enough to reduce EMI. Switch turn-on is when most EMI occurs since VSW rises rapidly. During switch turn-of, SW is discharged relatively slowly by the inductor current during the deadtime between high-side and low-switch on-times. In some cases it is desirable to reduce EMI further, at the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (<10Ω) resistance between BOOT and the external bootstrap capacitor. This will slow the high-side switch turn-on and VSW's rise. To remove the resistor from the capacitor charging path (avoiding poor enhancement due to undercharging the BOOT capacitor), use the external diode shown in Figure 5 to charge the BOOT capacitor and place the resistance between BOOT and the capacitor/diode connection. VPVDD Capacitor Selection Decouple VPVDD to PGND with a 1μF ceramic capacitor. High grade dielectric (X7R, or X5R) ceramic capacitors are recommended for their stable temperature and bias voltage characteristics. is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Thermal Considerations Recommendations for PCB Layout For continuous operation, do not exceed absolute maximum junction temperature. The maximum power dissipation depends on the thermal resistance of the IC package, PCB layout, rate of surrounding airflow, and difference between junction and ambient temperature. The maximum power dissipation can be calculated by the following formula : ` ` ` 1 Ounce Copper on Top Layer, plated-up through SMT PCB Mfg Process 1 Ounce Copper on Top Layer will improve Thermal performance Minimum 4 Layer PCB Stack up. Place the shape with 70mm2 as Figure 7 around the PSOP-8 Footprint to achieve best thermal performance. PD(MAX) = (TJ(MAX) − TA) / θJA where TJ(MAX) is the maximum junction temperature, TA is the ambient temperature, and θJA is the junction to ambient thermal resistance. For recommended operating condition specifications, the maximum junction temperature is 125°C. The junction to ambient thermal resistance, θJA, is layout dependent. For SOP-8 (Exposed Pad) package, the thermal resistance, θJA, is 49°C/W on a standard JEDEC 51-7 four-layer thermal test board. The maximum power dissipation at TA = 25°C can be calculated by the following formula : PD(MAX) = (125°C − 25°C) / (49°C/W) = 2.041W for SOP-8 (Exposed Pad) package Maximum Power Dissipation (W)1 The maximum power dissipation depends on the operating ambient temperature for fixed T J(MAX) and thermal resistance, θJA. The derating curve in Figure 6 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation. 3.0 Copper Area = 70mm2, θJA = 49°C/W Figure 7. PCB Copper Area ` Utilize Standard PTH (Plated Through Hole, 25mil diameter, as Figure 8) to Via down from Exposed Pad on Top Layer, to GND Plane on PCB. Figure 8. Standard PTH to GND Plane Four-Layer PCB 2.5 2.0 1.5 1.0 0.5 0.0 0 25 50 75 100 125 Ambient Temperature (°C) Figure 6. Derating Curve of Maximum Power Dissipation Copyright © 2013 Richtek Technology Corporation. All rights reserved. DS2858B-00 September 2013 is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT2858B Layout Consideration Follow the PCB layout guidelines for optimal performance of the RT2858B ` Keep the traces of the main current paths as short and wide as possible. ` Put the input capacitor as close as possible to the device pins (VIN and GND). The resistor divider must be connected as close to the device as possible. R2 GND C4 C5 SW node is with high frequency voltage swing and should be kept at small area. Keep sensitive components away from the SW node to prevent stray capacitive noise pickup. ` Connect feedback network behind the output capacitors. Keep the loop area small. Place the feedback components near the RT2858B feedback pin. ` The GND and Exposed Pad should be connected to a strong ground plane for heat sinking and noise protection. Input capacitor must be placed C1 as close to the IC as possible. VOUT R1 ` C2 EN 8 FB 2 PVCC SS 3 GND 4 9 SW should be connected to inductor by wide and short trace. Keep sensitive components away from this trace. VIN 7 BOOT 6 SW 5 GND C6 C7 L1 VOUT Figure 9. PCB Layout Guide Copyright © 2013 Richtek Technology Corporation. All rights reserved. www.richtek.com 18 is a registered trademark of Richtek Technology Corporation. DS2858B-00 September 2013 RT2858B Outline Dimension H A M EXPOSED THERMAL PAD (Bottom of Package) Y J X B F C I D Dimensions In Millimeters Dimensions In Inches Symbol Min Max Min Max A 4.801 5.004 0.189 0.197 B 3.810 4.000 0.150 0.157 C 1.346 1.753 0.053 0.069 D 0.330 0.510 0.013 0.020 F 1.194 1.346 0.047 0.053 H 0.170 0.254 0.007 0.010 I 0.000 0.152 0.000 0.006 J 5.791 6.200 0.228 0.244 M 0.406 1.270 0.016 0.050 X 2.000 2.300 0.079 0.091 Y 2.000 2.300 0.079 0.091 X 2.100 2.500 0.083 0.098 Y 3.000 3.500 0.118 0.138 Option 1 Option 2 8-Lead SOP (Exposed Pad) Plastic Package Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. DS2858B-00 September 2013 www.richtek.com 19