®
RT7294D
2.5A, 18V, 500kHz ACOTTM Synchronous Step-Down Converter
General Description
Features
The RT7294D is a synchronous step-down converter with
Advanced Constant On-Time (ACOTTM) mode control. The
4.3V to 18V Input Voltage Range
2.5A Output Current
Advanced Constant On-Time Control
Fast Transient Response
Support All Ceramic Capacitors
Up to 95% Efficiency
500kHz Switching Frequency
Adjustable Output Voltage from 0.6V to 8V
Cycle-by-Cycle Current Limit
Input Under-Voltage Lockout
Thermal Shutdown
TM
ACOT provides a very fast transient response with few
external components. The low impedance internal
MOSFET supports high efficiency operation with wide input
voltage range from 4.3V to 18V. The proprietary circuit of
the RT7294D enables to support all ceramic capacitors.
Ordering Information
RT7294D
Package Type
J6F : TSOT-23-6 (FC)
Applications
Lead Plating System
G : Green (Halogen Free and Pb Free)
Note :
Richtek products are :
RoHS compliant and compatible with the current require-
ments of IPC/JEDEC J-STD-020.
Suitable for use in SnPb or Pb-free soldering processes.
Pin Configurations
Marking Information
07= : Product Code
(TOP VIEW)
07=DNN
SW VIN EN
6
Industrial and Commercial Low Power Systems
Computer Peripherals
LCD Monitors and TVs
Green Electronics/Appliances
Point of Load Regulation for High-Performance DSPs,
FPGAs, and ASICs
5
4
2
3
DNN : Date Code
BOOT GND FB
TSOT-23-6 (FC)
Simplified Application Circuit
RT7294D
VIN
BOOT
VIN
SW
Enable
VOUT
FB
EN
GND
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RT7294D
Functional Pin Description
Pin No.
Pin Name
Pin Function
1
BOOT
Bootstrap Supply for High-Side Gate Driver. Connect a 0.1F ceramic capacitor
between the BOOT and SW pins.
2
GND
Power Ground.
3
FB
Feedback Voltage Input. The pin is used to set the output voltage of the converter
via a resistive divider. The converter regulates VFB to 0.6V
4
EN
Enable Control Input. Connect EN to a logic-high voltage to enable the IC or to a
logic-low voltage to disable. Do not leave this high impedance input unconnected.
5
VIN
Power Input. The input voltage range is from 4.3V to 18V. Must bypass with a
suitable large ceramic capacitor at this pin.
6
SW
Switch Node. Connect to external L-C filter.
Function Block Diagram
BOOT
VIN
VIN
Reg
PVCC
VIBIAS
VREF
Min off
PVCC
UGATE
OC
Control
SW
Driver
LGATE
OV
GND
PVCC
SW
Ripple
Gen.
+
Comparator
FB
GND SW
VIN
EN
On-Time
SW
EN
Operation
The RT7294D is a synchronous step-down converter with
advanced constant on-time control mode. Using the ACOT
control mode can reduce the output capacitance and fast
transient response. It can minimize the component size
without additional external compensation network.
UVLO Protection
Current Protection
Thermal Shutdown
The inductor current is monitored via the internal switches
cycle-by-cycle.
When the junction temperature exceeds the OTP
threshold value, the IC will shut down the switching
operation. Once the junction temperature cools down and
is lower than the OTP lower threshold, the converter will
autocratically resume switching.
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To protect the chip from operating at insufficient supply
voltage, the UVLO is needed. When the input voltage of
VIN is lower than the UVLO falling threshold voltage, the
device will be lockout.
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DS7294D-05 July 2015
RT7294D
Absolute Maximum Ratings
(Note 1)
VIN to GND ----------------------------------------------------------------------------------------------------SW to GND ---------------------------------------------------------------------------------------------------< 10ns ----------------------------------------------------------------------------------------------------------BOOT to GND ------------------------------------------------------------------------------------------------Other Pins -----------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C
TSOT-23-6 (FC) ----------------------------------------------------------------------------------------------- 1.429W
Package Thermal Resistance (Note 2)
TSOT-23-6 (FC), θJA ------------------------------------------------------------------------------------------ 70°C/W
TSOT-23-6 (FC), θJC ----------------------------------------------------------------------------------------- 15°C/W
Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------ 260°C
Junction Temperature ---------------------------------------------------------------------------------------- 150°C
Storage Temperature Range ------------------------------------------------------------------------------- −65°C to 150°C
ESD Susceptibility (Note 3)
HBM (Human Body Model) --------------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
−0.3V to 20V
−0.3V to (VIN + 0.3V)
−5V to 25V
(VSW − 0.3V) to (VSW + 6V)
−0.3V to 6V
(Note 4)
Supply Input Voltage, VIN ---------------------------------------------------------------------------------- 4.3V to 18V
Junction Temperature Range ------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range ------------------------------------------------------------------------------- −40°C to 85°C
Electrical Characteristics
(VIN = 12V, TA = 25°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Shutdown Current
ISHDN
VEN = 0V
--
--
4
A
Quiescent Current
IQ
VEN = 2V, VFB = 1V
--
0.5
--
mA
High-Side
RDS(ON)_H
VBOOTSW = 4.8V
--
150
--
Low-Side
RDS(ON)_L
VIN = 5V
--
90
--
Current Limit
ILIM
Valley Current
2.7
3.4
4
A
Oscillator Frequency
f SW
--
500
--
kHz
Maximum Duty Cycle
DMAX
--
90
--
%
Minimum On-Time
tON
--
60
--
ns
Feedback Threshold Voltage
VFB
591
600
609
mV
Switch-On
Resistance
EN Input Threshold
In CCM Mode
m
Logic-High
VEN_H
1.19
1.29
1.39
Logic-Low
VEN_L
0.93
1.03
1.13
3.55
3.9
4.25
V
--
340
--
mV
VIN Under-Voltage Lockout
Threshold
VIN Under-Voltage Lockout
Threshold Hysteresis
VUVLO
VIN Rising
V
Soft-Start Time
tSS
--
800
--
s
Thermal Shutdown Threshold
TSD
--
160
--
C
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RT7294D
Parameter
Symbol
Thermal Shutdown Hysteresis
TSD
VOUT Discharge Resistance
RDISCHG
Test Conditions
EN = 0V, VOUT = 0.5V
Min
Typ
Max
Unit
--
20
--
C
--
50
100
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may
affect device reliability.
Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC is
measured at the top of the package.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
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is a registered trademark of Richtek Technology Corporation.
DS7294D-05 July 2015
RT7294D
Typical Application Circuit
RT7294D
VIN
4.3V to 18V
5
CIN
10µF
BOOT
VIN
SW 6
4 EN
Enable
1
CBOOT
100nF
L
2µH
CFF
FB 3
R1
10k
R2
10k
GND 2
VOUT
1.2V
COUT
22µF x 2
Table 1. Suggested Component Values
VOUT (V)
R1 (k)
R2 (k)
L (H)
COUT (F)
CFF (pF)
5
110
15
4.7
22 x 2
39
3.3
115
25.5
3.6
22 x 2
33
2.5
25.5
8.06
3.6
22 x 2
NC
1.2
10
10
2
22 x 2
NC
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RT7294D
Typical Operating Characteristics
Efficiency vs. Load Current
100
90
90
80
80
70
VIN =
VIN =
VIN =
VIN =
VIN =
60
50
40
30
Efficiency (%)
Efficiency (%)
Efficiency vs. Load Current
100
4.5V
5V
9V
12V
18V
VIN = 12V
VIN = 15V
VIN = 18V
70
60
50
40
30
20
20
10
10
VOUT = 1.2V
0
0
0.25 0.5 0.75
1
1.25 1.5 1.75
2
VOUT = 5V
0
0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
2.25 2.5
2 2.2 2.4 2.6
Load Current (A)
Load Current (A)
Reference Voltage vs. Input Voltage
Reference Voltage vs. Temperature
0.65
0.620
Reference Voltage (V)
Reference Voltage (V)
0.615
0.610
0.605
0.600
0.595
0.590
0.585
0.63
0.61
0.59
0.57
VIN = 4.5V to 18V, VOUT = 1.2V, IOUT = 0A
VIN = 12V, IOUT = 0A
0.55
0.580
4
5
6
7
8
-50
9 10 11 12 13 14 15 16 17 18
-25
0
1.242
590
Output Voltage (V)
Switching Frequency (kHz)1
600
VIN = 18V
VIN = 12V
VIN = 9V
VIN = 4.5V
1.226
1.218
1.210
1.202
1.194
1.186
1.178
VIN = 4.5V to 18V, VOUT = 1.2V
1.170
0
0.25 0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5
Load Current (A)
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50
75
100
125
Switching Frequency vs. Input Voltage
Output Voltage vs. Load Current
1.250
1.234
25
Temperature (°C)
Input Voltage (V)
580
570
560
550
540
530
520
510
VIN = 4.5V to 18V, VOUT = 1.2V, IOUT = 0A
500
4
6
8
10
12
14
16
18
Input Voltage (V)
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RT7294D
Current Limit vs. Temperature
4.0
630
3.9
610
3.8
VIN = 6V
VIN = 12V
VIN = 18V
VIN = 4.5V
590
570
550
Current Limit (A)
Switching Frequency (kHz)1
Switching Frequency vs. Temperature
650
530
510
490
3.7
3.6
3.5
3.4
3.3
3.2
470
VOUT = 1.2V, IOUT = 0A
450
-50
-25
0
25
50
75
100
125
3.1
3.0
-50
-25
0
25
50
75
100
Temperature (°C)
Temperature (°C)
Load Transient Response
Load Transient Response
VOUT
(50mV/Div)
VOUT
(50mV/Div)
IOUT
(1A/Div)
IOUT
(1A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 1.25A to 2.5A
VIN = 12V, VOUT = 1.2V, IOUT = 0A to 2.5A
Time (250μs/Div)
Time (250μs/Div)
Switching
Switching
VOUT
(10mV/Div)
VOUT
(10mV/Div)
VSW
(5V/Div)
VSW
(5V/Div)
I SW
(1A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 1.25A
Time (1μs/Div)
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125
I SW
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (1μs/Div)
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RT7294D
Power On from EN
Power Off from EN
VEN
(2V/Div)
VEN
(2V/Div)
VOUT
(1V/Div)
VSW
(10V/Div)
VOUT
(1V/Div)
VSW
(10V/Div)
IOUT
(2A/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (2.5ms/Div)
Time (5ms/Div)
Power On from VIN
Power Off from VIN
VIN
(10V/Div)
VIN
(10V/Div)
VOUT
(1V/Div)
VSW
(10V/Div)
VOUT
(1V/Div)
VSW
(10V/Div)
IOUT
(2A/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (2.5ms/Div)
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VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
VIN = 12V, VOUT = 1.2V, IOUT = 2.5A
Time (5ms/Div)
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DS7294D-05 July 2015
RT7294D
Application information
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor value
is generally flexible and is ultimately chosen to obtain the
best mix of cost, physical size, and circuit efficiency.
Lower inductor values benefit from reduced size and cost
and they can improve the circuit's transient response, but
they increase the inductor ripple current and output voltage
ripple and reduce the efficiency due to the resulting higher
peak currents. Conversely, higher inductor values increase
efficiency, but the inductor will either be physically larger
or have higher resistance since more turns of wire are
required and transient response will be slower since more
time is required to change current (up or down) in the
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (ΔIL) about
20% to 40% of the desired full output load current.
Calculate the approximate inductor value by selecting the
input and output voltages, the switching frequency (fSW),
the maximum output current (IOUT(MAX)) and estimating a
ΔIL as some percentage of that current.
VOUT VIN VOUT
L=
VIN fSW IL
Once an inductor value is chosen, the ripple current (ΔIL)
is calculated to determine the required peak inductor
current.
VOUT VIN VOUT
VIN fSW L
I
IL(PEAK) = IOUT(MAX) L
2
IL
IL(VALLY) = IOUT(MAX)
2
IL =
Considering the Typical Operating Circuit for 1.2V output
at 2.5A and an input voltage of 12V, using an inductor
ripple of 1.08A (30%), the calculated inductance value is :
1.2 12 1.2
L=
= 2μH
12 500kHz 1.08
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The ripple current was selected at 1.08A and, as long as
we use the calculated 2μH inductance, that should be the
actual ripple current amount. The ripple current and required
peak current as below :
IL =
1.2 12 1.2
= 1.08A
12 500kHz 2μH
and 2.88μH = 2.5A 1.08 = 3.58A
2
Inductor saturation current should be chosen over IC's
current limit.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source and
to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS current
rating (and voltage rating, of course). The RMS input ripple
current (IRMS) is a function of the input voltage, output
voltage, and load current :
V
IRMS = IOUT(MAX) OUT
VIN
VIN
1
VOUT
Ceramic capacitors are most often used because of their
low cost, small size, high RMS current ratings, and robust
surge current capabilities. However, take care when these
capacitors are used at the input of circuits supplied by a
wall adapter or other supply connected through long, thin
wires. Current surges through the inductive wires can
induce ringing at the RT7294D input which could potentially
cause large, damaging voltage spikes at VIN. If this
phenomenon is observed, some bulk input capacitance
may be required. Ceramic capacitors (to meet the RMS
current requirement) can be placed in parallel with other
types such as tantalum, electrolytic, or polymer (to reduce
ringing and overshoot).
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled to
meet the RMS current, size, and height requirements of
the application. The typical operating circuit use 10μF and
one 0.1μF low ESR ceramic capacitors on the input.
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RT7294D
Output Capacitor Selection
The RT7294D is optimized for ceramic output capacitors
and best performance will be obtained using them. The
total output capacitance value is usually determined by
the desired output voltage ripple level and transient response
requirements for sag (undershoot on positive load steps)
and soar (overshoot on negative load steps).
Output Ripple
Output ripple at the switching frequency is caused by the
inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are similar
in amplitude and both should be considered if ripple is
critical.
VRIPPLE = VRIPPLE(ESR) VRIPPLE(C)
VRIPPLE(ESR) = IL RESR
VRIPPLE(C) =
IL
8 COUT fSW
For the Typical Operating Circuit for 1.2V output and an
inductor ripple of 1.08A, with 22μF output capacitance each
with about 5mΩ ESR including PCB trace resistance, the
output voltage ripple components are :
VRIPPLE(ESR) = 1.08A 5m = 5.4mV
VRIPPLE(C) =
1.08A
= 12mV
8 22μF 500kHz
VRIPPLE = 2.7mV 12mV = 14.7mV
Output Transient Undershoot and Overshoot
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load steps
up and down abruptly. The ACOT transient response is
very quick and output transients are usually small.
However, the combination of small ceramic output
capacitors (with little capacitance), low output voltages
(with little stored charge in the output capacitors), and
low duty cycle applications (which require high inductance
to get reasonable ripple currents with high input voltages)
increases the size of voltage variations in response to
very quick load changes. Typically, load changes occur
slowly with respect to the IC's 500kHz switching frequency.
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But some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings in
response to very fast load steps.
The output voltage transient undershoot and overshoot each
have two components : the voltage steps caused by the
output capacitor's ESR, and the voltage sag and soar due
to the finite output capacitance and the inductor current
slew rate. Use the following formulas to check if the ESR
is low enough (typically not a problem with ceramic
capacitors) and the output capacitance is large enough to
prevent excessive sag and soar on very fast load step
edges, with the chosen inductor value.
The amplitude of the ESR step up or down is a function of
the load step and the ESR of the output capacitor :
VESR _STEP = ΔIOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,the
input-to-output voltage differential, and the maximum duty
cycle. The maximum duty cycle during a fast transient is
a function of the on-time and the minimum off-time since
the ACOTTM control scheme will ramp the current using
on-times spaced apart with minimum off-times, which is
as fast as allowed. Calculate the approximate on-time
(neglecting parasitics) and maximum duty cycle for a given
input and output voltage as :
VOUT
tON
tON =
and DMAX =
VIN fSW
tON tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we can
neglect both of these since the on-time increase
compensates for the voltage losses. Calculate the output
voltage sag as :
VSAG =
L (IOUT )2
2 COUT VIN(MIN) DMAX VOUT
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR =
L (IOUT )2
2 COUT VOUT
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DS7294D-05 July 2015
RT7294D
Feed-forward Capacitor (Cff)
Enable Operation (EN)
The RT7294D is optimized for ceramic output capacitors
and for low duty cycle applications. However for high-output
voltages, with high feedback attenuation, the circuit's
response becomes over-damped and transient response
can be slowed. In high-output voltage circuits (VOUT > 3.3V)
For automatic start-up the EN pin can be connected to
VIN, through a 100kΩ resistor. Its large hysteresis band
makes EN useful for simple delay and timing circuits. EN
can be externally pulled to VIN by adding a resistorcapacitor delay (REN and CEN in Figure 2). Calculate the
delay time using EN's internal threshold where switching
operation begins (1.4V, typical).
transient response is improved by adding a small “feedforward” capacitor (Cff) across the upper FB divider resistor
{Figure 1), to increase the circuit's Q and reduce damping
to speed up the transient response without affecting the
steady-state stability of the circuit. Choose a suitable
capacitor value that following below step.
Get the BW the quickest method to do transient
response form no load to full load. Confirm the damping
frequency. The damping frequency is BW.
An external MOSFET can be added to implement digital
control of EN when no system voltage above 2V is available
(Figure 3). In this case, a 100kΩ pull-up resistor, REN, is
connected between VIN and the EN pin. MOSFET Q1 will
be under logic control to pull down the EN pin. To prevent
enabling circuit when VIN is smaller than the VOUT target
value or some other desired voltage level, a resistive voltage
divider can be placed between the input voltage and ground
and connected to EN to create an additional input under
voltage lockout threshold (Figure 4).
EN
VIN
BW
REN
EN
RT7294D
CEN
GND
Figure 2. External Timing Control
VOUT
R1
Cff
VIN
FB
RT7294D
R2
REN
100k
Q1
Enable
GND
Figure 1. Cff Capacitor Setting
EN
RT7294D
GND
Figure 3. Digital Enable Control Circuit
Cff can be calculated base on below equation :
Cff
1
2 3.1412 R1 BW 0.8
VIN
REN1
REN2
Internal Soft-Start (SS)
The RT7294D soft-start uses an internal soft-start time
800μs.
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EN
RT7294D
GND
Figure 4. Resistor Divider for Lockout Threshold Setting
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RT7294D
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected to
FB. The output voltage is set according to the following
equation :
VOUT = 0.6 x (1 + R1 / R2)
VOUT
resistance between BOOT and the external bootstrap
capacitor. This will slow the high-side switch turn-on and
VSW's rise. To remove the resistor from the capacitor
charging path (avoiding poor enhancement due to
undercharging the BOOT capacitor), use the external diode
shown in Figure 6 to charge the BOOT capacitor and place
the resistance between BOOT and the capacitor/diode
connection.
5V
R1
FB
RT7294D
R2
GND
BOOT
RT7294D
0.1µF
SW
Figure 5. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin. Choose
R2 between 10kΩ and 100kΩ to minimize power
consumption without excessive noise pick-up and
calculate R1 as follows :
R1
R2 (VOUT 0.6)
0.6
For output voltage accuracy, use divider resistors with 1%
or better tolerance.
External BOOT Bootstrap Diode
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode between
VIN (or VINR) and the BOOT pin to improve enhancement
of the internal MOSFET switch and improve efficiency.
The bootstrap diode can be a low cost one such as 1N4148
or BAT54.
External BOOT Capacitor Series Resistance
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low power
loss and good efficiency, but also slow enough to reduce
EMI. Switch turn-on is when most EMI occurs since VSW
rises rapidly. During switch turn-off, SW is discharged
relatively slowly by the inductor current during the
deadtime between high-side and low-side switch on-times.
In some cases it is desirable to reduce EMI further, at the
expense of some additional power dissipation. The switch
turn-on can be slowed by placing a small (<47Ω)
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Figure 6. External Bootstrap Diode
Over-Temperature Protection
The RT7294D features an Over-Temperature Protection
(OTP) circuitry to prevent from overheating due to
excessive power dissipation. The OTP will shut down
switching operation when junction temperature exceeds
160°C. Once the junction temperature cools down by
approximately 20°C, the converter will resume operation.
To maintain continuous operation, the maximum junction
temperature should be lower than 125°C.
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :
PD(MAX) = (TJ(MAX) − TA) / θJA
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications, the
maximum junction temperature is 125°C. The junction to
ambient thermal resistance, θJA, is layout dependent. For
is a registered trademark of Richtek Technology Corporation.
DS7294D-05 July 2015
RT7294D
TSOT-23-6 (FC) package, the thermal resistance, θJA, is
70°C/W on a standard four-layer thermal test board. The
maximum power dissipation at TA = 25°C can be calculated
by the following formula :
Layout Considerations
For best performance of the RT7294D, the following layout
guidelines must be strictly followed.
PD(MAX) = (125°C − 25°C) / (70°C/W) = 1.429W for
TSOT-23-6 (FC) package
The maximum power dissipation depends on the operating
ambient temperature for fixed T J(MAX) and thermal
resistance, θJA. The derating curve in Figure 7 allows the
designer to see the effect of rising ambient temperature
on the maximum power dissipation.
Input capacitor must be placed as close to the IC as
possible.
SW should be connected to inductor by wide and short
trace. Keep sensitive components away from this trace.
Maximum Power Dissipation (W)1
2.0
Four-Layer PCB
1.6
1.2
0.8
0.4
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
Figure 7. Derating Curve of Maximum Power Dissipation
SW should be connected to inductor by Wide and
short trace. Keep sensitive components away from
this trace. Suggestion layout trace wider for thermal .
Keep sensitive components away
from this trace. Suggestion layout
trace wider for thermal.
COUT
SW
Suggestion layout trace
wider for thermal.
COUT
VOUT
CS*
BOOT
VOUT R1
6
SW
GND
2
5
VIN
FB
3
4
EN
RS*
CIN
REN
CIN
VIN
R2
The REN component must
The feedback components must be
connected as close to the device as
possible.
GND
Input capacitor must be placed As close
to the IC as possible. Suggestion layout
trace wider for thermal.
be connected to VIN.
Suggestion layout trace
wider for thermal.
Figure 8. PCB Layout Guide
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS7294D-05 July 2015
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
13
RT7294D
Outline Dimension
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.700
1.000
0.028
0.039
A1
0.000
0.100
0.000
0.004
B
1.397
1.803
0.055
0.071
b
0.300
0.559
0.012
0.022
C
2.591
3.000
0.102
0.118
D
2.692
3.099
0.106
0.122
e
0.950
0.037
H
0.080
0.254
0.003
0.010
L
0.300
0.610
0.012
0.024
TSOT-23-6 (FC) Surface Mount Package
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot
assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be
accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
www.richtek.com
14
DS7294D-05 July 2015