INTERSIL ISL6224CAZ

ISL6224
I GNS
DE S T I S
W
E
PA R
OR N
E D F C EM E N T
D
N
Data Sheet
MME
PL A
ECO DED RE 269
R
T
EN
NO
ISL6
OM M
R EC
®
June 8, 2006
Single Output Mobile-Friendly PWM
Controller
Features
• Adjustable output voltgage: 0.9-5.5V
The ISL6224 provides power control and protection for a single,
adjustable output voltage required to power chip-sets and
memory banks in high-performance notebooks and PDAs. This
output voltage is adjustable in the range from 0.9-5.5V.
The hysteretic or PWM controller regulates the output
voltage from battery voltages ranging from 4V to 24V.
Synchronous rectification and hysteretic operation at light
loads contribute to a high efficiency over a wide range of
input voltages and loads. Efficiency is even further enhanced
by using MOSFET’s rDS(ON) as a current sense component.
Feed-forward ramp modulation, average current mode
control and internal feed-back compensation provide fast
and firm handling of transients when powering advanced
chip sets.
• High efficiency over wide load range
- Higher efficiency in hysteretic mode at light load
• Lossless current sense scheme
- Uses MOSFET’s rDS(ON)
- Optional current sense method higher precision
• Supply operation mode
- Wide VIN range: 4V-24V
- Single 5V system rail
• Input undervoltage lock-out on VCC pin (UVLO)
• Excellent dynamic response
- Combined voltage feed-forward and current mode
control
Two-stage conversion using system 5V voltage is possible at
a higher frequency (600kHz) to minimize the output filter size.
• Power-good indicator
The ISL6224 monitors the output voltage. A PGOOD (power
good) signal is issued when soft-start is completed and the
output is within ±10% of the set point.
• Thermal shut-down
A built-in overvoltage protection prevents output voltage
from going above 120% of the set point. Undervoltage
protection latches the chip off when the output drops below
70% of its setting value after soft-start sequence is
completed. The PWM controller’s overcurrent circuitry
monitors the output current by sensing the voltage drop
across the lower MOSFET. If higher precision sense
technique is required, an optional external current-sense
resistor may be used.
Applications
ISL6224CA
PART
MARKING
TEMP. (°C)
ISL6224CA
ISL6224CAZ 6224CAZ
(Note 1)
• 300/600kHz switching frequency
• Pb-free plus anneal available (RoHS compliant)
• Mobile PCs
• Graphic cards
• Hand-held portable instruments
Related Literature
• Application Note AN9983
Pinout
Ordering Information
PART
NUMBER
FN9042.8
PACKAGE
ISL6224 (SSOP)
TOP VIEW
PKG.
DWG. #
-10 to 85
16 Ld SSOP M16.15A
-10 to 85
16 Ld SSOP M16.15A
(Pb-free)
NOTES:
1. Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and
100% matte tin plate termination finish, which are RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations. Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the
Pb-free requirements of IPC/JEDEC J STD-020.
VIN
1
16
FCCM
PGOOD
2
15
BOOT
EN
3
14
UGATE
OCSET
4
13
PHASE
VOUT
5
12
ISEN
VSEN
6
11
VCC
SOFT
7
10
LGATE
GND
8
9
PGND
2. Add “-T” for Tape and Reel.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2004, 2006. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6224
Absolute Maximum Ratings
Thermal Information
Bias Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
Input Voltage, Vin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +27.0V
Phase and Isen Pins . . . . . . . . . . . . . . . . . . . . . GND -0.3V to +29.0V
BOOT and Ugate Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . + 32.0V
BOOT with respect to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . .+ 7.0V
All other pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V to 15V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance (Typical, Note 1)
θJA (°C/W)
SSOP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . .
112
Maximum Junction Temperature (Plastic Package) . . . . . . . . 150°C
Maximum Storage Temperature Range . . . . . . . . . . . -65°C to 150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 300°C
(SSOP - Lead Tips Only)
Recommended Operating Conditions
Bias Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.0V ±5%
Input Voltage, Vin . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.0V to +24.0V
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . .-10°C to 85°C
Junction Temperature Range. . . . . . . . . . . . . . . . . . .-10°C to 125°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
3. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
Operating Conditions: VCC = 5V, TA = 10°C to 85°C, Unless Otherwise Noted.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
850
1300
µA
-
5
15
µA
Rising Vcc Threshold
4.3
-
4.75
V
Falling Vcc Threshold
4.1
-
4.5
V
Vcc UVLO Hysteris
0.1
-
0.5
V
VCC SUPPLY
Bias Current
ICC
Shutdown Current
LGATE, UGATE Open, VSEN forced above
regulation point
ICCSN
VCC UVLO
VIN
Input Voltage Pin Current (Sink)
IVIN
VIN pin connected to the input voltage source
10
20
30
µA
Input Voltage Pin Current (Source)
IVIN
VIN pin connected to ground
-7
-15
-20
µA
Shutdown Current
IVIN
-
-
1
µA
OSCILLATOR
PWM Oscillator Frequency
Fc1
VIN = 3.5V - 24V
255
300
345
kHz
PWM Oscillator Frequency
Fc2
VIN ≤ 0.5V
510
600
690
kHz
Ramp Amplitude, pk-pk
VR1
VIN = 16V, By Design
-
2
-
V
Ramp Amplitude, pk-pk
VR2
VIN ≤ 5V, By Design
-
1.25
-
V
VROFF
-
0.5
-
V
VREF
-
0.9
-
V
-1.0
-
+1.0
%
Ramp Offset
REFERENCE AND SOFT-START
Internal Reference Voltage
Reference Voltage Accuracy
Soft-Start Current During Start-up
ISOFT
-
5
-
µA
Soft-Start Threshold
VSOFT
-
1.5
-
V
-1.0
-
+1.0
%
-
80
-
nA
PWM CONVERTER
Load Regulation
0.0mA < IVOUT1 < 3.0A; 5.0V < VIN < 24.0V
VSEN pin bias current
IVSEN
2
FN9042.8
June 8, 2006
ISL6224
Electrical Specifications
Operating Conditions: VCC = 5V, TA = 10°C to 85°C, Unless Otherwise Noted. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
40
55
65
kΩ
VOUT pin input impedance
IVOUT
Undervoltage Shutdown Level
VUV1
Fraction of the set point; ~3µs noise filter
70
-
80
%
VOVP1
Fraction of the set point; ~1µs noise filter
113
-
120
%
Overvoltage Protection
PWM CONTROLLER GATE DRIVERS
Upper Drive Pull-Up Resistance
R2UGPUP
-
8
15
Ω
Upper Drive Pull-Down Resistance
R2UGPDN
-
3.2
5
Ω
Lower Drive Pull-Up Resistance
R2LGPUP
-
8
15
Ω
Lower Drive Pull-Down Resistance
R2LGPDN
-
1.5
2.4
Ω
POWER GOOD AND CONTROL FUNCTIONS
Power-Good Lower Threshold
VPG-
Fraction of the set point; ~3µs noise filter
-14
-
-8
%
Power-Good Higher Threshold
VPG+
Fraction of the set point; ~3µs noise filter
10
-
15
%
VPULLUP = 5.5V
-
-
1
µA
IPGOOD = -4mA
-
-
0.5
V
EN - Low (Off)
-
-
0.8
V
EN - High (On)
2.0
-
-
V
-
Vcc/2
-
V
PGOOD Leakage Current
IPGLKG
PGOOD Voltage Low
FCCM -Hysteretic Operation Enabled
3
FN9042.8
June 8, 2006
Functional Block Diagram
FCCM
VIN
VCC
BOOT
GND
HGDR1
FCCM
4
FSET
CLK
RAMP
UGATE
HI
PHASE
SHUTOFF
POWER-ON
RESET (POR)
GATE LOGIC
CLK
GATE
CONTROL
POR
DEADT
VCC
PWM/HYST
LGDR
LGATE
PWM ON
POR
HYST ON
EN
LO
OVP
PGND
SDWN
REFERENCE
AND
-
CLK1
SOFT-START
+
ISL6224
SOFT
HYST COMP1
REF
OC COMP1
OC LOGIC1
+
PWM
LATCH 1 VCC
EA1
VSEN
- +
+
OUTPUT
VOLTAGE
MONITOR
+
Q D
R
Q <
Σ
REF
VOLTSECOND
CLAMP
FFBK
LGATE1
+
-
MODE CHANGE COMP
R1=20K
LGATE
ISEN
-
PGOOD
+
PWM/HYS
LOGIC
OCSET
VOUT
FN9042.8
June 8, 2006
ISL6224
Functional Pin Description
GND (Pin 8)
Signal ground for the IC.
VIN (Pin 1)
Provides battery voltage to the oscillator for feed-forward
rejection of the input voltage variation. Also, this pin
programs frequency of the internal clock and gain of the
ramp generator. When connected to the battery, which
voltage varies from 4V to 24V, the clock frequency is set to
300kHz and the ramp gain is set accordingly to
accommodate the wide input voltage range.
PGND (Pin 9)
For two step conversion from the system 5V power rail, the
Vin pin is connected to ground via a 150kΩ resistor. This
arrangement changes the gain of the ramp generator to
accommodate the lower input voltage but does not change
the clock frequency.
VCC (Pin 11)
When the Vin pin is connected to ground, the clock
frequency is set to 600kHz. The ramp generator gain is also
changed accordingly. This circuit arrangement enables the
designer to choose smaller output filter components.
This is the power ground connection for PWM converter.
This pin is connected to the lower MOSFET’s source
terminal.
LGATE (Pin 10)
This pin provides the gate drive for the lower MOSFET.
This pin provides power to the chip.
ISEN (Pin 12)
This pin is used to monitor the voltage drop across the lower
MOSFET for current feedback and overcurrent protection.
For precise current detection this input can be connected to
an optional current sense resistor placed in series with the
source of the lower MOSFET.
PHASE (Pin 13)
PGOOD (Pin 2)
PGOOD is an open collector output used to indicate the
status of the output voltage. This pin is pulled high when the
system output is within ±10% of its respective nominal
voltage.
Connect this pin to the PHASE node of the converter. The
PHASE node is the junction point of the upper MOSFET
source, output filter inductor, and lower MOSFET drain.
UGATE (Pin 14)
This pin provides the gate drive for the upper MOSFET.
EN (Pin 3)
This pin provides the enable/disable function for the chip. The
IC is enabled when this pin is pulled over 2V or left open.
Note: a pulldown resistance of 100kΩ or less is required to
disable the controller.
OCSET (Pin 4)
A resistor from this pin to GND sets the overcurrent
protection threshold.
VOUT (Pin 5)
This pin is used for feedback of the output voltage to
properly position output voltage during operational mode
change.
VSEN (Pin 6)
This pin is connected to the output via a resistive divider and
provides the voltage feedback signal for the PWM controller.
The PGOOD, UVP, and OVP circuits use this signal to report
output voltage status.
SOFT (Pin 7)
This pin provides soft-start of the PWM controller. When the
EN pin is pulled high, the voltage on the capacitor connected
to the soft-start pin is rising linearly due to the 5µA pull-up
current. The output voltage follows the voltage on the
capacitor until it reaches the value of 0.9V. The further rise of
the voltage on the soft-start capacitor does not affect the
output voltage.
5
BOOT (Pin 15)
This pin powers the upper MOSFET drivers of the PWM
converter. Connect this pin to the junction of bootstrap
capacitor with the cathode of the bootstrap diode. Anode of
the bootstrap diode is connected to the VCC pin.
FCCM (Pin 16)
This pin, when pulled to VCC, restrains hysteretic operation
in light loads.
General Description
Operational Overview
The ISL6224 is a single-channel PWM controller intended
for chipset, DRAM, or other low voltage power needs of
modern notebook and sub-notebook PCs. The IC integrates
control circuits and feedback compensation for a single
synchronous buck converter. The output voltage is set in the
range of 0.9–5.5V by an external resistive divider.
The synchronous buck converter can be configured for
either 300kHz or 600kHz switching frequencies. When
operated from battery, a switching frequency of 300kHz is
recommended. When operating from 5V, switching
frequencies of 300kHz or 600kHz are an option. For 300kHz
operation, pin 1 should be connected through a resistor
(150K) to gnd. For 600kHz operation, pin 1 should simply be
FN9042.8
June 8, 2006
ISL6224
grounded. Table 1. shows the configuration for different
modes of operation. Figure 1 below shows plots of the ramp
speed compensation.
of this resistor can be obtained from the following
expression:
11 • Risen
Rocset = --------------------------------Ioc • Rdson
300kHz CLOCK
600kHz CLOCK
Vin t
---------- --2 T
Vin- --t--------4 T
Vin t
---------- --8 T
Vo/4
Vo/8
FIGURE 1. RAMP SPEED COMPENSATION Vo = 2.5V
TABLE 1. CONFIGURATION FOR MODES OF OPERATION
OPERATION
PIN 1 CONNECTION
PIN 1 POTENTIAL
One-stage 300kHz
Vin
V1 > 4V
Two-stage 300kHz
150K-GND
1V < V1 < 2V
Two-stage 600kHz
GND
V1 < 0.5V
The synchronous converter light-load efficiency is enhanced
by a hysteretic mode of operation which is automatically
engaged at light loads when the inductor current becomes
discontinuous. As the filter inductor resumes continuous
current, the PWM mode of operation is automatically
restored.
The ISL6224 control IC employs an average current mode
control scheme with input voltage feedforward ramp
programming for better rejection of input voltage variations.
Current Sensing and Current Limit Protection
The PWM converter uses the lower MOSFET on-state
resistance, rDS(ON), as the current-sensing element. This
technique eliminates the need for a current sense resistor
and the associated power losses. If more accurate current
protection is desired, current sense resistors may be used in
series with the lower MOSFET’s source.
A current proportional signal is used to provide average
current mode control and overcurrent protection. The gain in
the current sense circuit is set by the resistor connected from
ISEN (pin 12) to the PHASE node of the buck converter. The
value of this resistor can be estimated by the following
expression:
where Ioc is the value of overcurrent. The resulting current
out of the ISEN pin through RISEN, is used for current
feedback and current limit protection. This is compared with
an internal current limit threshold. When a sampled value of
the output current is determined to be above the current limit
threshold, the PWM drive is terminated and a counter is
initiated. This limits the inductor current build-up and
essentially switches the converter into current-limit mode. If
an overcurrent is detected between 26ms to 53ms later, an
overcurrent shutdown is initiated. If during the 26ms to 53ms
period, an overcurrent is not detected, the counter is reset
and sampling continues as normal.
This current limit scheme has proven to be very robust in
applications like portable computers where fast inductor
current build-up is common due to a large difference
between input and output voltages and a low value of the
inductor.
Light-Load (Hysteretic) Operation
In the light-load (hysteretic) mode the output voltage is
regulated by the hysteretic comparator which regulates the
output voltage by maintaining the output voltage ripple as
shown in Figure 2. In hysteretic mode, the inductor current
flows only when the output voltage reaches the lower limit of
the hysteretic comparator and turns off at the upper limit.
Hysteretic mode saves converter energy at light loads by
supplying energy only at the time when the output voltage
requires it. This mode conserves energy by reducing the
power dissipation associated with continuous switching.
During the time between inductor current pulses, both the
upper and lower MOSFETs are turned off. This is referred to
as ‘diode emulation mode’ because the lower MOSFET
performs the function of a diode. This diode emulation mode
prevents the output capacitor from discharging through the
lower MOSFET when the upper MOSFET is not conducting.
NOTE: the PWM only operation can intentionally be forced by tying
pin 16, FCCM, to VCC.
Iomax • Rdson
Risen = ------------------------------------------- – 100
75µA
where Iomax is the maximum inductor current. The value of
RISEN should be specified for the expected maximum
operating temperature.
An overcurrent protection threshold is set by an external
resistor connected from OCSET (pin 4) to ground. The value
6
FN9042.8
June 8, 2006
ISL6224
VOUT
t
IL
t
PHASE
COMP
t
1 2 3 4 5 6 7 8
MODE
OF
OPERATION
due to the voltage drop on the output capacitor ESR. If the
decrease causes the output voltage to drop below the
hysteretic regulation level, the mode is changed to PWM on
the next clock cycle. This insures the full power required by
the increase in output current.
IL
t
HYSTERETIC
PWM
t
FIGURE 2. HYSTERETIC OPERATION MODE
Operation-Mode Control
The mode-control circuit changes the converter’s mode of
operation based on the voltage polarity of the phase node
when the lower MOSFET is conducting and just before the
upper MOSFET turns on. For continuous inductor current,
the phase node is negative when the lower MOSFET is
conducting and the converters operate in fixed-frequency
PWM mode as shown in Figure 3. When the load current
decreases to the point where the inductor current flows
through the lower MOSFET in the ‘reverse’ direction, the
phase node becomes positive, and the mode is changed to
hysteretic.
A phase comparator handles the timing of the phase node
voltage sensing. A low level on the phase comparator output
indicates a negative phase voltage during the conduction
time of the lower MOSFET. A high level on the phase
comparator output indicates a positive phase voltage.
When the phase node is positive (phase comparator high),
at the end of the lower MOSFET conduction time, for eight
consecutive clock cycles, the mode is changed to hysteretic
as shown in Figure 3. The dashed lines indicate when the
phase node goes positive and the phase comparator output
goes high. The solid vertical lines at 1,2,...8 indicate the
sampling time, of the phase comparator, to determine the
polarity (sign) of the phase node. At the transition between
PWM and hysteretic mode both the upper and lower
MOSFETs are turned off. The phase node will ‘ring’ based
on the output inductor and the parasitic capacitance on the
phase node and settle out at the value of the output voltage.
PHASE
NODE
t
1
2
3
4
5
6
7
8
PHASE
COMP
t
MODE
OF
OPERATION
PWM
HYSTERETIC
t
FIGURE 3. MODE CONTROL WAVEFORMS
Gate Control Logic
The gate control logic translates generated PWM control
signals into the MOSFET gate drive signals providing
necessary amplification, level shifting and shoot-through
protection. Also, it has functions that help optimize the IC
performance over a wide range of operational conditions.
Since MOSFET switching time can vary dramatically from
type to type and with the input voltage, the gate control logic
provides adaptive dead time by monitoring the gate-tosource voltages of both upper and lower MOSFETs. The
lower MOSFET is not turned on until the gate-to-source
voltage of the upper MOSFET has decreased to less than
approximately 1V. Similarly, the upper MOSFET is not turned
on until the gate-to-source voltage of the lower MOSFET has
decreased to less than approximately 1V. This allows a wide
variety of upper and lower MOSFETs to be used without a
concern for simultaneous conduction, or shoot-through.
The mode change from hysteretic to PWM can be caused by
one of two events. One event is the same mechanism that
causes a PWM to hysteretic transition. But instead of looking
for eight consecutive positive occurrences on the phase
node, it is looking for eight consecutive negative
occurrences on the phase node. The operation mode will be
changed from hysteretic to PWM when these eight
consecutive pulses occur. This transition technique prevents
jitter of the operation mode at load levels close to boundary.
The other mechanism for changing from hysteretic to PWM
is due to a sudden increase in the output current. This step
load causes an instantaneous decrease in the output voltage
7
FN9042.8
June 8, 2006
ISL6224
Soft-Start Operation
Soft-start of the Synchronous Buck Converter is
accomplished by means of a capacitor connected from pin 7,
SOFT to ground. The soft-start time can be obtained from
the following equation:
1.5V × Css
Tss = -----------------------------5.0µA
This ‘soft-crowbar’ and monitoring of the output, prevents the
output voltage from ringing negative as the inductor current
flows in the ‘reverse’ direction through the lower MOSFET
and output capacitors.
Component Selection Guidelines
Output Capacitor Selection
Figure 4 shows the soft-start initiated by the ENABLE pin
being pulled high with the VIN input at 5.6V and the resulting
3.3V output and PGOOD signal. While the ENABLE pin is
held low, prior to t0, the output is off. When the EN pin is
pulled high, at t0, the voltage on the capacitor connected to
the soft-start pin rises linearly due to the internal 5µA current
source starts charging the capacitor. The output voltage
follows the voltage on the capacitor till it reaches the value of
0.9V at t1. At this moment, t1, the output voltage started
regulation. The soft-start is complete when PGOOD pin is
high at t2 and further rise of the voltage on the soft-start
capacitor does not affect the output voltage.
The output capacitors have unique requirements. In general,
the output capacitors should be selected to meet the
dynamic regulation requirements including ripple voltage
and load transients.
Selection of the output capacitors is also dependent on the
output inductor so some inductor analysis is required to
select the output capacitors.
One of the parameters limiting the converter’s response to a
load transient is the time required for the inductor current to
slew to its new level. Given a sufficiently fast control loop
design, the ISL6224 will provide either 0% or 94% duty cycle
in response to a load transient. The response time is the
time interval required to slew the inductor current from an
initial current value to the load current level. During this
interval the difference between the inductor current and the
transient current level must be supplied by the output
capacitor(s). Minimizing the response time can minimize the
output capacitance required. If the load transient rise time is
slower than the inductor response time, as in a hard drive or
CD drive, this reduces the requirement on the output
capacitor.
The maximum capacitor value required to provide the full,
rising step, transient load current during the response time of
the inductor is:
I TRAN
L O × I TRAN
C OUT = ---------------------------------------------- × -------------------( V IN – V OUT ) × 2 DV OUT
t0
t1
t2
FIGURE 4. MODE CONTROL WAVEFORMS
Power Good Status
The ISL6224 monitors the output voltage. A single powergood signal, PGOOD, is issued when soft-start is completed
and the output is within 10% of it’s set point. After the softstart sequence is completed, undervoltage protection
latches the chip off when any of the monitored outputs drop
below 70% of its set point.
A ‘soft-crowbar’ function is implemented for an overvoltage
on the output. If the output voltage goes above 120% of its
nominal output level, the upper MOSFET is turned off and
the lower MOSFET is turned on. This ‘soft-crowbar’
condition will be maintained until the output voltage returns
to the regulation window and then normal operation will
continue.
8
Where: COUT is the output capacitor(s) required, LO is the
output inductor, ITRAN is the transient load current step, VIN
is the input voltage, VOUT is output voltage, and DVOUT is
the drop in output voltage allowed during the load transient.
High frequency capacitors initially supply the transient
current and slow the load rate-of-change seen by the bulk
capacitors. The bulk filter capacitor values are generally
determined by the ESR (equivalent series resistance) and
voltage rating requirements as well as actual capacitance
requirements. The output voltage ripple is due to the inductor
ripple current and the ESR of the output capacitors as
defined by:
V RIPPLE = ∆I L × ESR
where, ∆I L is calculated in the Inductor Selection section.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
FN9042.8
June 8, 2006
ISL6224
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load
circuitry for specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications, at 300kHz, for the bulk
capacitors. In most cases, multiple electrolytic capacitors of
small case size perform better than a single large case
capacitor.
The stability requirement on the selection of the output
capacitor is that the ‘ESR zero’, fZ, be between 1.2kHz and
30kHz. This range is set by an internal, single compensation
zero at 6kHz. The ESR zero can be a factor of five on either
side of the internal zero and still contribute to increased
phase margin of the control loop. Therefore:
1
C OUT = ------------------------------------------2 × π × ESR × f Z
Use a mix of input bypass capacitors to control the voltage
ripple across the MOSFETs. Use ceramic capacitors for the
high frequency decoupling and bulk capacitors to supply the
RMS current. Small ceramic capacitors can be placed very
close to the upper MOSFET to suppress the voltage induced
in the parasitic circuit impedances.
For board designs that allow through-hole components, the
Sanyo OS-CON® series offer low ESR and good
temperature performance.
For surface mount designs, solid tantalum capacitors can be
used, but caution must be exercised with regard to the
capacitor surge current rating. These capacitors must be
capable of handling the surge-current at power-up. The TPS
series available from AVX is surge current tested.
MOSFET Considerations
In conclusion, the output capacitors must meet three criteria:
1. They must have sufficient bulk capacitance to sustain the
output voltage during a load transient while the output
inductor current is slewing to the value of the load
transient
2. The ESR must be sufficiently low to meet the desired
output voltage ripple due to the output inductor current,
and
3. The ESR zero should be placed, in a rather large range,
to provide additional phase margin.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current and output capacitor(s) ESR. The ripple
voltage expression is given in the capacitor selection section
and the ripple current is approximated by the following
equation:
V IN – V OUT V OUT
∆I L = -------------------------------- × ---------------FS × L
V IN
where Fs is the switching frequency.
Input Capacitor Selection
The important parameters for the bulk input capacitor(s) are
the voltage rating and the RMS current rating. For reliable
operation, select bulk input capacitors with voltage and
current ratings above the maximum input voltage and largest
RMS current required by the circuit. The capacitor voltage
rating should be at least 1.25 times greater than the
maximum input voltage and 1.5 times is a conservative
guideline.
The logic level MOSFETs are chosen for optimum efficiency
given the potentially wide input voltage range and output
power requirements. One dual N-Channel or two N-Channel
MOSFETs are used in each of the synchronous rectified
buck converters for the outputs. These MOSFETs should be
selected based upon rDS(ON) , gate supply requirements,
and thermal management considerations.
The power dissipation includes two loss components;
conduction loss and switching loss. These losses are
distributed between the upper and lower MOSFETs
according to duty cycle (see the following equations). The
conduction losses are the main component of power
dissipation for the lower MOSFETs. Only the upper MOSFET
has significant switching losses, since the lower device turns
on and off into near-zero voltage.
2
I O × r DS ( ON ) × V OUT I O × V IN × t SW × F S
- + ---------------------------------------------------P UPPER = ----------------------------------------------------------V IN
2
2
I O × r DS ( ON ) × ( V IN – V OUT )
P LOWER = -------------------------------------------------------------------------------V IN
The equations assume linear voltage-current transitions and
do not model power loss due to the reverse-recovery of the
lower MOSFET’s body diode.
The gate-charge losses are dissipated by the ISL6224 and
do not heat the MOSFETs. However, a large gate-charge
increases the switching time, tSW which increases the upper
MOSFET switching losses. Ensure that both MOSFETs are
within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications.
The AC RMS input current varies with load. Depending on
the specifics of the input power and it’s impedance, most (or
all) of this current is supplied by the input capacitor(s).
9
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June 8, 2006
ISL6224
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting impedances
and parasitic circuit elements. The voltage spikes can
degrade efficiency, radiate noise into the circuit, and lead to
device overvoltage stress. Careful component layout and
printed circuit design minimizes the voltage spikes in the
converter. Consider, as an example, the turn-off transition of
one of the upper PWM MOSFETs. Prior to turn-off, the upper
MOSFET is carrying the full load current. During the turn-off,
current stops flowing in the upper MOSFET and is picked up
by the lower MOSFET. Any inductance in the switched
current path generates a voltage spike during the switching
interval. Careful component selection, tight layout of the
critical components, and short, wide circuit traces minimize
the magnitude of voltage spikes. See the Application Note
AN9983 for the evaluation board component placement and
the printed circuit board layout details.
There are two sets of critical components in a DC/DC
converter using an ISL6224 controller. The switching power
components are the most critical because they switch large
amounts of energy, and as such, they tend to generate
equally large amounts of noise. The critical small signal
components are those connected to sensitive nodes or
those supplying critical bias currents.
Power Components Layout Considerations
The power components and the controller IC should be
placed first. Locate the input capacitors, especially the highfrequency ceramic decoupling capacitors, close to the power
MOSFETs. Locate the output inductor and output capacitors
between the MOSFETs and the load. Locate the PWM
controller close to the MOSFETs.
A multi-layer printed circuit board is recommended. Dedicate
one solid layer for a ground plane and make all critical
component ground connections with vias to this layer.
Dedicate another solid layer as a power plane and break this
plane into smaller islands of common voltage levels. The
power plane should support the input power and output
power nodes. Use copper filled polygons on the top and
bottom circuit layers for the phase nodes, but do not
unnecessarily oversize these particular islands. Since the
phase nodes are subjected to very high dV/dt voltages, the
stray capacitor formed between these islands and the
surrounding circuitry will tend to couple switching noise. Use
the remaining printed circuit layers for small signal wiring.
The wiring traces from the control IC to the MOSFET gate
and source should be sized to carry 2A peak currents.
Small Components Signal Layout Considerations
The Vin pin 1 input should be bypassed with a 1.0µF
capacitor. The bypass capacitors for Vin and the soft-start
capacitor, should be located close to their connecting pins on
the control IC.
Refer to the Application Note AN9983 for a recommended
component placement and interconnections.
Figures 5, 6 and 7 show application circuits for the three modes
of operation. Mode 1 is operating from battery voltage and
operating at 300kHz switching frequency. Mode 2 is operating
off of 5V and operating at 300kHz switching frequency. Mode 3
is operating off of 5V and operating at 600kHz switching
frequency.
Insure the current paths from the input capacitors to the
MOSFETs, to the output inductors and output capacitors are
as short as possible with maximum allowable trace widths.
10
FN9042.8
June 8, 2006
ISL6224
ISL6224 DC-DC Converter Application Circuits
and circuit board description, see Application Note AN9983.
Also see Intersil’s web site (http://www.intersil.com) for the
latest information.
Figure 5 shows an application circuit of a DC/DC converter
for a notebook PC. The power supply provides +V2_5S from
either +4V–24VDC battery voltage or system +5V bus. For
detailed information on the circuit, including a bill of materials
+5.6-24VIN
C2
1µF
+
C1
56µF
GND
+5.0VCC
+
VIN
R2
OCSET
1
4
VCC
15
14
16
ISL6224
EN
13
BOOT
Q1
1/2 FDS6912A
C6
0.1µF
UGATE
PHASE
ISEN
3
C3
33µF
11
TBD
FCCM
CR1
Risen
+V2_5S
(3A)
L1 6.4µH
12
TBD
SOFT
7
C4
0.015µF
10
9
LGATE
C5
330µF
Q2
2/2 FDS6912A
+
PGND
R3
PGOOD
6
2
8
5
VSEN
VOUT
C6
R4
GND
FIGURE 5. APPLICATION CIRCUIT FOR ONE-STEP CONVERSION (MODE 1)
11
FN9042.8
June 8, 2006
ISL6224
+5.0VCC
+
R2
CR1
OCSET
4
BOOT
15
PHASE
ISEN
3
C4
0.1µF
UGATE
13
ISL6224
EN
Q1
1/2 FDS6912A
14
16
GND
VCC
11
TBD
FCCM
C1
33µF
Risen
L1
6.4µH
+V2_5S
(2A)
12
TBD
SOFT
7
C2
0.015µF
R3
C6
VSEN
6
2
+
PGND
9
PGOOD
C5
330µF
Q2
2/2 FDS6912A
LGATE
10
VOUT
8
1
5
R4
VIN
GND
R5
FIGURE 6. APPLICATION CIRCUIT FOR TWO-STEP 300kHz CONVERSION (MODE 2)
+5.0VCC
+
R2
CR1
OCSET
4
11
GND
C1
33µF
VCC
TBD
15
FCCM
14
16
13
ISL6224
EN
BOOT
Q1
1/2 FDS6912A
PHASE
ISEN
3
C4
0.1µF
UGATE
Risen
L1
6.4µH
+V2_5S
(2A)
12
TBD
SOFT
7
C2
0.015µF
10
9
PGOOD
6
2
LGATE
PGND
C5
330µF
Q2
2/2 FDS6912A
+
R3
C6
VSEN
VOUT
8
GND
5
1
R4
VIN
FIGURE 7. APPLICATION CIRCUIT FOR TWO-STEP 600kHz CONVERSION (MODE 3)
12
FN9042.8
June 8, 2006
ISL6224
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M16.15A
N
INDEX
AREA
H
0.25(0.010) M
16 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
E
-B1
2
INCHES
GAUGE
PLANE
3
0.25
0.010
SEATING PLANE
-A-
A
D
h x 45°
-C-
e
α
A2
A1
B
0.17(0.007) M
L
C
0.10(0.004)
C A M
B S
NOTES:
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.061
0.068
1.55
1.73
-
A1
0.004
0.0098
0.102
0.249
-
A2
0.055
0.061
1.40
1.55
-
B
0.008
0.012
0.20
0.31
9
C
0.0075
0.0098
0.191
0.249
-
D
0.189
0.196
4.80
4.98
3
E
0.150
0.157
3.81
3.99
4
e
0.025 BSC
0.635 BSC
-
H
0.230
0.244
5.84
6.20
-
h
0.010
0.016
0.25
0.41
5
L
0.016
0.035
0.41
0.89
6
8°
0°
N
1. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication Number 95.
MILLIMETERS
α
16
0°
16
7
8°
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
Rev. 2 6/04
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions.
Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable
dambar protrusion shall be 0.10mm (0.004 inch) total in excess
of “B” dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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13
FN9042.8
June 8, 2006