V15N3 - SEPTEMBER

LINEAR TECHNOLOGY
SEPTEMBER 2005
IN THIS ISSUE…
COVER ARTICLE
Breakthrough Buck-Boost Controller
Provides up to 10A from a Wide
4V–36V Input Range ...........................1
Theo Phillips and Wilson Zhou
Issue Highlights ..................................2
Linear Technology in the News….........2
DESIGN FEATURES
Dual, 1.4A and 800mA, Buck Regulator
for Space-Sensitive Applications .........7
Scott Fritz
µPower Precision Dual Op Amp
Combines the Advantages of
Bipolar and CMOS Ampliﬁers ..............9
Cheng-Wei Pei and Hengsheng Liu
High Voltage Micropower Regulators
Thrive in Harsh Environments ..........11
Todd Owen
Complete 2-Cell-AA/USB Power
Manager in a 4mm × 4mm QFN .........13
G. Thandi
Micropower Precision Oscillator
Draws Only 60µA at 1MHz ...............17
Albert Huntington
New Standalone Linear
Li-Ion Battery Chargers .....................20
Alfonso Centuori
Monolithic Buck Regulator Operates
Down to 1.6V Input; Simpliﬁes Design
of 2-Cell NiCd/NiMh Supplies .............22
Gregg Castellucci
Supply Tracking and Sequencing at
Point-of-Load: Easy Design without
the Drawbacks of MOSFETs ..............24
Scott Jackson
Versatile Controller Simpliﬁes High
Voltage DC/DC Converter Designs ......28
Tom Sheehan
VOLUME XV NUMBER 3
Breakthrough Buck-Boost
Controller Provides up to
10A from a Wide 4V–36V
Input Range
by Theo Phillips
and Wilson Zhou
Introduction
Many DC/DC converter applications
require an output voltage somewhere
within a wide range of input voltages. An everyday example would
be a well-regulated 12V output from
an automotive battery input, which
has a full charge voltage around 14V
and a ﬂuctuating cold crank voltage
under 9V.
There are a number of traditional
solutions to this problem, but all have
drawbacks, including low efﬁciency,
limited input voltage range or the use
bulky coupled inductors. Some even
produce output voltages of polarity
opposite to that of the input voltage.
A system designer must often decide
between an inefﬁcient topology or a
scheme that uses both a boost regula4-SWITCH BUCK-BOOST
TOPOLOGY YIELDS HIGH
EFFICIENCY AT HIGH POWER
VIN
A
CIN
Multichannel, 3V and 5V, 16-Bit ADCs
Combine High Performance, Speed,
Low Power and Small Size ................31
tor and a buck regulator, which adds
complexity with extra ﬁlter components and multiple control loops.
The LTC3780 offers a simpler solution with an approach that requires
neither cumbersome magnetics nor
additional control loops (see Figure 1).
This 4-switch controller takes the
form of a true synchronous buck or
boost, depending on the input voltage.
Transitions between modes depend on
duty cycle (Figure 2) and are quick and
automatic. The controller is versatile,
providing three modes of operation,
switching frequencies from 200kHz
to 400kHz, and output currents from
milliamps to tens of amps. The three
operating modes permit the designer
to choose between efﬁciency and low
continued on page 3
ONLY ONE INDUCTOR SIMPLIFIES
LAYOUT AND SAVES SPACE
SW2
L
SW1
B
D
SNS+
RSENSE
Ringo Lee
SNS–
DESIGN IDEAS
....................................................35–44
(complete list on page 35)
New Device Cameos ...........................45
VOUT
COUT
C
R1
SINGLE SENSE RESISTOR
KEEPS EFFICIENCY HIGH
LTC3780
SNS+
SNS–
R2
Design Tools ......................................47
Sales Ofﬁces .....................................48
Figure 1. Simpliﬁed diagram of the LTC3780 topology, showing how the
four power switches are connected to the inductor, VIN, VOUT and GND.
, LTC, LT, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD,
Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath,
PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, UltraFast and VLDO are trademarks of Linear
Technology Corporation. Other product names may be trademarks of the companies that manufacture the products.
EDITOR’S PAGE
Issue Highlights
M
any DC/DC converter applications require an output voltage
somewhere within a wide range
of input voltages. Traditional approaches to this problem can require
cumbersome magnetics or additional
control loops. The LTC®3780 offers
a simpler, much more compact and
high performance solution by taking
the form of a true synchronous buck
or boost, depending on the input
voltage. Transitions between modes
depend on duty cycle and are quick
and automatic.
See our cover article for more about
this breakthrough device.
The LTC3409 is a monolithic
synchronous step-down regulator
designed speciﬁcally to save space,
improve battery life and simplify
the design of 2-cell-alkaline, NiCd
and NiMh powered applications.
(Page 22)
The LTC2927 provides simple
tracking and sequencing solutions in
a tiny footprint, without the drawbacks
of series MOSFETs. (Page 24)
The LT3724 is a single-switch
DC/DC controller that can be used
in medium power step-down, step-up,
inverting and SEPIC converter topologies. (Page 28)
Featured Devices
High Performance ADCs
Many of today’s data acquisition applications require low-power and/or
remote sensing capabilities while occupying a minimum amount of board
space. Linear Technology introduces
a pin-compatible family of 12-bit and
16-bit A/D converters that meet these
requirements. (Page 31)
Below is a summary of the other devices featured in this issue.
Power Solutions
The LTC3417 is a dual synchronous,
step-down, current mode, DC/DC
converter designed for medium power
applications. It operates from an input
voltage between 2.25V and 5.5V and
switches at up to 4MHz, making it
possible to use low proﬁle capacitors
and inductors. (Page 7)
The LT3012 and LT3013 are high
voltage micropower regulators
designed for industrial and automotive environments. They offer an
input voltage range of 4V to 80V and
minimize power loss by running with
quiescent currents of 55µA (LT3012)
and 65µA (LT3013). Quiescent current
is reduced to just 1µA in shutdown.
(Page 11)
The LTC3456 is a complete
system power IC that seamlessly
manages power ﬂow between an AC
wall adapter, USB and 2-AA battery,
while complying with USB power
standards—all in a 4mm × 4mm QFN
package. (Page 13)
The LTC4061 and LTC4062 are
speciﬁcally designed to charge singlecell lithium-ion batteries from either
awall adapter or available USB power.
Both devices use constant current/
constant voltage algorithms to deliver
up to 1A of charge current. (Page 20)
2
Micropower Op Amps
The LTC6078 and LTC6079 are
dual and quad micropower, precision op amps that combine the low
offset and drift of traditional bipolar
ampliﬁers with the low bias current
of CMOS ampliﬁers. They include a
combination of features that allow
precision performance previously
available only through composite
ampliﬁers, manual offset trimming,
or calibration.(Page 9)
Precision Silicon Oscillator
The LTC6906 is a monolithic silicon
oscillator with signiﬁcant size, power,
cost and environmental sensitivity
advantages over other oscillators. It
requires only a single external resistor
to set the frequency over its full range
of 10kHz to 1MHz. (Page 17)
Design Ideas and Cameos
The Design Ideas start on page 35, including a discusion of Ni-based battery
chargers and a way to determine the
real resistance of a battery.
Linear Technology in
the News…
Linear Tops $1 Billion
On July 26, Linear Technology
Corporation announced ﬁnancial
results for its ﬁscal year 2005,
ending July 3, 2005. According to
Lothar Maier, CEO, “Fiscal 2005
was a good year for us in what
was generally described as a slow
growth environment, and our rate
of sales growth was greater than all
of our major competitors, allowing
us to achieve over $1 billion in
revenues for the ﬁrst time in the
Company’s history.”
Products in the News
Leading Edge… The May 12 edition of EDN featured the LT5527
400MHz to 3.7GHz High Signal
Level Downconverting Mixer in
the “Leading Edge” section of the
magazine. The “Leading Edge”
column focuses on what’s hot in
the design community.
Mixer Times Two… The June
issue of Wireless Design & Development featured the LT5527
in their “What’s Hot” section.
Electronic Products also featured
the LT5527 in the June Highlights
section.
Design Update… Electronics
Weekly (UK) featured Linear
Technology’s LTC2950 push button on/off controller chip, which
includes debounce, power supply
enable and processor interface
on the cover of the May 4 issue.
The part enables the power supply converter and releases the
processor once the supply is fully
powered up. When powering off,
the chip interrupts the system
processor to alert it to perform
housekeeping tasks. Once these
tasks are over, the processor can
command the LTC2950 to disable
power immediately.
Linear Technology Magazine • September 2005
DESIGN FEATURES
ripple at light loads. The frequency
can be selected by applying the proper
voltage to the PLLFLTR pin, or the
controller can be synchronized to an
external clock via an internal phaselock loop. The current sensing resistor
programs the current limit, freeing the
designer to choose among a broad array of power MOSFETs. Efﬁciency in a
typical application reaches 97%, and
exceeds 90% over more than a decade
of load current (Figure 3). The output
remains stable despite transients in
load current (Figure 4) and line voltage (Figure 5).
A 12V, 5A Converter
Operating from Wide
Input Voltage Range
Figure 6 shows a versatile LTC3780based converter providing 12V at up
to 5A with inputs from 5V to 32V; the
core circuit ﬁts in a cubic inch with
a footprint of only 2.5in2 as shown in
Figure 7. This converter can operate
with any of three light-load operating
modes, set at the three-state FCB pin:
continuous current mode, discontinu-
98%
DMAX
BOOST
DMIN
BOOST
DMAX
BUCK
3%
70
60
50
40
0.01
BOOST
VIN = 6V
VOUT = 12V
0.1
1
10
BUCK/BOOST REGION
D ON, C OFF
PWM A, B SWITCHES
BUCK REGION
ous current mode and Burst Mode®
operation (which becomes skip cycle
mode at higher input voltages). These
modes allow a designer to optimize
efficiency and noise suppression.
Continuous operation provides very
low output voltage ripple, since at
least one of the switch nodes is always
cycling at a constant, programmed
frequency. With at least one switch
always on, the lowest possible noise
is achieved since the output L-C ﬁlter
is not permitted to ring.
EFFICIENCY (%)
EFFICIENCY (%)
DISCONTINOUS
CURRENT MODE
CONTINOUS
CURRENT MODE
FOUR SWITCH PWM
Figure 2. The duty cycle determines the
operating mode, whether in continuous mode
(pictured) or in any of the power saving modes.
The power switches are properly controlled
so the transfer between modes is continuous.
When VIN approaches VOUT, the buck-boost
region is reached; the mode-transition time is
typically 300ns.
90
80
BOOST REGION
DMIN
BUCK
100
BURST MODE
OPERATION
A ON, B OFF
PWM C, D SWITCHES
In continuous operation, the power
switches’ operating sequence depends
on whether the input voltage is greater
than, nearly the same as, or less than
the desired output voltage. When the
input is well above the output (buck
mode), Switch D remains on and
switch C shuts off. When each cycle
begins, synchronous switch B turns
on ﬁrst and the inductor current is
determined by comparing the voltage
across RSENSE to an internal reference.
When the sense voltage drops below
the reference, synchronous switch B
turns off and switch A is turns on for
the remainder of the cycle. Switches
A and B turn on and off alternately,
behaving like a typical synchronous
buck regulator. The duty cycle of
switch A increases until the maximum
duty cycle of the converter in buck
mode reaches 94%–96%.
Figure 8a shows conceptual waveforms in this buck region. When
the input voltage comes close to the
output voltage, maximum duty cycle
is reached and the LTC3780 shifts to
buck-boost mode. Figures 8b and 8c
show the symmetrical, input voltage-
100
100
90
90
BURST MODE
OPERATION
80
EFFICIENCY (%)
LTC3780, continued from page 1
DISCONTINOUS
CURRENT MODE
70
CONTINOUS
CURRENT MODE
60
BUCK-BOOST
VIN = 12V
VOUT = 12V
50
40
0.01
0.1
1
10
80
SKIP CYCLE
MODE
DISCONTINOUS
CURRENT MODE
70
60
CONTINOUS
CURRENT MODE
50
40
0.01
ILOAD (A)
ILOAD (A)
0.1
BUCK
VIN = 18V
VOUT = 12V
1
10
ILOAD (A)
Figure 3. Efﬁciency is high throughout the range of load currents and operating modes.
VOUT
500mV/DIV
VOUT
500mV/DIV
VOUT
500mV/DIV
IL
5A/DIV
IL
5A/DIV
IL
5A/DIV
VIN = 12V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
VIN = 12V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
DISCONTINUOUS CURRENT MODE
VIN = 12V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
BURST MODE OPERATION
Figure 4. The LTC3780 provides excellent load transient response in any of its operating modes.
Linear Technology Magazine • September 2005
3
DESIGN FEATURES
VIN
10V/DIV
VIN
10V/DIV
VOUT
500mV/DIV
VOUT
500mV/DIV
IL
1A/DIV
IL
1A/DIV
VOUT = 12V
500µs/DIV
ILOAD = 1A
VIN STEP: 7V TO 20V
CONTINUOUS MODE
VOUT = 12V
500µs/DIV
ILOAD = 1A
VIN STEP: 20V TO 7V
CONTINUOUS MODE
Figure 5. The LTC3780 responds quickly to changing input voltages.
dependent behavior of the switches
in this region. If the cycle starts with
switches B and D turned on, switches
A and C turn on. Then, switch C turns
off, switch A remains on, and switch
D turns on for the remainder of the
cycle; but if the controller starts with
switches A and C turned on, switches
B and D turn on. Then, switch B
turns off, switch D remains on, and
switch A turns on for the remainder
of the cycle.
Figure 8d shows typical behavior
when the input is well below the output (boost mode). Here, switch A is
always on and synchronous switch B
is always off. When each cycle begins,
switch C turns on ﬁrst and the inductor current is monitored via RSENSE.
When the voltage across RSENSE rises
Figure 7. Typical LTC3780 layout. The four
MOSFETs are on the reverse side, with space
available on top for two dual MOSFETs.
RPU
2
CC2
47pF
CC1
0.1µF
R1
8.06k
1000pF
3
4
RC
100k
5
6
R2 113k
7
8
ON/OFF
INTVCC
9
10
11
12
PGOOD BOOST1
SS
LTC3780
SENSE+
SENSE–
ITH
TG1
SW1
VIN
EXTVCC
VOSENSE INTVCC
SGND
BG1
RUN
PGND
FCB
BG2
PLLFLTR
SW2
PLLIN
TG2
STBYMD BOOST2
CSTBYMD
0.01µF
INTVCC
VPULLUP
24
CA
0.22µF
23
22
D
Si7884DP
DA
1N5819HW
CF 0.1µF
21
COUT
3x
22µF
25V
X5R
D2
B320A
330µF
16V
C
Si7884DP
20
L
4.7µH
Toko
FDA1254
CVCC 4.7µF
19
D1
B340LA
18
RSENSE*
17
16
B
Si7884DP
15
DB
1N5819HW
14
22µF
35V
A
Si7884DP
13
10Ω
CB 0.22µF
CIN
3x
3.3µF
50V
X5R
VIN
5V TO 32V
1.24k
CCM
DCM
VOUT
12V
5A
+
1
+
CSS
0.022µF
above the reference voltage, switch
C turns off and synchronous switch
D turns on for the remainder of the
cycle. Switches C and D turn on and
off alternately, behaving like a typical
synchronous boost regulator.
The duty cycle of switch C decreases
until the minimum duty cycle of the
converter in boost mode reaches
4%–6%.
When this minimum duty cycle
is reached, the LTC3780 shifts into
buck-boost mode.
Like continuous current mode,
discontinuous current mode features
constant frequency and extremely
low ripple, and improves efﬁciency at
light loads by turning off the relevant
synchronous switch (B or D). In boost
mode, switch D remains off if the load
is light enough. In buck mode, switch B
turns on every cycle, just long enough
to produce a small negative inductor
current; this sequence maintains
constant frequency operation even at
no load (Figure 9).
Burst Mode (in boost operation,
Figure 10) and Skip Cycle mode (in
buck operation, Figure 11) provide the
highest possible light load efﬁciency.
In Burst Mode operation, switches C
and D operate in brief pulse trains
1.24k
18mΩ
BURST
*RSENSE =
18mΩ
Figure 6. An LTC3780-based DC/DC converter delivering 12V/5A from a 5V–32V input.
4
Linear Technology Magazine • September 2005
DESIGN FEATURES
CLOCK
CLOCK
SWITCH A
SWITCH A
SWITCH B
SWITCH B
0V
SWITCH C
SWITCH C
VOUT
SWITCH D
SWITCH D
I
IL
a. Buck mode (VIN > VOUT)
b. Buck-boost mode (VIN ≈ VOUT)
CLOCK
CLOCK
SWITCH A
SWITCH A
VIN
0V
SWITCH B
SWITCH B
SWITCH C
SWITCH C
SWITCH D
SWITCH D
I
IL
c. Buck-boost mode (VIN ≈ VOUT)
d. Boost mode (VIN < VOUT)
Figure 8. Power switch gate drive control in continuous conduction mode, in various regions of operation.
while holding switch A on. Skip Cycle
mode only turns on the synchronous
buck switch B when the inductor
current reaches a minimum positive
level, which does not happen every
cycle at very light loads. Since energy
devoted to switching dominates the
power loss picture at very light loads,
both of these switching arrangements
raise efﬁciency.
A single sense resistor placed
between ground and the source terminals of both synchronous MOSFETs
determines the current limit. It reliably
governs the valley of the inductor current in buck mode and the maximum
SWITCH A
inductor peak current in boost mode.
The LTC3780 monitors the current via
an internal comparator. This single
sense resistor structure dissipates
little power (compared with multiple
resistor sensing schemes) and provides consistent current information
for short circuit and over current
protection.
Flexible Power
Although the LTC3780 is ideal for applications where the range of possible
input voltages straddles the output
voltage in everyday operation, it is also
useful as a dedicated synchronous
buck or boost controller. Applications
requiring a ﬁxed output from a variety
of input rails can beneﬁt from the
simplicity of a single drop-in design.
At a minimum, the same layout can
be repeated, with power switches and
passive components scaled to the
particular input voltage and output
load requirements.
The LTC3780 is by itself an
outstanding synchronous boost controller. Dedicated boost controllers
typically have narrower input or output
voltage ranges than the LTC3780, and
nonsynchronous versions (the most
common type) suffer from signiﬁSWITCH A
SWITCH A
SWITCH B
SWITCH B
SWITCH B
SWITCH C
SWITCH C
SWITCH C
SWITCH D
SWITCH D
IL
IL
SWITCH D
IL
DISCONTINUOUS CURRENT MODE
BUCK MODE
NO LOAD
Figure 9. Switch operation in discontinuous
current mode, buck mode, no load. Switch B
turns on every cycle, until the inductor current
goes slightly negative. The inductor current
then free-wheels through the body diode of
switch B (or a Schottky diode in parallel with
it). Switches C and D occasionally trigger to
refresh switch D’s bootstrap capacitor.
Linear Technology Magazine • September 2005
BURST MODE
BOOST MODE
NO LOAD
Figure 10. Switch operation in Burst Mode
operation, boost mode, no load. Switches A
and B are toggled to connect the true boost
converter directly to the input rail, with
occasional refresh pulses for switch
A’s bootstrap capacitor. During the sleep
period between bursts, switches A, C, and D
remain off.
SKIP CYCLE MODE
BUCK MODE
NO LOAD
Figure 11. Switch operation in skip cycle
mode, buck mode, no load. Note the similarity
to discontinuous current mode, except switch
B is not turned on every cycle. In this way,
energy is saved by allowing the inductor
to discharge through the body diode of
switch B (or the Schottky diode across it,
if there is one).
5
DESIGN FEATURES
100
LTC3780
EFFICIENCY (%)
95
90
SEPIC
CONVERTER
85
12V/5A SEPIC
SOLUTION
80
12V/5A
LTC3780-BASED
SOLUTION
75
70
5
10
15
20
VIN (V)
Figure 13. They may be similar in functionality, but not even close in size.
The hulking inductor in the SEPIC on the left casts a big shadow on its
counterpart in the LTC3780-based 12V/5A application on the right.
Figure 12. The LTC3780 12V/5A converter
beats a SEPIC in efﬁciency across the board.
cant power loss in the free-wheeling
Schottky diode. Compared to a typical
non-synchronous boost converter,
the circuit of Figure 6 can yield an
increase of over 5% in efﬁciency at
moderate loads.
only less efﬁcient but quite a bit larger.
A SEPIC transformer would occupy
twice the footprint of the inductor in
our buck-boost example, and would
stand twice as high (Figure 13).
Even the large off-the-shelf coupled
inductor of Figure 13 would be insufﬁcient for the current levels seen
when boosting 5V to 12V at 5A—a
safe minimum input voltage would be
around 6V. To convert 32V to 12V, a
SEPIC would require a power switch
rated at 60V (the lowest prevailing
drain-to-source voltage > VIN + VOUT),
yet the output current would demand
a low RDS(ON), requiring multiple SO-8
MOSFETs or a much larger TO-220.
The coupling element would consist
of large, expensive, high voltage
ceramic capacitors, in addition to
VOUT
10V/DIV
Surpassing the SEPIC
SW2
20V/DIV
Whatever the operating mode, the
single inductor buck-boost structure
has high power density and high efﬁciency. Compared with a coupled
inductor SEPIC converter, its efﬁciency
can be 8% higher. Figure 12 shows
the efﬁciency comparison between a
typical LTC3780 12V/5A application
and a SEPIC converter, which is not
SW1
20V/DIV
IL
5A/DIV
20µs/DIV
Figure 14. Current foldback handles short
circuits without dragging down the input rail.
VIN, represented here by the peaks of SW2,
remains solid.
continued on page 46
RPU
2
CC2
47pF
CC1
0.01µF
68pF
RC
100k
R1
8.66k
R2 113k
4
5
6
7
ON/OFF
75k
3
INTVCC
INTVCC
8
9
10
11
12
CSTBYMD
0.01µF
DAC (VREF)
PGOOD BOOST1
SS
LTC3780
SENSE+
–
SENSE
ITH
TG1
SW1
VIN
EXTVCC
VOSENSE INTVCC
SGND
RUN
BG1
PGND
FCB
BG2
PLLFLTR
SW2
PLLIN
TG2
STBYMD BOOST2
24
CA
0.22µF
23
22
D
DA
1N5819HW
330µF
16V
CF 0.1µF
21
C
20
CVCC 4.7µF
19
18
L
4.7µH
Toko
FDA1254
RSENSE*
17
16
B
15
DB
1N5819HW
14
A
13
CB 0.22µF
10Ω
100Ω
100Ω
VREF = 2.33V TO 4.7V
VOUT = 13.28 – 1.5(VREF)
COUT
3x
22µF
25V
X5R
RENESAS
HAT2210WP
+
1
VPULLUP
22µF
25V
+
CSS
0.022µF
VOUT
6V–12V
4A
RENESAS
HAT2210WP
CIN
3x
22µF
25V
X5R
VIN
7V–15V
30mΩ
*RSENSE =
30mΩ
Figure 15. A compact, adjustable output supply
6
Linear Technology Magazine • September 2005
DESIGN FEATURES
Dual, 1.4A and 800mA, Buck
Regulator for Space-Sensitive
Applications
Introduction
The evolution of cell phones, PDAs,
palmtop PCs, digital cameras, PC
cards, wireless and DSL modems is
one of squeezing an increasing number
of features in ever-smaller devices.
As features increase, so do the number of required power supplies. The
problem is how to ﬁt more supplies
in less space. There are a number of
solutions, including: increasing the
switching frequency (allowing the use
of smaller and less costly capacitors
and inductors), integrating the switcher MOSFETs, or combining multiple
switchers into a single package. The
LTC3417 combines all of these.
A Small Package
Loaded with Features
The LTC3417 is a dual synchronous,
step-down, current mode, DC/DC
converter designed for medium power
applications. It operates from an input
voltage between 2.25V and 5.5V and
switches at up to 4MHz, making it
possible to use capacitors and inductors that are under 2mm in height. It
comes in a 3mm × 5mm, 16-lead DFN
or a 20-lead TSSOP. A complete dual
buck DC/DC switching regulator, using the LTC3417 in its small 16-lead
DFN package, can consume less than
0.45 square inches of board real estate,
as shown in Figure 1.
High Efﬁciency Dual Output
A typical application for the LTC3417
is shown in Figure 2. The two outputs of the LTC3417 are individually
adjustable from 0.8V to 5V. VOUT1
can provide up to 1.4A of continuous current while VOUT2 can provide
up to 800mA of continuous current,
both at efﬁciencies of as high as 96%.
OPTI-LOOP compensation allows the
transient response to be optimized
over a wide range of loads and output
capacitors.
Linear Technology Magazine • September 2005
by Scott Fritz
all the way up to 4MHz. With a 143k
resistor pulled from FREQ to ground,
the frequency of operation is 1MHz.
Figure 1. Dual buck regulator conserves space
Easy to Conﬁgure
The output voltages for the LTC3417
are set by the resistor dividers at the
VFB pins, where the feedback voltage is compared to an internal 0.8V
reference.
Major loop compensation adjustments are made with components at
the ITH pins. The placement of the
pole/zero combination is integral in
the loop dynamics of the device, and
consequently, different loop characteristics can be optimized with changes
in these components, such as turn-on
time, step response, and output ripple.
Furthermore, the feed forward capacitor connected from VOUT to VFB also
helps with step response and voltage
ripple. In all, the designer using the
LTC3417 has exceptional control over
the loop characteristics.
Constant Frequency up to 4MHz
for Noise Sensitive Applications
The LTC3417 uses a current mode,
constant frequency architecture
that beneﬁts noise sensitive applications—the constant frequency of the
oscillator simpliﬁes noise ﬁltering.
The frequency of operation is set using the FREQ pin. When the FREQ
pin is pulled high, to VIN, the internal
oscillator runs at 1.5MHz. Pulling the
FREQ pin low, through an external
resistor, allows the user to vary the
frequency anywhere between 600kHz
High Efﬁciency at Light Loads
Efﬁciency at light loads is important
in battery-powered applications since
many portable applications spend
most of their time in of standby or
sleep mode. The LTC3417 offers three
operating modes allowing the designer
to optimize light load efﬁciency and
noise: Burst Mode operation for the
highest efﬁciency at light loads, pulse
skipping mode for high efﬁciency and
simpliﬁed noise suppression, and
forced continuous mode for noise
sensitive applications. The operating
mode, for both outputs, is selected
through the MODE pin.
Figure 3 shows the efﬁciency vs
load current for all three modes for the
1.4A VOUT1 output. Figure 4 shows the
efﬁciency vs load current for all three
modes for the 800mA VOUT2 output.
The external components used to take
the data in Figures 3 and 4 are shown
in the typical application schematic
of Figure 2. In all modes, with no
load, the dual converter draws only
100µA. In dropout, when the output
voltage is within 100mV to 200mV
of the input voltage, the internal Pchannel MOSFET switch is turned on
continuously, thereby maximizing the
usable battery life. In shutdown, when
both outputs are turned off (RUN1
and RUN2 are pulled to Ground), the
LTC3417 draws less than 1µA, making
it ideal for low current, long battery
life, applications.
Burst Mode operation achieves
high efﬁciencies over a wide range of
load currents. Burst Mode is selected
for both outputs by pulling MODE to
VIN. In this mode gate charge losses
and internal quiescent current losses
are minimized at low load currents
thus achieving high efﬁciencies over
7
DESIGN FEATURES
VIN
2.25V TO 5.5V
CIN
10µF
CIN1
0.1µF
MODE
VIN
SW2
RUN1
VIN
RUN2
VFB1
R2
412k
PHASE
VOUT2
2.5V
800mA
C2 22pF
LTC3417
R1 511k
COUT1
22µF
L2
2.2µH
PGOOD
SW1
C1 22pF
R7
100k
VIN1 VIN2
L1
1.5µH
VOUT1
1.8V
1.4A
CIN2
0.1µF
R3 866k
VFB2
VIN
FREQ
ITH2
EXPOSED
GNDA PAD GNDD
COUT2
10µF
R4
412k
ITH1
R5
5.9k
R6
2.87k
C3
2200pF
C4
6800pF
L1: MIDCOM DUS-5121-1R5R
COUT1: KEMET C1210C226K8PAC
L2: MIDCOM DUS-5121-2R2R
COUT2, CIN: KEMET C1206C106K4PAC
Figure 2. Dual output converter produces 1.8V at 1.4A and
2.5V at 800mA, with ceramic input and output capacitors.
100
100
90
90
VIN = 3.6V
95 VOUT = 2.5V
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 2.5V
95 VOUT = 1.8V
85
80
75
Burst Mode
OPERATION
PULSE SKIP
FORCED
CONTINUOUS
70
65
60
0.001
0.01
0.1
1
85
80
75
Burst Mode
OPERATION
PULSE SKIP
FORCED
CONTINUOUS
70
65
10
60
0.001
LOAD CURRENT (A)
0.01
0.1
1
LOAD CURRENT (A)
Figure 3. 1.4A VOUT1 Efﬁciency.
Figure 4. 800mA VOUT2 Efﬁciency
a wide load current range. At low load
currents, the control loop turns off
all unnecessary circuitry, and stops
switching for short periods of time.
This generates variable frequency
VOUT ripple components that change
with load current. Of the three modes,
the output voltage ripple is highest in Burst Mode operation—up to
25mVP–P.
Where supply noise suppression
takes on more importance than
efﬁciency, especially in telecommunications devices, pulse skipping
mode can be selected by pulling the
MODE pin to ground. This mode does
not have the wide range of high efﬁciency that Burst Mode has, but the
voltage ripple is minimized and the
frequency components of that ripple
are controlled over a wider load current
range. At lower load currents, where
the output skips pulses, there can be
variable frequency components in the
voltage ripple, but the ripple is only
around 5mVP–P.
To reduce ripple noise even further use forced continuous mode.
This mode decreases the ripple noise
by sustaining the switching of the
MOSFETS over all load currents, which
results voltage ripple below 5mVP–P,
while trading off efﬁciencies at low
load currents. Since the MOSFETS are
always switching, the voltage ripple is
constant, allowing for better ﬁltering
of the voltage ripple noise. Forced
continuous mode is selected by setting
the MODE pin at VIN/2.
Out of Phase Operation Reduces
Ripple and Increases Efﬁciency
To help reduce noise on the input
voltage, and reduce the size of input
capacitor, the two outputs on the
LTC3417 can be selected to operate
out of phase. The second output,
when the PHASE pin is low, operates
180 degrees out of phase with the
ﬁrst channel. Out-of-phase operation
produces lower RMS current on VIN
and thus lowers RMS derating on the
capacitor on VIN.
A High Efﬁciency 2.25V Dual
Step-Down DC/DC Converter
with all Ceramic Capacitors
The low cost and low ESR of ceramic
capacitors make them a very attractive
choice for use in switching regulators. Unfortunately, the ESR is so
low that it can cause loop stability
problems. Solid tantalum capacitor
continued on page 27
VOUT1
20mV/DIV
VOUT1
20mV/DIV
VOUT1
20mV/DIV
IL
250mA/DIV
IL
250mA/DIV
IL
250mA/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 100mA
2µs/DIV
Figure 5. Burst Mode
8
VIN = 3.6V
VOUT = 1.8V
ILOAD = 100mA
2µs/DIV
Figure 6. Pulse Skipping
VIN = 3.6V
VOUT = 1.8V
ILOAD = 100mA
2µs/DIV
Figure 7. Forced Continuous
Linear Technology Magazine • September 2005
DESIGN FEATURES
µPower Precision Dual Op Amp
Combines the Advantages of
Bipolar and CMOS Ampliﬁers
by Cheng-Wei Pei and Hengsheng Liu
Introduction
The LTC6078 and LTC6079 are dual
and quad micropower, precision op
amps that combine the low offset and
drift of traditional bipolar ampliﬁers
with the low bias current of CMOS
ampliﬁers. Additionally, the LTC6078
features low supply current and low
noise, low supply voltage operation,
and rail-to-rail input and output
stages. This combination of features
allows precision performance previously available only through composite
ampliﬁers, manual offset trimming, or
calibration.
A Superior Op Amp
Traditional bipolar op amps can be
designed for excellent precision over
temperature, but bipolar ampliﬁers
lack the low bias currents of CMOS
ampliﬁers. In high source impedance
applications, a few nanoamps of input
bias current can equal millivolts or
more of input error, swamping the
ampliﬁer’s low VOS.
Traditional (and some non-traditional) CMOS ampliﬁers come with
their own set of strengths and weaknesses. Input bias current can be much
lower than their bipolar counterparts.
But VOS and VOS drift speciﬁcations
often limit the usefulness of these ampliﬁers in high-precision applications,
VDD
I1
R1
100Ω
presenting the opposite problem of
bipolar ampliﬁers. Chopper-stabilized
(also known as zero drift) ampliﬁers,
which are generally CMOS based, employ a synchronous offset cancellation
scheme to negate the DC imperfections of the ampliﬁer, so that VOS
and VOS drift of the ampliﬁer become
almost negligible. However, zero drift
An LTC6078 circuit with
passive high-impedance
sensors can run on two
alkaline AA batteries for
over 1.5 years.
ampliﬁers tend to draw much more
current than their continuous-time
counterparts, precluding their use
in low-power precision applications.
Additionally, zero drift ampliﬁers may
be much noisier at higher frequencies due to auto-zero circuit clock
harmonics.
The LTC6078 is a CMOS op amp
with a proprietary VOS trimming circuit
that yields 25µV maximum VOS and
0.7µV/°C maximum VOS drift, lowest
among all comparable bipolar and
CMOS op amps. Combined with the
50pA maximum input bias current
VDD
RSENSE
1Ω
+
2N7002
–
IL
LOAD
IBIAS
R2
VOUT =
•R
•I
R1 SENSE 1
R2
1k
0V < VOUT < VDD – VGS(MOSFET)
Figure 1. Precision, low-supply-voltage current sense ampliﬁer. The LTC6078 servos
the N-channel MOSFET drain current so that the voltage across R1 is the same as the
voltage across the sense resistor. The precision of the LTC6078 enables a small sense
resistor to be used for less power dissipation without sacriﬁcing DC accuracy.
Linear Technology Magazine • September 2005
over the entire temperature range,
the LTC6078 is ideal for all precision
or high-impedance instrumentation
applications. The low 54µA supply current and 2.7V minimum supply voltage
make the LTC6078 an excellent choice
for power-sensitive or hermetically
sealed circuits. An LTC6078 dual op
amp circuit with passive high-impedance sensors can run on two alkaline
AA batteries for over 1.5 years.
Precision Current
Sense and Control
The LTC6078’s rail-to-rail input and
output stages allow precision input
sensing right at VDD or VSS, which is
useful for simple high-side or low-side
current sensing. Figure 1 shows the
LTC6078 in a simple, precise high-side
current sensing application. The 25µV
precision translates to excellent current resolution with a very small sense
resistor, meaning more precision with
less power loss. Used in a feedback
loop, the LTC6078 can be used as a
precision current source/sink or as a
current servo.
Figure 2 shows the LTC6078 balancing the loads on two paralleled
LT1763 low dropout (LDO) voltage
regulators. A common practice when
paralleling two voltage regulators is to
simply tie the two outputs together.
However, internal voltage offsets cause
one regulator to handle the bulk (or
all) of the load current. In the case of
sink-source regulators, one may be
sourcing a great deal of current into
the other regulator!
Load sharing circuits work best
when the contribution to output
current is balanced between the
regulators. The LTC6078 compares
the voltage outputs of the two LDOs
and servos the feedback pin of the
second to balance them. The high
9
DESIGN FEATURES
For extremely low power applications
such as hermetically sealed batterypowered sensors, the 10-pin version of
the LTC6078 in the tiny DFN package
offers two shutdown pins (one for each
ampliﬁer). When in shutdown mode,
the low 54µA per ampliﬁer current
draw is reduced to a maximum of 1µA
(over the entire temperature range).
The fast 50µs turn-on and 2µs turnoff times ensure that minimal power
is dissipated during the transition
periods.
In applications where many inputs
need to be monitored and only a single
analog-to-digital converter is available,
the independent shutdown function of
the two ampliﬁers allows any number
of LTC6078 outputs to be multiplexed
together. The high-impedance output
of the LTC6078 in shutdown mode
does not load the output of an active
LTC6078. So long as two ampliﬁers
are not simultaneously active, there
IN
OUT
LT1763
SHDN
BYP
IN
LT1763
OUT
10µF
R1
2k
0.01µF
BYP
GND
FB
0A ≤ IOUT ≤ 1A
LOAD MATCHING TO WITHIN 1mA WITH 25mΩ OF
TRACE LENGTH (5 INCHES OF 28-GAGE STRANDED WIRE)
1.22V ≤ VOUT ≤ VDD (OF LTC6078)
IL
LOAD
LTC6078
Figure 2. The LTC6078 used as a current load balancing servo ampliﬁer. Short
lengths of copper wire or PCB trace can be used as ballast resistors due to the
LTC6078’s precision. VDD of the LTC6078 may be connected to VIN or VOUT,
as long as the minimum 2.7V supply voltage requirement is met.
is no need for external multiplexing
components.
Layout Considerations
In high source impedance applications
such as pH probes, photodiodes, strain
gauges, et cetera, the LTC6078’s low
input bias current (50pA maximum
over temperature) requires a clean
board layout to minimize additional
leakage current into a high-impedance signal node. A mere 100GΩ of
PC board resistance between a 5V
supply trace and an input trace adds
50pA of leakage current, which is
typically greater than the input bias
current of the LTC6078. For comparison, a bit of unwashed soldering ﬂux
OUT
IN–
5
4
IN+
4
V–
LEAKAGE
CURRENT
Figure 3. A sample layout using a low-impedance guard ring to shield a highimpedance signal trace from board leakage sources. The output pin can drive
the guard ring directly or through a low impedance (<100kΩ) feedback resistor.
The ampliﬁer is shown in a non-inverting gain conﬁguration.
10
R1
)
R2
1k
10k
R2
2k
NO LEAKAGE
CURRENT
GUARD
RING
VOUT = 1.22V (1 +
100Ω
0.01µF
LTC6078 CMS8
R
IDENTICAL LENGTH,
THERMALLY MATED
WIRE OR PCB TRACE
R2
2k
SHDN
6
NO SOLDER MASK
OVER THE GUARD RING
10µF
R1
2k
FB
GND
10µF
0.01µF
+
Shutdown Function
VIN
1.8V TO
20V
–
precision of the LTC6078 means that
discrete ballast resistors are unnecessary—short pieces of wire or PCB trace
are sufﬁcient to provide the ballast
resistance. With 25mΩ of resistance1,
the LTC6078 can balance the current
sharing of the LDOs to be within 1mA,
regardless of the absolute load current output. The feedback network
does not noticeably degrade the load
transient performance of the regulators, and Figure 2 can be expanded to
include as many paralleled regulators
as necessary.
can add a 1GΩ–10GΩ resistance. In
critical applications, or if leakage is
suspected, a guard ring around the
high-impedance input traces driven
by a low-impedance source to equal
the input voltage prevents such leakage problems. The guard ring should
extend as far as necessary to shield the
high-impedance signal from any and
all potential leakage paths. Figure 3
shows the recommended layout when
using a guard ring.
Conclusion
The LTC6078 offers all of the beneﬁts
of both bipolar and CMOS ampliﬁers,
as well as a slew of other features that
make it the ultimate choice for low
power, precision applications. The
combination of excellent offset, drift,
and input bias current speciﬁcations
is unmatched among both bipolar
and CMOS op amp offerings. For
applications requiring four precision
op amps, the LTC6079 is available in
16-pin surface-mount SSOP and DFN
packages.
Notes
1 A 25mΩ resistor is equal to approximately 5 inches
of AWG 28 gauge copper stranded wire or 1.25
inches of a 25 mil wide one-ounce copper PCB
trace at room temperature.
Authors can be contacted
at (408) 432-1900
Linear Technology Magazine • September 2005
DESIGN FEATURES
High Voltage Micropower Regulators
Thrive in Harsh Environments by Todd Owen
Introduction
Some of the harshest environments
for electronic circuits are found in
industrial and automotive applications, where high, unpredictable input
voltage transients, failing batteries and
disconnected supplies are an expected
part of doing business. A 12V car battery and a 48V industrial supply rail
offer many of the same power supply
design challenges. Input voltages can
spike to 75V on an industrial rail or
60V during an automotive load dump.
Cold cranking or overloaded lines can
drop the supply voltage to only a few
volts. Through all of this, the power
supply must be kept alive and remain
in regulation for critical circuitry, and
it cannot dissipate much quiescent
power—dead car batteries or lost industrial sensor data do not make for
happy customers.
Tough New Regulators
Provide Flexibility in
Rough Environments
Three new linear regulators provide
flexible options when running in
these environments. The LT3012
and LT3013 provide up to 250mA at
a dropout voltage of 360mV, and the
LT3014 provides up to 20mA at a
dropout voltage of 350mV.
IN
VIN
3.7V TO
80V
1µF
OUT
LT3014
SHDN
2.05M
ADJ
GND
VOUT
3.3V
20mA
0.47µF
1.21M
VSHDN OUTPUT
<0.3V
OFF
ON
>2.0V
NC
ON
Figure 1. LT3014 3.3V/20mA keep-alive supply with shutdown
The LT3012 and LT3013 offer an
input voltage range of 4V to 80V and
minimize power loss by running with
quiescent currents of 55µA (LT3012)
and 65µA (LT3013). Quiescent current
is reduced to just 1µA in shutdown.
Both are stable with only a small
3.3µF capacitor on the output. Small
ceramic capacitors can be used without any need for additional series
resistance as is common with other
regulators. The LT3013 adds a power
good ﬂag, PWRGD, to indicate that the
output is above 90% of nominal, with
a delay that is programmable using a
single capacitor. Both the LT3012 and
LT3013 are offered with adjustable
outputs and a 1.24V reference voltage.
The regulators are packaged in the
16-lead TSSOP and 12 pin low proﬁle
(0.75mm, 4mm × 3mm) DFN packages with exposed pads for enhanced
thermal handling capability. These
surface mount packages are capable
of handling 1W to 3W of continuous
dissipation and much higher peak
transients. See the data sheet for a
detailed discussion.
The LT3014 offers a 3V to 80V input
voltage range, and the 7µA operating
quiescent current (1µA in shutdown)
makes it an attractive choice for powercritical systems. The LT3014HV is a
higher voltage option of the regulator
that is rated to withstand 100V transients for up to 2ms. The regulator is
stable with only 0.47µF of capacitance
on the output—small ceramic capacitors can be used without any need for
added series resistance. It is available
as an adjustable part with a 1.22V
reference voltage and is packaged in
the 5-lead ThinSOT and 8-lead (3mm
× 3mm) DFN packages.
All of the regulators have internal
protection circuitry for reverse-bat-
Table 1. Linear Technology High Voltage Linear Regulator Portfolio
Part #
Output
Current
Maximum Input
Voltage
Dropout
Voltage
Quiescent
Current
Output
Capacitor
Added Features
LT3014
20mA
80V
350mV
7µA
0.47µF
5-lead ThinSOT Package or
8-lead (3mm × 3mm) DFN
LT3014HV
20mA
100V
(2ms Survival)
350mV
7µA
0.47µF
5-lead ThinSOT Package or
8-lead (3mm × 3mm) DFN
LT3010
50mA
80V
300mV
30µA
1µF
8-lead Thermally Enhanced
MSOP Package
LT3012
250mA
80V
360mV
55µA
3.3µF
16-lead TSSOP or 12-lead (4mm × 3mm)
DFN Thermally Enhanced Packages
LT3013
250mA
80V
360mV
65µA
3.3µF
All Features of the LT3012 Plus a PWRGD
Flag with Programmable Delay
Linear Technology Magazine • September 2005
11
DESIGN FEATURES
tery, current limit, thermal limit, and
reverse voltages from output to input.
Table 1 provides a summary feature
comparison of these and other high
voltage linear regulators.
Higher Output Current
Supply with PWRGD Flag
When higher output currents are
needed, Linear Technology offers
several options. The LT3010 regulator
can provide 50mA of output current
at a maximum dropout of 300mV and
12
LT3013
CIN
2.2µF
SHDN
CDELAY
22nF
CT
5V
250mA
OUT
ADJ
R1
750k
R2
249k
COUT
3.3µF
RPWRGD
100k
PWRGD
GND
PWRGD
Figure 2. LT3013 5V/250mA supply provides PWRGD ﬂag
a maximum input voltage of 80V. If a
higher output current is needed, the
LT3012 and LT3013 are now available.
Figure 2 shows a 5V supply that can
provide up to 250mA of current using
the LT3013. This supply operates at
only 65µA quiescent current and requires only 3.3µF of capacitance on
the output. An added feature that this
part has is a PWRGD ﬂag that indicates
when the ADJ pin is within 10% of its
nominal voltage of 1.24V.
The PWRGD ﬂag of the LT3013
can be used to provide a microprocessor reset signal or other logic ﬂag.
Figure 3 shows the block diagram
for the PWRGD ﬂag. The PWRGD pin
is an open-collector output, capable
of sinking 50µA when the output is
low; there is no internal pull-up on
the PWRGD pin, an external resistor
must be used. When the output rises
to within 10% of its ﬁnal value, a JK
ﬂip-ﬂop allows a 3µA current source to
begin charging the capacitor on the CT
pin. As the CT pin reaches its trip level
(approximately 1.6V at 25°C), the 3µA
current is shunted away to clamp the
capacitor voltage and set the PWRGD
ﬂag state to high impedance.
During normal operation, an internal glitch ﬁlter ignores short transients
(<15µs) on the output voltage. Longer
transients below the 10% low threshold
will reset the internal JK ﬂip-ﬂop. This
ﬂip-ﬂop ensures that the capacitor on
the CT pin is fully discharged to the
VCT(LOW) threshold before re-starting
the time delay. This is done to provide
a consistent time delay after the output
returns to within 10% of its regulated
voltage before the PWRGD pin switches
to the high impedance state.
If the PWRGD function of the
LT3013 is not needed for your application, the LT3012 can be used to provide
the same regulator performance. The
removal of the PWRGD circuitry allows
the LT3012 to operate at a lowered
quiescent current of 55µA.
Conclusion
Three new regulators satisfy the needs
of tough automotive and industrial
electrical environments. A range of
possible output currents and quiescent currents allow designers to
pick a solution that can limit power
dissipation during normal operation
while still providing the capability to
handle high voltage transients. The
most important feature of these devices
is that they can withstand the rigors
of the worst electrical environments,
thus ensuring a steady power supply
for critical circuits.
CT
PWRGD
ICT
3µA
ADJ
+
J
VREF • 90%
–
VCT(HIGH) – VBE
≈1.1V
Q
K
+
Figure 1 shows a typical application using the low current LT3014
to provide a 3.3V keep-alive supply,
such as a real-time clock, a security
system, or any other system that runs
constantly from a battery. The 7µA
quiescent current keeps the power
supply from being a signiﬁcant drain
on the battery.
The resistor divider is sized to
match the tiny operating currents of
the LT3014. Care must be exercised
when operating at such low currents,
since a small error can cause large
percentage shifts. Insufﬁcient cleaning of solder ﬂux after assembly can
provide resistances in the MΩ range,
leading to erroneous outputs.
The SHDN pin of the LT3014 can
be driven to GND by external circuitry
to turn the output of the regulator off
and reduce operating current. Leaving
the SHDN pin open or pulling it above
2V will turn the regulator on.
The output of the LT3014 needs only
the 0.47µF capacitor for stability. The
regulator is designed to be stable with
the widest possible range of output
capacitors—the ESR of the output
capacitor can be zero, as is common
with small ceramic capacitors, or
can be as high as 3Ω, a value found
more often with small tantalum or
aluminum electrolytic capacitors. The
minimum output capacitor coupled
with the micropower nature of the
LT3014 tends to give larger voltage
deviations with fast transients, so
larger values are recommended if there
are large current steps.
IN
–
High Voltage
Low Current Keep-Alive
Regulator Application
5.4V TO
80V
VCT(LOW)
≈0.1V
Figure 3. PWRGD circuit block diagram
Linear Technology Magazine • September 2005
DESIGN FEATURES
Complete 2-Cell-AA/USB Power
Manager in a 4mm × 4mm QFN
by G. Thandi
Introduction
One of the most popular battery solutions for consumer handheld devices
is the venerable two-cell AA (alkaline
or nickel-metal hydride) source,
especially in GPS navigators, digital
cameras and MP3 players. AA batteries are readily available, relatively
low cost and offer high power density.
Many of these same portable devices
supplement battery power with plugin wall adapter and offer a USB bus
(for data transfer). The USB bus can
also be used to provide power. The
problem is how to seamlessly switch
between these three disparate types
of supplies: 2-cell AA, wall and USB.
The solution is the LTC3456.
The LTC3456 is a complete system
power IC that seamlessly manages
power ﬂow between an AC wall adapter,
USB and 2-AA battery, while complying with USB power standards—all
in a 4mm × 4mm QFN package
(Figure1). The device generates two
separate power rails: a 3.3V (ﬁxed)
main supply and a 1.8V (adjustable) core supply. In addition, the
2 AA
CELLS
+
VCORE (1.8VADJ)
LTC3456
USB POWER
• SMART SELECTION
OF POWER SOURCE
AC ADAPTER
• USB POWER MANAGER
2 AA CELLS
• HIGH EFFICIENCY
DC/DC CONVERTERS
LTC3456 contains a fully featured USB
power manager, a Hot Swap output for
powering memory cards and an un-
The LTC3456 squeezes a USB
power manager, four high
efﬁciency DC-DC converters,
a Hot Swap controller,
a low-battery indicator
and much more into a
4mm × 4mm QFN package.
committed gain block suitable for use
as a low-battery comparator or an LDO
controller. The device also generates
an always-alive VMAX output, suitable
4.7µF
L2
4.7µH
VMAX
VBATT
About the LTC3456
The LTC3456 contains four high
efficiency 1MHz fixed frequency
switching regulators that operate
with efﬁciencies up to 92%. Figure 2
shows a typical LTC3456 application.
Most processors used in portable applications require dual power supply
voltages. These voltages can be 3.3V
for the I/O circuitry and 1.5V or, 1.8V
for the processor core. Additionally,
the processor might require that the
power supplies startup in a speciﬁc
sequence to prevent processor latch
22µF
SW2_BST
SW2_BK
VINT
1µF
1µF
Hot Swap OUTPUT
3.3V
50mA
AO
USB
CONTROLLER
L1
10µH
SUSPEND
LTC3456
USB
1Ω
MBRM120E
4.7µF
1Ω
HSO
USBHP
1k
ON/OFF
220pF
FB1
10µF
80.6k
WALLFB
4.32k
CORE OUTPUT
1.8V
10µF 200mA
100k
EXT_PWR
VEXT
11.3k
SW1
VINT
3.3V
MAIN OUTPUT
3.3V
150mA
VMAIN
80.6k
AC WALL
ADAPTER
(5V ±10%)
MEMORY CARD
REAL-TIME CLOCK
for supplying power to critical blocks
like the real time clock, which needs to
stay alive even during shutdown.
L3
10µH
AIN
4.7µF
DSP I/O
HOT SWAP (3.3V)
Figure 1. The LTC3456 is a complete system power
management IC available in a tiny 4mm×4mm package.
100k
USB POWER
(4.35V TO 5.5V)
DSP CORE
VMAIN (3.3V)
VMAX
PWRKEY
PBSTAT RESET MODE PWRON PGND AGND
1µF
VMAX
(POWERS
REAL-TIME CLOCK)
µP
Figure 2. A LTC3456-based, complete power solution, including a 1.8V output for processor core, a 3.3V output for the I/O, a 3.3V
Hot Swap supply for the memory card and a VMAX output for RTC. This design uses all ceramic capacitors with minimal parts count.
Linear Technology Magazine • September 2005
13
DESIGN FEATURES
90
80
80
70
70
EFFICIENCY (%)
100
90
EFFICIENCY (%)
100
60
50
40
30
20
VBATT = 2.4V
VCORE = 1.8V
10
0
1
PWM MODE
Burst Mode
OPERATION
10
100
LOAD CURRENT (mA)
1000
all discharged to ground in shutdown.
The VMAX output is the highest of
the VBATT, VINT,VEXT and USB voltages.
This output can be used to supply a
maximum of 1mA output current. The
VMAX output stays alive even when
the IC is in shutdown and is suitable
for supplying power to critical system
blocks like a real time clock.
VUSB = 5V
VUSBHP = 2V
60
50
40
30
20
1.8V OUTPUT
3.3V OUTPUT
10
0
1
10
100
LOAD CURRENT (mA)
PowerPath Control
1000
Figure 3. Core converter efﬁciency for
Figure 2’s circuit. LTC3456 is powered
from the battery. Efﬁciency is shown for
both PWM and Burst Mode operation.
Figure 4. Core and Main converter efﬁciency
for Figure 2’s circuit. The LTC3456 is powered
from the USB with the USB current limit set at
500mA (USBHP = High).
up or improper initiation. Usually the
core supply must come up before the
I/O supply.
The LTC3456 has in built power
supply sequencing for the core and
main outputs. At power-up, the VINT
output, a ﬁxed 3.3V supply, is the ﬁrst
one to power up. It supplies power
to most of the internal circuitry. The
amount of external loading at this
output should be limited (Refer to the
LTC3456 datasheet for more details).
The core output, adjustable from 0.8V
to 1.8V, comes up next followed by the
VMAIN output. The VMAIN output, a ﬁxed
3.3V supply, powers up with a delay
of 0.8ms (typ) after the core output
comes into regulation. The VMAIN output is generated from the VINT output
through an internal 0.4Ω (typ) PMOS
switch and can be used to power the
I/O circuitry. The 0.8ms delay gives
sufﬁcient time to the processor to
stabilize the system clock and load
internal registers before the peripheral
circuitry powers up.
The LTC3456 produces a Core
output, adjustable from 0.8V to 1.8V
suitable for powering new low voltage
processors (ARM and others). The
LTC3456 control scheme allows 100%
duty cycle operation for the core output. It provides low dropout operation
when the core output is powered from
the battery, thereby extending battery
life. Both Main and Core converters
offer Burst Mode operation (MODE
Pin selectable) when powered from
the battery resulting in high efﬁciency
at light loads as seen in Figure 3.
The Core converter features greater
than 92% efﬁciency when powered
from the battery. Burst Mode operation is disabled when powered from
USB/wall power. Figure 4 shows the
system efﬁciency when powered from
the USB. The Main converter achieves
up to 90% efﬁciency when powered
from the USB.
The LTC3456 has a built-in Hot
Swap output suitable for powering
ﬂash memory cards. The Hot Swap output features short-circuit and reverse
voltage blocking capability. It allows
memory cards to be hot swapped into
and out of the system. It has a built-in
120mA(typ) current limit suitable for
powering ﬂash memory cards.
The LTC3456 features short-circuit protection for both the main and
core outputs. It also provides output
disconnect for all the outputs with
the exception of the VMAX output. The
Core, Main and Hot swap outputs are
14
The LTC3456 contains a proprietary
PowerPath control scheme that seamlessly switches over the system power
from a 2-AA battery to USB/wall Power
and vice versa. Figure 5 shows a simpliﬁed block diagram of the internal
power-path. The AC adapter and the
USB bus supply power to the switching regulators via the VEXT pin. The
LTC3456 contains a full featured USB
power manager to control the ﬂow of
power from the USB pin via the state
of the USBHP and SUSPEND pins.
The current through the USB pin
is accurately limited to 100mA or,
500mA depending on the state of the
USBHP pin. All USB functionality can
be disabled by pulling the SUSPEND
pin high.
DC-DC conversion is a particularly
challenging task when the 2 AA battery voltage (1.8V to 3.2V) must be
boosted to generate 3.3V output, and
the USB/wall power (4V to 5.5V) must
be stepped down to generate the same
voltage. The LTC3456 accomplishes
this task via the BOOST and BUCK2
converters. This is the most efﬁcient
AC ADAPTER
5V ±5%
10µF
VEXT
USB
POWER
CORE
1.8V
USB
4.7µF
10µH
LTC3456
USB
MANAGER
SW2_BK
SW1
10µH
BUCK 2
BUCK 1
10µF
10µF
VINT
3.3V
VINT
BOOST
BUCK 3
VBATT
SW2_BST
4.7µH
2AA CELL
4.7µF
Figure 5. LTC3456’s simpliﬁed block diagram showing internal PowerPath.
Linear Technology Magazine • September 2005
DESIGN FEATURES
VMAIN
100mV/DIV
(AC COUPLED)
IL2
200mA/DIV
VCORE
100mV/DIV
(AC COUPLED)
IL1
100mA/DIV
IL3
100mA/DIV
SUSPEND
5V/DIV
SUSPEND
5V/DIV
VUSB = 5V
VUSBHP = 5V
VBATT = 2.4V
ICORE = 100mA
IMAIN = 100mA
200µs/DIV
a. Core output transient waveforms
VPWRON
5V/DIV
VMAIN
5V/DIV
VCORE
2V/DIV
RESET
5V/DIV
VUSB = 5V
VUSBHP = 5V
VBATT = 2.4V
ICORE = 100mA
IMAIN = 100mA
200µs/DIV
b. Main output transient waveforms
Figure 6. USB and 2AA Battery power supply switchover waveforms for Figure 2’s circuit.
The USB power is disconnected when the Suspend pin is taken high. Main and core
outputs both exhibit less than ±2% total deviation at the time of switchover.
way of generating the 3.3V power
rail. The LTC3456 achieves efﬁciency
greater than 90% when generating
3.3V output from the battery or USB/
wall adapter. The core output (1.8V) is
generated via BUCK1 (USB/wall Powered) and BUCK3 (Battery Powered)
converters. The unique topology of
LTC3456 generates the 1.8V rail via
a single inductor resulting in a cost
and space saving. It achieves efﬁciency
greater than 92% when generating
the 1.8V output from the battery. The
various operational modes of LTC3456
are summarized in Table 1.
Portable devices are required to
seamlessly switch-over from the
battery power to USB or wall power
and vice versa to ensure smooth
system operation. As an example, a
user is playing music on a portable
MP3 player with the USB cable connected. If the USB cable is suddenly
yanked off the device, the user should
be able to continue listening to the
music without any interruption. The
LTC3456 makes it possible through
seamless switchover of system power.
Figure 6 shows USB and 2-AA battery
power supply switchover waveforms
for Figure 2’s circuit. The USB power
is unavailable when the Suspend
pin is taken from low to high. Main
and core outputs both exhibit less
than ±2% total deviation at the time
of switchover making the switchover
seamless to the processor core and
the peripheral circuitry.
Easy Interfacing with
a Microprocessor
The LTC3456 simpliﬁes the task of
interfacing with a micro-processor.
The PWRON, PWRKEY, PBSTAT and
RESET pins provide all the required
system information to the processor
and simplify power sequencing. The
PWRON, PWRKEY and PBSTAT keys
simplify the task of orderly poweringup and shutting down the IC. The
datasheet contains the timing diagram
and gives detailed information about
their operation.
The LTC3456 also contains poweron reset circuitry (accessed via pin
RESET) that is active during both
VBATT = 2.4V
ICORE = 10mA
IMAIN = 10mA
100ms/DIV
Figure 7. Power-up and power-down waveforms
for Figure 2’s circuit. Both VMAIN and VCORE
outputs are discharged to ground during
shutdown. Power-on reset (RESET) is held
low for a delay of 262ms after VCORE comes
into regulation.
power-up and shutdown. The power-on reset is required to hold the
processor in its reset state at power-up
and it must keep the processor from
starting operation until all system
power supplies have stabilized. The
LTC3456’s built in power-on reset
circuitry monitors both the VINT (3.3V)
and Core (1.8V) voltages and interfaces
 E
 S
 E
 T
 pin. The
to the processor via the R
RESET pin is held low during initial
power-up. When both the Main and
Core outputs come into regulation, a
reset delay timer gets activated. There
is a full 262ms timeout before RESET
is released and the processor is allowed
to come out of reset and begin operation. The timeout delay of 262ms gives
sufﬁcient time for the processor to
initialize the internal registers/RAMs.
During power-off the RESET pin is
again pulled low. This prevents the
micro-processor from entering into
any random operational modes.
Figure 7 shows the power-up and
power down waveforms for the circuit
of Figure 2 in battery powered mode.
The RESET circuitry works similarly
Table 1. Summary of LTC3456 PowerPath operational modes and features
AC ADAPTER
USB POWER
Highest priority for powering the IC
Medium priority for powering the IC Lowest priority for powering the IC
Battery loading < 2µA
Battery loading < 2µA
Internal soft-start circuitry limits current
drawn from the adapter at start-up
Battery inrush current regulated during power-up.
USB pin current accurately limited to
Additionally, internal soft-start limits input current
100mA or 500mA
at start-up.
AC adapter (min) voltage set via the
WALLFB Pin
USB (min) voltage set to 4V
Linear Technology Magazine • September 2005
2 AA CELLS
Burst Mode operation (User Selectable) conserves
battery energy
Battery (min) voltage indicator set via the AIN Pin
15
DESIGN FEATURES
2 AA
CELLS
+
C1
4.7µF
L2
4.7µH
SW2_BST
SW2_BK
VBATT
VINT
VMAIN
USBHP
USB
CONTROLLER
C10
22µF
L3
10µH
C9
1µF
MAIN OUTPUT
3.3V
100mA
100k
Q1
A0
SUSPEND
49.9k
USB POWER
(4.35V TO 5.5V)
C2
4.7µF
20k
1Ω
VEXT
EXT_PWR
C4
10µF
D1
AC
ADAPTER
(5V ±10%)
LTC3456
HSO
C7
1µF
L1
10µH
1k
11.3k
SW1
VEXT
100k
PWRKEY
C6
10µF
CORE OUTPUT
1.8V
200mA
80.6k
VMAX
PBSTAT RESET MODE PWRON PGND AGND
1Ω
220pF
FLASH MEMORY CARD
3.3V
50mA
FB1
WALLFB
C3
4.32k
4.7µF
C8
2.2µF
AIN
USB
LCD LOGIC BIAS
2.8V
10mA
C5
1µF
VMAX
(TO REAL-TIME CLOCK)
MICROCONTROLLER
L1, L3: MURATA LQH32CN100K53
C1, C6 TO C10: X5R OR X7R, 4V
L2: MURATA LQH32CN4R7M53
C2 TO C5: X5R OR X7R, 6.3V
D1: ON SEMICONDUCTOR MBRM120E Q1: PHILIPS MMBT3906
Figure 8. A 2-AA-cell-powered, complete power supply for GPS navigation system. Note that the uncommitted
gain block (Pins AIN and AO) is conﬁgured as an LDO controller to generate an auxiliary 2.8V output.
when battery or externally powered.
The RESET pin is held low for a delay of 262ms after VCORE comes into
regulation. When the IC is shut-down,
both VMAIN and VCORE outputs are disconnected from the input power and
discharged to ground This prevents
the outputs from being stuck in an
indeterminate logic-level state and
adversely affecting the operation of
the microprocessor. It also ensures
that the outputs rise in a predictable
fashion during power-up.
Voltage Monitoring
The LTC3456 has an on-chip gain
block that can be used for low-battery
detection, with the low battery trip
point set by two resistors (Figure 2)
at the AIN pin. The nominal voltage
at AIN is 0.8V. The AO pin is an opendrain logic output that sinks current
whenever the voltage at the pin AIN
falls below 0.8V. The gain block can
also be conﬁgured to drive an external
PNP or PMOS transistor to generate
an auxiliary voltage.
16
In addition, the LTC3456 has on
board voltage comparator circuitry
to detect the presence of USB or
wall power, with a status output at
the EXT_PWR pin. The open-drain
logic output of EXT_PWR is capable
of sinking up to 5mA, suitable for
driving an external LED. The on-board
voltage detectors continuously monitor the status of the USB voltage and
AC adapter voltage (via the WALLFB
Pin). Whenever the USB or, wall power
is available and in regulation, the
EXT_PWR pin is pulled low.
Portable GPS Navigator
Power Supply
Today’s portable GPS navigators run
off two AA batteries or an AC adapter
and come equipped with a USB bus
(for data transfer). Long battery life
and small system size are the key
requirements for the power supply.
The microprocessor used in GPS
navigators usually require at least two
different voltage supplies: typically
3.3V for the I/O circuitry and 1.5V or
1.8V for the processor core. The navigator might also require an auxiliary
2.8V supply voltage to bias the LCD
display controller IC.
Figure 8 shows a complete, compact and efﬁcient power supply for
a portable GPS navigator. The VMAIN
(ﬁxed 3.3V) provides power to the I/O
circuitry. The power supply for the
processor core, VCORE, is set at 1.8V
and can be adjusted by changing the
feedback resistor ratio. The 3.3V Hot
Swap output powers ﬂash memory
cards. The LTC3456 contains an uncommitted gain block (Pins AIN and
AO) that can be used as a low-battery
indicator or an LDO controller. The
circuit in Figure 8 shows the gain
block being used as an LDO with an
external PNP to generate an auxiliary
2.8V output voltage from the Main
output. The auxiliary 2.8V supply is
being used to power an LCD controller
IC. The VMAX output of the LTC3456
stays alive even in shutdown and is
used to supply power to a real-time
clock.
continued on page 19
Linear Technology Magazine • September 2005
DESIGN FEATURES
Micropower Precision Oscillator
Draws Only 60µA at 1MHz by Albert Huntington
Device Description
The LTC6906 is a part of Linear
Technology’s line of resistor controlled
SOT-23 oscillators. These resistor
controlled oscillators use a single
inexpensive external resistor to accurately set the oscillator frequency,
and there is a simple linear relationship between the resistor value and
the output frequency.
The LTC6906 uses an innovative
low power architecture with a master
oscillator running between 100kHz
and 1MHz. A three state, divide pin
is provided which can engage an internal divider to decrease the output
frequency by a factor of 1, 3 or 10
Linear Technology Magazine • September 2005
to provide a total frequency range of
10kHz to 1MHz. For increased accuracy at the lower end of the frequency
range with very low bias currents, a
guard pin is provided for the frequency
setting resistor input.
The master oscillator frequency is
set by an external resistor connected
between the SET pin and ground. The
LTC6906 maintains the SET pin at
approximately 650mV above ground,
with a tempco of –2.2mV/°C. The
master oscillator frequency is related
to the SET resistor by:
fMASTER
 100kΩ 
= 1MHz • 
,
 RSET 
and is related only to the resistance
on the SET pin, without regard to the
exact SET pin current or voltage.
Low Power Dissipation
The LTC6906 uses only 10µA when
running at 100kHz (Figure 2). There
are three components to this current
draw. A static bias current of about
5µA is used by the internal reference
and bias circuits. A variable bias
current of about 6 times the current
in the SET resistor is used to power
and bias the internal oscillator. A load
current related to the load capacitance, power supply voltage and load
resistance makes up the remainder of
the dissipation equation. An approxi90
CL = 5pF
TA = 25°C
80
70
+
V = 3.6V
60
50
V+ = 2.25V
40
30
20
10
0
0
200
400
600
800
FREQUENCY (kHz)
1000
NO DECOUPLING
CAPACITOR
NEEDED
LTC6906
2.25V TO 3.6V
÷10
÷3
÷1
1200
Figure 2. The LTC6906 has
extremely low power dissipation.
V+
OUT
GND
GRD
DIV
SET
10kHz TO 1MHz
RSET
100k TO 1M
Figure 1. The LTC6906 requires
only a single external resistor.
mate expression for the total supply
current is:
ISUPPLY = 5µA + 6 • ISET +

V+ 
V+ • FOUT • (CLOAD + 5pF ) +
.
2 • RLOAD 

Figure 3 shows the relative magnitudes of these three components
over the frequency range in the case
of a load capacitance of 5pF, with no
resistive load.
Note that power dissipated in the
load ranges from 25% to over 40% of
the total power from 100kHz to 1MHz
operation. Any lessening in the load
capacitance or resistance can have
dramatic effects on the load current
portion of the power supply dissipation. Power dissipation as low at 7µA
at 100kHz is achievable with light
output loading. Decreasing the power
supply voltage also reduces the power
dissipated into the load.
CONTRIBUTION TO POWER DISSIPATION (%)
Traditionally, electronic clocks use
quartz crystals, ceramic resonators,
or discrete R, L or C elements as a
timing reference, but each of these
designs has several drawbacks that
make them unsuitable for a variety
of applications. Quartz crystals and
ceramic resonators can be power-hungry, and their accuracy is subject to
environmental stress. Crystal oscillators have the additional disadvantage
of being susceptible to damage from
shock or vibration. RC oscillators have
poor jitter and accuracy, or require
expensive precision components. A
more robust, and compact alternative
to all of these is an all silicon clock,
such as the LTC6906 micropower,
resistor-controlled oscillator.
The LTC6906 is a monolithic silicon
oscillator with signiﬁcant size, power,
cost and environmental sensitivity
advantages over other oscillators,
and it requires only a single external
resistor to set the frequency over its
full range of 10kHz to 1MHz (Figure 1).
Its 0.65% accuracy and jitter as low as
0.03% make it an excellent choice for
precision applications, and the power
and size advantages let the LTC6906
ﬁt in designs where a crystal oscillator
could never go.
POWER SUPPLY CURRENT (µA)
Introduction
60
50
STATIC BIAS
40
30
SET CURRENT
20
LOAD CURRENT
10
0
V+ = 3V
TA = 25°C
0
200
800
400
600
SET RESISTOR (kΩ)
1000
1200
Figure 3. Percentage contributions to
power dissipation of static bias, set
current and load currents. Data was
taken at 3V, 25ºC ambient temperature.
17
DESIGN FEATURES
Choosing a SET Resistor
The choice of a SET resistor is guided
by the desired frequency output. The
part is speciﬁed for master oscillator
frequencies between 100kHz and
1MHz, with possible DIV ratios of 1,
3 and 10. These DIV ranges overlap,
and some frequencies have multiple
valid combinations of DIV and SET
resistor values. The lowest power dissipation for a given frequency is always
obtained by setting the SET resistor
as high as possible and DIV as low as
possible. Generating 100kHz using
DIV = 10 and RSET = 100kΩ dissipates
much more power than using DIV = 1
and RSET = 1000kΩ.
The following equation relates the
desired master oscillator frequency to
the RSET value:
RSET =
1MΩ 100kHz
;
•
N
FOUT
where N is the divider ratio chosen
of 1,3 or 10, RSET is the SET resistor
value and fOUT is the desired output
frequency. For example, see Table 1
for valid RSET values to generate a
100kHz output frequency at the three
DIV settings. It is apparent from the
table that, depending on the DIV pin
setting, the current for a particular
output frequency could vary by a factor of up to 4.5.
There are tradeoffs to choosing the
largest possible SET resistor and the
smallest possible value of DIV. Jitter
increases at the smaller DIV values,
and frequency accuracy may suffer
Table 1. RSET values for 100kHz
Divider
RSET Value
Setting N
18
Approximate
Supply
Current
1
1MΩ
10µA
3
333.33KΩ
20µA
10
100kΩ
45µA
80
additional leakage from long traces, it
is recommended that the SET resistor
be located as close as possible to the
SET pin, and on the same side of the
PC board as the LTC6906.
V+ = 2.7V
70
POWER SUPPLY CURRENT (µA)
Engaging the internal divider has
larger effects on power dissipation
where the load current is higher at
higher frequencies, but little effect
where the internal bias currents
dominate at lower master oscillator frequencies, as illustrated in Figure 4.
60
÷1
50
÷3
40
÷10
Long Term Drift of
Silicon Oscillators
30
20
10
0
0
200
400
600
800
RSET (Ω)
1000
1200
Figure 4. The LTC6906 power supply
current vs DIV pin setting. All data
taken at 3V supply, 5pF load.
more with high RSET values due to
leakage at the SET pin, especially at
higher temperatures.
Layout Considerations
The LTC6906 is capable of frequency
accuracy of <0.65% over the commercial temperature range, and for best
accuracy, care must be exercised to
limit board leakage around the RSET
pin. A 1GΩ parasitic resistance to
ground can change the frequency by
0.1%, and the same resistance to the
positive supply could increase that
to 0.3%. A guard pin which is weakly
driven to the same DC voltage as
the SET pin has been provided, and
the guard signal should be routed
completely around the SET pin, on
the same side of the PC board as the
device, and should have no soldermask
(see Figure 5 ).
The guard ring is not be necessary in
all applications, especially those with
lower values of SET resistor and excellent assembly practices. The majority
of board leakage problems occur due
to insufﬁcient cleaning of ﬂux from the
board or from sloppy assembly. With
perfectly clean assembly, the guard
ring is completely unnecessary.
The LTC6906 uses a switched
current to drive the SET resistor, so
there may be some noise visible on
the SET line. Although this noise does
not contribute to jitter on the output
signal, it can inﬂuence the frequency
accuracy in the presence of parasitic
capacitance on the SET pin. Because
of this sensitivity to parasitic capacitance and because of the danger of
Long-term stability of silicon oscilla r , which is
tors is speciﬁed in ppm/√kH
typical of other silicon devices such as
operational ampliﬁers and voltage references. Because drift in silicon-based
oscillators is generated primarily by
movement of ions in the silicon, most
of the drift is accomplished early in the
life of the device and the drift can be
expected to level off in the long term.
The ppm/√kHr unit models this time
variant decay. Crystal oscillators are
occasionally speciﬁed with drift measured in ppm/year. This measurement
models a different drift mechanism,
and the decay proﬁle is not the same.
A comparison of various drift rates
over a ﬁve year time period is shown
in Figure 6.
When calculating the amount of
drift to be expected, it is important to
consider the entire time in the calculation, because the relationship to time
is not linear. The drift for 5 years is
not 5 times the drift for one year. A
sample calculation for drift over 5 years
at 300ppm/√kHr is as follows:
5 years • 365.25 days/year • 24 hours/day
= 43,830 hours = 43.830kHr
43.830kHr = 6.62 kHr
6.62 kHr •
300ppm
= 0.198% over 5 years
kHr
LTC6906
1
OUT
V+
6
GRD
2
GND
3
DIV
RSET
5
SET
4
NO SOLDER
MASK OVER
THE GUARD RING
GUARD
RING
NO LEAKAGE
CURRENT
LEAKAGE
CURRENT
Figure 5. The GRD ring should be routed on
the same side of the PC board as the LTC6906,
and should have the solder mask removed.
Linear Technology Magazine • September 2005
DESIGN FEATURES
0.45
0.4
0.35
600ppm/√kHr
DRIFT (%)
0.3
0.25
0.2
300ppm/√kHr
0.15
0.1
100ppm/√kHr
0.05
0
0
20
40
60
OPERATING TIME (MONTHS)
80
Figure 6. Comparison of 5-year drift
at 100ppm/√kHr, 300ppm/√kHr
and 600 ppm/√kHr
Drift calculations assume that the
part is in continuous operation during
the entire time period of the calculation. The movements of ions which
results in drift is usually aided by
electric ﬁelds in the operating parts,
and drift is substantially lower if the
parts are not powered up during the
entire period of drift. Conservative
calculations would use a tenth of the
drift speciﬁcation for time when power
is not applied to the part.
Switching the DIV Pin
The DIV input pin on the LTC6906,
similar in many ways to the DIV pin
on other LTC silicon oscillators, is a
three state input, capable of resolving three different states: high, open
and low. Three state input pins allow
greater functionality in low pin-count
packages, and are compatible with the
tri-state outputs of many microcontrollers. Static conﬁguration is easily
accomplished by tying the pin to either
the positive supply or ground, or leaving it ﬂoating.
In the OPEN state, the DIV pin of
the LTC6906 is reasonably immune to
noise commonly found on PC boards,
but care should be taken to avoid routing a long ﬂoating trace off the pin, or
routing the pin driving that trace next
to a line with strong AC signals. The
noise immunity of the DIV pin can be
easily improved by adding a capacitor
to ground, or a series resistor of up to
100kΩ placed near the DIV pin.
In normal operation, the DIV pin
uses a small current of about 1µA to
pull the DIV pin voltage close to half of
the power supply voltage. Therefore, if
the pin is left open, any extra capacitance on the pin slows its settling to
the OPEN state.
Applications that use the DIV pin
to switch frequency in real time need
to take into account that, because it
is designed for low power operation,
the DIV pin buffer circuit is slow, with
delays up to around 12µs between
activation of the DIV pin and changes
in the output of the LTC6906. This
switching delay must be accounted
for in the application, or an external
frequency divider can be substituted
for the internal frequency divider in
order to decrease the frequency change
response time.
Manipulating the SET Pin
The LTC6906 can be conﬁgured in
applications where the SET resistor
needs to be changed for operation at
different frequencies. When changing
the SET resistor, best performance and
accuracy is obtained by placing the
switching mechanism between the set
resistor and GND, not between the set
resistor and the SET pin (see Figure 7).
LTC6906
V+
V+
OUT
GND
GRD
DIV
SET
1MHz TO 100kHz
100k
VMOD
0V TO 0.65V
1M
Figure 8. Modulating the SET pin
current through a resistor provides
greater immunity to noise coupling.
LTC6906
V+
V+
OUT
GND
GRD
DIV
SET
1MHz TO 100kHz
100k
1M
Figure 7. Switching in different SET resistors
The SET pin is sensitive to interference
from external capacitance or signals,
and isolation through the SET resistor
reduces this sensitivity.
The LTC6906 is not ideally suited
to current modulation through the
SET pin because in order to save
power, the voltage on the SET pin is
not regulated over temperature or
load. This results in the modulation
of the frequency being a function of
the set pin voltage as well as the set
pin current. The frequency can still be
modulated through the SET pin, but
the relationship between the modulation current or voltage and the output
frequency is not very accurate since
it depends on the poorly deﬁned SET
pin voltage.
The circuit in Figure 8 shows a
modulation method that results in
low jitter and stable performance.
By modulating the SET pin current
through a resistor, the effects of
parasitic capacitance on the initial
frequency accuracy are reduced.
Conclusion
The LTC6906 is a micropower oscillator with 0.65% accuracy and very
low jitter. Its small size, simple conﬁguration and extremely low power
consumption make it ideal for low
power applications driving microcontrollers, FPGAs and providing a clock
reference for battery powered devices.
Authors can be contacted
at (408) 432-1900
LTC3456, continued from page 16
Conclusion
The LTC3456 is a complete system
power management IC that seamlessly
manages power ﬂow between an AC
adapter, USB cable and 2-AA battery
supply. A host of features, including
Linear Technology Magazine • September 2005
an integrated USB power manager,
high efﬁciency DC-DC converters, a
Hot Swap controller and a Low-Battery
Indicator, are squeezed into a 4mm
× 4mm QFN package. The external
components count and overall system
cost are minimized. Simplicity, design
ﬂexibility, a high level of integration
and small size makes LTC3456 an ideal
choice for powering many portable
USB devices.
19
DESIGN FEATURES
New Standalone Linear
Li-Ion Battery Chargers
Introduction
Rechargeable batteries are commonly
used to power portable devices such as
digital cameras, PDAs, mobile phones
and MP3 players. A wall adapter is
the most common source of charging
power, but an increasing number of
applications are tapping into available USB power. The LTC4061 and
LTC4062 are speciﬁcally designed to
charge single-cell lithium-ion batteries
from either of these sources.
Both devices use constant current/constant voltage algorithms to
deliver up to 1A of charge current
(programmable) with a ﬁnal ﬂoat voltage accuracy of ±0.35%. They include
an internal P-channel power MOSFET
and thermal regulation circuitry with
no blocking diode or external sense
resistor required—the basic charger
circuit requires only two external
components.
The LTC4061 and LTC4062 include both programmable time and
programmable current based charge
termination schemes. The open-drain
charge status pin, CHRG, can be
programmed to indicate the state of
the battery charge according to the
needs of the application. The LTC4061
provides an AC Power open-drain
status pin, ACPR, to indicate that
enough voltage is present at the input
to charge a battery. Additional safety
features designed to maximize battery
lifetime and reliability include Negative
Temperature Coefﬁcient, NTC, battery
temperature sensing (LTC4061) and
the SmartStart™ charging algorithm,
which extends the lifetime of the battery by preventing unnecessary charge
cycles.
In the LTC4062, a low Iq precision
comparator replaces the NTC and
A
 C
 P
 R
 functions of the LTC4061. Without input power applied, the LTC4062
internal low power comparator can
function while drawing just 10µA
from the battery. With input power
applied, LTC4061 and LTC4062 can
20
5V WALL
ADAPTER
ICHG = 800mA
USB POWER
ICHG = 500mA
by Alfonso Centuori
LTC4061
OR
LTC4062
D1
MP1
VCC
BAT
C/5
PROG
SYSTEM
LOAD
IDET
+
3.3k
1k
MN1
2k
Li-Ion
BATTERY
1.24k
Figure 1. LTC4061 and LTC4062 USB/wall adapter power Li-Ion
charger conﬁguration using charge current termination
be put into shutdown mode to reduce
the supply current to a very low value
(20µA) and the battery drain current
to less than 2µA.
Internal thermal feedback regulates the charge current to maintain
a constant die temperature during
high power operation or high ambient
temperature conditions.
Programmability
The LTC4061 and LTC4062 provide a
great deal of design ﬂexibility including programmable charge current and
programmable total time termination
or programmable current termination.
The maximum charge current is programmed using a single resistor from
the PROG pin to ground. The charge
current out of the BAT pin can be
determined at any time by monitoring
the PROG pin voltage and applying
the following equation:
IBAT =
VPROG
• 1000
RPROG
A current detection threshold, IDETECT, is set by connecting a resistor,
RDETECT, from IDET to ground. This
threshold is used to change the state
of the CHRG pin indicating that a battery is nearly full. Alternatively, this
threshold can be used as the termination current threshold completing the
charge cycle.
When using total time termination,
the charge time is set by connecting
a capacitor, CTIMER, from TIMER to
ground.
The TIMER pin controls which
method of termination the LTC4061
and LTC4062 uses. Connecting an
external capacitor to the TIMER pin
activates an internal timer that stops
the charger after the programmed time
period has elapsed. Grounding the
TIMER pin and connecting a resistor
to the IDET pin causes the charge cycle
to terminate once the charge current
falls below a programmed threshold
(IDETECT). Connecting the TIMER pin
to the input supply disables internal
termination, allowing the charger to
be manually shut down through the
enable, EN, input.
USB Compatibility
The C
 /
 5
 pin on LTC4061 and LTC4062
provides an easy method to choose between the two different power modes:
high power and low power. A logic high
on the C/5 pin sets the charge current
to 100% of the current programmed
by the PROG pin resistor (up to 1A),
while a logic low on the C/5 pin sets
the current limit to 20% of the current
programmed by the PROG pin resistor. A weak pull down on the C/5 pin
defaults to the low power state.
 / 5
 pin provides great ﬂexibility
The C
in applications that can automatically
choose between wall adapter or USB
power, as shown in Figure 1. If wall
adapter is present and its voltage is
Linear Technology Magazine • September 2005
DESIGN FEATURES
VIN
4.3V TO 8V
VCC
1.24k
BAT > 3V BAT < 3V
100k
EN
C/5
BAT
TIMER
PROG
IDET
IN+
GND
1µF
0.1µF
OUT
LTC4062
619Ω
800mA
715k
+
348k
SINGLE CELL
Li-Ion BATTERY
Figure 2. LTC4062 Li-Ion charger conﬁguration using time termination and battery detection
above the VTH of MP1, the power is
applied through the diode D1 and the
power available through the USB port
is not used since MP1 is in open state.
MN1 is closed and the 3.3kΩ and 2kΩ
resistors are in parallel, setting the
total maximum charge current up to
800mA (160mA if C/5 is set low). If
wall adapter is not present, the USB
powers the charger; MN1 is open
leaving only the 2kΩ resistor to set
the charge current up to maximum
500mA. Through the C/5 pin it is
possible to set the charge current to
100mA or 500mA as necessary by USB
applications.
Avoiding Unnecessary
Charge Cycles
LTC4061 and LTC4062 are designed
to avoid unnecessary charge cycles
to extend the life of Li-Ion batteries.
When power is ﬁrst applied or when
exiting shutdown, the LTC4061 and
LTC4062 check the voltage on the BAT
pin to determine its initial state. If the
BAT pin voltage is below the recharge
threshold of 4.1V (which corresponds
to approximately 80%–90% battery
capacity), the LTC4061 and LTC4062
enter charge mode and begin a full
charge cycle. If the BAT pin is above
4.1V, the battery is nearly full and the
VIN
5V
1µF
1k
5
0.1µF
Fault Detection
and Reporting
LTC4061 has an NTC (Negative Temperature Coefﬁcient) input to qualify
charge based on the temperature of
the battery, as shown in Figure 3.
When the battery temperature is
above or below safe levels, charging is
suspended, the internal timer is frozen
and the CHRG pin output blinks with
a square wave at either the frequency
set with CTIMER (if in timer mode) or
1.5Hz if in current or user termination mode (TIMER connected to GND
or to the input supply). The frequency
of the blinking using CTIMER is set by
the following formula:
1k
10
VCC
CHRG
charger does not initiate a charge cycle
and enters standby mode. When in
standby mode, the chargers continuously monitor the BAT pin voltage.
When the BAT pin voltage drops below
4.1V, the charge cycle is automatically
restarted and the internal timer is reset
to half the programmed charge time
(if time termination is being used).
These features eliminate the need
for periodic charge cycle initiations,
ensure that the battery is always fully
charged and reduce the number of
unnecessary charge cycles, prolonging
battery life.
BAT
1
6
C/5
3
TIMER
9
4
ACPR
PROG
LTC4061
1.24k
2
8
NTC
IDET
GND
619Ω
11
800mA
VIN
100k
+
SINGLE CELL
Li-Ion BATTERY
100k
NTC
Figure 3. LTC4061 fully featured (using time termination)
Linear Technology Magazine • September 2005
f CHRG =
0.1µF
• 1.5Hz
CTIMER
This feature can be disabled by
grounding the NTC pin.
While only the LTC4061 has the
ability to report a temperature fault,
both parts have the ability to report a
bad battery. When the BAT pin voltage is below the 2.9V trickle charge
threshold (VTRIKL), the charge current
is reduced to 10% of the programmed
value. If the battery remains in trickle
charge for more than 25% of the total
programmed charge time, the chargers
terminate charging and report that
the battery is defective. LTC4061 and
LTC4062 report this fault by driving
the CHRG output with a square wave.
The duty cycle of this oscillation is 50%
and the frequency is set by CTIMER.
An LED driven by the CHRG output
exhibits a blinking pattern, indicating to the user that the battery needs
replacing. A bad battery fault can be
cleared by toggling the EN input or
removing and reapplying power to
VCC. The defective battery detection
feature is only available when time
termination is being used.
Feature Differences between
LTC4061 and LTC4062
In addition to the NTC feature,
LTC4061 has an ACPR power supply status indicator. When sufﬁcient
voltage is present on VCC to charge a
battery, this pin is pulled low with an
open-drain NMOS device. Otherwise,
the pin assumes a high impedance
state.
In place of the NTC and ACPR
functions, the LTC4062 includes an
undedicated, precision, low power
comparator. The comparator is powered from the BAT pin and consumes
just 10µA. The open drain output,
OUT, is capable of driving an LED. Possible uses for this comparator include
precision low battery detection as
shown in Figure 2 and user programmable input supply monitoring.
Conclusion
LTC4061 and 4062 are complete linear
Li-Ion battery chargers for wall adaptcontinued on page 23
21
DESIGN FEATURES
Monolithic Buck Regulator Operates
Down to 1.6V Input; Simpliﬁes Design
of 2-Cell NiCd/NiMH Supplies
by Gregg Castellucci
Features
Soft Start
To reduce inrush currents at startup, the LTC3409 offers a soft start
function, which linearly ramps up
the output voltage in about 1ms. For
instance, the average output current
required during soft start to charge a
10µF output capacitor to 1.8V in 1ms
is 18mA. The total output current is the
sum of the output capacitor charging
current and the current delivered to
the load as VOUT ramps up.
80
3.1VIN
70
100
2.5VIN
60
50
40
10
30
10
10
100
LOAD CURRENT (mA)
40
1k
R2
133k
10
30
POWER LOSS AT 2.5VIN
10
0
VOUT = 1.5V
0.1
1
10
100
LOAD CURRENT (mA)
1k
1
Figure 2. Efﬁciency vs load current for
the LTC3409 in pulse skip mode.
Switching Frequency
Synchronization
The LTC3409 offers an internally compensated phase locked loop (PLL) for
switching frequency synchronization
from 1MHz to 3MHz in addition to ﬁxed
frequencies of 1.7MHz and 2.6MHz.
This high frequency range allows the
use of surface mount capacitors and
inductors.
The sync pin has three states: high,
where the LTC3409 operates at a ﬁxed
2.6MHz switching frequency; low,
where the LTC3409 operates at a ﬁxed
1.7MHz switching frequency; or as the
input to the PLL, when the sync pin is
toggled at a frequency of at least 1MHz
for greater than 100µs. The SYNC pin
threshold for PLL input is nominally
CIN
4.7µF
2.5VIN
50
1
Figure 1. Efﬁciency vs load current for
the LTC3409 in Burst Mode operation.
VIN
1.6V TO 5.5V
100
60
20
VOUT = 1.5V
1
3.1VIN
70
POWER LOSS AT 2.5VIN
20
1.8VIN
POWER LOSS (mW)
1.8VIN
80
0
0.1
1k
90
EFFICIENCY (%)
90
100
1k
100
EFFICIENCY (%)
The LTC3409 is a monolithic synchronous step-down regulator designed
speciﬁcally to save space, improve
battery life and simplify the design of
2-cell-alkaline, NiCd and NiMH powered applications. It operates from a
wide input voltage range, 1.6 to 5.5V,
without the complexity and accompanying loss of efﬁciency of competing
devices that require boost circuitry for
generating internal voltages greater
than VIN.
Space-saving features include an
available 3mm × 3mm DFN package
and a high, 1MHz to 3MHz, operating
frequency, which allows the use of surface mount capacitors and inductors.
To extend battery life, the LTC3409 offers two operating modes that improve
light load efﬁciency, including Burst
Mode operation, which consumes only
65μA of supply current at no load, and
pulse skipping mode, which offers
low ripple currents for noise-sensitive
applications. Both modes consume
less than 1μA quiescent current in
shutdown.
The LTC3409 also features soft
start, which limits inrush current at
start-up.
POWER LOSS (mW)
Introduction
0.63V, thus allowing compatibility to
low voltage logic interfaces.
Efﬁciency-Improving
Operating Modes
The Mode pin has two states corresponding to two operating modes that
improve efﬁciency at light loads: high
for pulse skip mode, and low for Burst
Mode operation. In pulse skipping
mode, constant-frequency operation
is maintained at lower load currents
to decrease the output voltage ripple,
and therefore reduce the chance of
interference with audio circuitry. If
the load current is low enough, cycle
skipping eventually occurs to maintain
regulation. Efﬁciency in pulse skipping
mode is worse than Burst Mode op-
LTC3409
VFB
SYNC
GND
RUN
VIN
SW
VIN
MODE
R1
191k
L1
2.2µH
VIN FOR PULSE SKIP MODE
GND FOR BURST MODE
COUT
10µF
CER
VOUT
1.5V
0.6A
L1: SUMIDA CDRH2D18/LD
C1
10pF
Figure 3. 1.5V/600mA step down regulator
22
Linear Technology Magazine • September 2005
DESIGN FEATURES
across the internal P-channel MOSFET
and the inductor resistance.
VOUT
100mV/DIV
ILOAD
500mA/DIV
INDUCTOR
CURRENT
500mA/DIV
20µs/DIV
Figure 4. LTC3409 transient response to a
50mA–600mA load step, pulse skip mode
eration at light loads, but comparable
when the output load exceeds 50mA
(see Figure 1 & 2).
In Burst Mode operation, the
internal power MOSFETs operate
intermittently based on load demand.
Short burst cycles of normal switching are followed by longer idle periods
where the load current is supplied
by the output capacitor. During the
idle period, the power MOSFETs and
any unneeded circuitry are turned
off, reducing the quiescent current
to 65µA. At no load, the output capacitor discharges slowly through the
feedback resistors resulting in very low
frequency burst cycles that add only
a few µA to the supply current. Burst
Mode operation offers higher efﬁciency
at low output currents than pulse skip
mode, but when activated, Burst Mode
operation produces higher output
ripple than pulse skip mode.
Output Voltage Programmability
The LTC3409 output voltage is externally programmed with two resistors
to any value above the 0.613V internal reference voltage, and is capable
of 100% duty cycle. In dropout, the
output voltage is determined by the
input voltage minus the voltage drop
LTC4061/62, continued from page 21
ers and USB sources. They extend
lifetime of the batteries by avoiding unnecessary charge cycles. The LTC4061
and LTC4062’s versatility of charge
Fault Protection
The LTC3409 protects against output
over-voltage, output short-circuit and
power over-dissipation conditions.
When an over-voltage condition at
the output (>10% above nominal) is
sensed, the top MOSFET is turned off
until the fault is removed. If the output
is shorted to ground, reverse current in
the synchronous switch is monitored
to prevent inductor-current runaway.
If the synchronous switch current is
too high, the top MOSFET remains off
until the synchronous switch current
falls to a normal level.
When the junction temperature
reaches approximately 160°C, the
thermal protection circuit turns off the
power MOSFETs allowing the part to
cool. Normal operation resumes when
the die temperature drops to 150°C.
1.5V/600mA Step-Down
Regulator Using
Ceramic Capacitors
Figure 3 shows an application of the
LTC3409 using ceramic capacitors.
This particular design supplies up
to a 600mA load at 1.5V with an input supply between 1.8V and 3.1V.
Ceramic capacitors have the advantages of small size and low equivalent
series resistance (ESR), allowing very
low ripple voltages at both the input
and output. Because the LTC3409’s
control loop does not depend on the
output capacitor’s ESR for stable
operation, ceramic capacitors can be
used to achieve very low output ripple
and small circuit size. Figures 4 and 5
show the transient response to a 50mA
terminations, low quiescent current,
simplicity, high level of integration and
small size makes them an ideal choice
for many portable USB applications.
VOUT
100mV/DIV
ILOAD
500mA/DIV
INDUCTOR
CURRENT
500mA/DIV
20µs/DIV
Figure 5. LTC3409 Transient response to a
50mA–600mA load step, Burst Mode operation
to 600mA load step for the LTC3409 in
pulse skip mode, and burst mode.
Efﬁciency Considerations
Figure 1 shows the efﬁciency curves
for the LTC3409 (Burst Mode operation
enabled) at various supply voltages.
Burst Mode operation signiﬁcantly
lowers the quiescent current, resulting in high efﬁciencies even with
extremely light loads. Figure 2 shows
the efﬁciency curves for the LTC3409
(pulse skipping mode enabled) at various supply voltages. Pulse skipping
mode maintains constant-frequency
operation at lower load currents. This
necessarily increases the gate charge
losses and switching losses, which
impact efﬁciency at light loads. Efﬁciency is still comparable to Burst
Mode operation at higher loads.
Conclusion
The LTC3409 operates over a wide,
1.6V to 5.5V, input range, which allows it to operate from various power
sources, from a 5V AC wall adapter
to two series alkaline batteries. This
ﬂexible device is available in a 3mm
× 3mm DFN package and includes a
number of features to improve battery
life and save space.
LTC4061 and LTC4062 are available
in a small 10-lead low proﬁle 3mm x
3mm DFN package.
For more information on parts featured in this issue, see
http://www.linear.com/designtools
Linear Technology Magazine • September 2005
23
DESIGN FEATURES
Supply Tracking and Sequencing at
Point-of-Load: Easy Design without
the Drawbacks of MOSFETs by Scott Jackson
Introduction
Multi-voltage electronics systems are
often saddled with complex power
supply voltage tracking or sequencing requirements, which, if not met,
can result in system faults or even
permanent failures in the ﬁeld. The
design difﬁculties in meeting these
requirements are often compounded
in distributed-power architectures
where point-of-load (POL) DC/DC
converters are scattered across PC
board space, sometimes on different
board planes. The problem is that
power supply circuitry is often the
last circuitry to be designed into the
board, and it must be shoehorned
into whatever little board real estate
EARLY
VIN
6V
is left. Centralized sequencing-tracking solutions can work well, but when
no signiﬁcant contiguous space is left
on a board and the system speciﬁcations are in ﬂux, one wishes for a
simple, drop-in, ﬂexible option. That
wish can be fulﬁlled with a tracking
and sequencing solution that installs
at the POL, and is tiny and versatile
enough to be easily dropped into the
board without disrupting the rest of
the system design.
Wish Granted
The LTC2927 provides a simple and
versatile solution in a tiny footprint for
VIN
0.1µF
RONB
487k
VCC
ON
RONA
100k
both tracking and sequencing without
the drawbacks of series MOSFETs.
Each POL converter that must
be tracked or sequenced can have a
single LTC2927 placed at point-of-load
as shown in Figure 1. By selecting
a few resistors and a capacitor, the
supplies are conﬁgured to ramp-up
and ramp-down with a variety of voltage proﬁles. Figure 2 shows various
tracking and sequencing scenarios,
including concurrent voltage tracking
(Figure 2a), offset tracking (Figure 2b),
ratiometric tracking (Figure 2c), and
supply sequencing (Figure 2d).
Many voltage tracking solutions use
series MOSFETs, which adds an in-
VCC
RAMP
MASTER
ON
CRAMP
0.47µF VIN
LTC2927
RAMP
LTC2927
VIN
IN
RAMPBUF
RTB1
IN
LTC1628
FB
FB = 0.8V
OUT
5V
SLAVE 1
RAMPBUF
RTB3
TRACK
RFB1
105k
FB = 0.8V
OUT
1.8V
SLAVE 3
OUT
2.5V
SLAVE 4
VCC
ON
RAMP
LTC2927
RAMPBUF
RTB2
RFB3
26.1k
VIN
VCC
ON
RFA3
20k
GND
RTA3
VIN
FB
RAMP
LTC2927
VIN
VIN
IN
IN
LTC1628
LTC3728
FB = 0.8V
RAMPBUF
OUT
3.3V
SLAVE 2
RTB4
FB
FB = 0.8V
TRACK
TRACK
RTA2
FB
TRACK
RFA1
20k
GND
RTA1
LTC3728
GND
RFA2
20k
RFB2
63.4k
RTA4
GND
RFA4
20k
RFB4
63.4k
Figure 1. Typical tracking application
24
Linear Technology Magazine • September 2005
DESIGN FEATURES
MASTER
SLAVE1
SLAVE2
a. Coincident tracking
resistors conﬁgures the behavior of a
slave supply relative to a master signal.
The choice of resistors can cause a
slave supply to track the master signal
exactly or with a different ramp rate,
voltage offset, time delay, or combination of these.
A master signal is generated by
tying a capacitor from the RAMP pin
to ground or by supplying another
ramping signal to be tracked as shown
in Figure 1.
Examples
MASTER
SLAVE1
SLAVE2
a. Offset tracking
MASTER
SLAVE1
SLAVE2
c. Ratiometric tracking
MASTER
SLAVE1
SLAVE2
Consider a complex tracking system.
The schematic in Figure 1 uses an
LTC1628 dual synchronous stepdown converter to produce 5.0V and
3.3V supplies and an LTC3728 dual
synchronous step-down converter to
produce 2.5V and 1.8V supplies from
a 6.0V input. Four LTC2927s connected to the feedback nodes control
the ramp-up and ramp-down behavior of these supplies. An early VIN is
supplied to the devices to guarantee
correct operation prior to tracking the
supplies.
The speciﬁcation calls for the 5.0V
and 3.3V supplies to track coincidently
at ~20V/s, the 1.8V supply should
ramp up quickly at 100V/s after the
3.3V supply reaches 2.0V, and the 2.5V
supply should ramp up at the same
rate as the 1.8V supply, but delayed
by 20ms. The LTC2927 data sheet
(available at www.linear.com) includes
a 3-step design procedure that is followed for each supply. When using
that procedure, use the following for
equation (1) in Step 1, with a master
signal ramp-rate SM of 20V/s:
CRAMP =
d. Supply sequencing
Figure 2. Types of power supply voltage tracking
herent voltage drop, additional power
consumption, and extra PC board
real estate. Instead, the LTC2927
controls supplies by injecting current
directly into the feedback nodes, thus
controlling supply outputs without
series MOSFETs. Figure 3 shows the
simple “tracking cell” used to inject this
Linear Technology Magazine • September 2005
current. Furthermore, power supply
stability and transient response remain
unaffected because the injected current
from the LTC2927 offsets the output
voltage without altering the power
supply control loop dynamics.
Power supply tracking is straightforward with the LTC2927. A pair of
10µA
≈ 0.47µF
20 V s
5V and 3.3V Supply
Coincident Tracking
Because the master ramp rate is chosen to be equal to the desired ramp rate
of the 5V and 3.3V supplies, coincident
tracking is selected. If the feedback
voltage of the switching power supply
is 0.8V, as it is on the LTC1628, then
coincident tracking can be conﬁgured
by setting the tracking resistors equal
to the feedback resistors (veriﬁed by
25
DESIGN FEATURES
VCC
5V
3.3V
2.5V
1.8V
1V/DIV
+
MASTER
–
RTB
+
–
FB
TRACK
5
0.8V
DC/DC
5
RTA
2927 F05
RFA
50ms/DIV
SLAVE
FB OUT
Figure 4. Output proﬁle of
the circuit in Figure 1
RFB
Figure 3. Simpliﬁed tracking cell
following Step 2 of the 3-Step Design
Procedure),
From equation (2) of the 3-Step Design
Procedure:
RTB1 = RFB1 = 105kΩ
RTB2 = RFB2 = 63.4kΩ
From Equation (3) of the 3-Step Design
Procedure:
RTA1ʹ = RFA1 = 20kΩ
RTA2ʹ = RFA2 = 20kΩ
1.8V and 2.5V Supply Sequencing
The 1.8V supply ramps up 2V below
the 3.3V supply but at a ramp rate
of 100V/s. Set the slave ramp rate to
100V/s in equation (2) to ﬁnd R TB3:
21.3 V s
≈ 56.2kΩ
100 V s
Complete Step 2 by solving for R TA3ʹ
using equation (3).
RTA3′ = −10.755kΩ
Step 3 adjusts R TA3 for the desired
delay between the 3.3V supply and the
1.8V supply. An offset of 2V results in
a delay of ~100ms for the ramp rate
chosen.
RTA 3″ = 2.09kΩ
= R ′ || R
R
TA 3
TA 3
21.3 V s
≈ 93.1kΩ
100 V s
RTA4′ = −28.052kΩ
R ″ = 28.8kΩ
TA 4
The tracking proﬁle for this system
is shown in Figure 4.
Note that not every combination
of ramp-rates and delays is possible.
Small delays and large ratios of slave
ramp rate to master ramp rate may
result in solutions that require negative resistors. In such cases, either the
delay must be increased or the ratio
of slave ramp rate to the master ramp
rate must be reduced. In addition,
the chosen resistor values should not
require more than 1mA to ﬂow from
EARLY
VIN
6V
″
TA 3
The 2.5V supply has the same
ramp rate as the 1.8V supply, but
VTRACK
RTA1 || RTB1
= 0.05mA < 1mA
VTRACK
ITRACK 2 =
RTA 2 || RTB2
= 0.05mA < 1mA
VTRACK
=
RTA 3 || RTB3
= 0.45mA < 1mA
VTRACK
ITRACK 4 =
RTA 4 || RTB4
= 0.24mA < 1mA
The connections between each
LTC2927 shown in Figure 1 allow
extra control for each supply. With
this system, the 3.3V supply uses the
5V supply as its master signal. If for
some reason the 5V supply should collapse, the 3.3V supply follows it down.
Likewise, the 1.8V and 2.5V supplies
use the 3.3V supply as their master
signal and track it up and down.
0.1µF
RONB
487k
VCC
ON
RONA
100k
RAMP
RAMPBUF
MASTER
VIN
CRAMP
0.1µF
LTC2927
DMMT5551
IN
RFA/2
34k
VREF = 1.25V
LTC3462
FB
RFA/2
34k
TRACK
RTA1
26.1k
ITRACK1 =
ITRACK 3
RTA 4 = RTA 4′ || RTA 4″
RTB1
137k
≈ 2.61kΩ
26
RTB4 = 43.2kΩ
≈ 3.24kΩ
In the 3-step design procedure R TAʹ
represents the value of R TA that produces no delay or offset. Since no delay
is desired, R TA = R TAʹ, and Step 3 of the
Design procedure is unnecessary.
RTB3 = 26.1kΩ
is delayed another 20ms. Repeating
Step 2 and Step 3 for the 2.5V supply
results in:
the TRACK and FB pins. Therefore,
conﬁrm that less than 1mA ﬂows from
TRACK when VMASTER is at 0V.
GND
FB = 0V
OUT
–5V
SLAVE
RFB
274k
Figure 5. Supply tracking of GND referenced negative regulator
Linear Technology Magazine • September 2005
DESIGN FEATURES
0V
SLAVE
–VMASTER
0V
1V/DIV
1V/DIV
–5V SLAVE
–VMASTER
–5V SLAVE
–VMASTER
10ms/DIV
10ms/DIV
a. Tracking error due to current
mirror pull-down limitation
b. Tracking without current
mirror pull-down limitation
Figure 6. Output proﬁle of circuit of Figure 9
Negative Supply Tracking
It is possible to track negative voltage
regulators with the LTC2927. Figure 5
shows a tracking example using a
LT3462 inverting DC/DC converter to
produce a –5V supply. This converter
has a ground-based reference, which
allows current to be pulled from a
node where RFA has been divided in
two. To properly pull current from the
LT3462 FB network, a current mirror
must be placed between the LTC2927
and the converter. The 3-Step design
procedure remains the same with
minor modiﬁcations to equations (2)
and (3):
LTC3417, continued from page 8
ripple at VOUT1 and the current through
the inductor while the LTC3417 is
in Burst Mode operation. The ripple
voltage in this example was taken at
an ILOAD of 40mA and is only 15mVP–P.
The worst case output voltage ripple
occurs just before the part switches
from bursting to continuous mode,
which occurs at about 250mA. At his
point, the VOUT ripple can be as high
as 25mVP–P.
Figure 6 shows the VOUT1 ripple and
the current through the inductor when
the part is in Pulse Skipping Mode.
Notice that the current through the
inductor does go slightly negative, and
then produces some high frequency
components. The higher frequency
components are due to the switching
MOSFETS turning off. At lower currents, the part starts skipping pulses,
and thus produces some lower frequency components. In this case, the
voltage ripple does indeed show some
higher frequency components, yet the
ripple itself is at about 5mVP–P.
Figure 7 shows the voltage ripple
at VOUT1 and the inductor current
ESR generates a loop zero at 5kHz to
50kHz that is instrumental in giving
acceptable loop phase margin. Ceramic capacitors remain capacitive to
beyond 300kHz and usually resonate
with their ESL before ESR becomes
effective. Also, ceramic caps are prone
to temperature effects, requiring the
designer to check loop stability over
the operating temperature range. For
these reasons, great care must be
taken when using only ceramic input
and output capacitors. The LTC3417
helps solve loop stability problems
with its OPTI-LOOP phase compensation adjustment, allowing the use of
ceramic capacitors. For details, and a
process for optimizing compensation
components, see Linear Technology
Application Note 74 (AN76).
Although the LTC3417 is capable
of operating at 4MHz, the frequency in
this application is set for 1.5MHz by
connecting the FREQ pin to VIN.
Figures 5 through 7 show the trade
off between mode and VOUT ripple
noise. Figure 5 shows the voltage
Linear Technology Magazine • September 2005
RTB =
RTA′ =
RFB SM
•
2 SS
VTRACK
2VREF VTRACK
−
RFA
RTB
All other equations remain the
same.
Figure 6a shows the tracking proﬁle
of Figure 5 with a ramp rate of 100V/s.
VMASTER is positive, but the inverse is
shown for clarity. The –5V slave does
not pull all the way up to 0V at VMASTER = 0V. This is because the ground
referenced current mirror cannot pull
its output all the way to ground. If the
converter has a FB reference voltage
greater than 0V or if a negative supply
is available for the current mirror, the
error can be removed. The resulting
waveform is shown in Figure 6b.
Conclusion
The LTC2927 simpliﬁes power supply
tracking and sequencing by offering superior performance in a tiny
point-of-load area. A few resistors can
conﬁgure simple or complex supply
behaviors. Series MOSFETs are eliminated along with their parasitic voltage
drops and power consumption. The
LTC2927 offers all of these features
in a tiny 8-lead ThinSOT™ and 8-lead
(3mm × 2mm) DFN package.
when the part is in Forced Continuous mode. Notice that the current
through the inductor goes negative.
At no time, during Forced Continuous
doe the MOSFETS actually turn off,
they keep switching. Therefore, the
frequency component of the voltage
ripple stays constant at the operating
frequency. The voltage ripple therefore
looks constant and stays below 5mV
over all load currents.
Conclusion
The LTC3417 is a dual synchronous,
step-down, current mode, DC/DC converter designed to ﬁt in the tight spaces
afforded by today’s portable devices.
Switching MOSFETS are integrated
into the device, and high frequency
operation enables the use of small
sized components. It is also designed
with versatility in mind with external
components for loop compensation,
variable frequency operation and different operating modes to optimize
efﬁciency and noise.
27
DESIGN FEATURES
Versatile Controller Simpliﬁes High
Voltage DC/DC Converter Designs
by Tom Sheehan
Introduction
The LT3724 is a single-switch DC/DC
controller that can be used in medium
power step-down, step-up, inverting
and SEPIC converter topologies. It
offers simple solutions to regulating
system voltages at high efﬁciencies
over a wide input voltage range
(4V–60V) and wide load range.
VIN
30V TO
60V
CIN
68µF
1M
52.3k
200k
12
90
10
EFFICIENCY (%)
80
6
75
4
LOSS
2
VIN = 48V
0
10
Figure 2. 30V–60V to 24V, 75W DC/DC
converter efﬁciency and power loss
28
POWER LOSS (W)
8
1
LOAD CURRENT (A)
Burst_EN
120pF
4.99k
1000pF
0.22µF
Si7852
0.025Ω
SW
47µH
SS3H9
BAS19
VCC
+
VOUT
24V
75W
COUT
330µF
1µF
PGND
VC
SENSE+
SGND
SENSE–
93.1k
Figure 1. 30V–60V to 24V 75W DC/DC converter with input
UVLO and full time usage of on board high voltage regulator
exceeds the maximum current sense
threshold, pulse skipping occurs.
The LT3724 also incorporates a
programmable soft-start that controls
the slew rate of the converter output
voltage during startup to reduce supply inrush currents and output voltage
overshoot.
The gate driver is capable of driving large, low RDS(ON), standard level,
n-channel MOSFETS without the need
for a gate drive buffer. The driver uses a
bootstrapped supply rail which allows
it to drive either a high side MOSFET,
as found in buck converters, or a
low side MOSFET, as found in boost
converters.
On-Board Regulator
EFFICIENCY
85
680pF
TG
CSS
40.2k
95
0.1
LT3724
VFB
The LT3724 uses a 200kHz ﬁxed-frequency current-mode architecture. An
internal high voltage bias regulator
allows for simple startup and biasing,
and it can be back driven by the output
to increase supply efﬁciency and lower
power dissipation in the IC.
User selectable Burst Mode operation can maintain high efﬁciency over
a wide load range. In Burst Mode operation quiescent current is reduced
to under 100µA, making the LT3724
ideal for use in applications with
supply maintenance requirements
or light load and no-load conditions.
A precision shutdown pin threshold
allows for easy supply under voltage
lockout where quiescent currents are
reduced to less than 10µA.
Supply short circuit control is via an
external sense resistor, through which
the LT3724 continuously monitors inductor current. If the inductor current
65
BOOST
SHDN
LT3724 Features
70
VIN
The LT3724’s internal 8V linear
regulator eliminates the need for an
external regulator or a slow-charge
hysteretic start scheme. This regulator
generates the local supply that powers
the IC (VCC), from the converter input
supply, VIN.
The on-board regulator can operate the IC continuously, provided the
input voltage and/or FET gate charge
currents are low enough to avoid excessive power dissipation in the part.
Common practice uses the on board
regulator during startup and then
back drives the VCC pin above its 8V
regulated voltage during operation.
This reduces the power dissipation
in the IC and increases converter efﬁciency. The LT3724 has a start-up
requirement of VIN ≥ 7.5V. This assures
that the on-board regulator brings
the VCC pin above its undervoltage
lockout threshold of 6.25V. If VCC is
maintained using an external supply,
such as the converter output, the
LT3724 can continue to operate with
VIN as low as 4V.
Burst Mode Operation
The LT3724 employs low-current
Burst Mode operation to maximize ef-
VOUT
5V/DIV
2.5ms/DIV
Figure 3. 30V–60V to 24V, 75W DC/DC
converter output soft-start waveform
Linear Technology Magazine • September 2005
DESIGN FEATURES
VIN
4.5V TO 20V
60V TRANSIENT
CIN1
22µF
2x
25V
CIN2
25V
1µF RA
100k
1
3
C1
390pF
R3
200k
4
5
R2
130k
6
7
R1
14.7k
R4
47k
VIN
BOOST
TG
SHDN
CSS
VCC
BURST_EN
VFB
VC
8
R5
SGND
10k
C3
2200pF
C2
120pF
SW
LT3724
PGND
SENSE+
SENSE–
16
15
D1B
GSD2004
L1
20µH
C7
0.1µF
•
C5
22µF
3x
25V
M1
14
R6
10Ω
12
C4
1µF
25V
11
10
L1
20µH
C6
56pF
9
COUT1
330µF
16V
•
COUT2
22µF
25V
RSENSE
0.010Ω
R7
10Ω
VOUT
12V AT 25W
D2
D1A
GSD2004
C5, CIN1, COUT2 = TDKC453X7R1E226M
COUT1 = SANYO, OS-CON 16SVP330M
D2 = ON SEMI, MBRD660
L1 = COILCRAFT VERSAPAC VP5-D83
M1 = VISHAY, Si7852DP
D3
D1N4148
Figure 4. 15V to 12V 25W SEPIC DC/DC converter
The LT3724 SHDN pin is used for
precision shutdown in analog monitoring applications, as well as logic-level
controlled applications. Input supply undervoltage lockout for supply
1
CIN
33µF ×2
25V
C1
1500pF
0.1µF
25V
R3
4.7M
3
RCSS
200k
4
5
R2
187k
6
7
R1
10k
C2
120pF
R6
40.2k
8
VIN
BOOST
LT3724
SHDN
TG
SW
VIN = 20V
91
VIN = 15V
EFFICIENCY (%)
90
89
88
3.0
2.0
1.5
LOSS
VIN = 15V
86
85
3.5
2.5
VIN = 10V
87
Precision Shutdown Threshold
D1
BAV99
VIN
8V TO16V
92
1.0
POWER LOSS (W)
drive is provided for VCC, all VCC bias
currents originate from the VIN pin,
giving a total VIN current of 100µA.
An internal negative-excursion clamp
on the VC pin is set at 100mV below
the switch disable threshold, limiting
the negative excursion of the pin voltage and minimizing converter output
ripple during Burst Mode operation.
ﬁciency during light-load and no-load
conditions. Burst Mode is enabled by
shorting the BURST_EN pin to SGND
and can be disabled by shorting
BURST_EN to VFB.
When the required switch current,
sensed via the VC pin voltage, is below
15% of programmed current limit,
the Burst Mode function is engaged.
During the Burst interval, switching
ceases and all internal IC functions
are disabled, with the exception of
the VCC regulator, error ampliﬁer, and
bandgap reference. Current at the VIN
pin is reduced to 20µA and VCC current is reduced to 80µA. If no external
0.5
0.1
1
LOAD CURRENT (A)
0
10
Figure 5. 15V to 12V 25W SEPIC DC/DC
converter efﬁciency and power loss
sequencing or start-up over-current
protection is easily achieved by driving
RSENSE
0.015Ω
16
L1
10µH
D2
SBM540
15
14
VOUT
24V AT 50W
CSS
BURST_EN
VFB
VCC
PGND
VC
SENSE+
SGND
SENSE–
M1
12
11
10
C4
1µF
25V
9
C3
4700pF
COUT1
330µF
35V
COUT2
2.2µF x3
50V
CIN = SANYO, 25SVP33M
L1 = VISHAY, IHLP-5050FD-011
M1 = SILICONIX, Si7370DP
COUT1 = SANYO, 35CV330AXA
COUT2 = TDK, C4532X7R1H225K
D2 = DIODESINC., SBM540
RSENSE = IRC LRF2512-01-R0I5-F
Figure 6. 12V to 24V/50W boost converter
Linear Technology Magazine • September 2005
29
DESIGN FEATURES
Continuous High-Side
Inductor Current Sensing
The LT3724 uses a wide commonmode input range current sense
ampliﬁer that operates over a 0V to 36V
range. This current sense ampliﬁer
provides continuous inductor current
sensing via an external sense resistor.
This scheme does not require blanking intervals or a minimum on-time
to monitor current, an advantage over
schemes that sense switch current.
The sense ampliﬁer monitors inductor
current independent of switch state,
so the main switch is not enabled
unless the inductor current is below
the current that corresponds to the
VC pin voltage. This “turn-on” decision is performed at the start of each
cycle, and individual switch cycles are
skipped should an over-current condition occur. This eliminates many of the
potential over-current dangers caused
by minimum on-time requirements,
such as those that can occur during
startup, short-circuit, or abrupt input
transients.
Current Mode Control
The LT3724 uses current mode control
architecture enabling a higher supply
bandwidth thereby improving line and
load transient response. Current mode
control also requires fewer compensation components than voltage mode
control architectures, making it much
easier to compensate over all operating
conditions.
100
LOSS
VIN = 12V
EFFICIENCY (%)
98
VIN = 16V
96
VIN = 12V
94
92
VIN = 8V
90
88
3.0
2.5
2.0
1.5
1.0
POWER LOSS (W)
the SHDN pin with a resistor divider
from the VIN supply, such that the
divider output is 1.35V when VIN is
at the desired undervoltage lockout
rising threshold voltage. 120mV of
input hysteresis on the SHDN pin allows the IC to withstand almost 10%
of input supply droop before disabling
the converter. The SHDN pin has a
secondary threshold of 0.5V, below
which the IC operates in an ultralow-current shutdown mode with IVIN
< 10 µA. The shutdown function can
be disabled by connecting the SHDN
pin to VIN through a large value pullup resistor.
0.5
0.1
1
LOAD CURRENT (A)
0
10
Figure 7. 12V to 24V/50W boost converter
efﬁciency and power loss
Soft Start
The LT3724 employs an adaptive softstart scheme that directly controls the
rising rate of DC/DC converter output
voltage. Output voltage overshoot and
inrush current are well controlled with
this method. This rising rate of the
output voltage is programmed with a
capacitor connected to the converter
output, where:
2µA = COUT • (Desired ΔV/Δt)
The soft-start function maintains
this desired output rising rate up to
95% of the regulated output voltage.
The soft-start function is re-enabled
if the converter output droops below
70% regulation, so converter recovery is graceful from a short-duration
shutdown or an output short-circuit
condition.
Applications
The applications here present only a
small sample of what can be accomplished with the LT3724. See the data
sheet at www.linear.com for more,
including an inverting converter.
30V–60V to 24V, 75W
DC/DC Converter
Figure 1 shows a 30V–60V to 24V,
75W converter conﬁgured for supply
input undervoltage lockout and full
time usage of the onboard high voltage bias regulator. This application
demonstrates how a high efﬁciency
supply can be built inexpensively
and with fewer than 20 components.
Figure 2 shows the converter efﬁciency
and power loss vs load current.
Power for the IC is obtained directly from VIN through the LT3724’s
internal VCC regulator. VIN UVLO is
programmed via a resistor divider
to enable the LT3724 at 90% of the
speciﬁed low end of VIN range, or 27V,
which corresponds to the SHDN pin
voltage exceeding 1.35V. The SHDN
input has 120mV of hysteresis, so
the converter is disabled if VIN drops
below 24V.
The LT3724 soft-start function
controls the rising slew rate of the
output voltage at startup such that
the current through the soft start
capacitor is 2µA, so the converter
output rises at a controlled rate of
2µA/1nF, or 2V/mS. Figure 3 shows
the soft start ramp.
4V–60V to 12V, 2A
SEPIC Converter
In LT3724 converter applications with
output voltages in the 9V to 20V range,
back-feeding VCC from the converter
output is accomplished by connecting
a diode from the supply output to the
VCC pin. Figure 4 shows a 15V to 12V,
2A SEPIC converter conﬁgured to use
the 12V output voltage to back drive
VCC. This application also shows the
versatility of the LT3724 by conﬁguring it to control a SEPIC converter.
SEPIC converters are used where the
input voltage can be both less than or
greater than the output voltage, such
as a battery powered application.
In some DC/DC converter applications, the converter must withstand
or operate through intermittent input
voltage excursions. This is typical
of automotive battery-voltage applications, where high voltage line
transients such as load-dump or low
voltage transients such as startup
must be accommodated. This converter design is optimized to operate
with a 15V nominal input voltage but
can regulate the output voltage over a
wide input range of 4V to 60V. Figure 5
shows the converter efﬁciency and
power loss vs load current.
This converter also uses an external
current limit fold-back scheme. This
fold-back circuit consists of a single
1N4148 diode (D2) and a resistor (R5).
continued on page 34
30
Linear Technology Magazine • September 2005
DESIGN FEATURES
Multichannel, 3V and 5V, 16-Bit ADCs
Combine High Performance, Speed,
by Ringo Lee
Low Power and Small Size
Introduction
Many of today’s data acquisition applications require low-power and/or
remote sensing capabilities while occupying a minimum amount of board
space. Linear Technology introduces
a pin-compatible family of 12-bit and
16-bit A/D converters that meet these
requirements. The ﬂagship device of
this new family is the LTC1867. It
consists of an 8-channel analog input
multiplexer (MUX), a high performance
and lower-power 16-bit switched
capacitor A/D converter, a simple
serial I/O, and ﬁts in a small 16-pin
narrow SSOP package (5mm × 6mm
footprint).
Product Features
❑ Sample Rate: 200ksps (LTC1867);
175ksps (LTC1867L)
❑ 16-Bit No Missing Codes
❑ 8 Single-Ended or 4 Differential
Channels
❑ SPI/MICROWIRE™ Serial I/O
❑ On-Board or External Reference
❑ Low Power Operation: 1.3mA
(LTC1867); 0.75mA (LTC1867L)
❑ Automatic Nap and Sleep Modes
❑ 16-Pin Narrow SSOP Package
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7/COM
1
2
3
4
5
6
7
8
ANALOG
INPUT
MUX
9
REFCOMP
Figure 1. The simpliﬁed block diagram of the LTC1863 and LTC1867,
which include a programmable 8-channel input multiplexer, 16-bit
switched capacitor ADC and an on-board reference.
Table 1 identiﬁes the differences
between the four members of this
new family of high performance A/D
converters. The members are classiﬁed according to supply voltage and
resolution. The 16-bit LTC1867 and
the 12-bit LTC1863 operate on a single
5V supply while sampling at 200ksps.
The LTC1867L and the LTC1863L operate on a reduced 3V supply with the
same features of the LTC1867 and the
LTC1863, respectively. The LTC1867L
and LTC1863L key specifications
are guaranteed for 2.7V operation.
In addition, all four parts provide an
automatic Nap mode, Sleep mode,
unipolar and bipolar operation, and
an internal bandgap reference. Fig-
Table 1. LTC1867 Family Members
Part Number
LTC1867
LTC1863
LTC1867L
LTC1863L
VDD
5V
5V
3V
3V
Resolution
16-Bit
12-Bit
16-Bit
12-Bit
fSAMPLE
200ksps
200ksps
175ksps
175ksps
Input Ranges
0V–4V, ±2V
0V–4V, ±2V
0V–2.5V,
±1.25V
0V–2.5V,
±1.25V
Supply Current
1.3mA
1.3mA
0.75mA
0.75mA
INL (Unipolar)
±2LSB
±1LSB
±3LSB
±1LSB
INL (Bipolar)
±2.5LSB
±1LSB
±3LSB
±1LSB
No Missing
Codes
YES
YES
YES
YES
Linear Technology Magazine • September 2005
LTC1863/LTC1867 16 V
DD
15
GND
14
SDI
13
+ 12-/16-BIT
SDO
SERIAL
200ksps
12
PORT
–
SCK
ADC
11
CS/CONV
10
VREF
INTERNAL
2.5V REF
ure 1 shows a block diagram for the
LTC1867/LTC1863.
MUX Conﬁguration
The eight-channel analog input multiplexer can be selected either in 4
differential pairs, 8 single-ended, 7
single-ended channels versus COMMON pin (pin 8, CH7/COM acts as
COMMON MINUS) or combinations
thereof. These conﬁgurations are set
up by a 7-bit input word deﬁned in Table 2 through the SDI/SCK serial port.
The channel-to-channel matching for
offset and gain error are excellent. For
the LTC1867, the offset error match
and gain error match are both speciﬁed as ±2LSB (max). The crosstalk
between channels is typically better
than 110dB. Figure 2 illustrates the
ﬂexibility of the 8-channel MUX.
Unipolar and Bipolar Mode
These A/D converters can sample
the difference of positive input (+VIN)
and negative input (–VIN) at the same
instant either in unipolar or bipolar
modes depending on UNI bit of the
Input Word. For the LTC1867 and
LTC1863, the input ranges (i.e. +VIN
minus –VIN) are 0V to 4V in unipolar
mode and ±2V in bipolar mode. For
example, the +VIN can swing from 0V to
4V if –VIN is tied to Ground in unipolar
mode, and the +VIN can swing from
31
DESIGN FEATURES
8 Single-Ended
4 Differential
CH0
CH1
+ (–)
– (+) {
+ (–)
– (+) {
CH4
CH5
+
+
+
+
+
+
+
+
CH2
CH3
CH6
CH7/COM
GND (–)
Combinations
of Differential
and Single-Ended
7 Single-Ended
to CH7/COM
+
+
+
+
+
+
+
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7/COM (–)
+
–{
CH0
CH1
–
+{
+
+
+
+
CH2
CH3
1st Conversion
+
–{
+
–{
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7/COM
CH4
CH5
CH6
CH7/COM
GND (–)
2nd Conversion
CH2
CH3
–
+
{
CH2
CH3
CH4
CH5
+
+
{
CH4
CH5
CH7/COM
(UNUSED)
CH7/COM (–)
Figure 2. These examples show the ﬂexibility
of the LTC1867 8-channel input multiplexer.
0.5V to 4.5V if –VIN is tied to 2.5V in
bipolar mode.
Figure 3 depicts the unipolar and
bipolar operation for the LTC1867.
On the upper half of Figure 3, the
LTC1867 samples and digitizes a sinewave on Channel 1 (CH1) that swings
from 0V to 4V and measures a DC voltage of 2.5V on Channel 0 (CH0), both
in unipolar mode. On the lower half of
Figure 3, the ADC performs a bipolar
operation with CH0 versus CH1, and
Figure 3. The upper two traces show the digitized results from the LTC1867 operating in the
unipolar mode with signals applied to Channel 1 and Channel 0. The lower traces demonstrate
how the results are changed when LTC1867 is conﬁgured to operate in the bipolar mode with the
same signals applied to Channel 1 and Channel 0.
vice versa. Under these conditions the
differential input signals applied to the
ADC exceed the bipolar input range of
±2V. For example, in the case of CH0
minus CH1 the resulting digitized
input range will be –1.5V to 2.0V.
In both unipolar and bipolar operation, +VIN and –VIN are sampled
simultaneously; so common mode
noise on both inputs is rejected by
the truly differential inputs. In the
unipolar mode, the +VIN range of the
LTC1867 is from 0V to VDD and the
–VIN range is from 0V to VDD/2. Some
competing devices only allow the
minus input to reach a few hundred
millivolts to one volt above ground.
By comparison the LTC1867 offers
about twice the range on the minus
input. This makes the LTC1867 a
great choice for remote applications
SD OS S1
S0
COM
UNI
SLP
2. OS
= SINGLE/DIFFERENTIAL BIT
 N
 BIT
= ODD/SI G
3. S1
1. SD
4. S0
Simple Serial I/O
Allows Easy Isolated
or Remote Applications
The serial I/O employed by the
LTC1867 and its other family members is compatible with the industry
standard SPI/MICROWIRE interface.
The shift clock (SCK) synchronizes
the data transfer with each bit being
transmitted on the falling SCK edge
and captured on the rising SCK edge
in both the transmitting and receiving systems. After the conversion is
complete, the input word is shifted into
the ADC through the SDI/SCK ports,
and at the same time, data bits are
2.0
2.0
1.5
1.5
1.0
1.0
0.5
0.5
0
– 0.5
0
– 0.5
– 1.0
– 1.0
= ADDRESS SELECT BIT 1
– 1.5
– 1.5
= ADDRESS SELECT BIT 0
– 2.0
5. COM = CH7/COM CONFIGURATION BIT
 O
 L A
 R
 BIT
6. UNI = UNIPOLAR/BI P
7. SLP = SLEEP MODE BIT
32
INL (LSB)
Table 2. 7-Bit Input Word (SD = MSB)
where large common mode voltages
can be present. In the bipolar mode,
both +VIN and –VIN can accept signals
from 0V to VDD.
DNL (LSB)
+ (–)
– (+) {
+ (–)
– (+) {
0
16384
49152
32768
OUTPUT CODE
65536
Figure 4. The DC accuracy of the LTC1867
is shown in the INL curve. The accuracy is
achieved with capacitor matching, which is
very stable over time and temperature.
– 2.0
0
16384
32768
49152
OUTPUT CODE
65536
Figure 5. No missing codes and very accurate
capacitor matching found in the LTC1867 DAC
is illustrated by the DNL curve.
Linear Technology Magazine • September 2005
DESIGN FEATURES
2.5V
5V
R1
6k
10 VREF
BANDGAP
REFERENCE
VIN
2.2µF
4.096V
LT1019A-2.5
VOUT
9 REFCOMP
10
2.2µF
REFERENCE
AMP
10µF
9
+
R2
10µF
0.1µF
15
R3
15 GND
VREF
LTC1863/
LTC1867
REFCOMP
GND
LTC1863/LTC1867
Figure 6. LTC1863/LTC1867 reference circuit. The internal reference can be overdriven
by an external Reference, LT1019A-2.5 for better drift and/or accuracy performance.
Outstanding
DC and AC Performance
Operating from a 5V supply and sampling up to 200ksps, the LTC1867
2.0
SUPPLY CURRENT (mA)
released through the SDO/SCK ports.
The A/D converter starts to acquire
the analog input signals after reading
in the 7-bit Input Word. These ADCs
have an internally trimmed conversion clock which allows the sampling
frequency to approach DC without
affecting the conversion results. The
4-wire interface allows the LTC1867
and its siblings to ﬁt well with isolated
or remotely located applications.
delivers 16-bit, no missing codes
performance with an accurate INL
speciﬁcation of ±2LSB(max) in unipolar mode and ±2.5LSB(max) in
bipolar mode. Typical INL and DNL
plots for LTC1867 versus output code
are shown in Figures 4 and 5. This
performance is achieved with capacitor matching which is very stable over
time and temperature.
Along with outstanding DC performance, the LTC1867 also has very
good AC performance. The signal-tonoise ratio (SNR) is typically 89dB with
an input range of 4V and improves to
90.5dB when an external reference
VDD = 5V
1.5
1.0
0.5
0
1
10
100
fSAMPLE (ksps)
1000
Figure 7. The LTC1867 features an Automatic
Nap mode that cuts the power dissipation as
the sampling frequency is reduced.
1/fSCK
CS/CONV
tCONV
NAP MODE
NOT NEEDED FOR LTC1863
SCK
SDI
SDO
(LTC1863)
SDO
(LTC1867)
DON'T CARE
1
2
3
4
SD
0S
S1
S0
D9
D8
5
6
7
COM UNI
SLP
8
9
10
11
12
14
15
16
DON'T CARE
Hi-Z
MSB
D11 D10
D6
D5
D4
D3
D2
D1
D0
Hi-Z
MSB
D15 D14 D13 D12 D11 D10
D9
D8
D7
D6
D5
D4
D7
13
D3
D2
D1
D0
Figure 8. The Automatic Nap mode provides power reduction at reduced sample rates. This
feature is activated when CS/CONV remains high after the conversion is completed.
tACQ
CS/CONV
NOT NEEDED FOR LTC1863
SCK
SDI
DON'T CARE
1
2
3
4
5
6
7
COM UNI
SLP
8
9
10
11
12
SD
0S
S1
S0
MSB = D11
D10
D9
D8
D6
D5
D4
D3
D2
D1
D0
MSB = D15
D14 D13 D12 D11 D10
D9
D8
D7
D6
D5
D4
13
14
15
16
DON'T CARE
t CONV
SDO
(LTC1863)
Hi-Z
SDO
(LTC1867)
Hi-Z
D7
t CONV
D3
D2
D1
D0
Figure 9. The Automatic Nap mode is not activated if the CS/CONV pulse is shorter
than the conversion time. After the conversion the ADC remains powered up.
Linear Technology Magazine • September 2005
33
DESIGN FEATURES
voltage of 5V is applied to the REFCOMP pin (tie VREF pin to 0V to turn
off internal reference buffer).
Internal Reference
This family has an on-chip, temperature compensated, curvature
corrected, bandgap reference that
is factory trimmed to 2.5V for the
LTC1867 and the LTC1863, and 1.25V
for the LTC1867L and the LTC1863L.
The reference is internally connected
to a reference ampliﬁer and is available at VREF (Pin 10). A 6kΩ resistor
in the LTC1867 and the LTC1863
(3kΩ resistor for the LTC1867L and
the LTC1863L) is in series with the
output so that it can be easily overdriven by an external reference if better
drift and/or accuracy are required as
shown in Figure 6. The reference ampliﬁer gains the VREF voltage by 1.638
to 4.096V at REFCOMP (Pin 9). This
reference ampliﬁer compensation pin,
REFCOMP, must be bypassed with a
10µF ceramic or tantalum in parallel
with a 0.1µF ceramic for best noise
performance.
Low Power
Improves Battery Life
The LTC1867 and LTC1863 consume
only 1.3mA at a sampling rate of
200ksps. As the sampling frequency is
reduced, the converters use even less
LT3724, continued from page 30
The current limit fold-back circuit
provides additional control during the
ﬁrst few switch cycles of start-up, and
provides reduced short-circuit output
current. When the output is at ground,
the diode/resistor clamp the VC pin to
a value that corresponds to 25% of the
programmed maximum current. This
circuit is only active with VOUT close
to ground, and becomes completely
disabled once the output voltage rises
past about 10% regulation.
8V–16V to 24V, 50W
Boost Converter
The 24V Boost converter shown in
Figure 6 achieves over 95% conversion efﬁciency at 50W with less than
34
supply current with the automatic Nap
feature. For example, the parts draw
only 760µA and 200µA at sampling
frequencies of 100ksps and 10ksps,
respectively.
Automatic Nap mode is active when
the CS/CONV pulse width is longer
than the conversion time of the A/D
converter. The part goes to Nap mode
automatically right after a conversion
is completed and remains powered
down (the ADC draws 150µA in Nap
mode) as long as the CS/CONV stays
HIGH after conversion. The internal
reference, however, is still active and
provides a 2.5V output. In this way,
the LTC1867/LTC1863 requires no
additional wake up time before the next
conversion is started. Figure 7 shows
how the supply current is greatly reduced as the sample rate is decreased
when using this feature.
The ADCs can also go into the Sleep
mode during long inactive periods.
In sleep mode the internal reference
is also powered down, thus reducing
the draw to leakage currents of less
than 1μA. The wake up time out of
the sleep mode is determined by how
fast the reference bypass capacitors
can be charged. The wake up time
can be estimated with the values of
bypass capacitor on VREF and the
on-chip resistor between the internal
reference and VREF pin. For the 16-bit
20 components. Because this is a
boost converter, VCC is driven by the
input voltage to improve efﬁciency
and lower power dissipation. Figure 7
shows efﬁciency and power loss vs
load current.
Conclusion
The LT3724 is a feature packed DC/
DC controller that is versatile enough
to be conﬁgured to control multiple
converter topologies. It offers a simple
and inexpensive solution to regulating
system voltages at high efﬁciencies
over a wide input voltage range and
wide load range.
The integrated high voltage regulator facilitates true single-supply
operation. Burst Mode operation
LTC1867L, the wake up time can be
estimated as:
(resistor value) • (bypass capacitor value)
• (number of time constant needed to
settle to 16-bit accuracy)
or (3k • 2.2µF • 11).
Typically with bypass capacitors
of 2.2µF and 10µF on the VREF and
REFCOMP pins, this takes about 80ms
for LTC1867L. However, if an external
reference is used, the wake up time is
less than 10ms.
 S
 /CONV pulse is shorter
When the C
than the conversion time, the ADCs
stay powered up and the Automatic
Nap mode is not activated. In this
conﬁguration, the digital output, SDO,
becomes active after the conversion
is completed. Figures 8 and 9 show
the timing diagrams for the two cases
described.
Conclusion
The LTC1867 family packs an 8channel analog input multiplexer,
low-power A/D converter, serial I/O,
and an internal reference in a narrow
16-pin SSOP package. With outstanding DC and AC performance, and
equipped with the automatic Nap and
Sleep modes for power reduction, these
complete A/D converters can be used
in many space-sensitive as well as low
power applications.
improves efﬁciency during light load
and no load operation. The current
mode control architecture allows for
simple design of the power supply
control loop and excellent transient
response. Continuous current sensing
protects the supply from being damaged during an over current or short
circuit fault condition. The innovative
soft start function limits output voltage
overshoot and inrush current during
startup, brownout or short circuit
recovery.
for
the latest information
on LTC products,
visit
www.linear.com
Linear Technology Magazine • September 2005
DESIGN IDEAS
Linear Charger for Nickel Cadmium
or Nickel Metal Hydride Batteries
by Fran Hoffart
Minimizes Parts Count
Introduction
A Primer on Charging
NiCd/NiMH Batteries
The various methods for charging
Nickel based batteries are categorized
by speed: slow, quick and fast. The
simplest type of charger is a slow
charger, which applies a timer-controlled, relatively low charge current
for about 14 hours. This may be too
long for many portable applications.
For shorter charge times, quick and
fast chargers apply a constant current
while monitoring the battery voltage
and/or temperature to determine
when to terminate or stop the charge
cycle. Charge times typically range
anywhere from 3 to 4 hours (quick
charge) to around 0.75–1.5 hours
(fast charge).
Fast and quick chargers force a
constant charge current and allow
the battery voltage to rise to the level
it requires (within limits) to force this
current. During the charge cycle, the
charger measures the battery voltage
at regular intervals to determine when
to terminate the charge cycle. During
the charge cycle, the battery voltage
Linear Technology Magazine • September 2005
6.5
6.4
BATTERY VOLTAGE (V)
Although rechargeable Lithium Ion
and Lithium Polymer batteries have
lately been the battery of choice in high
performance portable products, the
old workhorse nickel cadmium (NiCd)
and the newer nickel metal hydride
(NiMH) batteries are still important
sources of portable power. Nickel
based batteries are robust, capable
of high discharge rates, good cycle life
and are relatively inexpensive. NiMH
batteries are replacing NiCd in many
applications because of the higher
capacity ratings (40 to 50% higher) and
because of environmental concerns of
the cadmium contained in NiCd cells.
This article covers NiCd/NiMH battery
charging basics, and introduces the
LTC4060 linear battery charger.
–∆V
6.3
6.2
BATTERY
VOLTAGE
6.1
6.0
CHARGE CURRENT
5.9
2A
1A
5.8
5.7
0
10
20
30 40 50 60
CHARGE TIME (MIN)
70
80
Figure 1. Typical charge proﬁle for a
2000mAHr NiMH 4-cell battery pack
charged at a 1C rate.
rises as it accepts charge (see Figure 1).
Near the end of the charge cycle, the
battery voltage begins to rise much
faster, reach a peak, then begins to fall.
When the battery voltage has dropped
a ﬁxed number of mV from the peak
DESIGN IDEAS
Linear Charger for Nickel Cadmium
or Nickel Metal Hydride Batteries
Minimizes Parts Count ......................35
Fran Hoffart
Determine the Real Internal
Resistance of a Battery .....................38
Jim Williams
Digitally Programmable Output
Monolithic Buck Regulator with
Built-In DAC and I2C Interface...........39
Earl Barber
Connect High Impedance Sensors
Directly to an Easy Drive™
Delta Sigma ADC ...............................40
Mark Thoren
Dual Switching Converter Provides
Two Outputs of Any Polarity .............41
Jesus Rosales
Micropower SOT-23 Inverting DC/DC
Converter Extends Battery Life in
Space-Sensitive Applications ............42
Eric Young
OLED Driver with Output Disconnect
and Automatic Burst Mode Improves
Standby Mode Efﬁciency ...................44
David Kim
(–ΔV), the battery is fully charged and
the charge cycle ends.
The battery has an internal safeguard against overcharge. While the
cell voltage is dropping from its peak,
the battery temperature and internal
pressure quickly rise. If fast charging
continues for a signiﬁcant amount of
time after full charge is reached, the
battery pressure seal may momentarily open causing gas to vent. This
is not necessarily catastrophic for the
battery, but when a cell vents, some
electrolyte is also released. If venting
occurs often, the cell will eventually
fail. In addition, after venting, the
seal may not close correctly and the
electrolyte can dry out.
Differences Between
NiCd and NiMH Batteries
The open circuit voltage (nominal 1.2V)
and the end-of-life voltage (0.9V to 1V)
are almost identical between the two
battery types, but the charging characteristics differ somewhat. All NiCd cells
can be trickle charged continuously,
but some NiMH cells cannot, and may
be damaged if the trickle charge is
continued after reaching full charge.
Also, the battery voltage proﬁle during
a fast charge cycle differs between the
two battery types.
For NiMH cells, the decrease in
battery voltage (–ΔV) after reaching a
peak is approximately one half that
of NiCd cells, thus making charge
termination based on –ΔV slightly
more difﬁcult. In addition, the NiMH
battery temperature rise during the
charge cycle is higher than NiCd,
and the higher temperature further
reduces the amount of –ΔV that occurs when full charge is reached. For
NiMH cells, –ΔV is almost non-existent at high temperatures for charge
rates less than C/2. (See sidebar for
the deﬁnition of “C”). Older batteries
35
DESIGN IDEAS
and cell mismatching further reduce
the already minute drops in battery
voltage.
Other differences between the two
chemistries include higher energy
density and greatly reduced voltage
depression or “memory effect” for NiMH
cells, although NiCd is still preferred
for high current drain applications.
NiCd cells also enjoy lower self-discharge characteristics, but NiMH
technology has room to improve in
this regard, while NiCd technology is
fairly mature.
The LTC4060 NiCd/NiMH
Battery Charger Controller
The LTC4060 is a complete NiCd or
NiMH linear battery charger controller
that provides a constant charge current and charge termination for fast
charging up to four series-connected
cells. Simple to use and requiring a
minimum of external components, the
IC drives an inexpensive external PNP
transistor to provide charge current.
The basic conﬁguration requires only
ﬁve external components, although
additional functions are included such
as, NTC input for battery temperature
qualiﬁcation, adjustable recharge voltage, status outputs capable of driving
an LED and shutdown and pause inputs. Selecting the battery chemistry
and the number of cells to charge is accomplished by strapping pins, and the
charge current is programmed using a
standard value resistor. With adequate
thermal management, charge current
up to 2A is possible, and even higher
current when using an external current sense resistor in parallel with the
internal sense resistor.
What’s Important When
Designing a Charger
Using the LTC4060?
Once the battery chemistry and
number of cells are set, one must
determine the correct charge current.
The LTC4060 is designed for fast
charging nickel-based batteries and
uses –ΔV as the charge termination
method. Battery temperature can
also be monitored to avoid excessive
battery temperature during charging,
and a safety timer shuts down the
36
About Battery Capacity and Charge Current
The correct charge current is always related to a battery’s capacity, or simply “C”. The letter “C” is a term used to indicate the manufacturers stated
battery discharge capacity, which is measured in mA • Hr. For example, a
2000mAHr rated battery can supply a 2000mA load for one hour before the
cell voltage drops to 0.9V or zero capacity. In the same example, charging
the same battery at a C/2 rate would mean charging at 1000mA (1A).
The correct charge current for fast charging NiCd or NiMH batteries is between approximately C/2 and 2C. This current level is needed for the cell
to exhibit the required –ΔV inﬂection that occurs when the cell reaches full
charge, although charging at 2C may cause excessive battery temperature
rise, especially with small, high capacity NiMH cells. Because of chemical
differences between the two battery chemistries, NiMH cells generate more
heat when fast charging.
near the battery to pause the charge
cycle allowing the battery to cool down
before resuming the charge cycle.
charger if charge termination does
not occur. The typical fast charge voltage proﬁle (the rapid rise, then drop
in battery voltage (–ΔV) near the end
of the charge cycle) only occurs at a
relatively high charge current. If the
charge current is too low, the battery
voltage does not produce the required
drop in battery voltage after reaching
a peak, which is necessary for the
LTC4060 to terminate the charge cycle.
At very low charge current, –ΔV does
not occur at all. On the other hand,
if the charge current is too high, the
battery may become excessively hot
requiring an NTC thermistor located
A Typical LTC4060
Charge Cycle
With sufﬁcient input voltage applied,
no battery connected and the correct charge current, charge time and
thermistor connections in place, the
charger’s output voltage is very close
to the input voltage. Connecting a
discharged battery to the charger pulls
down the charger’s output voltage
below 1.9 • VCELL (VCELL is the total
battery voltage divided by the number
OPTIONAL POWER PATH
COMPONENTS
VIN
9V TO 10V
LED
1µF
LED
1k
1k
4.42k
14
VCC
15
11
CHRG
NTC
LTC4060
13
3
ACPR SENSE
100k
DRIVE
5
OFF ON
INPUTS
ON PAUSE
6
68k
BAT
1
D2
Q2
THERMISTOR
10k
Q1
ICHRG
2
10µF
10V
PAUSE
+
4 CELLS
NiMH
2000mAhr
LTC4060
10
9
4
PROGRAMS
1.5nF
SAFETY TIMER
SHDN
SYSTEM
LOAD
D1
SEL1
PROG
7
158Ω
SEL0
TIMER ARCT
CHEM
12
1.5V
GND
16
8
TOTAL RESISTANCE
PROGRAMS CONSTANT
CHARGE CURRENT
1.18V
590Ω
PROGRAMS RECHARGE
THRESHOLD VOLTAGE
Q1: MJD210 (MUST BE SOLDERED TO GENEROUS AMOUNTS OF COPPER)
Q2: FDN306P
D1: B220A
D2: MBRM120LT3
Figure 2. 4-cell 2A NiMH battery charger with NTC thermistor and power path control
Linear Technology Magazine • September 2005
DESIGN IDEAS
of cells being charged) thus starting
a charge cycle.
If the battery temperature, as
measured by the NTC thermistor, is
outside a 5°C to 45°C window, the
charge cycle pauses and no charge
current ﬂows until an acceptable
temperature is reached. When the
battery temperature is within limits,
the battery voltage is measured and
must be below the max limit.
If VCELL is below 900mV, the charger begins a trickle charge of 20% of the
programmed charge current until the
voltage exceeds 900mV, at which point
the full programmed charge current
begins. Several hundred milliseconds
after the charge cycle begins, if the
battery voltage exceeds 1.95V, the
charge cycle stops. This overvoltage
condition usually means the battery
is defective requiring that the charger
be manually reset by replacing the
battery, toggling the shutdown pin, or
removing and reapplying power.
Once the programmed constant
charge current starts ﬂowing, a period
of time known as “hold-off-time” begins. This hold-off-time ranges from 4
minutes to 15 minutes depending on
the charge current and charge time
settings. During the hold off time,
the –ΔV termination is disabled to
prevent false charge termination. A
battery that is deeply discharged or
has not been charged recently may
exhibit a drop in battery voltage during
the early portion of the charge cycle,
which could be mistaken for a valid
–ΔV termination.
During the charge cycle, the battery voltage slowly rises. When the
battery approaches full charge, the
battery voltage begins to rise faster,
reaches a peak, then begins to drop.
The charger continuously samples the
For further information on any
of the devices mentioned in this
issue of Linear Technology, use
the reader service card or call the
LTC literature service number:
1-800-4-LINEAR
Ask for the pertinent data sheets
and Application Notes.
Linear Technology Magazine • September 2005
battery voltage every 15 to 40 seconds,
depending on charge current and timer
settings. If each sampled voltage reading is less than the previous reading,
for four consecutive readings, and the
total drop in battery voltage exceeds
8mV/cell for NiMH or 16mV/cell for
NiCd, the charge current stops, ending the charge cycle. The open drain
output pin “CHRG”, which was pulled
low during the charge cycle, now be-
The LTC4060 is a complete
NiCd or NiMH linear
battery charger controller
that provides a constant
charge current and charge
termination for fast
charging up to four seriesconnected cells.
comes high impedance.
A user programmable recharge
feature starts a new charge cycle if
the battery voltage drops below a set
voltage level because of self-discharge
or a load on the battery. Also, if a fully
charged battery greater than 1.3V is
connected to the charger, the –ΔV termination detection circuit is enabled
immediately with no hold-off-time,
thus shortening the charge cycle for
a battery that is already close to full
charge.
If the battery reaches approximately
55°C during the charge cycle, the
charger pauses until the temperature
drops to 45°C, then resumes charging
until the –ΔV termination ends the
charge cycle. If no –ΔV termination
takes place, the safety timer stops
the charge cycle. If the timer stops the
charge cycle, it is considered a fault
condition and the charger must be
reset by removing and replacing the
battery, toggling the SHDN pin or toggling the input power to the charger.
Watch Out for These Pitfalls
Don’t connect a load directly to the
battery when charging. The charge
current must remain relatively constant for the –ΔV charge termination
to be effective. Loads with changing
current levels result in small changes
in battery voltage which can trigger
a false –ΔV charge termination. For
applications that require a load, refer
to the power path components shown
in Figure 2. When the input voltage is
present, the load is powered from the
input supply through Schottky diode
D1 and the battery is isolated from
the load. Removing the input voltage
pulls the gate of Q2 low, turning it on
providing a low resistance current path
between the battery and the load.
Minimize the DC resistance between the charger and the battery.
Some battery holders have springs
and contacts that have excessive resistance. The increased resistance in
series with the battery can prevent a
charge cycle from starting because of
a battery overvoltage condition once
the full charge current begins. Poorly
constructed battery holders can also
produce false charge termination if
battery movement generates a premature –ΔV reading.
Unlike Lithium Ion cells that can be
paralleled for increased capacity, NiCd
or NiMH cells should not be paralleled,
especially when fast charging. Interaction between the cells prevents proper
charge termination. If more capacity
is required, select larger cells.
Not all NiCd or NiMH batteries
behave the same when charging.
Manufacturers differ in materials and
construction resulting in somewhat
different charge voltage proﬁles or
amount of heat generated. A battery
can be designed for general purpose
use, or optimized for high capacity,
fast charge rate, or high temperature
operation. Some batteries may not be
designed for high current (2C) charge
rates resulting in high cell temperature when charging. Also, most new
cells are not completely formed and
require some conditioning before they
reach their rated capacity. Conditioning consists of multiple charge and
discharge cycles.
A thermistor mounted near the battery pack, preferably making contact
with one or more of the cells, is highly
recommended, both as a safety feature
and to increase battery lifetime. Unlike
continued on page 43
37
DESIGN IDEAS
Determine the Real Internal
Resistance of a Battery
Introduction
An accurate measure of a battery’s true
internal resistance can reveal much
about its condition or its suitability
for an application, but measurement
is not as easy as hooking up a precision ohmmeter. Inherent capacitance
of a battery reduces the accuracy of
measurements taken with a common
AC-based milliohmmeters operating
in the kHz range. Figure 1, a very
simplistic battery model, shows a
resistive divider with a partial shunt
capacitive term. This capacitive term
introduces error in AC based measurement. Also, the battery’s unloaded
internal resistance can signiﬁcantly
differ from its loaded value. A realistic
determination of internal resistance
must be made under loaded conditions
at or near DC.
Figure 2’s circuit meets these
requirements, permitting accurate
internal resistance determination of
batteries up to 13V over a range of
0.001Ω to 1.000Ω. A1, Q1 and associated components form a closed loop
current sink which loads the battery
9V
IN
LTC1798
2.5V
10k
1%
1k
1%
USER
TERMINALS
Figure 1. Simplistic model shows battery
impedance terms include resistive and
capacitive elements. Capacitive component
corrupts AC based measurement attempts
to determine internal DC resistance. More
realistic results occur if battery voltage drop
is measured under known load.
via Q1’s drain. The 1N5821 provides
reverse battery protection.
The voltage across the 0.1Ω resistor, and hence the battery load, is
determined by A1’s “+” input voltage.
This potential is alternately switched,
via S1, between 0.110V and 0.010V
derived from the 2.5V reference
driven 3-resistor string. S1’s 0.5Hz
square wave switching drive comes
from the CD4040 frequency divider.
The result of this action is a 100mA
1N5821
SWITCHED
CURRENT SINK
2.5V
OUT
237k
1%
INTERNAL
IMPEDANCE
TERMS
S1
0.010V
6
9
9V
+
–
A1
LT1077
Q1
IRLR-024
+
–
0.001µF
1
S2
3
10k
MODULATOR
0.1Ω
1%
14 LTC6943
FORCE
≈0.5Hz SQUARE WAVE
FREQUENCY 10µF
DIVIDER AND
CHARGE PUMP
BAT85
OUTPUT
0 – 1.000V =
0 – 1.000Ω
A2
LTC1150
V–
–
4
1
≈1kHz SQUARE WAVE
200pF
16
2
S3
9V
15
SINGLE POINT GND AT 0.1Ω RESISTOR
# = LTC6943 PIN NUMBER
HEAT SINK Q1
+
1µF
BATTERY
UNDER
TEST
SENSE 1µF 10k
9V
9V
5
10k
SENSE
2.7k
biased, 1A, 0.5Hz square wave load
applied to the battery. The battery’s
internal resistance causes a 0.5Hz
amplitude modulated square wave to
appear at the Kelvin-sensed, S2-S3A2 synchronous demodulator. The
demodulator DC output is buffered by
chopper stabilized A2, which provides
the circuit output. A2’s internal 1
kHz clock, level shifted by Q2, drives
the CD4040 frequency divider. One
divider output supplies the 0.5Hz
square wave; a second 500Hz output
activates a charge pump, providing a
–7V potential to A2. This arrangement
allows A2’s output to swing all the way
to zero volts.
The circuit pulls 230µA from its
9V battery power supply, permitting
about 3000 hours battery life. Other
speciﬁcations include operation down
to 4V with less than 1mV (0.001Ω)
output variation, 3% accuracy and
battery-under-test range of 0.9V–13V.
Finally, note that battery discharge
current and repetition rate are easily
varied from the values given, permitting observation of battery resistance
under a variety of conditions.
SYNCRONOUS
DEMODULATOR
FORCE 1µF
7
0.110V
by Jim Williams
NC
+V CD4040
÷2048
CLK
÷2
R
13
9V
100k
200k
≈ –7V
Q2
2N3904
BAT85
10µF
Figure 2. Battery internal resistance is determined by repetitively stepping calibrated discharge current and reading resultant voltage drop.
S1 based modulator, clocked from frequency divider, combines with A1-Q1 switched current sink to generate stepped, 1A battery discharge
cycles. S2-S3-A2 synchronous demodulator extracts modulated voltage drop information, provides DC output calibrated in Ohms.
38
Linear Technology Magazine • September 2005
DESIGN IDEAS
Digitally Programmable Output
Monolithic Buck Regulator with
Built-In DAC and I2C Interface by Earl Barber
Introduction
Minimal Space
Figures 1 and 2 show the LTC3447
powered from a single Lithium-Ion
battery. To minimize critical board
real estate, only two ceramic capacitors, a single inductor, and a single
resistor are required for operation.
The LTC3447 regulator is internally
Linear Technology Magazine • September 2005
VIN
Li-Ion
2.5V TO 5.5V
I2C
PWREN
VCCD
10k
C2
4.7µF
CERAMIC
C3
4.7µF
10k
SDA
SCL
VIN
RUN
RPU1
20k
VIN
PGOOD
LTC3447
VCCD
SW
SDA
VOUT
SCL
FB
L1
3.3µH
C1
10µF
* 600mA AT VIN = 3V
VOUT
0.69V
TO 2.05V
AT 600mA*
R1
100k
R2
49.9k
GND
EXPOSED PADDLE
TO GROUND
Figure 1. A typical Li-Ion-to-programmable-output
application suitable for powering a microprocessor
Figure 2. Very little space is needed for a
programmable output solution. This circuit
includes optional start-up resistors and I2C
pull-up resistors.
compensated to further reduce the
need for additional external components. Optional external resistors
can be used when a start-up voltage
other than 1.38V is desired. When
using the optional start-up resistors,
the regulated voltage can be set to a
value outside of the normal DAC output range. Once the internal DAC is
changed, the regulated output voltage
remains between 0.69V and 2.05V.
The Efﬁciency Advantage
In an effort to extend battery life, many
µProcessors use a variety of power
modes. Reducing the supply voltage to
circuits not in use and then increasing the supply voltage when in use is
a common technique. The LTC3447
is designed to easily accomplish such
tasks thru its I2C interface. The 6-bit
DAC allows the designer to easily
change the supply voltage level from
0.69V to 2.05V. Another technique
is to simply reduce the current load
of the µprocessor. The LTC3447 can
sense light load conditions and enter
power-saving Burst Mode operation
for further power savings. Using
the LTC3447 to combine both these
techniques can greatly extend the
life of the battery. Figure 3 shows the
efﬁciency of the LTC3447. Notice the
jump in efﬁciency for light load currents when Burst Mode operation is
enabled.
100
BURST MODE OPERATION
90
80
EFFICIENCY (%)
A small package and high efﬁciency
make Linear Technology’s new
LTC3447 buck regulator an ideal
choice for portable devices using lithium-ion batteries. The tiny 3mm × 3mm
DFN package supplies up to 600mA
of current over an I2C programmable
output range of 0.69V to 2.05V. An
internal 6-bit DAC gives the designer
the ﬂexibility needed to control the
supply voltage for various modes of
operation.
LTC3447 Features Include:
❑ Soft Start — Limits peak inductor
current for a short period when
the regulator is ﬁrst enabled.
❑ Frequency Foldback — Reduces
oscillator frequency when the
regulated voltage is below the
desired operating point. This allows time for the inductor current
to discharge fully and prevent
thermal runaway.
❑ Over Temperature Protection
— Turns off internal switching
FETs until the operating temperature returns to a normal level.
❑ Power Good Reporting — Reports
when the regulated voltage is
either under-voltage or over-voltage. This feature can be disabled
via the I2C interface.
❑ Burst Mode Operation — Improves efﬁciency at light loads to
improve battery life. When a light
load is detected, the regulator enters a highly efﬁcient mode whose
quiescent current is 33µA.
70
DAC = 2.05V
60
50
40
30
DAC = 0.69V
20
10
0
VIN = 4.2V
0.1
1
10
100
LOAD CURRENT (mA)
1k
Figure 3. Efﬁciency of the circuit in Figure 1
39
DESIGN IDEAS
Connect High Impedance Sensors
Directly to an Easy Drive
Delta Sigma ADC
by Mark Thoren
Delta Sigma ADCs are accurate and
have high noise immunity, making
them ideal for directly measuring many
types of sensors. Nevertheless, input
sampling currents can overwhelm
high source impedances or low-bandwidth, micropower signal conditioning
circuits. The LTC2480 family of Delta
Sigma converters solves this problem
by balancing the input currents, thus
simplifying or eliminating the need for
signal conditioning circuits.
A common application for a delta
sigma ADC is thermistor measurement. Figure 1 shows the LTC2480
connections for direct measurement of
thermistors up to 100kΩ. Data I/O is
through a standard SPI interface, and
the sampling current in each input is
approximately
5V
C8
1µF
IIN+ = IIN–
5
IN+
IN–
3
REF
2
VCC
CS
SCK
LTC2480 SDO
SDI
GND GND FO
8
6
9
7
1
10
11
1.6Ω due to the slight shift in common
mode voltage; far less than the 1% error
of the reference resistors themselves.
No ampliﬁer is required, making this
an ideal solution in micropower applications.
The LTC2480 family of
Delta Sigma converters
balances input sampling
currents, thus simplifying
or eliminating the need for
signal conditioning circuits.
It may be necessary to ground one
side of the sensor to reduce noise
pickup or simplify wiring if the sensor
is remote. The varying common mode
5V
5V
102k
+
10k–100k
0.1µF
4-WIRE
SPI INTERFACE
Figure 1. LTC2480 connections
 VREF 

 – VCM
V + +V –
 2 
IN
, where VCM = IN
2
1.5MΩ
or about 1.67μA when VREF is 5V and
both inputs are grounded.
Figure 2 shows how to balance the
thermistor such that the ADC input
current is minimized. If the two reference resistors are exactly equal, the
input current is exactly zero and no
errors result. If the reference resistors
have a 1% tolerance, the maximum
error in the measured resistance is
4
C7
0.1µF
LT1494
1k
–
TO IN+
0.1µF
1k
TO IN–
0.1µF
5V
R1
51.1k
C4
0.1µF
C3
0.1µF
TO IN+
IIN+ = 0
R3
10k–100k
TO IN–
R4
51.1k
IIN– = 0
Figure 2. Centered sensor
voltage produces a 3.5kΩ full-scale error in the measured resistance if this
circuit is used without buffering.
Figure 3 shows how to interface
a very low power, low bandwidth op
amp to the LTC2480. The LT1494
has excellent DC specs for an ampliﬁer with 1.5µA supply current—the
maximum offset voltage is 150µV and
the open loop gain is 100,000—but its
2kHz bandwidth makes it unsuitable
for driving conventional delta sigma
ADCs. Adding a 1kΩ, 0.1µF ﬁlter
solves this problem by providing a
charge reservoir that supplies the
LTC2480’s instantaneous sampling
current, while the 1kΩ resistor isolates
the capacitive load from the LT1494.
Don’t try this with an ordinary delta
sigma ADC—the sampling current
from ADCs with speciﬁcations similar
to the LTC2480 family would result in
a 1.4mV offset and a 0.69mV full-scale
error in the circuit shown in Figure 3.
The LTC2480’s balanced input current allows these errors to be easily
cancelled by placing an identical ﬁlter
at IN–.
for
the latest information
on LTC products,
visit
www.linear.com
Figure 3. Grounded, buffered sensor
40
Linear Technology Magazine • September 2005
DESIGN IDEAS
Dual Switching Converter Provides
Two Outputs of Any Polarity by Jesus Rosales
Op amps, CCD imagers, LCDs,
medical diagnostic equipment and
a host of other circuits require dual
power supplies. Both supplies may be
positive, both negative, or they may
be opposite polarity. Dual supplies
are implemented in a variety of ways,
including using two converters, tapping off the switch with capacitors and
diodes, or using multi-winding transformers. Each of these solutions adds
unnecessary cost, size and complexity,
especially now that there is a simple,
single device solution available with
the LT3471.
Simplicity and versatility are
two features that are often at odds
in the world of switching regulators—simplicity usually means fewer
components; versatility more. The
LT3471 turns this idea on its head by
offering a versatile feature set with a
minimal number of components in a
dual-output converter.
Consider the circuit in Figure 1.
With a Li Ion battery input, this converter provides two positive supplies:
a 5V output and a 12V output. Only
one input ﬁlter capacitor is required.
Both outputs are independently controlled, but the same clock runs both
switchers to avoid any interference
(beat frequency) between them.
The circuit in Figure 2 provides two
output voltages of opposite polarity:
a 15V output and a –8V output. This
design uses very small inductors,
measuring 3.2mm by 2.5mm with a
proﬁle of 1.7mm. In applications where
circuit size is critical, this circuit can
ﬁt in an area as small as 0.16 square
inches. The saturation current in the
inductors used dictates the maximum
available current to the values shown
in the schematic. If more current is
needed, the LT3471 can deliver by
choosing bigger inductors. As in Figure 1, the circuit in Figure 2 provides
two outputs requiring only one input
ﬁlter capacitor.
Linear Technology Magazine • September 2005
VIN
VIN
2.6V TO 4.2V
C1
4.7µF
6.3V
GND
C3
VOUT1
5V(1) 22µF
6.3V
R1
20k
1%
L2
6.8µH
10
C7
100pF
SW1
1
R2
4.99k
1%
L1
3.3µH
D1
MBRM120E
VIN
D2
MBRM120E
8
6
VIN
SW2
FB1N
FB2N
5
9
C1: TDK C2012X5R0J475MT
C3: TDK C2012X5R0J226M
C4: TDK C3225X7R1C106M
L1: SUMIDA CR43-3R3
L2: SUMIDA CR43-6R8
SHDN/SS1
GND
SHDN/SS2
7
FB1P VREF FB2P
11
2
3
4
CSS1
0.33µF
C2
0.1µF
C4
V
10µF OUT2
12V(1)
16V
R4
4.99k
1%
VIN
LT3471
RSS1
4.7k
R3
54.9k
1%
C8
220pF
RSS2
4.7k
VOUT2 (12V)
VOUT1 (5V)
630mA IF VIN = 3.3V 210mA IF VIN = 3.3V
560mA IF VIN = 3.0V 190mA IF VIN = 3.0V
425mA IF VIN = 2.6V 145mA IF VIN = 2.6V
CSS2
0.33µF
Figure 1. A 1.2MHz, Li-ion to 5V and 12V Converter
VIN
VIN
2.6V TO 4.2V
GND
VOUT1
16V
55mA C3
4.7µF
16V
GND
R1
71.5k
1%
C1
4.7µF
6.3V
C7
100pF
SW1
VIN
RSS1
4.7k
C1: TDK C2012X5R0J475MT
C2: TAIYO YUDEN EMK212BJ105MG
C3, C4: KEMET C1206C106K4PAC
L1, L3: MURATA LQH32CN4R7M53
L2: MURATA LQH32CN2R2M53
8
6
VIN
SW2
FB2N
5
FB1N
LT3471
9
C2
1µF
16V
L2
2.2µH
10
1
R2
4.99k
1%
L1
4.7µH
D1
MBRM120E
SHDN/SS2
7
L3
4.7µH
D2
MBRM120E
RSS2
4.7k
CSS2
0.33µF
SHDN/SS1
GND
11
FB1P VREF
2
3
CSS1
0.33µF
C2
0.1µF
FB2P
C8
56pF
R3
121k
1%
VIN
VOUT2
–8V
110mA
C4
10µF
10V
GND
4
R4
15k
1%
Figure 2. A 1.2MHz, Li-ion to 15V at 55mA and –8V at 110mA converter
In situations where inrush current
is a problem, the LT3471 contains a
capacitor-programmable soft start
feature that allows the designer to
individually program the ramp rate of
each output. Figure 3 shows a typical
layout.
for
the latest information
on LTC products,
visit
www.linear.com
Figure 3. A compact, 1.2MHz dual output
converter for Li-ion to 5V and 12V.
41
DESIGN IDEAS
Micropower SOT-23 Inverting DC/DC
Converter Extends Battery Life in
Space-Sensitive Applications by Eric Young
Introduction
Low power negative bias supplies are
commonly used in many of today’s
handheld products for imaging and
display modules. As is the case with
all portable products, small size and
efﬁcient operation are top requirements. The LT3483 steps in to ﬁll
this need with a minimum footprint,
low proﬁle negative supply that yields
long battery life.
One of the strengths of the LT3483
is its versatility. It can be used for
inverting step-up (boost) or for inverting step-down applications. It features
an input range of 2.5V to 16V, so
the device works well with a range of
battery types and conﬁgurations. Its
internal 40V switch and integrated 40V
Schottky rectiﬁer allow it to generate
output voltages to ±38V.
The LT3483 also includes features
to maximize battery run time. At no
load conditions, the device draws only
36µA of battery current to maintain the
output or outputs in regulation. The
current limited ﬁxed off-time control
scheme delivers power-on-demand
to achieve high efﬁciency operation
over a wide range of load currents.
A shutdown pin disables the device
and reduces quiescent current to
less than 1µA. During operation, the
shutdown pin draws only 5µA from a
3.6V supply.
Simple, Accurate
Negative Regulators
It is easy to set the negative output
voltage of the LT3483 inverting converter, because there is no need to
compensate for a variable FB input
bias current. The FB input is referenced to GND and features a 2%
accurate, temperature compensated
10µA reference source current. An
external resistor between FB and the
negative output sets the output voltage
within 2% plus resistor tolerances. By
42
VIN
3.6V
C2
0.22µF
L1
10µH
D1
10Ω
SW
VIN
D
LT3483
C1
4.7µF
5pF
SHDN
FB
GND
VOUT
–8V
25mA
R1
806k
C3
2.2µF
C1: MURATA GRM219R61A475KE34B
C2: TAIYO YUDEN LMK107BJ224
C3: MURATA GRM219R61C225KA88B
D1: PHILIPS PMEG2005EB
L1: MURATA LQH2MCN100K02L
Figure 1. Low proﬁle 3.6V to –8V inverting converter in 50mm2
eliminating the untrimmed current
sourced by the negative FB (NFB) pin
of other inverting regulators, calculation of the feedback resistor has been
simpliﬁed as follows:
VOUT = –10µA • R
The resulting output voltage is
therefore more accurate and less current ﬂows into the feedback divider.
–8V at 25mA in 50mm2
The 200mA current limit and 300ns
off-time allow the use of tiny low proﬁle
inductors and low proﬁle ceramic capacitors. Figure 1 shows a bias supply
useful for CCD and OLED applications
that produces a well regulated –8V
supply at up to 25mA from 3.6V using as little as 50mm2 of board space.
All components in this design are less
than 1mm in height. While the inductor
usually dominates board area and pro-
VOUT
20mV/DIV
ISW
100mA/DIV
2µs/DIV
Figure 2. Output ripple of the 3.6V
to –8V inverter at 15mA is 40mV.
ﬁle, regulators built with the LT3483
are able to take maximum advantage
of smaller size low proﬁle inductors
such as the Murata LQH2 series—with
minor reductions in output power
capability and efﬁciency. The resulting converter circuits squeeze the
most performance out of the smallest
spaces. The –8V converter also uses
low proﬁle ceramic capacitors for the
input, output and ﬂying capacitors.
Figure 2 shows that the output voltage
ripple of the –8V converter at 15mA
is about 40mV. Switching at no load,
the converter circuit draws 79µA from
the battery.
±15V at 5mA in 90mm2
A typical LCD application requires
both a positive and a negative voltage.
The LT3483 circuit shown in Figure 3
provides a 15V and a –15V output
from a 3.6V supply. The –15V rail is
generated using an inverting charge
pump and is regulated through the
feedback resistor. The quasi-regulated
15V is generated by a charge pump
tapped from the switch node.
With this circuit conﬁguration, it
is straightforward to generate other
complementary pairs of regulated
outputs besides ±15V. All components
in this design are low proﬁle (<1mm)
and the circuit makes efﬁcient use of
Linear Technology Magazine • September 2005
DESIGN IDEAS
C2
0.1µF
D2
VIN
C4
1µF
D1
SW
VOUT1
–15V
5mA
D
LT3483
CIN
4.7µF
6.3V
1.5M
SHDN
VOUT2
15V
5mA
70
EFFICIENCY (%)
C1
0.1µF
L1
10µF
VIN
2.7V TO
4.2V
75
D3
60
C3
1µF
FB
55
0.01
GND
Figure 3. Compact, high efﬁciency LCD power supply yields 5mA at ±15V in less than 90mm2.
lead-acid battery as a standby power
supply. Figure 5 shows the LT3483
in a robust step-down backup supply, which uses a small, low proﬁle
1:1 coupled inductor in an inverting
ﬂy-back conﬁguration.
One of the strengths of the
LT3483 is its versatility. It
can be used for inverting
step-up or for inverting stepdown applications.
–5V at 100mA from 12V
The LT3483 can also regulate a negative output voltage that is smaller in
magnitude than the input voltage,
useful for systems that employ a 12V
The LT3483 can be always active,
ready if primary power fails, drawing
only 45µA from the battery. If the
normal power supply fails, the backup
L1A
10µH
•
The LT3483 provides a very compact,
low quiescent current step-up or stepdown DC/DC inverter solution for a
wide input voltage range of 2.5V to 16V
and outputs to –38V, making it a good
ﬁt for a variety of portable or battery
backup applications.
75
70
•
SW
C1
4.7µF
Conclusion
L1B
10µH
VIN
VOUT
–5V
D
LT3483
22pF
511k
FB
SHDN
GND
C2
10µF
C1: TAIYO YUDEN EMK316BJ475ML
C2: TAIYO YUDEN JMK316BJ106ML
L1A, L1B: WURTH 744876100
Figure 5. –5V step-up/step-down converter
LTC4060, continued from page 37
Lithium Ion batteries that exhibit very
little temperature rise when charging,
Nickel based batteries will heat up during the charge cycle, especially NiMH
batteries. Minimizing the length of time
the battery is exposed to elevated temperature extends battery lifetime.
Linear Technology Magazine • September 2005
10
circuit using the LT3483 immediately
delivers up to 100mA at –5V. In the dual
inductor conﬁguration, the LT3483
is also protected against grounding
of the output. A proprietary current
limiting scheme prevents the buildup
of excessive switching currents which
could cause damage to components in
the power path.
Conclusion
NiCd and NiMH batteries are ideal
sources of rechargeable power for
many portable products and backup
applications. This article helps to
familiarize the user with some of the
charging characteristics of nickel
EFFICIENCY (%)
VIN
2.5V TO 16V
1
0.1
LOAD CURRENT (mA)
Figure 4. Efﬁciency of ±15V
converter at VIN = 3.6V.
C1, C2: TAIYO YUDEN UMK212BJ104KG
C3, C4: TAIYO YUDEN TMK212BJ105KG
D1, D2, D3: PHILIPS PMEG2005EB
L1: MURATA LQH2MCN100
board space. The additional components for the charge pump are offset
by the internal feedback resistor and
integrated Schottky diode. During
shutdown, both the positive and negative loads are disconnected from the
battery, which increases battery run
time. Switching with no load, the circuit draws 135µA from a 3.6V supply.
The advantages offered by this circuit
are low quiescent current, low parts
count, and small footprint.
65
VIN = 5V
65
VIN = 12V
60
55
0.1
1
10
LOAD CURRENT (mA)
100
Figure 6. Efﬁciency of –5V
step-up/step-down converter
based batteries and how they apply
to the LTC4060 charger. Charging
NiCd and NiMH batteries correctly and
safely is simpliﬁed using the LTC4060
linear battery charger controller.
43
DESIGN IDEAS
OLED Driver with Output Disconnect
and Automatic Burst Mode Improves
by David Kim
Standby Mode Efﬁciency
Introduction
The LT3473 is a micropower step-up
DC/DC converter designed to drive
self-luminous organic light-emitting
diode or OLED display. The LT3473
features an integrated output disconnect switch that prevents leakage
from OLED display during standby
or shutdown mode by isolating the
OLED display from input supply. The
LT3473 also features an automatic
burst mode, which allows outputs to
be regulated with minimum circuit
operation to maximize the light load
efﬁciency. The small DFN package
(3mm × 3mm), high level of integration
and constant switching frequency yield
a tiny solution size.
Some OLED applications require
intermediate bias voltages for enhancing the display refreshing rate, such
as in passive matrix OLED displays,
the LT3473A includes two NPN transistors for generating two additional
bias voltages.
PGOOD
44
SW
CIN
4.7µF
LT3473
CAP
VIN
20k
2M
FB
SHDN
GND
100nF
100k
CINT
0.47µF
CIN: TAIYO YUDEN JMK107BJ475
CINT: TAIYO YUDEN GMK212BJ474
COUT: TAIYO YUDEN GMK325BJ225
L1: TOKO A915AY-6R8M (TYPE D53LC)
Figure 1. Space saving OLED bias supply
voltage with any lower value, allowing
full control of the output voltage.
Power Good indication is also integrated in the LT3473 solution. When
The small DFN package
(3mm × 3mm), high level of
integration and constant
switching frequency yield a
tiny OLED solution size.
OLED Bias Supply
the output voltage reaches 90% of the
set value, the open collector logic at
power good pin starts to sink current
to indicate that output voltage has
reached power good stage.
80
VIN = 3.6V
75
EFFICIENCY (%)
Figure 1 shows an OLED bias supply
solution ideally suited for handheld
and other battery powered portable devices. Using the internal 1A switch, the
circuit is capable of delivering 25V at
up to 80mA from a Li- Ion cell (3~4.2V)
input. An LT3473-based OLED bias
supply requires only a few external
components, because most functions
are integrated into the part, including:
the power switch, a Schottky diode,
the output disconnect switch, a reference override, power good indication
and optimized loop compensation.
As a result, the circuit in Figure 1
only requires less than 50mm2 of PC
board space.
For simple dimming or contrast
adjustment, the LT3473 solution has
an auxiliary reference input (CTRL
pin) that allows the user to override
the internal 1.25V feedback reference
CTRL
L1 6.8µH
VIN
3V TO 4.2V
VOUT
25V
COUT 80mA
2.2µF
OUT
VOUT = 15V
VOUT = 25V
VOUT = 20V
70
Conclusion
The LT3473 offers highly integrated
solution for OLED bias applications.
Key features include output disconnect, automatic burst mode for light
load, reference override and auxiliary
intermediate bias output (LT3473A) for
overall efﬁciency and performance of
OLED bias applications. The resulting
small circuit size and high efﬁciency
makes LT3473 an ideal solution for
space-conscious portable device applications such as cellular phones and
other handheld applications.
For further information on any
of the devices mentioned in this
issue of Linear Technology, use
the reader service card or call the
LTC literature service number:
65
60
55
The efﬁciency shown in Figure 2
remains above 75% over a load current
range of 20mA–70mA. Figure 2 also
shows efﬁciency for 15V and 20V outputs. For a 15V output, the circuit is
capable of generating 100mA of output
current above 75% efﬁciency.
1-800-4-LINEAR
0
20
60
80
40
LOAD CURRENT IO (mA)
100
Figure 2. Efﬁciency of OLED bias supply
Ask for the pertinent data sheets
and Application Notes.
Linear Technology Magazine • September 2005
NEW DEVICE CAMEOS
New Device Cameos
Dual, Low Voltage, No RSENSE
Synchronous Controller in
4mm x 4mm QFN
The LTC3736-2 is the newest addition to the LTC3736 family of dual,
2-Phase, No R SENSE ™ step-down
DC/DC synchronous controllers with
output tracking. The LTC3736-2 uses
a constant frequency, peak current
mode architecture with MOSFET VDS
sensing, improving efﬁciency and
eliminating the need for current sense
resistors. The two controllers are
operated 180° out of phase, reducing
the input ripple current and required
input capacitance. The LTC3736-2
can regulate output voltages as low as
0.6V from input supplies from 2.75V
to 9.8V, making it ideal for 3.3V and
5V rails, as well as many different
battery chemistries.
Compared to the original LTC3736,
the LTC3736-2 features a 0.6V ±1%
voltage reference (over the full temperature range from –40°C to +85°C).
It also has a higher adjustable current
limit range to allow it to be used with
a larger selection of power MOSFETs.
The LTC3736-2 can be conﬁgured
for pulse skipping (discontinuous) or
forced continuous operation at light
loads.
The LTC3736-2 keeps many of
the other popular features of the
LTC3736 family, including selectable
frequency up to 750kHz, synchronizable frequency from 250kHz to 850kHz
using its PLL, a power good output
voltage indicator, supply tracking,
internal soft-start, optional external
soft-start, and 9µA shutdown current.
The LTC3736-2 is available in the tiny
4mm × 4mm QFN package.
Dual/Quad 18MHz,
Low Noise, Rail-to-Rail,
CMOS Op Amps
The LTC6241 and LTC6242 are dual
and quad low noise, low offset, rail-torail output, unity gain stable CMOS op
amps that feature 1pA of input bias
current. The 0.1Hz to10Hz noise of
only 550nVP–P, along with an offset of
just 125µV make them uncommon
Linear Technology Magazine • September 2005
among traditional CMOS op amps.
Additionally, noise is guaranteed to
be less than 10nV/√Hz at 1kHz. An
18MHz gain bandwidth, and10V/µs
slew rate, along with the wide supply
range and low input capacitance, make
them perfect for use as fast signal
processing ampliﬁers.
These op amps have an output stage
that swings within 30mV of either
supply rail to maximize the signal
dynamic range in low supply applications. The input common mode range
extends to the negative supply. They
are fully speciﬁed on 3V and 5V, and
an HV version guarantees operation
on supplies up to ±5.5V. The LTC6241
is available in the 8-pin SO, and for
compact designs it is packaged in the
tiny dual ﬁne pitch leadless (DFN)
package. The LTC6242 is available in
the 16-PinSSOP as well as the 5mm ×
3mm DFN package.
Inductorless Multi-Mode High
Current LED Charge Pump
Delivers 700mA of Current
with Over 90% Efﬁciency
The LTC3215 is a fractional charge
pump, high current white LED driver
that delivers up to 700mA of LED current. Its high efﬁciency multi-mode
architecture automatically switches
between 1x, 1.5x or 2x boost modes
by monitoring the voltage across the
LED current source and switching
modes only when ILED dropout is
detected. This enables the LTC3215
to maximize efﬁciency (up to 92%)
throughout the entire Li-Ion operating
range. A 900kHz switching frequency
and a low external parts count (two
ﬂying capacitors, two programming
resistors and two bypass capacitors
at VIN and CPO) provide a very tiny
footprint and cost-effective solution,
ideally suited for video and ﬂash applications in camera phones and other
portable lighting applications.
Built-in soft-start circuitry prevents
excessive inrush current during startup. High switching frequency enables
the use of small external capacitors.
LED current is programmed with an
external resistor. The LED is disconnected from VIN during shutdown.
An ultralow dropout current source
maintains accurate LED current at
very low ILED voltages. Automatic
mode switching optimizes efﬁciency
by monitoring the voltage across the
LED current source and switching
modes only when ILED dropout is
detected. The LTC3215 is available
in a low proﬁle 3mm × 3mm 10-Lead
DFN package.
1.5GHz to 2.4GHz
High Linearity Direct
Quadrature Modulator
The LT5528 is a direct I/Q modulator
designed for high performance wireless applications, including wireless
infrastructure. It allows direct modulation of an RF signal using differential
baseband I and Q signals. It supports
PHS, GSM, EDGE, TD-SCDMA, CDMA,
CDMA2000,W-CDMA and other systems. It may also be conﬁgured as an
image reject up-converting mixer, by
applying 90° phase-shifted signals to
the I and Q inputs. The I/Q baseband
inputs consist of voltage-to-current
converters that in turn drive doublebalanced mixers. The outputs of these
mixers are summed and applied to an
on-chip RF transformer, which converts the differential mixer signals to
a 50Ω single-ended output. The four
balanced I and Q baseband input ports
are intended for DC coupling from a
source with a common-mode voltage
level of about 0.5V. The LO path consists of an LO buffer with single-ended
input, and precision quadrature generators that produce the LO drive for
the mixers. The supply voltage range
is 4.5V to 5.25V.
Low Power 125Msps 14-bit
Wideband ADC Improves
Base Station Power Efﬁciency
and Battery Life for Portable
Electronics
The LTC2255 is a 125Msps, 14-bit
Analog to Digital Converter (ADC) that
features excellent AC performance and
extremely low power. Outperforming
45
NEW DEVICE CAMEOS
its nearest 14-bit competitor, the
LTC2255 consumes 49% less power
at just 395mW, signiﬁcantly lowering
the power budget and thermal considerations required for multiple channel
devices. This provides a signiﬁcant
advantage in applications where efﬁciency and cooling is critical, such as
satellite receivers, wireless base stations and portable electronics. As part
of an extensive pin-compatible family,
the LTC2255 comes in a conveniently
small 5mm × 5mm QFN package with
integrated bypass capacitors, requiring only a small number of tiny external
components. The LTC2255 eliminates
the need for large and costly decou-
pling capacitors, affording the smallest
solution size available, which eases
PCB space constraints and allows for
more compact, cost effective designs.
With its small dimensions, low power
and reduced external component
requirement, designers can easily ﬁt
four LTC2255 ADCs where just one
competing solution would ﬁt.
The LTC2255 is well placed to
meet the needs of 3G and emerging
4G technologies, WiMAX and other
wideband wireless applications where
high performance ADCs play a key role
in handling the demands of increasing network trafﬁc. For wireless base
station system designers, reduced
LTC3780, continued from page 6
Short Circuit Protection
The basic boost regulator topology
provides no short circuit protection.
When the output is pulled low, a large
current can ﬂow from the input to the
output. Nevertheless, if an overload
causes an LTC3780 circuit to reach
current limit, current foldback prevents the overload from carrying over
to the input without shutting down
the whole circuit. Figure 14 shows
the result: the converter is forced into
buck mode, and the duty cycle of SW2
is reduced such that the voltage at
SW2 continues to swing between VIN
and ground. VIN remains solid since
current foldback limits the inductor
current, so the supply only draws
100mA more than it would without
any load. A power good output opendrain logic output signals whether the
output voltage is in or out of regulation.
When the overload disappears, the
output voltage returns to its normal
value—there is no need to shut down
and restart the LTC3780.
Keep Alive
LTC3780 applications often work
alongside related subsystems requiring very little current. The LTC3780’s
46
100
75kΩ resistor, the output varies from
12V down to 6V. The proper external
voltage can be approximated from the
equation VOUT = 13.28V – 1.5(VREF).
Naturally, this implementation of the
LTC3780 could be applied to many
other ranges of input/output voltages
and currents.
15VIN
EFFICIENCY (%)
those required for input and output
decoupling. The LTC3780 allows the
designer to avoid these expensive,
space-wasting complexities while
increasing efﬁciency.
95
11VIN
7VIN
90
85
Conclusion
LOAD = 4A
6
7
10
8
9
OUTPUT VOLTAGE (V)
11
power consumption is an important
design consideration in helping to
lower overall system operation costs.
In addition, the combination of high
sampling rate, low current and 14-bit
resolution make it ideally suited to battery powered, high performance test
and instrumentation equipment.
The LTC2255 offers exceptional
low-level input signal performance due
to its high linearity, and it is designed
with good margin relative to the sample
rate for reliable performance over a
wide temperature range. At 125Msps
sampling rate, it achieves excellent AC
performance with 72.1dB SNR and
85dB SFDR at 70MHz.
12
Figure 16. Efﬁciency for the adjustable output
supply is consistently in the mid-90s.
STDBYMD pin allows the internal low
dropout regulator to remain functional
even when the RUN pin disables all
other functions of the controller. The
LDO then provides 6V at up to 40mA
at the INTVCC pin for neighboring
“wake-up” circuitry.
Compact, Efﬁcient Regulator
with Programmable VOUT
With an external voltage applied to
its VOSENSE pin through a resistor, the
LTC3780 can control a supply capable
of providing a 4A, 6V–12V output from
a 7V–15V input (Figure 15). Efﬁciency
is in the mid-90 percent range throughout a wide range of inputs and load
currents, as Figure 16 illustrates. Dual
MOSFETs with integrated Schottky
diodes keep the footprint to a minimum. With the application of 0.85V to
4.9V to the feedback node through a
It is not a trivial task to deliver high
current with tight regulation when
the input voltage can be more than,
less than, or equal to the output
voltage. The LTC3780’s proprietary
architecture shoulders the complexity and simpliﬁes the power supply
designer’s job. It is the ﬁrst buckboost controller to provide extremely
high efﬁciency, seamless transitions
between operating modes, and wide
input voltage range, all without resorting to cumbersome magnetics or
multiple control loops.
A converter designed around the
LTC3780 naturally has a wide input
voltage range, which gives it unparalleled versatility. A single converter
design can be powered by any of a
number of rails with the high efﬁciency
of a true synchronous buck or boost
converter. Its unique advantages over
common designs make the LTC3780
ideal for automotive, telecom, industrial, and battery-powered applications.
Linear Technology Magazine • September 2005
DESIGN TOOLS
DESIGN TOOLS
CD-ROM
Product Information
The September 2005 CD-ROM contains product data
sheets, application notes and Design Notes released
through August of 2005. Use your browser to view
product categories and select products from parametric
tables or simply choose products and documents from
part number, application note or design note indexes.
Linear Technology offers high-performance analog
products across a broad product range. Current
product information and design tools are available at
www.linear.com. Our CD-ROM product selector tool,
which is updated quarterly, and our most recent databook
series, published in 2004, can be obtained from your
local Linear Sales ofﬁce (see the back of this magazine)
or requested from www.linear.com.
www.linear.com
Product information and application solutions are
available at www.linear.com through a powerful search
tools, which yield weighted results from our data sheets,
application notes, design notes, Linear Technology
magazine issues and other LTC publications. The LTC
website simpliﬁes the product selection process by
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Search methods include a text search for a particular part
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Applications Handbooks
Linear Applications Handbook, Volume I — Almost a
thousand pages of application ideas covered in depth by
40 Application Notes and 33 Design Notes. This catalog
covers a broad range of real world linear circuitry. In
addition to detailed, systems-oriented circuits, this
handbook contains broad tutorial content together with
liberal use of schematics and scope photography. A
special feature in this edition includes a 22-page section
on SPICE macromodels.
Linear Applications Handbook, Volume II — Continues
the stream of real world linear circuitry initiated by Volume
I. Similar in scope to Volume I, this book covers Application Notes 40 through 54 and Design Notes 33 through
69. References and articles from non-LTC publications
that we have found useful are also included.
Linear Applications Handbook, Volume III —
This 976-page handbook includes Application Notes 55
through 69 and Design Notes 70 through 144. Subjects
include switching regulators, measurement and control
circuits, ﬁlters, video designs, interface, data converters,
power products, battery chargers and CCFL inverters.
An extensive subject index references circuits in Linear
data sheets, design notes, application notes and Linear
Technology magazines.
Linear Technology Magazine • September 2005
Brochures
Power Management & Wireless Solutions for Handheld
Products — The solutions in this product selection guide
solve real-life problems for cell phones, digital cameras,
PDAs and other portable devices, maximizing battery
run time and saving space. Circuits are shown for LiIon battery chargers, battery managers, USB support,
system power regulation, display drivers, white LED
drivers, photoﬂash chargers, DC/DC converters and
RF PA power supply and control.
Automotive Electronic Solutions — This selection guide
features high performance, high reliability solutions for
a wide range of functions commonly used in today’s
automobiles, including telematics, infotainment systems,
body electronics, engine management, safety systems
and GPS navigation systems.
Industrial Signal Chain — This product selection guide
highlights analog-to-digital converters, digital-to-analog
converters, ampliﬁers, comparators, ﬁlters, voltage
references, RMS-to-DC converters and silicon oscillators designed for demanding industrial applications.
These precise, ﬂexible and rugged devices feature
parameters fully guaranteed over the –40°C to 85°C
temperature range.
Software
SwitcherCAD™ III/LTC SPICE — LTC SwitcherCAD III is
a fully functional SPICE simulator with enhancements
and models to ease the simulation of switching regulators. This SPICE is a high performance circuit simulator
and integrated waveform viewer, and also includes
schematic capture. Our enhancements to SPICE result
in much faster simulation of switching regulators than is
possible with normal SPICE simulators. SwitcherCAD III
includes SPICE, macromodels for 80% of LTC’s switching
regulators and over 200 op amp models. It also includes
models of resistors, transistors and MOSFETs. With this
SPICE simulator, most switching regulator waveforms
can be viewed in a few minutes on a high performance
PC. Circuits using op amps and transistors can also be
easily simulated. Download at www.linear.com
FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aided design program for creating ﬁlters with Linear Technology’s
ﬁlter ICs. FilterCAD is designed to help users without
special expertise in ﬁlter design to design good ﬁlters
with a minimum of effort. It can also help experienced
ﬁlter designers achieve better results by playing “what if”
with the conﬁguration and values of various components
and observing the results. With FCAD, you can design
lowpass, highpass, bandpass or notch ﬁlters with a
variety of responses, including Butterworth, Bessel,
Chebychev, elliptic and minimum Q elliptic, plus custom
responses. Download at www.linear.com
SPICE Macromodel Library — This library includes LTC
op amp SPICE macromodels. The models can be used
with any version of SPICE for analog circuit simulations.
These models run on SwitcherCAD III/LTC SPICE.
Noise Program — This PC program allows the user to
calculate circuit noise using LTC op amps, determine the
best LTC op amp for a low noise application, display the
noise data for LTC op amps, calculate resistor noise and
calculate noise using specs for any op amp.
Databooks
Ampliﬁers (Book 1 of 2) —
• Operational Ampliﬁers
Ampliﬁers (Book 2 of 2) —
• Operational Ampliﬁers
• Instrumentation Ampliﬁers
• Application Speciﬁc Ampliﬁers
References, Filters, Comparators, Special
Functions, RF & Wireless —
• Voltage References • Special Functions
• Monolithic Filters
• RF & Wireless
• Comparators
• Optical Communications
• Oscillators
Monolithic Switching Regulators —
• Micropower Switching Regulators
• Continuous Switching Regulators
Switching Regulator Controllers (Book 1 of 2) —
• DC/DC Controllers
Switching Regulator Controllers (Book 2 of 2) —
• DC/DC Controllers
• Digital Voltage Programmers
• Off-Line AC/DC Controllers
Linear Regulators, Charge Pumps,
Battery Chargers —
• Linear Regulators
• Charge Pump DC/DC Converters
• Battery Charging & Management
Hot Swap Controllers, MOSFET Drivers, Special
Power Functions —
• Hot Swap Controllers
• Power Switching & MOSFET Drivers
• PCMCIA Power Controllers
• CCFL Backlight Converters
• Special Power Functions
Data Converters (Book 1 of 2) —
• Analog-to-Digital Converters
Data Converters (Book 2 of 2) —
• Analog-to-Digital Converters
• Digital-to-Analog Converters
• Switches & Multiplexers
Interface, System Monitoring & Control —
• Interface — RS232/562, RS485,
Mixed Protocol, SMBus/I2C
• System Monitoring & Control — Supervisors,
Margining, Sequencing & Tracking Controllers
Information furnished herein by Linear Technology Corporation
is believed to be accurate and reliable. However, no responsibility
is assumed for its use. Linear Technology Corporation makes
no representation that the interconnection of its circuits, as
described herein, will not infringe on existing patent rights.
47
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© 2005 Linear Technology Corporation/Printed in U.S.A./30K
www.linear.com
Linear Technology Magazine • September 2005

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