LINEAR TECHNOLOGY DECEMBER 2005 IN THIS ISSUE… COVER ARTICLE A Better Way to Push Your Buttons......1 Victor Fleury Issue Highlights ..................................2 Linear Technology in the News….........2 DESIGN FEATURES Fast CMOS Op Amp Challenges Bipolar Amps on All Key Specs............5 John Wright and Glen Brisebois Photoflash Capacitor Chargers Keep Up with Shrinking Cameras .......9 Mike Negrete Fully Differential Amplifier with Rail-to-Rail Outputs Offers 16-Bit Performance at 1MHz on a Single 2.5V Supply ............................13 Arnold Nordeng Negative High Voltage Hot Swap Controllers Incorporate an Accurate Supply Monitor and Power Module Sequencing ..........16 Kevin Wong Simplify High-Resolution Video Designs with Fixed-Gain Triple Multiplexers ...........................20 Jon Munson High Efficiency, Monolithic Synchronous Buck-Boost LED Driver Drives up to 1A Continuous Current .....................23 Aspiyan Gazder Constant Current from 3A DC/DC Converter with 2 Rail-to-Rail Current Sense Amplifiers ..................25 Daniel Chen 4-Channel I2C Multiplexer Provides Address Expansion, Bus Buffering and Fault Management .....................28 John Ziegler DESIGN IDEAS ....................................................32–44 (complete list on page 32) New Device Cameos ...........................45 Design Tools ......................................47 Sales Offices .....................................48 VOLUME XV NUMBER 4 A Better Way to Push Your Buttons Introduction by Victor Fleury Is there a better way to debounce the current (6µA) is an insignificant drain on/off push button of a handheld on the battery. The device is available device? Some designers use dis- in space saving 8-lead 3mm × 2mm crete logic, flip-flops, resistors and DFN and ThinSOT packages. capacitors. Others use an on-board microprocessor, which requires constant More than Just a De-Bouncer power—even after the handheld device The LTC2950 is not just a low power, has been turned off. Additionally, for high voltage push button de-bouncer. multi-cell battery applications, a high The debounced push button input voltage LDO is needed to drive the low toggles an open drain enable outvoltage logic and microprocessor. All put. This low leakage output can be this extra circuitry used to control not only increases the shutdown pin board space, but of a DC/DC conHas this happened to you? also drains the verter, and thus Your PDA or laptop has battery when the allow manual frozen—not responding to handheld device control of system any input. You try to restart has been turned power. The part the device by pressing off. also contains a the on/off button. Nothing The LTC2950 simple microprofamily of parts cessor interface happens. The unresponsive eliminates all of that provides inpush button is probably these problems. telligent power the result of an on/off push The part incorpoup and power button that was de-bounced rates the flexible down sequencby an unresponsive µP— timing needed ing. During power evidenced by the crash. The to debounce the up, an internal push button intimer ensur es LTC2950 eliminates this put during system that the system common fault. power on and syswill not power tem power off. The into a short and LTC2950’s wide input voltage range drain the battery. During power (2.7V to 26.4V) is designed to operate down, the LTC2950 interrupts the from single cell to multi-cell battery microprocessor 1024ms before de-asstacks, thus eliminating the need for serting the enable output. This gives an LDO. The part’s set of features al- the microprocessor time to perform lows the system designer to turn off housekeeping tasks (such as saving to system power to all circuits except the memory) before power is turned off. LTC2950, whose very low quiescent continued on page 3 , LTC, LTM, LT, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView, µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products. EDITOR’S PAGE Issue Highlights W hen a push button is pressed, the voltage on the pin does not seamlessly switch from the pull-up voltage to ground. The voltage fluctuates as the push button makes and breaks contact, potentially causing the microprocessor to see a series of on/off events. The LTC2950 solves this problem by ignoring all the noise and driving the enable pin high 32ms after the push button stops bouncing. See our cover article for more about this breakthrough device. Featured Devices Below is a summary of the other devices featured in this issue. Power Solutions The LTC3454 is a synchronous buckboost DC/DC converter, designed for driving a single high power LED with regulated currents up to 1A from a single Li-Ion battery. (Page 23) The LT3477 combines a traditional voltage feedback loop and two unique current feedback loops to operate as a constant-current, constant-voltage source. It is a current mode, 3A DC/ DC converter with dual rail-to-rail 100mV current sense amplifiers that can be configured as a buck mode or buck-boost mode LED driver. It is versatile enough to also be configured as an input-output current limited boost, SEPIC or inverting converter. (Page 25) Hot Swap and 2-Wire Bus Solutions The LTC4253A and LTC4253A-ADJ facilitate safe board insertion and removal from a live backplane by applying power in a controlled manner. Running off a simple, fast responding shunt regulated supply that allows Linear Technology in the News… Linear Announces New Line of Power Modules In October, Linear Technology announced a new line of high-density power modules. The new product line provides designers with simple, compact and reliable power supplies for a broad range of applications. Using these compact, board-ready products, designers can significantly accelerate time-to-market and reduce risk in implementing high performance power systems. The first product in the family, the LTM®4600 µModule™ is a 10A switchmode DC/DC step-down power supply in a small, surface mount package. This new product line leverages the company’s core strengths in power management, resulting in a highly integrated module with record power density. According to Don Paulus, Vice President and General Manager for Power Management Products, “Designers today are challenged to develop systems at ever-higher 2 power densities with relentless time to market pressure. With power supplies becoming more complex, designers increasingly require sophisticated power design expertise. Using the LTM4600 µModule, the power design is virtually done, freeing system designers to focus on their core expertise.” Expanded Test Facilities Last month, Linear Technology announced the opening of the company’s second semiconductor test facility in Singapore. This expansion will allow the company to more than double its current production capacity, strengthening its ability to meet the growing global demand for Linear’s high performance analog circuits. This growing facility, combined with Linear’s two US wafer fabrication plants and its assembly facility in Malaysia, ensure that customers receive the highest quality ICs with fast delivery times. very high voltage operation, they are uniquely suited for applications on the –48V bus. (Page 16) The LTC4306 4-channel 2-wire bus multiplexer/switch with bus buffers addresses a variety of capacitive buffering, addressing and Hot Swap issues. (Page 28) Op Amps The LTC6241 dual and LTC6242 quad CMOS op amps compete head-on with bipolar op amps in noise, speed, offset voltage, and offset drift, while maintaining superior low input bias and noise current. (Page 5) What sets the LT1994 apart from other fully differential amplifiers are its low noise, low distortion, rail-to-rail output, and an input common mode range that extends to ground on power supplies as low as 2.5V. This eliminates the need for a negative power supply, and makes the LT1994 uniquely able to interface to differential input ADCs while sharing the same power supply. (Page 13) The LT6555 and LT6556 triple video multiplexers offer up to 750MHz performance in compact packages, requiring no external gain-setting resistors to establish a gain of two or unity. A single integrated circuit, in a choice of either 24-lead SSOP or 24-contact QFN (4mm × 4mm), performs fast switching between a pair of three-channel video sources, such as RGB or component HDTV. (Page 20) Photoflash Capacitor Chargers The LT3484 and LT3485 photoflash capacitor chargers squeeze high performace xenon flash technology into the small spaces afforded to cameras in cell phones and PDAs. (Page 9) Design Ideas and Cameos Design Ideas start on page 32, including a discusion of Li-Ion-based battery chargers and an op amp selection guide. A cameo about the exciting new LTM4600 µModule DC/DC converter appears on page 45. Linear Technology Magazine • December 2005 DESIGN FEATURES LTC2950, continued from page 1 VIN 3V – 26V Watch the Push Button Bounce SHDN VIN PB GND Typical Power On/Off Timing Sequence Figure 3 shows a typical LTC2950-1 power on and power off sequence. A high to low transition on PB (t1) initiates the power on sequence. This diagram does not show any bounce on PB. In order to assert the enable output, the PB pin must stay low continuously (PB high resets timers) for a time controlled by the default 32ms and the external ONT capacitor (t2 – t1). Once EN goes high (t2), an internal 512ms blanking timer is started. This blanking timer is designed to give sufficient time for the DC/DC converter to reach its final voltLinear Technology Magazine • December 2005 R1 10k EN LTC2950-1 ONT INT INT KILL KILL µP/µC OFFT 2950 TA01 CONT* 0.033µF COFFT* 0.033µF *OPTIONAL EN 2V/DIV Need Longer Debounce Times? Turn On Debounce Time = 32ms + (6.7 • 106) • CONT Turn Off Debounce Time = 32ms + (6.7 • 106) • COFFT VOUT DC/DC BUCK When a push button is pressed, the voltage on the pin does not seamlessly switch from the pull-up voltage to ground. The voltage fluctuates as the push button makes and breaks contact. Figure 1 shows an application with significant bounce on the push button pin. The LTC2950 ignores all the noise and drives the enable pin high 32ms after the push button stops bouncing. The scope trace shows the turn on debounce time of 32ms—that is, no external capacitor at the ONT pin. This application requires only one external component (R1). It is no problem to extend the debounce time of the push button input. The power on and power off debounce times can be extended independently by placing an external capacitor on the ONT and OFFT pins, respectively. Figure 2 shows the turn on timing with an external 0.033µF capacitor on the ONT pin (~250ms). The following equations describe the relationship between total debounce time and external capacitors: VIN PB 10ms/DIV Figure 1. Typical circuit and de-bounce timing age, and to allow the µP enough time to perform power on tasks. The KILL pin must be pulled high within 512ms after the EN pin went high. Failure to do so results in the EN pin going low 512ms after it went high (EN = low, see Figure 4). Note that the LTC2950 does not sample KILL and PB until after the 512ms internal timer has expired. The reason PB is ignored is to ensure that the system is not forced off while powering on. Once the 512ms turn on blanking timer expires (t4), the release of the PB pin is then de-bounced with an internal 32ms timer. The system has now properly powered on and the LTC2950 monitors PB and KILL (for EN a turnoff command) while consuming only 6µA of supply current. A high to low transition on PB (t5) initiates the power off sequence. PB must stay low continuously (PB high resets de-bounce timer) for a period controlled by the default 32ms and the external OFFT capacitor (t6–t5). At the completion of the OFFT timing (t6), an interrupt (INT) is set, signifying that EN will be switched low in 1024ms. Once a system has finished performing its power down operations, it can set KILL low (t7) and thus immediately set EN low), terminating the internal 1024ms timer. The release of the PB pin is then de-bounced with an internal 32ms timer. The system is now in its reset state: where the LTC2950 is in low power mode (6µA). PB is monitored for a high to low transition. What if the DC/DC Converter is Faulty at Power Up? 1V/DIV PB 50ms/DIV Figure 2. PB turn on de-bounce time increased with an external 0.033µF capacitor When a user turns on a handheld device, the LTC2950 EN output pin enables a DC/DC converter. The output of the converter can then power a µP, which in turn drives the KILL pin (see Figure 1). If there is a system fault 3 DESIGN FEATURES t1 t2 PB t3 t4 t5 t6 PB & KILL IGNORED tDB, ON tONT t7 PB IGNORED tKILL, ON BLANK < tKILL, OFF DELAY tDB, OFF ONT tOFFT OFFT INT <tKILL, OFF DELAY KILL EN 2950 F01 Figure 3. Typical Power On/Off Timing Sequence for LTC2950-1 (shorted DC/DC output, for example) that prevents the µP from driving the KILL input high within 512ms, the LTC2950 automatically releases its enable output. This in turn shuts off the converter and prevents the handheld device from turning on. Figure 4 depicts an aborted power on sequence. tABORT PB tDB, ON + tONT 512ms INTERNAL TIMER POWER ON TIMING Protect Against µP Hang Ups Has this happened to you? Your PDA or laptop has frozen—not responding to any input. You try to restart the device by pressing the on/off button. Nothing happens. In frustration you resort to unplugging the device and removing any batteries to shut it down. The unresponsive push button is probably the result of an on/off push button that was de-bounced by an unresponsive µP—evidenced by the crash. The LTC2950 eliminates this common fault. The LTC2950 always responds to the push button in some way. It does this by initiating a power down sequence (in response to the user pressing the push button) by asserting INT low and starting an internal 1024ms timer. This event alerts the µP of the impending power down. If the KILL pin remains high (µP not respond4 POWER TURNED OFF EN µP FAILED TO SET KILL HIGH KILL 2950 F02 Figure 4. Aborted power on sequence for LTC2950-1 ing) at the end of the 1024ms timeout period, the LTC2950 automatically releases its enable pin, thus shutting off system power. This fault protection feature makes sure that a user is always capable of turning off system power, even when the rest of the system is faulty or not responding. PB Pin Survives Minor Lightning Strike The PB and VIN pins are both high voltage pins (33V absolute maximum). Additionally, high ESD strikes (±10kV, HBM) will not damage the PB pin. FigFigure 5. ESD Strikes PB Pin continued on page 46 Linear Technology Magazine • December 2005 DESIGN FEATURES Fast CMOS Op Amp Challenges Bipolar Amps on All Key Specs by John Wright and Glen Brisebois Introduction The LTC6241 dual and LTC6242 quad CMOS op amps compete headon with bipolar op amps in noise, speed, offset voltage, and offset drift, while maintaining superior low input bias and noise current. Crucial advances in these amplifiers’ parameters translate to tighter system specs, lower complexity, and a wider supply voltage operating range than previous CMOS op amps. These extremely low input bias current op amps are optimized for high impedance transducer applications such as photodiode transimpedance amplifiers, TIAs, though they are also well suited to a variety of precision applications. The LTC6241 and LTC6242 do not employ complicated post-package schemes to reduce offset voltage, yet their 125µV offset voltage and 2.5µV/°C offset drift are among the best CMOS amplifiers available. The 18MHz gain bandwidth and very low noise further distinguishes them from a field of mediocre amplifiers. They are fully specified on 3V, and 5V, with an HV version that guarantees operation to ±5.5V. Supply current consumption is 2.2mA/amplifier maximum. Table 1 summarizes the conservative specs for these op amps. The LTC6241 is available in the SO8, and for compact designs it is packaged in the tiny dual fine pitch leadless (DFN) package. The LTC6242 is available in a 16-Pin SSOP as well as a 5mm × 3mm DFN package. CMOS with Low 1/f Noise? What about Noise Current? CMOS op amps have traditionally had much higher 1/f noise than bipolar amplifiers. It is common to find CMOS amplifiers with a 1/f corner above several kilohertz, but the LTC6241 rivals the best bipolar op amps with a 1/f noise corner of only 40Hz. This exceptionally low noise translates Linear Technology Magazine • December 2005 to just 550nVP–P in a 0.1Hz to 10Hz bandwidth, and represents the lowest 1/f noise available in a non-autozero CMOS op amp. In I-to-V applications such as photodiode amplifiers, where the amplifier is operated inverting, noise current dominates at high frequency. CMOS op amp noise current has two sources. The first is the input device channel thermal noise coupling through the gate-to-source and gate-to-drain capacitances. The second noise current is derived from the op amp’s input capacitance, and capacitance associated with the input transducer. This input referred noise current (CV noise) is due to the amplifier’s noise voltage, VN, impressed across the total input capacitance, CT, causing a current of magnitude 2πfCTVN to flow through the feedback resistor. The way to make CMOS or bipolar low noise amplifiers is with large input transistors. The problem is that big input structures carry the burden of high input capacitance. High input capaci- Table 1. LTC6241/LTC6242 Performance: Ta = 25°C, VS = 5V/0V unless otherwise specified. The ● denotes specifications that apply over –40°C to 85°C. Parameter Conditions Offset Voltage VCM = 0 S8, LTC6241 GN16, LTC6242 DD, DHC, LTC6241/42 Min Typ Max Units 40 50 100 125 150 550 µV µV µV TC VOS ● 0.6 2.5 µV/°C Input Bias Current ● 1 10 75 pA pA Noise Voltage f = 1kHz f = 0.1Hz to 10Hz 7 550 Noise Current f = 100kHz 110 fA/� Input Capacitance f = 100kHz CDM CCM 0.5 3 pF pF Large Signal Gain RL = 1kΩ to VS/2 90 215 V/mV 80 105 dB CMRR VCM = –V to +V – 1.5V ● 2.8 2.8 10 Operating Supply Range LTC6241/42 LTV6241HV/42HV ● ● VOUT Low ISINK = 5mA ● 190 VOUT High ISOURCE = 5mA ● 4.81 4.675 Supply Current per amplifier ● 1.8 Slew Rate AV = –2, RL = 1kΩ, ● 5 10 Gain Bandwidth Product RL = 1kΩ ● 13 18 nV/� nVP–P 6 11 V V 325 mV 2.2 V mA V/µs 5 DESIGN FEATURES ITAIL V– VIN VIN – I1 V+ I2 M3 CM V+ V+ DESD2 DESD1 + I1 DESD5 RT2 RT1 M1 DIFFERENTIAL DRIVE GENERATOR M2 VO DESD6 C1 V– V V– DESD4 DESD3 V+ Q2 Q1 BIAS R1 – M4 R2 V– Figure 1. Simplified schematic tance increases high frequency noise current, as well as reduces overall op amp speed. An uncommon feature of the LTC6241 is its low differential input capacitance of just 0.5pF, which is a major benefit in I-to-V amplifier designs. This input capacitance is 8 to 10 times lower than than that of other CMOS amps. Simple Architecture Yields Low Noise and DC Precision Figure 1 is a simplified schematic of one half of the LTC6241, which has a pair of low noise input transistors M1 and M2. A simple folded cascode Q1, Q2, and R1, R2 allow the input stage to swing to the negative rail, while performing level shift to the differential drive generator. Transistors M1 and M2 along with current sources 90 16 VS = ±2.5V 80 SO-8 PACKAGE 12 60 NUMBER OF UNITS NUMBER OF UNITS 70 50 40 30 10 8 6 20 4 10 2 0 VS = ±2.5V 2 LOTS –55°C TO 125°C 14 –70 –50 –30 –10 10 30 50 INPUT OFFSET VOLTAGE (µV) 0 70 –1.0 –0.6 –0.2 0.2 0.6 1.0 DISTRIBUTION (µV/°C) 1.4 1.8 I1 and I2 have been optimized for low noise and consume over 30% of the die area. Low offset is achieved by laser trimming resistors R T1 and R T2. Stresses that occur during package assembly have minimal affect on this simple, stable architecture, and consequently, complicated post-package trim schemes that adjust offset voltage and drift are unnecessary. The LTC6241 and LTC6242 were intentionally designed without a rail-to-rail input stage as to not compromise their noise specs. Many CMOS rail-to-rail input amplifiers show large offset shift and higher noise when the common mode voltage is operating in this top side transition region, limiting their usefulness. The LTC6241 and LTC6242 have reverse-biased ESD protection diodes on all inputs and outputs as shown in Figure 1. These diodes protect the amplifiers from ESD strikes up to 1.7kV. No current flows into the gate on a DC basis, but these ESD protection diodes are the source of input bias current specified on the data sheet. These diodes have leakage current that doubles approximately every 7°C, but input current typically remains below 10pA up to 85°C ambient. Capacitor C1 reduces the unity cross frequency and improves the frequency stability without degrading the gain bandwidth of the amplifier. Capacitor CM sets the overall amplifier gain bandwidth. The differential drive generator supplies signal to transistors M3 and M4 that swing the output from rail-to-rail. Figure 2. VOS distribution and VOS temperature coefficient distribution 60 VS = 5V, 0V 100 TA = 85°C 10 TA = 25°C 1 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 COMMON MODE VOLTAGE (V) 1000 TA = 25°C VS = ±2.5V VCM = 0V 50 NOISE CURRENT (fA/√Hz) TA = 125°C NOISE VOLTAGE (nV/√Hz) INPUT BIAS CURRENT (pA) 1000 40 30 20 TA = 25°C VS = ±2.5V VCM = 0V 100 10 1 10 0 1 10 100 1k FREQUENCY (Hz) 10k 100k 0.1 100 1k 10k FREQUENCY (Hz) 100k Figure 3. Input bias current vs common mode voltage and voltage and current noise vs frequency 6 Linear Technology Magazine • December 2005 DESIGN FEATURES VIN C 0.1µF R 1M – VS + 1/2 LTC6241 + – VOUT = –∫ VIN/2πRCdt t=0 VIN R1 1M Figure 4. A textbook integrator is inverting Figure 2 shows the distribution of offset voltage and offset voltage drift. Figure 3 shows the input bias current vs common mode voltage as well as the noise voltage and current spectrum. Applications Non-Inverting Integrator Integrators are used widely in feedback control systems and filters. CMOS input amplifiers like the LTC6241 are preferred for this function because the low input bias current allows the use of large value resistors and small capacitors for a given integration time constant. The most common form of integrator is the inverting form, shown in Figure 4. It has a transfer function of: ⌠ VIN – dt ⌡ 2πRC t =0 If inversion is not desired in the feedback control loop using the circuit in Figure 4, a designer must add another op amp to invert again. A simpler overall solution produces a non-inverting integrator using just one op amp. Figure 5 shows the circuit. At low frequencies, R1 • C1 does not attenuate, and the non-inverting integration function is provided by the op amp gain and its feedback components C2 and R2. At higher frequencies, C2 becomes a short circuit so the op amp goes to a gain of one, and the integration function is provided by R1 and C1. If the time constants are matched, the integrator conformance is excellent. Matching is not easy. In most loops, to guarantee that the phase of the integrator does not exceed 90 degrees, the time constants can be intentionally skewed so that R1 • C1 < R2 • C2. For an example Linear Technology Magazine • December 2005 VS+ VOUT = ∫ VIN/2πRCdt t=0 1/2 LTC6241 + C1 0.1µF VS – VOUT = C2 0.1µF R2 1M LET R1 • C1 = R2 • C2 VS– Figure 5. A non-inverting integrator can be very simple. Ideally, R1 • C1 = R2 • C2, but mismatch is inevitable. To avoid any phase buildup from a mismatch, the time constants may be skewed so that R1 • C1 < R2 • C2. of a specific closed loop utilization of a non-inverting integrator, see LTC Design Note DN254. under physical acceleration. Figure 6 shows the classical “charge amplifier” approach. The op amp is in the inverting configuration so the sensor looks into a virtual ground. All of the charge generated by the sensor is transferred across the feedback capacitor by the op amp action. Because the feedback capacitor is 100 times smaller than the sensor, the output is forced to a voltage 100 times what would have Piezoelectric Accelerometers: Inverting vs Non-Inverting Figures 6 and 7 show two different approaches to amplifying signals from a capacitive sensor using the LTC6241. The sensor in both cases is a 770pF piezoelectric shock sensor accelerometer, which generates charge SHOCK SENSOR VS + MURATA-ERIE PKGS-00LD 770pF + 1/2 LTC6241HV – VS CABLE CAPACITANCE VARIATIONS OKAY VOUT – 7.7pF BIAS RESISTOR 1GΩ VISHAY-TECHNO CRHV2512AF1007G (OR EQUIVALENT) VOUT = 110mV/G VS = ±1.4V TO ±5.5V Figure 6. Classical inverting charge amplifier. Variations in cable capacitance (i.e. length) do not affect the signal gain. Use this circuit when the accelerometer is remote from the amplifier and the cable length is unspecified. Drawbacks are that gain is set by the low valued feedback capacitor and low frequency performance is set by the bias resistor working into the same. SHOCK SENSOR + MURATA-ERIE PKGS-00LD 770pF – VS + 1/2 LTC6241HV 100 1k 10k VOUT 1k + 1/2 LTC6241HV BIAS RESISTOR 1GΩ VISHAY-TECHNO CRHV2512AF1007G (OR EQUIVALENT) – 100 VS – 10k VOUT = 110mV/G VS = ±1.4V TO ±5.5V BW = 0.2Hz TO 10kHz Figure 7. Non-inverting charge amplifier offers several advantages. Stages can be paralleled for lower voltage noise. Bias resistor works into higher capacitance for better low frequency response. 7 DESIGN FEATURES CF 8pF drawback to this circuit is that the parasitic capacitance at the input reduces the gain slightly. This circuit is favored in cases where parasitic input capacitances such as traces and cables are relatively small and invariant. Consider making the bias resistor larger than bandwidth calculations would suggest. This actually reduces the noise floor at low frequency. For example, to support frequencies down to 10Hz at –3dB, the bias resistor would calculate to: RF 1M IPD – HAMAMATSU LARGE AREA PHOTODIODE S1227-1010BQ CPD = 3000pF + 5V 1/2 LTC6241HV VOUT = 1M • IPD –5V Figure 8. Large area photodiode amplifier provides about 25kHz bandwidth. DCs are good but output is noisy. 1 = 20MΩ 2π • 10Hz • 770pF been the sensor’s open circuit voltage. Thus, the circuit gain is 100. The benefit of this approach is that the signal gain of the circuit is independent of any cable capacitance introduced between the sensor and the amplifier, making this a good solution for remote accelerometers where the cable length may vary. Difficulties with the circuit are inaccuracy of the gain setting with the small capacitor, and low frequency cutoff due to the bias resistor working into the small feedback capacitor. Figure 7 shows a non-inverting amplifier approach. This approach has many advantages. First, the gain is set accurately with resistors rather than with a small capacitor. Second, the low frequency cutoff is dictated by the bias resistor working into the large 770pF sensor, rather than into a small feedback capacitor, for lower frequency response. Third, the noninverting topology can be paralleled and summed (as shown) for scalable reductions in voltage noise. The only 5V –5V PHILIPS BF862 JFET D 4.99k S At 10Hz, the 20M resistor would contribute 580nV/√Hz of noise, and be 3dB down just like the signal. Making the resistor 1GΩ as shown, its 4000nV/√Hz voltage noise would be attenuated down to effectively 80nV/√Hz by the accelerometer capacitance, while the signal would barely be attenuated at all. That’s an easy seven-fold improvement in the signal-to-noise ratio. Large Area Photodiode Amplifiers Figure 8 shows the LTC6241 used as a transimpedance amplifier for a high capacitance large area photodiode. The circuit has unity noise gain at DC, so resolution is entirely noise limited. The bandwidth rolls due to the fact that the photodiode impedance drops with frequency raising the effective gain (the noise gain), which the op amp looks into. This severely limits the bandwidth and increases the output noise. The –3dB bandwidth for this CF 0.5pF IPD HAMAMATSU LARGE AREA PHOTODIODE S1227-1010BQ CPD = 3000pF – The LTC6241 and LTC6242 combine the low noise, offset, and drift of the best bipolar op amps with low input bias and noise current of CMOS op amps. These amplifiers operate from 2.7V to ±5.5V and represent all-in-one solutions for fast, low noise signal processing. 5V 1/2 LTC6241 + VOUT = 1M • IPD 300nV/� per DIV –5V Figure 9. A simple bootstrap circuit drastically improves the ACs while leaving the DCs excellent. Output noise is now 221nV/√Hz at 10kHz, and bandwidth is 220kHz. Rise time is 1.58µs from a 3000pF photodiode at 1MΩ of gain! 8 Conclusion 3µV/� RF 1M G circuit was measured at 25kHz, and the output noise density at 10kHz was measured at 1.6µV/√Hz. That may be good enough for many applications. If it’s not good enough, keep reading. The main problem with the previous circuit is the large capacitance of the photodiode. The perfect thing to do is to bootstrap that capacitance with a low noise JFET. Figure 9 shows the circuit. The low noise JFET source follower runs about 1mA down through the 4.99k resistor, with the source sitting about 0.6V above ground. Now the effective input voltage noise placed across the photodiode capacitance is the 1nV/√Hz of the JFET rather than the 8nV/√Hz of the op amp. The op amp is looking into its own 3pF of input capacitance plus the 2pF of gate-drain capacitance, plus parasitics. That’s a much better situation than looking into 3000pF! The effects of this simple modification are drastic. The compensation capacitor CF can be reduced, and bandwidth is improved to 220kHz (1.58µs rise time). Output noise density at 10kHz is reduced to 221nV/√Hz, as shown in Figure 10. DC performance remains excellent because the JFET is not involved; it simply provides a slight reverse bias to the photodiode. 0nV/� f = 1kHz to 100kHz, 10kHz/DIV Figure 10. Output noise spectral density of the bootstrap circuit of Figure 9 Linear Technology Magazine • December 2005 DESIGN FEATURES Photoflash Capacitor Chargers Keep Up with Shrinking Cameras by Mike Negrete Introduction Camera-phones have come a long way since the first generation of integrated cameras offered low-resolution CMOS images through the eye of a plastic lens. Now PDAs and high-end cell phones include high quality cameras with 2 megapixel resolutions and glass optics. Since these devices are carried by most users at all times, size is of the utmost importance. LED flashes were introduced in early model cell phone cameras, but they cannot produce enough light and lack the spectral quality required for higher-end cameras. Although xenon flashes are an optimal source of light for photography, they required substantially more board space than LED flashes until 10 COUT = 50µF CHARGE TIME (SECONDS) 9 8 7 6 LT3484-1 5 LT3484-2 4 3 2 LT3484-0 1 0 2 3 4 5 VIN (V) 6 7 8 Figure 2. Charge time for the LT3484 VBAT 1.8V TO 8V T1 1:10.2 C1 4.7µF D1 1 4 2 5 320V + 4, 5 SW 6 VIN 2.5V TO 8V C2 0.1µF DONE CHARGE 3 VBAT VIN R1 100k 1 2 LT3484-0 GND COUT PHOTOFLASH CAPACITOR 7 DONE CHARGE C1: 4.7µF, X5R OR X7R, 10V T1: KIJIMA MUSEN PART# SBL-5.6-1, LPRI = 10µH, N = 10.2 D1: VISHAY GSD2004S DUAL DIODE CONNECTED IN SERIES R1: PULL UP RESISTOR NEEDED IF DONE PIN USED Figure 1. Compact, 320V photoflash capacitor charging circuit needs no external Schottky diode the LT3468 allowed xenon flashes to fit into the spaces of cell phones and PDAs. The LT3484 and LT3485 photoflash capacitor chargers improve upon the LT3468. The LT3484 and LT3485 are based on the LT3468’s patented control scheme, providing well controlled battery current, fast charge times and high efficiency. Both series of parts use the same tiny, low-profile transformers as the LT3468. Available in a 6-Lead 2mm × 3mm DFN, the LT3484 reduces the board space significantly with its smaller package and total solution size compared to the LT3468. The LT3484 has also added an additional pin, VBAT, to allow it to operate from two alkaline cells. For xenon photoflash applications with an IGBT, the LT3485 decreases the solution size further with the same photoflash functionality as the LT3484 and an integrated IGBT driver in its 10-Lead 3mm × 3mm DFN package. The LT3485 also features an output voltage monitor pin. Overview A typical application circuit for the LT3484 is shown in Figure 1. With a high level of integration inside the part, Table 1. Photoflash capacitor charger features LT3484-0 LT3484-1 LT3484-2 LT3485-0 LT3485-1 LT3485-2 LT3485-3 Peak SW Current (A) 1.4 0.7 1.0 1.4 0.7 1.0 2.0 Average Input Current (mA) (VIN = 3.6V, VOUT = 225V) 500 250 400 500 250 400 750 Charge Time Coefficient Kijima (τ) 0.65 0.30 0.50 0.75 0.34 0.51 NA Charge Time Coefficient TDK (τ) 0.62 0.32 0.51 0.73 0.37 0.51 1.10 Minimum Battery Voltage(V) 1.8 1.8 Integrated IGBT Drive + VOUT Monitor No Yes External Schottky Diode Required No No Package 2mm × 3mm DFN 6L 3mm × 3mm DFN 10L Linear Technology Magazine • December 2005 9 DESIGN FEATURES DANGER HIGH VOLTAGE — OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY 1:10.2 320V SEE TABLE 2 1 4.7µF 2 • • 5 SW VBAT DONE 1M 150µF PHOTOFLASH CAPACITOR TRIGGER T 1 LT3485-0 VIN 0.22µF IGBTPWR IGBTIN 5 A 2.2µF 600V CHARGE VCC 5V 6 4 GND VMONT 2 FLASHLAMP 3 TO MICRO C CHARGE TIME (SECONDS) VBAT 2 AA OR 1 TO 2 Li-Ion LT3485-1 4 LT3485-2 3 2 1 LT3485-0 IGBT IGBTOUT 0 Figure 3. Compact, 320V photoflash capacitor charging circuit with integrated IGBT drive the application circuit only requires a tiny, low-profile transformer, a high voltage diode, and an input bypass capacitor to charge any size output capacitor to 320V. Despite requiring only 70mm2 of valuable board space, the patented control scheme with its high power, integrated low resistance NPN power switch produces fast charge times shown in Figure 2. There are three versions of the LT3484 depending on charge time and input current requirements. The LT3484-0 has the highest input current at 500mA, while the LT3484-1 has the lowest average input current at 225mA. The LT3484-2 has an input current at 375mA. A typical application circuit for the LT3485 is shown in Figure 3. In addition to the photoflash capacitor charging circuitry, the LT3485 integrates an IGBT drive and a voltage output monitor. The integrated IGBT drive saves valuable board space and cost by eliminating several external components. The voltage output monitor provides a solution to monitor the output voltage without resorting to a resistor divider on the output, which would drain the output capacitor. Along with identical current level versions of the LT3484, the LT3485 series features a high input current part, the LT3485-3, at 750mA. Typical charge times are shown in Figure 4. Operation Figure 5 shows a block diagram for the LT3484 and LT3485, which have identical operation except for the IGBT drive and voltage output monitor in the LT3485—highlighted in the diagram. A low-to-high transition on the CHARGE pin initiates the part. An edge triggered one-shot triggered by the CHARGE pin 2 4 3 LT3485-3 5 VIN (V) 6 7 8 Figure 4. Charge time for the LT3485 puts the various latches inside the part into the proper state. The part begins charging by turning on the power NPN transistor Q1. With Q1 on, the current in the primary of the flyback transformer increases. When it reaches the current limit, Q1 is turned off and the secondary of the transformer delivers current to the photoflash capacitor via diode D1. During this time, the voltage on the SW pin is proportional to the output voltage. Since the SW pin is higher than VBAT by an amount roughly equal to (VOUT + 2 • VD)/N, the output of the discontinuous conduction (DCM) mode comparator is high. In this equation, VOUT is the photoflash capacitor voltage, VD is the rectifying diode forward drop, and N is the turns ratio of the transformer. Once the current in the secondary of the transformer decays to zero, Table 2. Pre-designed transformers — typical specifications unless otherwise noted For Use With Transformer Name Size (W × L × H) mm LPRI (µH) LPRILeakage (nH) N RPRI (MΩ) RSEC (Ω) Vendor LT3484/5-0 LT3484/5-2 LT3484/5-1 SBL-5.6-1 SBL-5.6-1 SBL-5.6S-1 5.6 × 8.5 × 4.0 5.6 × 8.5 × 4.0 5.6 × 8.5 × 3.0 10 10 24 200 Max 200 Max 400 Max 10.2 10.2 10.2 103 103 305 26 26 55 Kijima Musen Hong Kong Office 852-2489-8266 (ph) [email protected] (email) LT3484/5-0 LT3484/5-1 LT3484/5-2 LT3485-3 LDT565630T-001 LDT565630T-002 LDT565630T-003 LDT565630T-041 5.8 × 5.8 × 3.0 6 5.8 × 5.8 × 3.0 14.5 5.8 × 5.8 × 3.0 10.5 5.8 × 5.8 × 3.0 4.7 200 Max 500 Max 550 Max 150 Max 10.4 10.2 10.2 10.4 100 Max 240 Max 210 Max 90 Max 10 Max 16.5 Max 14 Max 16.4 Max TDK Chicago Sales Office (847) 803-6100 (ph) www.components.tdk.com LT3485-0 LT3485-1 LT3485-1 LT3485-3 T-15-089 T-15-089 T-15-083 T-17-109A 6.4 × 7.7 × 4.0 6.4 × 7.7 × 4.0 8.0 × 8.9 × 2.0 6.5 × 7.9 × 4.0 400 Max 400 Max 500 Max 300 Max 10.2 10.2 10.2 10.2 211 Max 27 Max 211 Max 27 Max 675 Max 35 Max 78 Max 18.61 Max 10 12 12 20 5.9 Tokyo Coil Engineering Japan Office 0426-56-6262 (ph) www.tokyo-coil.co.jp Linear Technology Magazine • December 2005 DESIGN FEATURES PRIMARY C1 TO VIN C2 DONE 9 10 3 Q3 SAMPLE AND HOLD CHARGE TO VIN IGBTIN R 2 R2 60k CHIP POWER Q2 ENABLE R3 4k R1 2.5k R4 120k DCM COMPARATOR ONESHOT + – + + – 60mV A2 IGBT DRIVER POWER VOUT COMPARATOR – 1.25V REFERENCE 7 DRIVER R IGBT DRIVER COUT PHOTOFLASH CAPACITOR A3 LT3485 ONLY ONESHOT 1 8 Q S VOUT SECONDARY SW 4, 5 VMONT Q D1 T1 TO BATTERY S Q Q1 20Ω + ONESHOT 20k RSENSE A1 – +– GND 11 20mV 6 LT3485 ONLY LT3485-3: RSENSE = 0.010Ω LT3485-0: RSENSE = 0.015Ω LT3485-2: RSENSE = 0.022Ω LT3485-1: RSENSE = 0.030Ω TO GATE OF IGBT Figure 5. Block diagram for the LT3484 and the LT3485 the voltage on the SW pin collapses to VBAT, or lower. As a result, the output of the DCM comparator goes low, which triggers the one-shot. This leads to Q1 turning on again and the cycle repeats. Output voltage detection is accomplished via comparator A2. When the SW pin is 31.5V higher than VBAT on any cycle, the output of A2 goes high. This resets the master latch and the part stops delivering power to the photoflash capacitor. Power delivery can only restart by taking the CHARGE pin low and then high. Note that the flux in the flyback transformer is brought to zero on each switching cycle. This is generally referred to as boundary mode operation since the transformer is operated in between continuous conduction mode and discontinuous conduction mode (CCM and DCM respectively). When the CHARGE pin is forced low at any time, the LT3484/LT3485 ceases power delivery and goes into shutdown mode, thus reducing quiescent current to less than 1µA. Figure 6 shows some typical waveforms for the LT3484 and LT3485. Voltage Output Monitor VSW 10V/DIV IPRI 1A/DIV VIN = 3.6V VOUT = 300V 1µs/DIV Figure 6. A LT3485 switching waveform at 300V output Linear Technology Magazine • December 2005 Camera manufacturers continue to try to differentiate their product with novel features such as strobe shots and sequential shots. These new features rely on fast capacitor charging to be done in the time between shots. If the capacitor is not fully charged, is the voltage high enough to produce a flash? The LT3485 addresses this problem by including a 1V full-scale output, VMONT, proportional to the capacitor voltage. This output can easily be read by a microcontroller with an ADC. Figure 7 shows the measured output of VMONT. Because of the high speed nature of the circuit and the high dV/dt of the switch pin, there is a small amount of ripple on the VMONT output, which can be reduced by adding a 0.1µF capacitor to the output or by using the ADC to sample the VMONT output multiple times and taking the average. CHARGE 2V/DIV VOUT 100V/DIV VMONT 200mV/DIV 100ms/DIV Figure 7. Voltage output monitor waveform during charging 11 DESIGN FEATURES IGBT Drive Most camera flashes are capable of redeye reduction and light-feedback flashing. These features quench, or stop, the flash before the capacitor drains completely. This added level of control requires a high current, high voltage Insulated Gate Bipolar Transistor (IGBT). An IGBT has the advantage of a BJT’s high voltage and high current capabilities but does not need base current since it has a MOSFET gate as the input. The tradeoff for these two advantages is speed. Since a flash is on the order of milliseconds, speed is not an issue in this application and an IGBT fits perfectly for the role. Like a MOSFET, the gate acts like a capacitor. The IGBT driver’s job is to charge and discharge the gate. The IGBT driver does not need to be fast, and actually a fast driver can potentially destroy the device. The IGBT turns on when the IGBTIN pin is above 1.5V and turns off when the IGBTIN pin is below 0.3V. When the input is high, the driver draws a small amount of current to hold the gate high with a PNP. When the input is low, the driver has zero quiescent current. During transitions the driver is capable of delivering 150mA of current. The speed of the driver needs to be carefully controlled or the IGBT may be destroyed. The IGBT driver does not need to pull up the gate fast because of the inherently slow nature of the IGBT. A rise time of 2µs is sufficient to charge the gate of the IGBT and create a trigger pulse. With slower rise times, the trigger circuitry does not have a fast enough edge to create the required 4kV pulse. The fall time of the IGBT drive is critical to the safe operation of the IGBT. The IGBT gate is a network of resistors and capacitors. When the gate terminal is pulled low too quickly, the capacitance closest to the terminal goes low but the capacitance further from the terminal remains high, causing a small portion of the IGBT device to handle the full 100A of current which quickly destroys the device. The pull down circuitry therefore needs to be slower than the internal RC time constant in the gate of the IGBT. To slow down the driver, a 20Ω series resistor is integrated into the LT3485. Which Part to Use The LT3484 and LT3485 families of photoflash capacitor chargers suit about any photoflash need. The basic photoflash functionality in each part is identical and both parts are capable of operating from 2AA cells. The integrated IGBT drive and voltage output monitor differentiate the LT3485 from the LT3484, along with its higher current capabilities. The LT3484 is the smallest solution available if quenching the bulb is not needed. When using an IGBT to trigger the flash, the LT3485 offers valuable board space savings over the LT3484 by eliminating several external components. Table 1 shows the major functional differences between these seven parts. Once the decision is made on the integrated IGBT driver, choosing a current option is a matter of balancing the inherent trade-off between input current and charge time. For a given photoflash capacitor size, the device which results in the highest input current offers the fastest charge time. The limit on how much current the photoflash charger can draw is usually set by the battery technology used, and how much load they LT6555/56, continued from page 22 Demonstration Circuits Available The LT6555 and LT6556 have Demo Boards available that make evaluation of these parts a simple plug-and-play operation. To evaluate the LT6555 ask for DC858A (SSOP-24 package) or DC892A-A (QFN package). To evalu12 ate the LT6556 ask for DC892A-B (in QFN package). All three of these demo circuits have high-quality 75Ω BNC connections for best performance and illustrate high-frequency layout practices that are important to obtaining the best performance from these super-fast amplifiers. can handle. The LT3485-3 offers the fastest charge times of the chargers discussed here. The following equation predicts the charge times (T) in seconds for the seven parts: T= ( COUT • VOUT(FINAL)2 – VOUT(INIT)2 τ • VIN ) where COUT is the value of the photoflash capacitor in Farads, VOUT(FINAL) is the target output voltage, VOUT(INIT) is the initial output voltage, VIN is the battery voltage to which the flyback transformer is connected, and τ is the charge time coefficient listed in Table 1. The charge time coefficients for each part are different depending on the transformer due to differences in efficiency and average input current. The charge time coefficients are given for Kijima Musen and TDK transformers, with part numbers and typical specifications for these transformers listed in Table 2. Conclusion The LT3484 and LT3485 provide simple, efficient capacitor charging solutions for digital still cameras and integrated digital cameras in cell phones. The high level of integration reduces the amount of external components while also producing tightly controlled output voltage and average input current distributions. The three current limits in the LT3484 family and the four current limits in the LT3485 family allow for flexibility in the trade-off between input current and charge time. The LT3485 saves even more space for some applications by integrating an IGBT driver and voltage output monitor. For further information on any of the devices mentioned in this issue of Linear Technology, visit www.linear.com, use the reader service card or call the LTC literature service number: 1-800-4-LINEAR Linear Technology Magazine • December 2005 DESIGN FEATURES Fully Differential Amplifier with Rail-to-Rail Outputs Offers 16-Bit Performance at 1MHz on a by Arnold Nordeng Single 2.5V Supply Introduction With increasing levels of IC integration, and shrinking transistor geometries, A/D converter supply voltages have decreased and their inputs have been designed to process signals differentially to maintain good dynamic range. These ADCs typically run from a single low voltage supply with an optimal common mode input somewhere near mid-supply. The LT1994 facilitates interfacing to these ADCs by providing differential conversion and amplification, common mode translation of wide band, ground referenced, single-ended or differential input signals. It comes in an 8-pin MSOP or DFN package, which is pin-for-pin compatible with other commercially available fully-differential amplifiers. What sets the LT1994 apart from other fully- differential amplifiers are its low noise, low distortion, rail-to-rail output, and an input common mode range that extends to ground on power supplies as low as 2.5V. This eliminates the need for a negative power supply, and makes the LT1994 uniquely able to interface to differential input ADCs while sharing the same power supply. This saves the user system cost, and power. RI 499 3V 1.5V 0.1 F – + 2 VOCM 8 2.5V 2.75V 1.5V 0.25V 4 LT1994 VOUT 5 + – 7 RI 499 0.1 F 3 1 6 2.75V 1.5V 0.25V RF 499 VIN 5VP-P 0V –2.5V Figure 1. Common mode translation of VIN using the LT1994 Performance of LT1994 The first advantage of the LT1994 is that it can convert and level-shift ground referenced, single-ended or differential signals to VOCM pin referenced, differential output signals. Figure 1 shows how. A single-ended 5VP–P ground referenced signal (which swings 2.5V below the supply of both the ADC and the LT1994) is translated by the LT1994 from being a ground referenced signal to a differential mid-supply referenced signal. This is accomplished within the LT1994 by two feedback loops: a differential feedback loop, and a common mode Table 1. LT1994 key specifications Parameter Typical Specification Supply Current at 3V 13.3mA en – Input referred Voltage Noise 3nV/� HD2 at VIN =2VP–P, 1MHz –94dBc HD3 at VIN =2VP–P, 1MHz –108dBc Gain-Bandwidth 70MHz Slew Rate 65V/µs 0.01% Settling on a 2V step 120ns Linear Technology Magazine • December 2005 RF 499 feedback loop. Both loops have high open loop gain, around 100dB. The common mode feedback loop forces the instantaneous average of the two outputs to be equal to the voltage on the VOCM pin. Its feedback loop is internal to the LT1994. The differential feedback loop works similarly to traditional op amps forcing the difference in the summing node voltages to zero. As a result, the differential output is simply governed by the equation: VOUT = VOUT+ – VOUT – ≈ RF • VIN RI By eliminating the need for a negative supply, the LT1994 gives the user maximum dynamic range at minimal power. Since each output of the LT1994 is capable of swinging rail-to-rail, and with the LT1994’s 3nV/√Hz input referred voltage noise (see Figure 2 for the LT1994’s noise spectral density plot), applications such as the one shown in Figure 1 have a signal-to-noise ratio approaching 96dB in a 10MHz noise bandwidth. This represents a 6dB increase in dynamic range compared to single ended output rail-to-rail amplifiers 13 100 100 VS = 3V TA = 25°C 10 INPUT CURRENT NOISE DENSITY (pA/√Hz) INPUT REFERRED VOLTAGE NOISE DENSITY (nV/√Hz) DESIGN FEATURES 10 en in 1 10 100 1k 10k FREQUENCY (Hz) 1 1M 100k Linearity is enhanced using a fully differential architecture allowing the cancellation of even order harmonics. To see how this works, a pure single tone sine wave input is applied to the LT1994 as shown in Figure 1. The outputs of the LT1994 can be represented by a Taylor series expansion: across the power supplies with short traces with the V– tied directly to a low-impedance ground plane. On split supplies, additional 0.1µF high quality, low ESR, surface-mount bypass caps should be used to bypass each supply separately to a low-impedance ground plane. V V 2 VOUT+ = K 1 IN + K 2 IN + 2 2 Interfacing to ADCs 3 V V K 3 IN + K 4 IN + 2 2 Figure 2. LT1994 input referred noise spectral density V V 2 VOUT – = K 1 – IN + K 2 – IN + 2 2 –40 VS = 3V VIN = 2VP-P (SINGLE ENDED) –50 R = R = 499Ω F I V 3 V 4 K 3 – IN + K 4 – IN + 2 2 DISTORTION (dB) –60 –70 3RD HARMONIC –80 VOUT is the difference: 2ND HARMONIC –90 VOUT = VOUT+ – VOUT – V V 3 = 2K 1 IN + 2K 3 IN + 2 2 –100 –110 100k 4 1M FREQUENCY (Hz) 10M Figure 3. LT1994 disortion vs frequency with the similar noise floors. Some of the LT1994’s key specifications are tabulated in Table 1. Another benefit of fully-differential signal processing is that interference such as ground noise or power supply noise appear as common mode signals and are rejected by the internal matching and balance of the amplifier. Power supply rejection and common-mode rejection becomes limited primarily by internal transistor matching and are typically around 100dB. , leaving just the odd harmonic terms. Figure 3 shows a plot of distortion vs frequency with the LT1994 configured in the closed-loop unity gain configuration shown in Figure 1. With a 2VP–P, 1MHz, single-ended input, the 2nd harmonic measures –94dBc, and the 3rd harmonic measures –108dBc. Getting the best distortion out of the LT1994 requires careful layout, paying close attention to symmetry and balance. In single supply applications, it is recommended that high quality, low ESR, surface mount 1µF and 0.1µF caps be paralleled and tied directly The sampling process of ADCs create a sampling glitch caused by switching in the sampling capacitor on the ADC front end which momentarily “shorts” the output of the amplifier as charge is transferred between the amplifier and the sampling cap. The amplifier must recover and settle from this load transient before this acquisition period ends for a valid representation of the input signal. In general, the LT1994 settles faster from these periodic load impulses than from a 2V input step, but it is a good idea to place a small RC filter network between the output of the LT1994 and the input of the ADC to help absorb the charge injection that comes out of the ADC from the sampling process (Figure 4 shows an example of this). The capacitance of this decoupling network serves as a charge reservoir to provide high frequency charging during the sampling process, while the two resistors of the decoupling network are used to dampen and attenuate any charge kickback from the ADC. The selection of the RC time constant is trial and error for a given ADC, but the following general guidelines are recommended: Too large a resistor in the decoupling network leaving 50Ω 475Ω 499Ω 3V 50Ω 1 2 0.1µF 8 – + VOCM 24.9Ω 4 LT1994 5 + – 6 0.1µF 0.1µF 3 7 499Ω 3V AIN+ 47pF 24.9Ω VDD LT1403A-1 AIN– GND SDO CONV SCK VREF 50.4MHz 10µF 499Ω Figure 4. ADC buffering with common mode translation and differential conversion 14 DIFFERENTIAL OUTPUT MAGNITUDE (dB) LOW DISTORTION SIGNAL SOURCE 0 f = 2.8Msps –10 fSAMPLE IN = 1.001MHz –20 INPUT = 2VP-P SINGLE ENDED –30 SFDR = 93dB –40 –50 –60 –70 –80 –90 –100 –110 –120 0 0.175 0.35 0.525 0.7 0.875 1.05 1.225 1.4 FREQUENCY (MHz) Figure 5. 4096 sample FFT of the LT1994 driving a 14-bit ADC Linear Technology Magazine • December 2005 DESIGN FEATURES R3 464Ω 3V 1 C1 270pF 2 0.1µF 8 R3 464Ω 0.1µF 3 – + VOCM 4 LT1994 5 + – 7 R1 232Ω point FFT. The spurious free dynamic range is about 93dB and is limited by the non-linearities of the ADC rather than the LT1994 (The SFDR of the LTC1403A-1 is specified around 86dB at 1.4MHz). This shows that the LT1994 has no problem settling and accommodating the LTC1403A’s 39ns acquisition times. R2 232Ω C2 68pF 6 Single 3V Supply, 2.5MHz, 2nd Order Fully Differential Butterworth Filter C2 68pF R2 232Ω Figure 6 shows a low noise, single supply, butterworth active filter with a 2.5MHz bandwidth suitable for antialiasing applications. The differential output spot noise at 50kHz is about 7nV/√Hz, and the amplifier provides about 40dB of stopband rejection at 25MHz. The filter’s frequency response is shown in Figure 7. The filter’s low frequency gain is set by the ratio of R2 to R1. If a different cutoff frequency is desired, the capacitors C1, and C2 can easily be scaled inversely with cutoff frequency. Figure 6. Low noise differential active RC filter insufficient settling time creates a voltage divider between the dynamic input impedance of the ADC and the decoupling resistors. Too small of a resistor possibly prevents the resistor from properly dampening the load transient caused by the sampling process, prolonging the time required for settling. Start with 25Ω on each output to decouple the ADC input capacitance. Then choose a capacitance (taking account of the sampling capacitance), which gives the amplifier time to settle to desired accuracy during the acquisition period. In 16-bit applications, this typically requires a minimum of 11 RC time constants. The capacitor chosen should have a high quality dielectric (for example, C0G multi-layer ceramic). Figure 4 shows the LT1994 driving the LTC1403A-1, a 14-bit ADC, sampling at 2.8MHz on a single 3V supply. Figure 5 shows its 4096- Gain-of-2 Amplifier (No resistors required) Figure 8 shows the LT1994 configured in circuit configuration in which the output consists of an in-phase and an out-of-phase representation of the input signal. The circuit has the benefit of having high input impedance. The input-to-output transfer function is governed by the equation: VOUT = 2 • VIN 0.1µF 1V 2 8 1V 0V 3 0V – + VOCM 4 LT1994 5 + – 7 VIN VS = 3V –10 –20 –30 –40 –50 –60 0.001 0.01 0.1 1 FREQUENCY (MHz) 10 100 Figure 7. Differential filter response The circuit works well enough, but the consequence of such a configuration is that it reflects the performance of the common mode path, rather than the differential path. Because of this, the output does not have the benefit of the differential noise (3nV/√Hz), but rather is swamped by the common mode noise of 15nV/√Hz gained up by a factor of two (30nV/√Hz). This is a consequence of mismatch in feedback factors from the LT1994 outputs to their respective inputs. In fact, whenever the two feedback paths from the output to the input mismatch, and to the degree they mismatch, common mode noise is converted to differential noise at the output. eNO(DIFF ) = 2eN( VOCM) (βF1 – βF 2 ) (βF1 + βF 2 ) where βF1, and βF2 are the two feedback factors from each output to their respective input. Conclusion 5V 1 0 GAIN (dB) R1 232Ω 6 VOUT –1V 1V 0V 0.1µF –1V –1V –5V The LT1994’s low noise, low distortion, and high performance make it an ideal amplifier for interfacing with single supply ADCs. Its rail-to-rail outputs, low distortion, and 3nV/√Hz input referred voltage noise maximize dynamic range, and its ability to common mode to ground eliminates the need for a negative supply in single supply systems, saving cost and power. Figure 8. Gain of two (no resistors required) For more information on parts featured in this issue, see http://www.linear.com/designtools Linear Technology Magazine • December 2005 15 DESIGN FEATURES Negative High Voltage Hot Swap Controllers Incorporate an Accurate Supply Monitor and Power Module Sequencing by Kevin Wong Introduction Typical Application When a circuit board is inserted into a live backplane slot, discharged supply bypass capacitors on the board can draw large transient currents from the system supplies. In high-voltage systems like the –48V backplanes prevalent in high reliability telecom systems, such transients can reach hundreds of amps and damage connector pins, PCB traces and board components. In addition, current spikes can cause voltage glitches on the power bus, causing other boards in the system to reset. This is particularly unacceptable in telecom systems where the ability to safely Hot Swap modules is a primary system requirement. The LTC4253A and LTC4253A-ADJ facilitate safe board insertion and removal from a live backplane by applying power in a controlled manner. Running off a simple, fast responding – 48V RTN (LONG PIN) shunt regulated supply that allows very high voltage operation, they are uniquely suited for applications on the –48V bus. User programmable, high accuracy undervoltage and overvoltage detectors act as supply monitors and ensure the supply is stable and within tolerance before applying power to the load. An inrush current control loop then takes over, resulting in a controlled startup current profile. When the external pass transistor is fully enhanced, Power Good status outputs then allow time adjustable or load feedback enabled sequencing of up to four load modules. Short circuits or excessive supply current events trigger protective circuits which quickly isolate the fault to prevent glitching of the backplane supply. With all these features, these devices offer a comprehensive solution for –48V Hot Swap applications. R3 22k RIN 10k 20k(1/4W)/2 VIN1 R5 2.2k R4 2.2k CIN 1µF PUSH RESET – 48V RTN (SHORT PIN) Q2 FZT857 Figure 1 shows a typical –48V Hot Swap application using the LTC4253A. The LTC4253A floats on the negative rail and uses an internal shunt regulator that together with RIN and CIN, regulates VIN to about 13V above the negative rail. The MOSFET N-channel transistor Q1 is placed in the power path to control turn on and turn off with input from resistor RS which senses the load current. RC and CC provides compensation for the current limit loop. R1 and R2 form a resistive divider that allows MOSFET turn on only when the –48V supply is between the user programmed undervoltage and overvoltage thresholds. The resistive divider R1/R2 connects to the –48V RTN rail via a short pin so that during plug in, the MOSFET is held off by the undervoltage condition until the longer power pins are properly seated. The five opto-couplers form an CL 100µF R6 2.2k + POWER MODULE 1 POWER MODULE 2 EN EN POWER MODULE 3 EN CSS 33nF UV PWRGD1 OV PWRGD2 RESET PWRGD3 EN2 SS DRAIN R9 47k CSQ 0.1µF TIMER CT 0.68µF VEE EN2 VIN1 RD 1M Q1 IRF530S GATE SENSE POWER MODULE 2 OUTPUT EN3 EN3 SQTIMER – 48V (LONG PIN) † † LTC4253A C1 10nF R1 30.1k 1% † VIN R2 392k 1% RC 10Ω CC 10nF RS 0.02Ω VIN1 R7 POWER MODULE 1 OUTPUT R8 † † †MOC207 Figure 1. –48V/2.5A Hot Swap controller with opto-isolated Power Good sequencing 16 Linear Technology Magazine • December 2005 DESIGN FEATURES –48RTN (SHORT PIN) R3 392k 1% OV SHUTDOWN = 71.8V OV RECOVERY = 70.3 UV RECOVERY =38V UV SHUTDOWN = 34V LTC4253A UV R2 4.32k 1% C1 10nF OV R1 30.1k 1% –48V (LONG PIN) VEE Figure 2. Undervoltage and overvoltage resistive divider connection to the LTC4253A electrically isolated interface between LTC4253A and the load modules for power sequencing. Undervoltage and Overvoltage Detection The LTC4253A and LTC4253A-ADJ have 1% accurate undervoltage and overvoltage threshold detectors that can be set to any desired power supply range. This level of accuracy and flexibility allow these parts to be easily designed to conform to any operating UVL UV 1 VIN ranges specified by the various prevailing –48V standards. In the LTC4253A, an UV hysteretic comparator detects undervoltage conditions at the UV pin, with the following thresholds (with respect to VEE): ❑ UV low-to-high (VUV) = 3.08V ❑ UV low-to-high hysteresis (VUVHST) = 0.324V An OV hysteretic comparator detects overvoltage conditions at the OV pin, with the following thresholds(with respect to VEE): ❑ OV low-to-high (VOV) = 5.09V ❑ OV low-to-high hysteresis (VOVHST) = 0.102V The undervoltage recovery and overvoltage shutdown thresholds are designed to match the standard telecom operating range of 43V to 71V with the UV and OV pins shorted as in Figure 1. The undervoltage shutdown and overvoltage recovery thresholds are then 38.5V and 69.6V respectively. The UV and OV pins can also be separated for implementing different operating ranges as shown in Figure 2. UVIN – 3.08V + OVL UVD 2 3 OV 4 VLKO 1 VIN 36V (UNDERVOLTAGE SHUTDOWN VOLTAGE) 38V (UNDERVOLTAGE RECOVERY VOLTAGE) (–48V RTN) SHORT PIN UVD OVD UVLO UNDERVOLTAGE SHUTDOWN NORMAL OPERATION a. Undervoltage comparator + OVD 3 71V (OVERVOLTAGE SHUTDOWN VOLTAGE) OVIN UNDERVOLTAGE SHUTDOWN OVIN 4 OV 0VL UVLO NORMAL OPERATION 69V (OVERVOLTAGE RECOVERY VOLTAGE) VOVLO 5.08V VOVHI 5.09V OVL OV UV – VLKO (–48V RTN) SHORT PIN UVL UV UVL 5.09V 2 VUVLO 3.08V VUVHI 3.08V UVIN The LTC4253A-ADJ offers additional flexibility in allowing the user to implement any required undervoltage recovery, undervoltage shutdown, overvoltage recovery and overvoltage shutdown thresholds. It achieves this by having two extra pins UVL and OVL connected to the internal comparators as shown in Figure 3. The undervoltage comparator has multiplexed inputs from UVL and UV, which is tapped off a resistive string across the power supply as in Figure 4. When comparator output UVD is high, UV is multiplexed to the comparator input UVIN. When UVD is low, UVL is multiplexed to UVIN. The overvoltage comparator similarly implements the overvoltage function. The various thresholds to note are (with respect to VEE): ❑ UV low-to-high (VUVHI) = 3.08V ❑ UVL high-to-low (VUVLO) = 3.08V ❑ OV low-to-high (VOVHI) = 5.09V ❑ OVL high-to-low (VOVLO) = 5.08V By tapping UVL, UV, OVL and OV off a resistive string across the power supply, undervoltage recov- OVERVOLTAGE SHUTDOWN NORMAL OPERATION b. Overvoltage comparator Figure 3. The LTC4253A-ADJ UV/OV detector block Linear Technology Magazine • December 2005 17 DESIGN FEATURES – 48V RTN (LONG PIN) + RIN 10k 20k(1/4W)/2 – 48V RTN (SHORT PIN) 294k 1% 2.74k 1% CIN 1µF R5 VIN EN2 EN3 LTC4253A-ADJ UVL R4 2.37k 1% R3 2.1k 1% VIN UV OVL PWRGD1 OV PWRGD3 R2 20k 1% CSQ 0.1µF POWER MODULE 1 EN POWER MODULE 2 EN POWER MODULE 3 EN † POWER MODULE 4 EN † R10 3k † RD 1M VOUT SS TIMER VEE Q1 IRF530S GATE SENSE SEL RC 10Ω CC 10nF CT 0.68µF – 48V (LONG PIN) R9 100k C3 0.1µF R8 100k DRAIN SQTIMER R1 20k 1% R7 100k PWRGD2 RESET C1 10nF CSS 33nF R6 100k C2 100µF RS 0.02Ω † FMMT493 Figure 4. A –48V/2.5A Hot Swap controller with transistor enabled Power Good sequencing ery = 43V, undervoltage shutdown = 39V, overvoltage recovery = 78V and overvoltage shutdown = 82V are implemented in Figure 4. Any required supply operating range can thus be implemented with great accuracy. dI/dt Soft Start The LTC4253A offers a current soft start pin (SS) that acts as the reference for the analog current limit amplifier (VACL = VSS/20). By attaching a capacitor at the SS pin, the analog current limit threshold ramps up in an exponential profile with an RC time constant equal to 50kΩ • CSS. The analog current limit amplifier forces the inrush current to follow this profile when the GATE pin rises above the external MOSFET threshold and turns on the MOSFET. In this way, inrush current ramps up with a controlled slew rate (dI/dt) that is approximately fixed and adjustable by CSS (Figure 5a). Controlling the load current slew rate reduces system EMI and disturbances to the supply rail during startup. The LTC4253A-ADJ offers an additional mode when the SEL pin is held low (it has an internal pullup to VIN of 20μA ). In this mode, the SS pin is servoed from the time the GATE pin is released until it clears the external MOSFET threshold and turns the MOSFET on. The result is that the LTC4253A-ADJ enters analog current limit with VACL ramping up from close to zero. The resultant inrush current GATE 10V GATE 10V SS 1V SS 1V SENSE 50mV SENSE 50mV VOUT 50V VOUT 50V 1ms/DIV 1ms/DIV a. dI/dt inrush current control using the LTC4253A or LTC4253A-ADJ with SEL = 1 (Figure 1 circuit) b. Enhanced dI/dt inrush current control using the LTC4253A-ADJ with SEL = 0 (Figure 4 circuit) Figure 5. dI/dt soft-start waveforms 18 profile presents a smooth ramp up from zero and the load current slew rate is able to maintain an approximately fixed dI/dt gradient right from turn on (Figure 5b). This dI/dt gradient is similarly adjustable by CSS. Power Good Sequencing The LTC4253A has three sequenced PWRGD outputs and two enable (EN) inputs. This allows three load modules to be enabled sequentially, minimizing any sudden load power demand on the backplane supply. The three load modules can be timer sequenced as in Figure 4 where the EN pins are enabled by tying them to VIN. The three PWRGD signals assert sequentially with a fixed time delay adjustable by capacitor CSQ (approximate TD = 600ms • (CSQ/1µF)). The load modules can also be load feedback sequenced as in Figure 1 where the load modules control the EN pin inputs. In this way when Load Module 1 is enabled by PWRGD1 and fully started up, it can signal back via the EN2 input to enable Load Module 2, which is enabled after one SQTIMER delay ramp. Load Module 2 can similarly enable Load Module 3 when it is ready. This mode of sequencing is shown in Figure 6. The interface between the Hot Swap controller and the load modules is implemented with opto-couplers as in Figure 1 to take care of the differing Linear Technology Magazine • December 2005 DESIGN FEATURES signal common. If the load modules’ EN inputs have sufficient protection against negative bias current, a simpler NPN interface can be implemented as in Figure 4. Figure 7 highlights an additional feature of the LTC4253A-ADJ. The PWRGD1 signal only activates after one SQTIMER ramp delay from the time GATE goes high and DRAIN goes low. This feature can be exploited to provide an additional EN1 signal so up to four load modules can be sequenced as in Figure 4. VOUT 50V/DIV VOUT 50V/DIV GATE 10V/DIV GATE 10V/DIV SQTIMER 5V/DIV SQTIMER 5V/DIV PWRGD1 10V/DIV EN1 2V/DIV PWRGD1 10V/DIV EN2 10V/DIV Short Circuit Operation Current faults are controlled in three stages using three thresholds: 50mV for a timed circuit breaker function, 60mV for an analog current limit loop and 200mV for a fast comparator that limits peak current in the event of a catastrophic short-circuit. This three-stage fault current minimizes backplane supply disturbances due to current faults. A voltage across the SENSE resistor (RS) of greater than 50mV triggers TIMER to source 200µA into a timing capacitor CT. CT eventually charges to a 4V threshold and the part latches off. If the fault goes away before CT reaches 4V, CT slowly discharges (5µA). A low impedance short can glitch the voltage across RS above 200mV. This triggers a fast comparator that asserts a hard pulldown on the MOSFET gate to quickly bring the voltage across RS PWRGD2 10V/DIV PWRGD2 10V/DIV PWRGD3 10V/DIV EN3 10V/DIV 50ms/DIV PWRGD3 10V/DIV 50ms/DIV Figure 6. The LTC4253A controlling the turn on of three load modules using Load Feedback Sequencing back below 200mV. This effectively limits the initial transient fault current. An analog current limit loop then controls the voltage across RS to 60mV until TIMER reaches 4V (see Figure 8). RD in Figure 1 allows the MOSFET drain to pump current into the DRAIN pin internally clamped at around SUPPLY RING OWING TO MOSFET TURN-OFF –48V RTN 50V SENSE 200mV SUPPLY RING OWING TO CURRENT OVERSHOOT TRACE 1 ONSET OF OUTPUT SHORT CIRCUIT TRACE 2 GATE 10V FAST CURRENT LIMIT ANALOG CURRENT LIMIT TIMER 5V CTIMER RAMP TRACE 3 TRACE 4 LATCH OFF 0.5ms/DIV Figure 8. Output short-circuit waveforms Linear Technology Magazine • December 2005 Figure 7. The LTC4253A-ADJ controlling the turn on of four load modules using Timing Sequencing 6V. This current is multiplied up by eight times and added to the 200μA circuit breaker TIMER pullup current. This adds a component to the circuit breaker timeout period that is linearly proportional to the VDS of the MOSFET, thus allowing the MOSFET to be designed to function closer to its SOA limits under different conditions. Conclusion The LTC4253A and LTC4253A-ADJ inherit the proven capabilities of Linear Technology’s –48V Hot Swap family and add enhanced features. Chief among these is a highly flexible and 1% accurate undervoltage and overvoltage detection capability. Additional features include enhanced slew rate controlled inrush current profile and the ability to sequence up to four load modules. The LTC4253A is available in a 16-pin SSOP package and is completely pin compatible to the LTC4253. The LTC4253A-ADJ is available in a 20-pin SSOP package as well as a 20-pin 4mm × 4mm QFN package. Authors can be contacted at (408) 432-1900 19 DESIGN FEATURES Simplify High-Resolution Video Designs with Fixed-Gain by Jon Munson Triple Multiplexers Introduction V+ The LT6555 and LT6556 triple video multiplexers offer up to 750MHz performance in compact packages, requiring no external gain-setting resistors to establish a gain of two or unity. A single integrated circuit, in a choice of either 24-lead SSOP or 24-contact QFN (4mm × 4mm), performs fast switching between a pair of three-channel video sources, such as RGB or component HDTV. The LT6555 provides a built-in gain of two that is ideal for driving back-terminated cables in playback or signal routing equipment. The LT6556 provides a unity-gain function, in the same footprints, that is ideal as an input selector in high-performance video displays and projectors. The three video channels exhibit excellent isolation between themselves (50dB typical at 100MHz) and the inactive inputs (70dB typical at 100MHz) for the highest quality video transmission. Excellent channel-to-channel gain-matching preserves high fidelity color balance. The increasing popularity of the UXGA professional graphics format (1600 × 1200), which generates a whopping 200-megapixel-per-second flow, has put exceptional demands on the frequency response of video amplifiers. For instance, pulse-amplitude RINA GINA BINA LT6555 75Ω 75Ω 75Ω 75Ω AGND 100kHz Figure 2. Wide frequency response of circuit in Figure 1 20 1GHz 75Ω ×2 RINB GINB BINB 75Ω ROUT GOUT 75Ω 75Ω ×2 75Ω 75Ω BOUT SELECT A/B 75Ω ENABLE DGND V – 6555 TA01a Figure 1. The LT6555 in an RGB cable driving multiplexer circuit waveforms like those of RGB baseband video, generally require reproduction of high-frequency content to at least the 5th harmonic of the fundamental frequency component, which is 2.5 times the video pixel rate, accounting for the 2-pixels-per-fundamentalcycle relationship. This means that UXGA requires a flat frequency response to beyond 0.5GHz! The wide bandwidth performance of the LT6555 and LT6556 makes them ideally suited to such high performance video applications. response and crosstalk anomalies can plague the circuit development process. The LT6555 and LT6556 conveniently solve these problems by providing internal factory-matched resistors and an efficient 3-channel, 2-input group, flow-through layout arrangement. Figure 1 shows the typical RGB cable driver application of an LT6555, and its excellent frequency and time response plots are shown in Figures 2 and 3 (as implemented on demo 1.8 Easy Solution for MultiChannel Video Applications Baseband video generated at these higher rates is processed in either native red, green and blue (RGB) domain or encoded into component luma plus blue and red chroma channels (YPbPr); three channels of information in either case. With frequency response requirements extending to beyond 500MHz, amplifier layouts that require external resistors for gain setting tend to be real-estate inefficient, and frequency 1.6 1.4 1.2 OUTPUT (V) 0dB 3dB/DIV 75Ω ×2 1.0 0.8 0.6 0.4 0.2 VIN = 0V TO 700mV VS = ±5V RL = 150Ω TA = 25°C 0 –0.2 –0.4 0 2 4 6 8 10 12 14 16 18 20 TIME (ns) Figure 3. Fast pulse response of circuit in Figure 1 Linear Technology Magazine • December 2005 DESIGN FEATURES V+ V+ BIAS V EN 1k 40k VREF TO OTHER OUTPUT STAGES 40k + V 46k 770Ω – OUT VREF SEL INA V+ V+ INB 100Ω 100Ω V– 360Ω 360Ω 360Ω 360Ω AGND V– DGND VREF VREF SELECT TO OTHER INPUT STAGES V– V– Figure 4. Simplified internal circuit functionality of the LT6555 and LT6556 circuit 892A-A). Frequency markers in Figure 2 show the small-signal –0.5dB response beyond 500MHz and –3dB response above 600MHz. The LT6556, when used to drive high impedances, provides bandwidth to 750MHz, though the LT6556 demo circuit 892AB uses 75Ω back termination (rather than 1kΩ), resulting in performance similar to the LT6555. Taking a Look at the Internal Details The LT6555 and LT6556 integrate three independent sections of circuitry that form classic current-feedback amplifier (CFA) gain blocks, but with switchable input sections, all implemented on a very high-speed fabrication process. The diagram in Figure 4 shows the equivalent internal circuitry (one LT6555 section shown). Feedback resistors are provided on-chip to set the closed-loop gain to either unity or two, depending on the part. The nominal feedback resistances are chosen to optimize flat frequency response. The LT6555 is intended to drive back-terminated 50Ω or 75Ω cables (for effective loading of 100Ω to 150Ω respectively), while the LT6556 is designed to drive ADCs or other high impedance loads (characterized with 1kΩ as a reference loading condition). Linear Technology Magazine • December 2005 Common to all three CFAs in each part is a bias control section with a power-down command input. The input select logic steers bias current to the appropriate input circuitry, enabling the input function of the selected signal. The shutdown function includes an internal on-chip pull-up resistance to provide a default disable command, which when invoked, reduces typical power consumption to less than 125µA for an entire threechannel part. During shutdown mode the amplifier outputs become high impedance, though in the case of the LT6555, the feedback resistor string to AGND is still present. The parts come into full-power operation when the enable input voltage is brought within 1.3V above the DGND pin. The typical on-state supply current of about 9mA per amplifier provides for ample cable-drive capacity (>40mA) and ultra-fast 2.2V per nanosecond slew rate performance. Expanding MUX Input Selection The power-down feature of the LT6555 and LT6556 may be used to control multiple ICs in a configuration that provides additional input selections. Figure 5 shows a simple 4-input RGB selecting cable driver using two LT6555 devices with the enable pins driven by complementary logic signals. The shared-output connections between the devices need to be kept as short as possible to minimize printed-circuit parasitics that might affect frequency response. This circuit would be ideal in an A/V control-unit for driving the component-video output, for example. The same basic expansion concept applied to an LT6556 pair would be ideal at the input section of a four-source HD video display. Operating with the Right Power Supplies The LT6555 and LT6556 require a total power supply of at least 4.5V, but depending on the input and output swings required, may need more to avoid clipping the signal. The LT6556, having unity gain, makes the analysis simple—the maximum output swing is (V+ – V- – 2.6)VP–P and governed only by the output saturation voltages. This means a total supply of 5V is adequate for standard video (1VP–P). For the LT6555, extra allowance is required for load-driving, so the output swing is (V+ – V- – 3.8)V. This means a total supply of about 6V is required for the output to swing 2VP–P, as when driving cables. For best dynamic range along with reasonable power consumption, a good choice of supplies would be ±3V for the LT6556 and +5V/–3V for the LT6555. 21 DESIGN FEATURES RED 1 GREEN 1 BLUE 1 75Ω V+ LT6555 #1 IN1A IN1B 5V OUT1 ×2 75Ω 75Ω RED 2 GREEN 2 BLUE 2 IN2A OUT2 ×2 IN2B 75Ω 75Ω IN3A 75Ω OUT3 ×2 IN3B AGND DGND SEL V RED 3 GREEN 3 BLUE 3 75Ω VREF EN 75Ω – 75Ω –3V 5V 75Ω V LT6555 #2 IN1A 75Ω OUT1 ×2 IN1B 75Ω + 75Ω ROUT GOUT BOUT 75Ω 75Ω RED 4 GREEN 4 BLUE 4 IN2A OUT2 ×2 IN2B 75Ω 75Ω IN3A 75Ω OUT3 ×2 IN3B AGND DGND SEL VREF EN SEL0 V– NC75Z14 SEL1 SEL1 SEL0 OUTPUT 0 0 1 0 1 2 1 0 3 1 1 4 –3V Figure 5. A 4-to-1 video multiplixer using the shutdown feature for expansion Since many systems today lack a negative supply rail, a small LTC19833 solution can be used to generate a simple –3V rail for local use, as shown in Figure 6. The LTC1983-3 solution is more cost effective and performs at high frequencies better than ACcoupling and resistor network biasing techniques that might otherwise be employed. For example, Figure 7 shows the typical AC-coupling networks used when operating from a single supply. With six input networks and three large output capacitors required, the AC-coupled method uses more board space and adds parasitics to the signal path that can degrade frequency response. continued on page 12 OFF ON VOUT VIN LTC1983-3 (SOT23-6) SHDN VOUT = –3V IOUT = UP TO 100mA COUT 10µF GND C– C+ 7V TO 12V INPUT 22µF* IN 80.6Ω CFLY 1µF Figure 6. Generating a local –3V supply with four tiny components 22 6.8k 2.2k AGND LT6555 OR LT6556 OUT 75Ω 220µF** + VIN 3V TO 5.5V CIN 10µF * AVX 12066D226MAT ** SANYO 6TPB220ML 75Ω NOTE: ONLY ONE INPUT AND ONE OUTPUT SHOWN Figure 7. AC-coupling techniques for single-supply operation Linear Technology Magazine • December 2005 DESIGN FEATURES High Efficiency, Monolithic Synchronous Buck-Boost LED Driver Drives up to 1A Continuous Current by Aspiyan Gazder Introduction The LTC3454 is a synchronous buckboost DC/DC converter, designed for driving a single high power LED with regulated currents up to 1A from a single Li-Ion battery. Switching converters are typically used to regulate a voltage, but LEDs require constant current to generate predictable light output. The LTC3454 uses an autozero transconductance error amplifier in its regulation loop to accurately control LED current. The LED current can be set to one of four levels, including shutdown, using two external resistors and dual enable pins. In shutdown no current is drawn. The wide VIN range of a Lithium-Ion battery (2.7V to 4.2V) requires that a converter be able to both step-up and step-down the input voltage when the LED forward voltage is within the range of the battery discharge profile. The LTC3454 LED driver efficiently performs step-up and step-down conversion via four internal switches. The regulator operates in synchronous buck, synchronous boost or buckboost mode, depending on VIN and the LED forward voltage. Transitions between modes are automatic and smooth. The LTC3454 operates at a high fixed frequency of 1MHz, which re- L2 SW1 VIN 2.7V TO 5.5V C6 R9 10µF 20.5k 1% C7 10µF D1 LED EN2 ISET2 VC C8 D1: LXCL-LW3C L2: CDRH6D28-5R0NC EN1 EN2 0 1 0 1 0 0 1 1 ILED 0 (SHUTDOWN) 150mA 850mA 1000mA Figure 1. LTC3454 used in a typical application duces inductor size and eases output filtering. Application Figure 1 shows the LTC3454 driving a high power LED in torch and flash modes. Only six external components are required in this application. Efficiency, PLED/PIN, greater than 90% is possible over the entire usable range of a Li-Ion battery (see Figure 2). The LTC3454 has two enable pins that control two current setting amplifiers. A resistor connected from an ISET pin to GND programs the LED current to: ILED = 3850 • 0.8 , RISET when the current setting amplifier is enabled via its EN pin. When both enable pins are asserted, the net LED AI = 3850 + INTERNAL CURRENT SETTING AMPLIFIER 1 I SET1 I ILED = 3850 • I Σ ISET2 – ILED = 150mA RISET1 90 EFFICIENCY (%) ISET1 LTC3454 EXPOSED PAD 100 ISET1 85 80 VOUT EN1 R8 3.65k 1% 800mV 95 SW2 VIN ILED = 1mA 800mV + INTERNAL CURRENT SETTING AMPLIFIER 2 75 70 – 65 T = 25°C A EFFICIENCY = (VOUT – VLED)ILED/VIN • IIN 60 3.5 2.7 3.1 3.9 4.3 4.7 5.1 RISET2 5.5 VIN (V) Figure 2. Efficiency for circuit of Figure 1 Linear Technology Magazine • December 2005 ISET2 Figure 3. Two current setting amplifiers give the user the flexibility to choose more than one non-zero current level. 23 DESIGN FEATURES VIN COUT CIN ILED AUTO ZERO gm ISET R – PWM AND GATE MULTIPLEXING LOGIC + – R C/D PAIR PWM COMPARATOR + VC CVC + – A/B PAIR PWM COMPARATOR Figure 4. An auto-zeroing transconductance amplifier maintains loop regulation. current is the sum of each individually programmed current. Figure 3 shows schematically how the LED current is programmed. Autozeroing Transconductance-AmplifierBased Current Regulation The LTC3454 employs an auto-zeroing transconductance amplifier in its regulation loop, as shown in Figure 4. The autozero amplifier topology nullifies any offset at its input, allowing an accurate LED current to be achieved with very low common mode input voltage levels, resulting in high PLED/ PIN efficiency. The regulation voltage present at the LED pin can be as low as 100mV at 100mA of LED current. Synchronous Buck-Boost DC/DC Converter The LTC3454 can drive an LED at up to 1A continuous current. LEDs that can be driven with such high current typically have forward voltage drops of 3.3V – 3.6V. When powered from a single Li-Ion battery (2.7V to 4.2V), as in the case of handheld battery powered applications, neither a pure buck nor a pure boost solution can efficiently regulate the LED current. A pure buck would dropout at lower battery voltages, causing a lower than programmed LED current to flow. At high battery voltages, a pure boost converter would regulate a higher output voltage than necessary, result24 ing in low efficiency. The buck-boost converter can efficiently regulate LED current over the entire Li-Ion battery range. The autozero amplifier topology nullifies any offset at its input, allowing an accurate LED current to be achieved with very low common mode input voltage levels, resulting in high PLED/PIN efficiency. The control voltage, VC, determines the region of operation of the buckboost converter. The gate drives of the internal power switches A, B, C and D are controlled by the logic block (Figure 4). A patented gate drive multiplexing scheme enables smooth SW1 2V/DIV transition between buck and boost modes and through the four-switch region. In buck mode, the duty cycles on gate drives of switches A and B are controlled while switch D is turned on continuously. In boost mode, duty cycles of switches C and D are controlled, while switch A is on continuously. Using synchronous rectifier switches B and D instead of catch diodes helps improve efficiency. This scheme requires that the synchronous rectifier switch and the main switch are not turned on simultaneously. A cross conduction delay prevents this condition from occurring. The LTC3454 has a break before make time of approximately 30ns. During this time the current conduction path is completed through the body diodes of the switches. In the case of forward current flow from the SW1 pin to the SW2 pin through the inductor, the body diode of NMOS switch B conducts in buck mode. The SW1 node is pulled a diode drop below ground. Likewise, in boost mode, the body diode of PMOS switch D conducts during the switch C and switch D switching, but node SW2 now flys above VOUT by a diode drop. Body diodes of the main switches A and C conduct during reverse current flow. Figure 5 shows the switch waveforms in the buck-boost mode. The LTC3454 has both forward and reverse current limiting—requiring no external sense resistors. If the peak input current exceeds approximately 3.4A, forward current limit is tripped and switches B and D are turned on for the rest of the cycle. The reverse current limit is tripped when current flowing from switch D through the inductor to the SW1 node exceeds approximately 250mA and switches A and C are turned on for the rest of the cycle. Robust Design: Can Tolerate Open and Shorted LED Conditions 0V SW2 2V/DIV 0V VIN = 3.6V VOUT = 3.5V Figure 5. Switching waveforms in buck-boost mode If the LED faults as an open circuit, the regulation loop drives VC higher, which has the effect of raising the output voltage. A safety amplifier—a continued on page 46 Linear Technology Magazine • December 2005 DESIGN FEATURES Constant Current from 3A DC/DC Converter with 2 Rail-to-Rail by Daniel Chen Current Sense Amplifiers Introduction Traditional DC/DC converters use voltage feedback for constant output voltage regulation. There are many applications, however, that need to regulate a constant output current. Driving LEDs in series is one such application. The LT3477 combines a traditional voltage feedback loop and two unique current feedback loops to operate as a constant-current, constant-voltage source. It is a current mode, 3A DC/DC converter with dual rail-to-rail 100mV current sense amplifiers that can be configured as a buck mode or buck-boost mode LED driver. It is versatile enough to also be configured as an input-output current limited boost, SEPIC or inverting converter. Both current sense voltages can be adjusted independently using the IADJ1 and IADJ2 pins. With two identical precision current sense amplifiers, the LT3477 can provide an accurate input current limit as well as an accurately regulated output current. With an input voltage range of 2.5V to 25V, the LT3477 works from a variety of input sources. The 42V switch rating allows an output voltage of up to 41V to be generated, easily The unique feature of the three-feedback-loop topology (two current and one voltage) is that it can support constant voltage and/or constant current applications. VADJ – + + A1 VADJ – + + A2 IA1 – + IA2 – IADJ2 FBP + FBN – VREF A3 – + A4 R S VA Σ SLOPE VREF 1.25V Q1 Q – ISN2 SW VC + IADJ1 ISP2 Figure 1 shows a block diagram of the LT3477. The voltage error amplifier has both FBP and FBN pins to allow a positive or negative output configuration. With the addition of two current feedback control loops, amplifier A3 becomes a summing point for three feedback loops. Depending on configuration, any of the loops can take over feedback control by sourcing or sinking current at the VC node. The unique feature of the three-feedbackloop topology (two current and one voltage) is that it can support constant voltage and/or constant current applications. + ISN1 How It Works driving up to ten white LEDs in series. The buck mode LED configuration is capable of driving multiple ten-LED strings in parallel if external current mirroring circuitry is added. The switching fequency is adjustable from 200kHz to 3.5Mhz, set by SS ISP1 a single resistor. The available high operating frequencies allow the use of low profile inductors and capacitors—important in applications where space is a premium. The wide available range makes it possible to optimize size and efficiency for your application. OSCILLATOR SHDN VIN RT Figure 1. LT3477 block diagram Linear Technology Magazine • December 2005 25 DESIGN FEATURES 120 PVIN 32V VCM = 10V C1 2.2µF R1 0.1Ω VOLTAGE SENSE (mV) 100 80 D2 1A 60 D5 • • • LED STRING C2 1µF 40 L1 20 0 0 100 200 300 400 500 600 700 800 IADJ VOLTAGE (mV) VIN 3.3V Figure 2. Current sense amplifier voltage sense level vs IADJ pin voltage R2 1k D2 ISN1 R1 0.1Ω D1 R5 200k SW1 ISN2 FBP RT GND SS C4 33nF R3 22k Schottky diode is connected between the SW and PVIN nodes. With high side current sense, the boost converter is effectively converted into a buck LED converter, which increases the part’s power handling capability. In addition, the VIN pin, which provides the chip operating current, can be tied to a lower voltage level such as 3.3V. As a result, the power consumption on the chip itself is also reduced, thus improving overall efficiency. Over 90% efficiency can be readily achieved with a wide range of inductor and frequency selections. FBN 90 85 LT3477 ISP2 ISN2 VREF RT GND C3 10nF SS C4 33nF VIN = 8V 80 EFFICIENCY (%) VC FBP ISP2 VREF LED drivers use a grounded current sense resistor to regulate current, but the LT3477 current sense amplifiers work in a high side sense scheme, so the sensed voltage for current feedback no longer needs to be ground referred. In buck mode configuration, the sense resistor is placed right at the input supply. The LEDs are placed between the sense resistor and the inductor and the VIN IADJ1 IADJ2 SHDN LT3477 Figure 3. Buck mode high current LED driver L1 4.7µH SHDN R6 10k C1: NIPPON UNITED CHEMICON NTS40X5R1H225M C2: TAIYO YUDEN GMK316BJ105ML C3: TAIYO YUDEN LMK316BJ475 L1: TOKO D1OFA814AY-330M D1: DIODES INC DFLS140 D3 LED BRIGHTNESS CONTROL 0mV TO 650mV FBN C5 4.7nF Buck Mode High Current LED Driver Figure 3 shows a typical application to drive high current LEDs. Traditionally, ISP1 VIN IADJ1 IADJ2 SHDN R5 309k SW ISN1 VC Applications C1 3.3µF C3 3.3µF SHDN Current sense levels are adjustable via sense resistors at the IADJ1 and IADJ2 pins. The default sense voltage is 100mV for each current sense amplifier if the IADJ1 and IADJ2 pins are tied to a potential higher than 650mV. If the potentials at the IADJ1 and IADJ2 pins are lower than 625mV, the LT3477 linearly adjusts the current sense level. Figure 2 shows the voltage sense level vs the IADJ pin voltage. For LED drivers, IADJ1 and IADJ2 pins can be used to adjust LED current levels. Rail-to-rail current sense amplifiers allow flexible current sense schemes. VIN 2.7V TO 16V ISP1 D1 R3 18k R6 10k C2 4.7µF 75 VIN = 4.2V 70 65 60 55 C1: TAIYO YUDEN LMK316BJ335ML C2: MURATA GRM31CR71E475KA88L D1: DIODES, INC. B320A L1: TOKO FDV0630-4R7M Figure 4. Buck-boost LED driver 26 50 0 0.2 0.4 0.6 IOUT (A) 0.8 1.0 Figure 5. Buck-boost LED driver efficiency Linear Technology Magazine • December 2005 DESIGN FEATURES L2 10µH C1 3.3µF D1 VIN IADJ1 IADJ2 R1 10k LT3477 ISP2 VC RT GND FBP SS C3 33nF C4 4.7nF 330mA R6 0.3Ω ISN2 VREF R4 1k 80 FBN SHDN SHDN 85 C2 3.3µF R2 200k SW ISN1 ISP1 90 EFFICIENCY (%) VIN 5V R3 22k LED1 55 LED2 50 LED3 LED4 VIN 3V TO 16V C1 3.3µF ISP1 ISN1 VIN IADJ1 IADJ2 SHDN SHDN 0.4 0.3 R4 0.15Ω 5.5V 670mA R5 34.8k L2 4.7µH 5.5V SEPIC Converter with Short-Circuit Protection Certain applications demand a converter output that is DC-isolated from the input. SEPICs (single-ended primary inductance converters) provide the solution. Figure 8 is an implementation which provides a 5.5V output with complete short-circuit protection. The current sense amplifier used for current sense not only provides excellent short-circuit protection, but also helps soft start the output. The accurate output current limit ensures the maximum current is set at 670mA. When the load demands more, the output voltage will droop while the 670mA output current is maintained. Efficiency is shown in Figure 9. Cuk Converter The LT3477 provides pins for both inputs to the voltage error amplifier, which enables negative output voltages. Figure 10 is an implementation continued on page 40 FBN LT3477 0.2 Voltage feedback is used for open LED protection. D1 SW 0.1 Figure 7. 4W LED driver efficiency 330mA LED Driver with Open LED Protection LT3477 can also be used for LED driver applications using a conventional boost topology with the current sense amplifier for current regulation. Figure 6 shows a typical application circuit, and Figure 7 shows the efficiency. Figure 6 uses a high side current sense configuration for feedback control. The current sense amplifier could also be used for a grounded current sense for this application, if desired, so the output can be tied to the LED string directly. ISP2 would be tied to the cathode side of the LEDs, and ISN2 is tied to ground. C2 10µF 0 IOUT (A) Figure 6. 4W LED driver L1 4.7µH 70 65 60 C1: TAIYO YUDEN LMK316BJ335ML C2: TAIYO YUDEN TMK325BJ335MN D1: DIODES INC. DFLS120L L1: TOKO A915AY-100M Buck-Boost LED Driver In some applications, the input voltage might be comparable to the total LED voltage drop or the input voltage might fluctuate to higher or lower than the total LED voltage drop. A buckboost LED driver works well in this type of application. Figure 4 shows the LT3477 buck-boost LED driver. The cathode end of the LED string is tied back to the input voltage, which allows it to operate from a wide input voltage range. R5 and R6 in Figure 4 are used for open LED protection. Figure 5 is the efficiency measured for this circuit. 75 90 VIN = 3V 85 ISP2 ISN2 VREF R2 1k FBP RT GND SS C5 4.7nF C4 33nF R3 18.2k C3 10µF C1: TAIYO YUDEN LMK316BJ335ML C2: TAIYO YUDEN LMK325BJ106MN C3: TAIYO YUDEN LMK316BJ106ZL D1: DIODES INC. DFL5120L L1, L2: TOKO FDV0630-4R7M Figure 8. 5.5V SEPIC converter with short-circuit protection Linear Technology Magazine • December 2005 R6 10k EFFICIENCY (%) 80 VC 75 70 65 60 55 50 0 0.1 0.2 0.3 0.4 IOUT (A) 0.5 0.6 0.7 Figure 9. 5.5V SEPIC converter with short-circuit protection efficiency 27 DESIGN FEATURES 4-Channel I2C Multiplexer Provides Address Expansion, Bus Buffering by John Ziegler and Fault Management Introduction As data processing, mass storage and communications systems have grown, the size and complexity of the subsystems employed to transfer information such as temperature, fan speed, system voltages and Vital Product Data (VPD, board identification, for example) have grown in proportion. This information is most often transferred through two-wire serial buses, such as I2C or SMBus. Several practical problems can arise in the design of these systems, especially as they become large. First, many devices, such as Small Form Factor Pluggable optical modules (SFPs) have hard-wired I2C addresses, preventing the use of multiple such devices due to address conflict. Second, as the variety of devices increases and more I/O cards are hot-swapped into and out of a system, the likelihood of an I2C device becoming confused and holding the bus low increases. Third, bus timing specifications become difficult to meet with increasing equivalent bus capacitance. In addition to these large system issues, cycling power whenever a new I/O card is installed is not an option in uninterruptible systems of any size. The LTC4306 4-channel 2-wire bus multiplexer/switch with bus buffers addresses all of these issues (see Table 1 for a short list of features). A master on the upstream 2-wire bus (SDAIN, SCLIN) can connect to any combination of downstream buses through the LTC4306’s bus buffers and multiplexers/switches. As a result, the same device address can be used on multiple downstream buses. The buffers provide capacitive isolation between the upstream and downstream buses, allowing for partitioning of the system loading. Rise time accelerators further aid in overcoming capacitance limitations. Stuck Low Timeout circuitry 28 Table 1. Some features of the LTC4306 Feature Benefits 4 Selectable Downstream Buses ❏ Maximum flexibility of bus configurations Disconnect from Stuck Bus ❏ Frees masters to resume upstream communications 2-Wire Bus Buffers ❏ Breaks up capacitance Buffer Supply Independence ❏ Level-shifting: 2-Wire buses can be pulled up to supply voltages ranging from 2.2V to 5.5V, independent of the LTC4306 VCC voltage Slew Limited Rise Time Accelerators ❏ Aid in reducing rise time ❏ Nested addressing when used a MUX ❏ Allow larger bus pull-up resistors for better noise margin ❏ Drive long cables with no reflection issues 2-Wire Bus Hot Swap ❏ Prevents 2-wire bus corruption during live insertion and removal from backplane Fault Reporting ❏ Helps master find and resolve system faults efficiently Mass Write Address ❏ Issue one command to all LTC4306s at the same time disconnects the upstream bus from the downstream buses when the bus is low for a programmed length of time, freeing the upstream bus to resume communications. Finally, any of the LTC4306’s 2-wire bus pins can be hotswapped into and out of a live system without corrupting it. The LTC4306 works with supply voltages ranging from 2.7V to 5.5V. General Operation A block diagram for the LTC4306 is shown in Figure 1, and a description of its register contents is given in Table 2. The UVLO comparator prevents the LTC4306 from receiving commands until the VCC voltage rises above 2.5V (typical). This ensures that the LTC4306 does not try to function until it has sufficient bias voltage. When ENABLE is brought below 1V, the LTC4306 is reset to its default high-impedance state and ignores any attempts at communication on its 2wire buses. When ENABLE is brought back above 1.1V, masters may resume communication with the LTC4306. Disconnecting from a Stuck Bus The LTC4306 disconnects the upstream bus from the downstream buses when the 2-wire bus is stuck low for a programmed period of time. Masters are then free to resume communications on the upstream bus, assuming the source of the problem resides on a downstream bus. The Stuck Low Timeout circuitry monitors the two common internal nodes of the downstream SDA and SCL switches and runs a timer whenever either of the internal node voltages is below 0.52V. The timer is reset whenever both internal voltages are above 0.6V. Linear Technology Magazine • December 2005 Linear Technology Magazine • December 2005 GPIO1 GPIO2 ENABLE INACC SDAIN 1.1V/1V + – + – 1.6V/1.52V SCLIN SLEW RATE DETECTOR 2.5V/2.35V VCC READY SCLIN SDAIN INACC SLEW RATE DETECTOR 2pF VCC VCC 1µs FILTER 1V 1V UVLO 100ns GLITCH FILTER + – + – + – 50k PORB VCC 100ns GLITCH FILTER + – UPSTREAM DOWNSTREAM BUFFERS UPSTREAM DOWNSTREAM BUFFERS OUTACC 2-WIRE DIGITAL INTERFACE AND REGISTERS STUCK LOW TIMEOUT CIRCUITRY 5 4 OUTACC INACC ADDRESS FIXED BITS “10” AL1-AL4 4 BUS1_LOG-BUS4_LOG FAILCONN_ATTEMPT CONN_REQ 4 CH1CONN-CH4CONN TIMEOUT_LATCH TIMEOUT_REAL TIMSET0 TIMSET1 4 FET1-FET4 CONN STUCK LOW 0.52V COMPARATORS SLEW RATE DETECTOR OUTACC SLEW RATE DETECTOR CONNECTION CIRCUITRY I2C ADDR FET1 FET2 FET3 FET4 5 4 4 4 UVLO FET1-FET4 AL1-AL4 1 OF 27 ALERT LOGIC ALERT 1V THRESHOLD COMPARATORS DOWNSTREAM 1V THRESHOLD COMPARATORS ADR0 ADR1 ADR2 GND ALERT ALERT4 ALERT3 ALERT2 ALERT1 SCL4 SCL3 SCL2 SCL1 SDA4 SDA3 SDA2 SDA1 DESIGN FEATURES Figure 1. A block diagram of the LTC4306 29 DESIGN FEATURES cables. In addition, given the strong drive provided by the accelerators, system designers can choose large resistor pull-ups to minimize bus logic low voltages, thereby maximizing logic low noise margin. Table 2. LTC4306 Register Contents Register Contents 0 Gives logic state of ALERT1#–ALERT4# pins, and present and latched states of Stuck Low Timer. Indicates whether upstream bus is connected to any downstream buses and whether any failed attempts at connection occurred. 1 Activates/deactivates upstream and downstream rise time accelerators. Fault Information Aids Diagnosis Writes and reads logic states of GPIO pins. 2 Configures behavior mode of GPIOs. Enables/disables Mass Write feature. Programs Stuck Low Time. Sets requirements on downstream bus logic states for connection to upstream bus. 3 Connects upstream bus to any combination of 4 downstream buses. Masters can read logic state of the downstream buses before connecting to them. Using register 2, masters can set times of 7.5ms, 15ms, or 30ms, or they can choose to disable the timeout feature. 2-Wire Bus Buffers and Multiplexer Switches Provide Capacitance Buffering and Level Shifting Masters write to register 3 to connect to any combination of downstream channels. The 2-Wire Bus Buffers provide capacitive isolation between the upstream SDAIN, SCLIN bus and the downstream buses. Thanks to this feature, masters can include LTC4306s at various points in their system to break one large bus into several smaller buses. When any downstream bus is connected, the LTC4306 allows the READY pin to be pulled to a logic high by an external resistor. By default, the LTC4306 only connects to downstream buses that are high. Attempts to connect to a low downstream bus fail and cause the LTC4306 to pull the ALERT# pin low to indicate a fault. Masters can override this feature by writing to register 2 and instructing the LTC4306 to execute connection commands regardless of the downstream logic state. The upstream and downstream bus pull-up supply voltages can range from 2.2V to 5.5V, independent of the LTC4306 VCC voltage—the LTC4306 therefore provides level-shifting between buses having different pull30 After a fault occurs and the LTC4306 pulls the ALERT# pin low, the LTC4306 works with the master to resolve the fault simply and quickly. The LTC4306 stores specific fault information in read-only register 0. Faults stored include a stuck low bus, faults on the downstream buses, and a failed attempt to connect to a downstream channel. If the source of the problem is on a connected downstream bus, the master can communicate directly with the offending device. In this case, the LTC4306 acts transparently, with the master and offending device communicating directly via the LTC4306’s bus buffers. In all other cases, the LTC4306 communicates with the master on the upstream 2-wire bus to resolve the fault. After the master broadcasts the Alert Response Address (ARA), the LTC4306 responds with its address on SDAIN and releases ALERT#. The LTC4306 also releases ALERT# if it is addressed by the master. The master determines the source of the fault by reading register 0. After the master solves the problem, it writes a dummy byte to register 0 (which is a read-only register) to reset the fault detection circuitry. up voltages. To guarantee proper operation when connecting multiple downstream channels at once, make sure that the LTC4306 VCC voltage is less than or equal to all downstream pull-up voltages to maintain channel-to-channel isolation during logic highs. Rise Time Accelerators Reduce Rise Times By writing to Register 2, masters may activate the rise time accelerators on the upstream bus, downstream bus, neither or both. When activated, the accelerators turn on in a controlled manner and source current into the buses to make them rise at a typical rate of 100V/µs during positive bus transitions. These strong pull-up currents allow users to build large, heavily capacitive systems while still meeting rise time specifications, but are also slew limited for driving long 3.3V 2.5V 0.01µF 10k 10k 10k 10k 10k 10k VCC MICROCONTROLLER SCLIN SDAIN ALERT SCL1 SDA1 ALERT1 SFP MODULE 1 ADDRESS = 1111 000 • • • LTC4306 ADR2 ADR1 ADR0 GND SCL4 SDA4 ALERT4 5V 10k 10k 10k SFP MODULE 4 ADDRESS = 1111 000 ADDRESS = 1000 100 Figure 2. A circuit illustrating the nested addressing and level shifting features of the LTC4306 Linear Technology Magazine • December 2005 DESIGN FEATURES Nested Addressing and Level-Shifting VCC = 3.3V 0.01µF The circuit shown in Figure 2 illustrates the nested addressing, level-shifting and capacitance buffering features of the LTC4306. For simplicity, only channels 1 and 4 are shown. Note that the backplane, card 1 and card 4 are pulled up to three different supply voltages. Also, the SFP modules have the same address, but no conflict occurs as long as channels 1 and 4 are never active at the same time. 2-Wire Bus Hot Swapping with the LTC4306 Located on the Backplane 10k 10k 10k 10k 10k 10k VCC SCL1 SDA1 ALERT1 SCLIN SDAIN ALERT µP TEMP SENSOR 10k VCC ADR2 ADR1 ADR0 GND I/O CARD 3.3V LTC4306 10k 10k SCL4 SDA4 ALERT4 VOLTAGE MONITOR ADDRESS = 1010 000 BACKPLANE CARD CONNECTOR CONNECTOR Figure 3. A 2-Wire Bus hot-swapping application circuit with the LTC4306 resident on the backplane Figure 3 shows a circuit with the LTC4306 located on the backplane and an I/O card plugging into downstream channel 4. Again, channels 2 and 3 are omitted for simplicity. Before plugging and unplugging the card, make sure that channel 4 is not connected to the upstream bus, so that any transaction occurring on the upstream bus is not disturbed. The pull-up resistors on SDA4 and SCL4 are shown on the backplane, but they may be located on the I/O card, as long as masters on the backplane do not connect to channel 4 when no card is present. The pull-up resistor on ALERT4# must be located on the backplane, to prevent false fault reporting when the I/O card is not present. 2-Wire Bus Hot Swapping with the LTC4306 Located on an I/O Card In Figure 4 the LTC4306 resides on the edge of an I/O card having four separate downstream buses. Connect a 200kΩ resistor from ENABLE to ground and make ENABLE the shortest pin on the connector. This ensures continued on page 42 VCC = 3.3V R1 10k R2 10k C1 0.01µF R3 10k VCC SCLIN R4 10k R5 10k R6 10k SCL1 CARD_SCL1 SDA1 CARD_SDA1 CARD_ALERT1 ALERT1 SDAIN PIC MICROCONTROLLER ALERT VCC R11 200k ENABLE LTC4306 VCC ADR2 OPEN ADR1 ADR0 GND R15 10k R16 10k R17 10k SCL4 CARD_SCL4 SDA4 CARD_SDA4 CARD_ALERT4 ALERT4 R10 10k READY BACKPLANE CONNECTOR CARD CONNECTOR ADDRESS = 1010 000 Figure 4. A 2-Wire Bus hot-swapping application circuit with the LTC4306 resident on the I/O card Linear Technology Magazine • December 2005 31 DESIGN IDEAS Lithium Ion Battery Charger Allows Choice of Termination Method and Includes 100mA Adjustable Low Dropout Regulator by Fran Hoffart Introduction Lithium Ion Batteries Are Simple to Charge There are several recommended methods for charging Li-Ion cells. One method is to apply a current limited constant voltage to the battery for three hours, then stop. Using this method, the battery will be 100% charged after 3 hours, provided the charge current is set between approximately C1 and C/2. A second, similar method is to apply a current limited constant voltage to the battery while monitoring the charge current. During the first portion of the charge cycle, the charger is in constant current mode, with the battery voltage slowly rising as the 32 1000 CHARGE CURRENT 900 CHARGE CURRENT (mA) 800 4.40 BATTERY VOLTAGE 700 4.20 4.10 600 500 100% 400 80% 300 60% 200 CHARGE CAPACITY CHARGE SIGNAL 3 HOUR TIMER ENDS 100 0 4.30 0 20 40 4.00 3.90 3.80 3.70 BATTERY VOLTAGE (V) Lithium ion batteries, including lithium ion polymer, come relatively close to being the perfect battery: high energy density, lightweight, low selfdischarge, high voltage (compared to other cells), no memory problem, low maintenance, and best of all, they are simple to charge. Of course, there are some disadvantages too, but let us leave that for later in this article. Since many hand held products can operate from a single Li-Ion cell, many single cell chargers use a linear, rather than a switching topology. Linear chargers are simpler than switchers and comparably efficient at the low input-to-output voltage differential typical of portable devices. This article presents a simple standalone 1A battery charger that combines many desirable charger features and an LDO regulator in a tiny 3mm × 3mm low profile DFN package. Also, a brief discussion on lithium ion battery pros and cons, and charging methods are discussed. 3.60 3.50 3.40 60 80 100 120 140 160 180 TIME (MINUTES) Figure 1. Charge cycle of a 900mAHr Li-Ion cell charged at 1C using timer termination battery accepts charge. As the battery voltage approaches the programmed constant (float) voltage, the charge DESIGN IDEAS Lithium Ion Battery Charger Allows Choice of Termination Method and Includes 100mA Adjustable Low Dropout Regulator .....................32 Fran Hoffart Low Ripple Micropower SOT-23 Buck Regulator with Integrated Boost and Catch Diodes Accepts Inputs to 40V .......................34 Leonard Shtargot Taking Full Advantage of Very Low Dropout Linear Regulators .........35 Joe Panganiban Op Amp Selection Guide for Optimum Noise Performance .............37 Glen Brisebois Multi-Output Supply Drives White LEDs, Provides LCD or OLED Bias in a 3mm × 3mm DFN Package ..........39 Gurjit Thandi Single Cell Step-Up DC/DC Converter Features 400mA Switch Current in an SC70 Package .............41 Dave Salerno Tiny DC/DC Buck Controller Provides High Efficiency and Low Ripple ........43 Theo Phillips current begins to drop exponentially. When the charge current drops to a sufficiently low value, the charger stops charging. Depending on the minimum charge current selected, the battery is between 95% and 100% charged. Since Li-Ion batteries are unable to absorb an overcharge, all charge current must stop when the battery becomes fully charged. A Charger and an LDO Regulator in One Small DFN Package The LTC4063 is a complete single cell Li-Ion battery charger that provides the user a choice of charge termination methods and includes an adjustable low dropout 100mA linear regulator. In addition to the usual constantcurrent/constant-voltage charge algorithm, other desirable features include power limiting that reduces the charge current under high ambient temperature and/or high power dissipation conditions. This allows the charger to provide higher charge currents under normal conditions and still provide safe charging under abnormal conditions such as high ambient temperature, high input voltage or low battery voltage. The LTC4063 contains many common features of other Li-Ion chargers including trickle charge for low battery, auto recharge, charge current monitor, charge status output, capable of charging from USB power, low battery drain current when VIN is removed and precision (±0.35%) battery float voltage accuracy. What sets this linear charger apart from other single cell chargers is the selectable charge termination and the onboard voltage regulator. Linear Technology Magazine • December 2005 DESIGN IDEAS Termination can be based on either total time, which is programmable, or minimum charge current which is also programmable, or the charge cycle can be stopped by the user via the charge enable pin. The low dropout regulator, which is powered from the battery, is adjustable from 1V to almost 4.2V and can provide up to 100mA to a load. A low 15µA operating quiescent current and 2.5µA shutdown current extend battery life. Charge Termination Methods: Which One to Use The first portion of a charge cycle consists of forcing a constant current (typically 1C) into the battery until the cell voltage approaches the programmed float voltage (typically 4.2V ±1% or better) at which time the charge current begins to drop. For a depleted battery this occurs after approximately 30 minutes with the battery state of charge at approximately 55% of full capacity. Since the charge current drops rather quickly in the constant voltage portion of the charge cycle, the battery requires another 2 hours to bring the battery up to a 100% charge level. Unfortunately, there is not much that can be done to speed up this portion of the charge cycle without exceeding the recommended charge voltage. Some chargers utilize a Negative Temperature Coefficient (NTC) thermistor that is located near or inside the battery pack to measure battery temperature. This protects the battery by not allowing a charge cycle to begin if the battery temperature is less than 0°C or greater than 50°C. During a normal charge cycle, there is very little temperature rise for LiIon batteries. Figure 1 shows a LTC4063 charge cycle for a 900mAHr Li-Ion polymer battery charging at a 1C rate. The curves show the relationship between the charge current, battery voltage, charge capacity and the CHRG output signal. Since the timer termination method was selected, the charge cycle ended after approximately 172 minutes with the battery at 100% charge Linear Technology Magazine • December 2005 VIN 4.2V TO 8V LED 330Ω TERMINATION METHOD C/X – GND TIMER – CAP EXT. – VCC MONITOR CHARGE CURRENT 1V FOR FULL CURRENT 10 1µF 6 7 0.1µF (3 HOURS) 9 VCC CHGEN CHRG LDOEN TIMER BAT 5 CHARGER 1 OUT ICH = IPROG • 1000 EXCEPT WHEN PINS 9 AND 10 ARE CONNECTED TOGETHER, THEN ICH = IPROG • 500 2k FB IDET GND 11 EXPOSED PAD + 4.2V Li-Ion 2 1.1k 8 ON OFF 900mA 2.2µF LTC4063-4.2 PROG ON OFF 4 REGULATOR 3 464k 1% 169k 1% 2.2µF SHUTDOWN INPUTS LDO REGULATOR OUTPUT 3V 100mA REGULATOR OUTPUT ADJUSTABLE FROM 1V TO 4.2V Figure 2. Complete single cell Li-Ion charger with timer termination, 50mA minimum charge current detection and 3V 100mA LDO voltage regulator level. (Note: the charge current near the end of the charge cycle is a very low 6mA). Also shown in Figure 1 is the CHRG open drain output signal, which was programmed to go high when the charge current dropped below 50mA (IDETECT threshold) or approximately C/20. Had the minimum charge current termination method been selected rather than the timer method, the charge cycle would have ended when the CHRG signal went high (after 105 minutes). At that point the battery is approximately 97% charged, and it would take another hour of charging for the last 3%. The programmable IDETECT current threshold level of the LTC4063 has excellent accuracy, even at current levels as low as 5mA. Programming a low IDETECT current and selecting minimum current termination would result in the charge cycle ending at approximately the same time as timer termination. Which termination is better? From the previous paragraph, it appears that it may not make much difference because by selecting a low IDETECT current level, the two methods can be made virtually identical. Minimum charge current termination can have an advantage in a situation where different charge current levels may need to be selected during a charge cycle, or when charging a battery that still has a partial charge, the charge cycle can be very short. But timer termination may be better if a load that is greater than the programmed IDETECT current level is permanently connected to the battery. In that situation, the charge cycle may never terminate. Also, in timer termination, if the battery does not reach the recharge threshold of 4.1V when the timer ends, the timer is reset and a new charge cycle begins. A Quick Primer on Rechargeable Li-Ion Batteries Within the lithium ion family of batteries there are several formulations: mainly lithium cobalt oxide or lithium manganese oxide as the positive electrode, and either coke or graphite as the negative electrode. The electrolyte is a liquid in cylindrical cells or a solid or a gel in Li-Ion polymer cells. Since no liquid is used in the polymer cells, the cell package can consist of an inexpensive lightweight foil pouch continued on page 44 About Battery Capacity and Charge Current The correct charge current is always related to a battery’s capacity, or simply “C”. The letter “C” is a term used to indicate the manufacturers stated battery discharge capacity, which is measured in mAHr. For example, a 900mAHr rated battery can supply a 900mA load for one hour before the cell is depleted. In the same example, charging the battery at a C/3 rate would mean charging at 300mA. 33 DESIGN IDEAS Low Ripple Micropower SOT-23 Buck Regulator with Integrated Boost and Catch Diodes Accepts Inputs to 40V by Leonard Shtargot Introduction The LT3470 is a micropower buck regulator that integrates a 300mA power switch, catch diode and boost diode into a low profile 8-Pin ThinSOT package (see Figure 1). The combination of single cycle Burst Mode and continuous operation allows the use of tiny inductor and capacitors while providing a low ripple output to loads of up to 200mA. With its wide input range of 4V to 40V and low quiescent current of 26µA (12V in to 3.3V out) the LT3470 can regulate a wide variety of power sources, from 2-cell Li-Ion batteries to unregulated wall transformers and lead acid batteries. 3 BIAS 7 + – BOOST 500ns ONE SHOT R Q′ S Q SW – ENABLE 5V, 200mA from 40V Consumes Less than 1mW at No Load SHDN VREF 1.25V 6 5 + BURST MODE DETECT 2 NC 1 Figure 2 shows a 5V, 200mA supply that accepts inputs from 5.5V to 40V. While the output is in regulation and with no load the power loss is lower than 1mW. The LT3470 can also be put in a shutdown mode that reduces the input current to <1µA by pulling the SHDN pin low. When always-on operation is desired, the SHDN pin can be tied to VIN. The LT3470 uses a control system that offers low (<10mV) ripple at the VIN gm FB GND 8 4 Figure 1. The block diagram of the LT3470 shows the integrated boost and catch Schottky diodes. Inductor current is kept under control at all times by monitoring the VIN current as well as the catch diode current, thereby providing short circuit protection even if VIN = 40V. output while keeping quiescent current to a minimum. When output load is light, the LT3470 remains in sleep mode while periodically waking up for single switch cycles to keep the output in regulation. The current limit of these single switch cycles is about 100mA, which keeps output ripple to a minimum. At greater output loads the LT3470 no longer enters sleep mode, and instead servos the peak switch current limit (up to 300mA) to regulate the output. See Figure 3 for operating waveforms. continued on page 36 90 1000 VIN = 12V 80 BOOST LT3470 OFF ON SHDN 0.22µF 33µH SW BIAS 22pF 2.2µF VOUT 5V 200mA GND FB 604k 1% 200k 1% 22µF 100 60 50 10 40 30 POWER LOSS (mW) VIN 70 EFFICIENCY (%) VIN 5.5V TO 40V 1 20 10 0.1 1 10 LOAD CURRENT (mA) 100 0.1 Figure 2. The LT3470 uses a minimum of board space and external components while delivering wide onput range and high efficiency. This buck regulator supplies up to 200mA at 5V from inputs up to 40V. Input power loss is below 1mW when there is no output load. 34 Linear Technology Magazine • December 2005 DESIGN IDEAS Taking Full Advantage of Very Low Dropout Linear Regulators by Joe Panganiban Introduction 70 60 DROPOUT VOLTAGE (mV) Linear regulators are generally considered inefficient step-down DC/DC converters, but low dropout linear regulators (LDOs) can be a good fit in many handheld battery applications where low power and efficient power conversion are critical. The lower the dropout voltage, the more efficient the LDO solution. Generally, LDOs with a very small dropout voltage come at the expense of increased package size and higher quiescent current. The LTC3035 overcomes these tradeoffs by offering a very low dropout voltage without sacrificing small solution size or low power. The LTC3035 is a micropower, VLDO™ (very low dropout) linear regulator, which operates from input voltages between 1.7V and 5.5V. The device is capable of supplying 300mA of continuous output current with an ultra-low dropout voltage of 45mV typical (see Figure 1). The output voltage is externally adjustable over a wide voltage range, spanning between 0.4V and 3.6V. The LTC3035 is ideal for batterypowered applications where low power, low dropout, low noise, and small solution size are essential. Under noload conditions, the chip draws only 50 TA = 125°C 40 TA = 25°C 30 20 TA = –40°C 10 0 0 50 100 200 150 IOUT (mA) 250 300 Figure 1. Typical dropout voltage versus load current 100µA from the VIN supply, and drops to 1µA when in shutdown. The LDO is stable for all ceramic capacitors down to 1µF. Other features include output short-circuit protection, reverse output current protection, and thermal overload protection, all available in a tiny 3mm × 2mm DFN package. Low Dropout from an NMOS Pass Device Conventional LDOs integrate a P-type transistor (either PNP or PMOS) as the power pass device to deliver current from the input supply to its output. The LTC3035, instead, incorporates an NMOS transistor as its pass element in a source-follower configuration. This architecture allows for several performance advantages over conventional P-type LDOs, such as greater VIN power supply rejection, lower dropout voltage, and better transient response characteristics, while maintaining a smaller solution size. Using an NMOS pass device is not entirely transparent. In order to achieve low dropout performance using an NMOS pass device, the LDO circuitry must be capable of driving the NMOS gate above the VIN supply. This implies that a separate higher voltage supply is necessary to power the LDO circuitry. For many applications, the luxury of an extra higher supply is simply unavailable. The LTC3035 overcomes this problem by including a built-in charge pump that generates a higher BIAS supply from the VIN input to power its LDO circuitry. The charge pump requires only a 0.1µF flying capacitor and a 1µF bypass capacitor for operation. The value of the generated BIAS supply is adaptively controlled to provide sufficient gate drive over the full VIN operating range, optimizing the current carrying capabilities and dropout characteristics of the VLDO regulator. 0.1µF 10µH S S 3 VIN = 2.7V TO 4.2V LTC3440 7 S VIN 8 Li-Ion + 10µF 2 * 1 RT 60.4k S OFF ON SW2 SW1 VOUT SHDN/SS FB MODE/SYNC VC RT GND S 3.4V 600mA 6 CM CP 1µF BIAS IN 1µF 4 LTC3035 SHDN OUT S GND ADJ S S 22µF S 15k 5 200k S 1µF VOUT = 3.3V IOUT ≤ 300mA 40.2k 1.5nF 10 S 294k 357k 9 S S S S *1 = Burst Mode OPERATION 0 = FIXED FREQUENCY C1: TAIYO YUDEN JMK212BJ106MG C2: TAIYO YUDEN JMK325BJ226MM L1: SUMIDA CDRH6D38-100 Figure 2. A high-efficiency and low-noise lithium-ion to 3.3V solution Linear Technology Magazine • December 2005 35 DESIGN IDEAS 0.1µF LTC3035 INTPUT AC 20mV/DIV DUAL ALKALINE BATTERY 1µF 1µF LTC3035 OFF ON LTC3035 OUTPUT AC 20mV/DIV CM BIAS CP IN SHDN OUT 140k 1µF VOUT = 1.8V IOUT ≤ 300mA ADJ GND 40.2k 20µs/DIV Figure 3. Input and output waveforms to the LTC3035 in the Li-Ion to 3.3V application, showing its excellent ripple rejection (IOUT = 25mA, LTC3440 in Burst Mode®) High Efficiency, Low Noise Li-Ion to 3.3V Figure 2 shows a high efficiency and low noise lithium-ion to 3.3V solution. The LTC3440, a buck-boost converter, converts the Li-Ion battery voltage to an efficient intermediate voltage (3.4V) at the input of the VLDO. The LTC3035 then regulates this intermediate voltage down to 3.3V, providing a lower noise output voltage. Figure 3 shows the input and output waveforms of the LTC3035 at 25mA of output current, illustrating its excellent power supply rejection characteristics for a lower noise solution. For optimum total efficiency, the input to output voltage differential across the LDO should be as small as possible, since the magnitude of the dissipated power equals the product of the voltage differential and the output current. Because of the LTC3035’s LT3470, continued from page 34 The fast cycle-by-cycle current limit of the LT3470 keeps the switch and inductor currents under control at all times. In addition, the LT3470 uses hysteretic mode control where the switching frequency automatically adjusts to accommodate variations in Figure 4. A very low dropout dual-alkaline to 1.8V application very low dropout voltage, its input voltage can be programmed to only 100mV above the 3.3V output and still maintain regulation at 300mA. Conventional LDOs with higher dropout voltages force greater input and output voltage differentials, effectively reducing efficiency by the same ratio. Double Alkaline to 1.8V LDO Handheld applications using two alkaline batteries in series demand low power solutions that use as much of the battery’s operating voltage range as possible. In Figure 4, two series alkaline batteries are regulated down to provide a 1.8V supply taking advantage of the LTC3035’s excellent dropout characteristics. The dropout voltage and maximum output current capabilities of typical low power LDOs using P-type transistors suffer as the input voltage supply decreases, since the power transistor’s overdrive reduces. With input and VIN and VOUT. This means that the part switches at a slower frequency when the output is in short circuit or when VIN/VOUT ratio is high. This ensures that the LT3470 can handle a short circuit at the output even if VIN = 40V and the inductor value is small. It NO LOAD 10mA LOAD VOUT 20mV/DIV VOUT 20mV/DIV IL 100mA/DIV IL 100mA/DIV 1ms/DIV 5µs/DIV Figure 3. Operating waveforms show the output voltage ripple remains at 10mV in BurstMode operation, while requiring only a 22µF ceramic output capacitor. 36 output voltages near 1.8V, conventional low power LDOs may have dropout voltages over 200mV, if they can deliver 300mA of output current at all. Using the LTC3035, the battery voltage can discharge much further to only about 50mV above the 1.8V output before the LDO begins to drop out at 300mA. Allowing the battery to discharge longer essentially extends the battery life for the application when compared to solutions that use higher dropout LDOs. Conclusion The very low dropout characteristics of the LTC3035 can be exploited in battery-powered applications to obtain higher efficiency and increased battery life. Its very low dropout voltage, excellent power supply rejection, lowquiescent current, and small solution size make the LTC3035 an ideal choice for many low power, handheld battery applications. is, however, important to choose an inductor that does not saturate excessively at currents below 400mA to guarantee short circuit protection. Conclusion The LT3470 is a small buck regulator with a unique combination of features that make it a great choice in applications requiring small size, high efficiency across a wide range of currents, and low output ripple. It can deliver up to 200mA from inputs as high as 40V using only an inductor, four small ceramic capacitors, and two resistors while consuming only 26µA during no load operation. Linear Technology Magazine • December 2005 DESIGN IDEAS Op Amp Selection Guide for Optimum Noise Performance by Glen Brisebois Introduction Linear Technology continues to add to its portfolio of low noise op amps. This is not because the physics of noise has changed, but because low noise specifications are being combined with new features such as rail-to-rail operation, shutdown, low voltage, and low power operation. Op amp noise is dependent on input stage operating current, device type (bipolar or FET) and input circuitry.This selection guide is intended to help you identify basic noise tradeoffs and select the best op amps, new or old, for your application. Quantifying Resistor Thermal Noise and Op Amp Noise The key to understanding noise tradeoffs is the fact that resistors have noise. At room temperature, a resistor R has an RMS voltage noise density (or “spot noise”) of VR = 0.13√R noise in nV/√Hz. So a 10k resistor has 13nV/√Hz and a 1M resistor has 130nV/√Hz. Rigorously speaking, the noise density is given by the equation VR = √4kTR, where k is Boltzman’s constant and T is the temperature in degrees Kelvin. This dependency on temperature explains why some low noise circuits resort to super-cooling the resistors. Note that the same resistor can also be considered to have a noise current of IR = √4kT/R, or a noise power density PR = 4kT = 16.6 • 10–21W/Hz = 16.6 zeptoWatts/Hz independent of R. Selecting the right amplifier is simply finding which one will add the least amount of noise above the resistor noise. Don’t be alarmed by the strange unit “/√Hz”. It arises simply because noise power adds with bandwidth (per Hertz), so noise voltage adds with the square root of the bandwidth (per root Hertz). To make use of the specification, simply multiply it by the square Linear Technology Magazine • December 2005 RS VR(EQ) VN + IN – R1 VN AND IN ARE THE NOISE VOLTAGE AND NOISE CURRENT DENSITIES OF THE OP AMP FROM THE DATA SHEET R2 REQ = EQUIVALENT SOURCE RESISTANCE = RS + R1||R2 VR(EQ) = 0.13 REQ IS RESISTOR THERMAL NOISE IN nV EXPRESS VN, VR(EQ) AND IN•REQ IN nV VN(TOTAL) = 2 VN + VR(EQ)2 ( + IN • REQ ) Hz Hz 2 = THE TOTAL INPUT REFERRED NOISE IN nV Hz Figure 1. The op amp noise model. VN and IN are op amp noise sources (correlated current noise is not shown). VR(EQ) is the voltage noise due to the resistors. root of the application bandwidth to calculate the resultant RMS noise within that bandwidth. Peak-to-peak noise, as encountered on an oscilloscope for example, will be about 6 times the total RMS noise 99% of the time (assuming Gaussian “bell curve” noise). Do not rely on the op amp to limit the bandwidth. For best noise performance, limit the bandwidth with passive or low noise active filters. Op amp input noise specifications are usually given in terms of nV/√Hz for noise voltage, and pA/√Hz or fA/√Hz for noise current, and are therefore directly comparable with resistor thermal noise. Due to the fact that noise density varies at low frequencies, most op amps also specify a typical peak-to-peak noise within a “0.1Hz to 10Hz” or “0.01Hz to 1Hz” bandwidth. For the best ultra low frequency performance, you may want to consider an zero drift amplifier like the LTC2050 or LTC2054. Summing the Noise Sources Figure 1 shows an idealized op amp and resistors with the noise sources presented externally. The equation for the input referred RMS sum of all the noise sources, VN(TOTAL), is also shown. It is this voltage noise density, multiplied by the noise gain of the circuit (NG = 1 + R1/R2) that appears at the output. From the equation for VN(TOTAL) we can draw several conclusions. For the lowest noise, the values of the resistors should be as small as possible, but since R1 is a load on the op amp output, it must not be too small. In some applications, such as transimpedance amplifiers, R1 is the only resistor in the circuit and is usually large. For low REQ, the op amp voltage noise dominates (as VN is the remaining term), while for very high REQ the op amp current noise dominates (as IN is the coefficient of the highest order REQ term). At middle values of REQ, the resistor noise dominates and the op amp contributes little significant noise. This is the ROPTIMUM of the amplifier and can be found by taking the quotient of the op amp’s noise specs: VN/IN = ROPT. Selecting the Best Op Amps Figure 2 shows plots of voltage noise density of the source resistance and of various op amps at three different 37 DESIGN IDEAS frequencies. Each point labelled by an op amp part number is that part’s voltage noise density plotted at its ROPT. Use the graph with the most applicable frequency of interest. Find your source resistance on the horizontal axis, and mark that resistance at the point where it crosses the resistor noise line. This is the “source resistance point.” The best noise performance op amps are under that point, the further down the better. For all candidate op amps, draw a horizontal line from your source 1k resistance point all the way to the right hand side of the plot. Op amps beneath that line will give good noise performance, again the lower the better. Draw another line from the source resistance point down and to the left at one decade per decade. Op amps below that line are also good candidates. If you still can’t find any candidates, then you have a very low source impedance and should use op amps that are closest to the bottom. In such cases, paralleling of low noise op amps is also an option. Conclusion Noise analysis can be a daunting task at first and is unfamiliar territory for many design engineers. The greatest influence on overall noise perfomance is the source impedance associated with the signal. This selection guide helps the designer, whether novice or veteran, choose the best op amps for a given source impedance. f = 10Hz LT1494 100 LTC1992 LT1490 VOLTAGE NOISE DENSITY (nV Hz ) LT1122 LT1211 LT1112 LT1097 LT6010 LT1880 LT1113 LT1001 LT1881 LT1884 LT6013 10 LTC6078 LT1022, LT1055 LT1169 LTC6241 LT1793 LT1012 LT1792 LT1468 LT1213 LT6233 LT1677, LT1678 RESISTOR NOISE LT1007 1 LT1124 LT1028 0.1 10 1k 100 100k 10k 1M 10M REQ FOR RESISTOR NOISE; ROPT FOR OP AMPS (Ω) 1k 1k f = 1kHz 100 100 LT1800 LT1815,8 10 LT1806 LT1722 LT1222 RESISTOR NOISE LT1567 LT6230 1 LT6200 LT1226 LT1028 100 1k LT1215 LT6010 LT1880 LT1097 LT1112 LT1012 LT1881 LT1363 LT1360 LT6220 Hz ) LT1490 LTC1992 LT1122 LT1022,55 LT1213 LT1884 LT6013 LT1793 LT1213 LT1468 LT1169 LT1113 LTC6241 LT1677 LT1124 LT1007 LT6202 VOLTAGE NOISE DENSITY (nV Hz ) VOLTAGE NOISE DENSITY (nV LT1211 LT1001 LT6220 LT1357 0.1 10 f = 100kHz LT1812 LT1800 10 RESISTOR NOISE LT1567 67 1 LT1028 LT6230 LT6200 LT6233 10k LT1215 T1215 LT121 LT1211 1 LT1213 3 LT121 LT1357 T1357 LT1815, LT1818 LT1793 LT179 3 LT1468 LTC6241 C6241 LT LT1722 2 LT1806 LT1792 92 LT17 LT1677 LT167 7 LT1226 6 LT1222 202 LT6233 LT6202 100k REQ FOR RESISTOR NOISE; ROPT FOR OP AMPS (Ω) 1M 10M 0.1 10 100 SHADED ENCLOSES S HADED AREA EN CLOSES CANDIDATE OP AMPS C ANDIDATE O P AMP S FFOR OR NOISE OISE AT REEQ LLOW OW N Q = 100k 1k 10k 100k 1M REQ FOR RESISTOR NOISE; ROPT FOR OP AMPS (Ω) Figure 2. Use these three plots to find the best low noise op amps for your application. See “Selecting the Best Op Amps” in the text. 38 Linear Technology Magazine • December 2005 DESIGN IDEAS Multi-Output Supply Drives White LEDs, Provides LCD or OLED Bias in a 3mm × 3mm DFN Package by Gurjit Thandi Introduction Many of today’s cell phones, PDAs and digital still cameras contain a high-resolution TFT-LCD display and sometimes an additional secondary OLED (Organic Light-Emitting Diode) display. OLED displays are fast becoming the secondary display of choice because they are brighter, thinner and more responsive than equivalent LCDs. The LT3466-1 is a dual switching regulator designed to meet the power supply requirements of small displays, including LCD-bias, white LED backlight and OLED displays. The LT3466-1 integrates a full featured white LED driver and a boost converter in a low profile 3mm × 3mm DFN package. It provides space and component savings with integrated 44V power switches and Schottky diodes. The LED driver can be configured to drive up to 10 white LEDs in series from a single Li-Ion battery. The white LED driver features a low 200mV reference for programming the LED current, thereby minimizing the power loss in the current setting resistor for better efficiency. The boost converter can be used for generating the main LCD bias voltages, or for providing the OLED bias supply. The boost converter achieves ±1.5% out- 3V TO 5V L1 33µH 6 LEDs COUT1 1µF EFFICIENCY (%) VIN SW2 VOUT1 LT3466-1 VOUT2 FB2 SHUTDOWN AND DIMMING CONTROL 1 RT GND 63.4k CIN 1µF COUT1 1µF VIN 5 10 15 20 25 30 OUTPUT CURRENT (mA) Figure 2. Efficiency versus load current for the circuit in Figure 1 Linear Technology Magazine • December 2005 –15V 10mA COUT3 1µF 15V 10mA VOUT2 LT3466-1 OFF ON D1 SW2 VOUT1 CTRL1 RT RFB1 13.3Ω C1 0.1µF L2 33µH L1 33µH 70 0 a 200kHz to 2MHz range. Additional features include output overvoltage protection, internal compensation and internal soft-start. The LT3466-1 operates from a wide input voltage range of 2.7V to 24V, making it suitable for a variety of applications. FB1 VIN = 3.6V 6 LEDs VOUT2 = 16V R2 24.9k 3V TO 5V BOOST CONVERTER 55 R1 475k CTRL2 SHUTDOWN AND DIMMING CONTROL 2 put voltage accuracy by the use of an internal precision 0.8V reference. The LT3466-1 also provides independent dimming and shutdown control of the two converters. The operating frequency of LT3466-1 can be set with an external resistor over LED DRIVER 60 16V 30mA Figure 1. The LT3466-1 powers a main LCD backlight and a secondary OLED display. It provides a 20mA drive for the six-white-LED LCD backlight and a 16V output for the OLED display. SW1 65 COUT2 1µF CIN: TAIYO YUDEN JMK107BJ105 COUT1, COUT2: TAIYO YUDEN GMK316BJ105 L1, L2: 33µH TOKO D52LC 75 50 SW1 CTRL1 RFB1 10Ω 8 LEDs 80 1µF FB1 90 85 L2 33µH 475k COUT2 1µF FB2 CTRL2 63.4k 26.7k OFF ON CIN: TAIYO YUDEN JMK107BJ105 COUT1, COUT2, COUT3: TAIYO YUDEN GMK316BJ105 C1: TAIYO YUDEN UMK212BJ104 L1, L2: 33µH TOKO D52LC D1: PHILIPS BAT54S Figure 3. High efficiency, Li-Ion powered complete TFT-LCD supply (bias and backlighting). The LT3466-1 drives eight white LEDs at 15mA to provide the backlight and generates dual, ±15V, outputs for the LCD-bias. 39 DESIGN IDEAS Dual Display Power Supply for Cell Phones A typical application for the LT3466-1 is as a driver for dual displays in cell phones. Present day, clam-shell cell phones typically use a color TFT-LCD main display and a secondary OLED display. Figure 1 shows the LT34661 powering the main LCD backlight and the secondary OLED display. The 86 VIN = 3.6V 8 LEDs +15V/10mA –15V/10mA EFFICIENCY (%) 84 Low Cost, Complete LCD Bias and White LED Backlighting Solution for Small TFT Displays 82 80 Small, active-matrix, TFT-LCD displays, used in cell phones, PDAs and other handheld devices generally require four to ten white LEDs for providing the backlight and fixed +15V and –15V supply voltages to bias the LCD. Figure 3 shows LT3466-1 powered complete TFT-LCD supply with minimal external components 78 76 74 0 2.5 LT3466-1 drives 6 white LEDs at 20mA for backlighting the main LCD panel and generates 16V output for powering the OLED. The LT3466-1 allows for independent dimming control of the main and secondary displays via the respective CTRL1 and CTRL2 pins. Figure 2 shows the efficiency versus output current for both the LED driver and the boost converter. The typical efficiency at 3.6V input supply is 84% with the white LEDs and the OLED driven at 20mA. 5 7.5 10 LED CURRENT (mA) 12.5 15 Figure 4. Efficiency versus LED current for the circuit in Figure 3. The circuit achieves greater than 83% efficiency driving eight LEDs at 15mA from 3.6V input. and high efficiency. The LT3466-1 drives eight white LEDs at 15mA and generates 15V boost output powered from a single Li-Ion supply. A discrete charge pump produces the secondary output of –15V. As seen in Figure 4, the circuit achieves greater than 83% efficiency driving eight LEDs at 15mA from 3.6V input. Conclusion The LT3466-1 integrates a full featured white LED driver and a boost converter in a space saving 3mm × 3mm DFN package. Integrated power switches and Schottky diodes reduce the overall system cost and size making it an excellent fit for handheld applications. Features like internal compensation, soft-start, Open LED protection enables LT3466-1 to provide complete TFT-LCD supply (bias and white LED backlight) for handheld devices with minimal external components and high efficiency. LT3477, continued from page 27 C1 3.3µF ISP1 ISN1 SW VIN IADJ1 IADJ2 SHDN SHDN D1 R2 402k LT3477 ISP2 75 RLOAD R4 0.2Ω ISN2 VREF FBP –5V FBN VC 4.75k L2 10µH EF(FICIENCY (%) R1 0.05Ω VIN 5V 85 C2 0.47µF L1 10µH C5 3.3µF 55 45 RT GND 65 SS 0 100 200 300 400 LOAD CURRENT (mA) 500 Figure 11. Efficiency of the Cuk converter. C3 22nF R3 100k 100pF C4 33nF R5 18.2k R6 10k C1, C5: TAIYO YUDEN LMK316BJ335ML D1: DIODES INC. DFL5120L L1, L2: TOKO A915AY-100M (D53LC SERIES) Figure 10. Negative output voltage Cuk converter. using a Cuk topology for 5V to –5V conversion. The first current sense amplifier is used for input current limit, and the second current sense amplifier is used for ground rail current sense to accurately limit the load current at 500mA. Even though the two current sense amplifiers are used, 40 efficiency up to 81% at 500mA output load can still be achieved. Figure 11 shows the efficiency. Conclusion The rail-to-rail constant-current/constant-voltage operation of the LT3477 makes the device an ideal choice for a variety of constant-current designs, including negative outputs. The dual current-sense amplifiers allow flexible configuration for input current limit, constant output current and fail-safe protection, along with excellent output voltage regulation. A wide input voltage range and the ability to produce outputs up to 42V make the LT3477 extremely versatile. Authors can be contacted at (408) 432-1900 Linear Technology Magazine • December 2005 DESIGN IDEAS Single Cell Step-Up DC/DC Converter Features 400mA Switch Current in an by Dave Salerno SC70 Package Introduction Small But FullFeatured Solution Despite its diminutive SC70 package, the LTC3525 includes many sophisticated features, such as: 100 LTC3525-3.3 90 100 3.0VIN 90 2.4VIN 80 EFFICIENCY (%) The LTC3525 raises the bar for boost converter performance and power capability in an SC70 package. It is an inductor-based synchronous step-up (boost) DC/DC converter that operates from input voltages as low as 1V, boosting them to 3.3V or 5V. Its powerful internal 400mA switch allows the LTC3525 to deliver up to 150mA of load current with efficiency up to 94%. To further save space, it requires only three external components (two small ceramic capacitors and a small inductor), so a complete solution fits into spaces previously reserved only for charge pump designs. The LTC3525-3.3 and LTC3525-5 are both packaged in the 2mm × 2mm × 1mm SC70 package, and operate over an input range of 0.8V to 4.5V. This flexibility makes them suitable for compact applications powered by 1 to 3 alkaline/NiMH cells, or a single Liion battery. The 3.3V version can even maintain regulation with input voltages exceeding the output voltage. 1.2VIN 70 60 50 40 30 20 10 0 0.01 Figure 1. Typical application using the LTC3525-3.3/-5 0.1 1 10 100 LOAD CURRENT (mA) 1k Figure 2. Efficiency versus load for the LTC3525-3.3 output disconnect, inrush current limiting, low output voltage ripple, synchronous rectification, single cell capability, anti-ring control and less than 1µA shutdown current. It also features overcurrent protection and thermal shutdown, enabling it to sustain an indefinite short circuit without damage. External component selection is easy, since most applications require just a 1µF ceramic input capacitor for local decoupling, a 10µF ceramic output filter capacitor and a 10µH inductor (although any value from 4.7 to 15µH can be used). Be sure to use only X5R or X7R style capacitors, keeping them close to the pins of the IC. The LTC3525 is enabled by pulling the SHDN pin up to any voltage between 1V and 5V, regardless of input or output voltage. High Efficiency Over a Wide Range of Input Voltages & Load Currents The LTC3525 uses a proprietary, patent pending technique of adaptively adjusting peak inductor current as a function of load and input voltage. This technique provides optimum efficiency at light to medium loads, while enabling it to supply heavier load currents that are beyond the capability of other solutions of this size. The LTC3525’s low quiescent current of only 7µA on VOUT allows it to maintain impressive efficiency down to extremely light loads, as shown in Figure 2, and over a broad range of input voltages. By comparison, the efficiency of a charge pump design varies widely as the battery voltage changes, as illustrated in the graph in EFFICIENCY (%) 80 6.8µH* 70 60 CHARGE PUMP 50 1V to 1.6V 3 40 1 30 20 10 0 2 1µF IOUT = 20mA 0.5 1 1.5 2 VIN (V) 2.5 3 LTC3525-3.3 VIN SW SHDN VOUT GND GND 6 4 5 VOUT 3.3V 60mA 10µF** 6.3V 3.5 Figure 3. Comparison of efficiency versus input voltage for LTC3525-3.3 and and an equivalent charge pump based boost circuit Linear Technology Magazine • December 2005 * COILCRAFT LPO3310-682MXD ** MURATA GRM219R60J106KE19D Figure 4. Single cell to 3.3V converter delivers 60mA of load current in a 1mm profile 41 DESIGN IDEAS L1* 10µH VIN Li-Ion OFF ON 1µF IOUT = 100mA LTC3525-3.3 3V to 4.5V 50mV/DIV SW SHDN VOUT GND GND VOUT 5V 175mA IOUT = 10mA 22µF** 6.3V 10µs/DIV Figure 6. Output voltage ripple of the 5V converter at min and max load * MURATA LQH32CN100K53 ** MURATA GRM31CR60J226KE19L Figure 5. Li-ion to 5V converter delivers 175mA of load current with <0.5% ripple Single Cell to 3.3V Converter with 1mm Profile A single alkaline or nickel cell to 3.3V converter, using the LTC3525-3.3, is shown in Figure 4. This application uses an inductor and output capacitor chosen to achieve a 1mm profile. It delivers 60mA of load current from a single cell, and 140mA from two cells, while fitting into a 5mm × 7mm footprint. The ability of the converter to operate with input voltages below 1V allows it to use all the available energy in the battery, and also prevents the converter from shutting off in the event that a load transient causes a momentary drop in input voltage. LTC4306, continued from page 31 that ENABLE remains at a constant logic low while all other pins are connecting, so that the LTC4306 remains in its default high impedance state and ignores connection transients on SDAIN and SCLIN during connection. In addition, make the ALERT# connector pin shorter than the VCC pin, so that VCC establishes solid contact with the I/O card pull-up supply pin and powers the pull-up resistors on ALERT1#–ALERT4# before ALERT# 42 Li-ion/3-Cell to 5V Converter Delivers Over 175mA with Low Output Ripple The LTC3525 has been designed for very low output ripple with minimal output capacitance. In most applications, a 10µF ceramic capacitor will yield less than 1% peak-to-peak output ripple. By using a 22µF capacitor, the output ripple can be reduced to less than 0.5% of VOUT, making it suitable for many noise sensitive applications that previously required a larger, more expensive fixed frequency converter. The circuit in Figure 5, which occupies a space of just 6mm × 6mm, supplies 5V at 175mA or more from a Li-ion battery (or three alkaline or nickel cells). With a 22µF output capacitor, the output ripple is only 22mVP-P at light load, and less than 50mVP-P at full load, as shown in Figure 6. The efficiency peaks at 93% and remains above 85% over three decades of load current, as shown in Figure 7. This solution could also be used to provide 5V at 200mA in a 3.3V powered system. The entire solution fits in a 1.8mm profile. makes contact. When disconnecting, ENABLE breaks contact first, resetting the LTC4306 to its default state, so that it causes minimal disturbance on the SDAIN and SCLIN bus as the card disconnects. Conclusion The LTC4306 eases the practical design issues associated with large 2-wire bus systems. It serves as a multiplexer to provide nested addressing. It disconnects buses when they are 90 3.6VIN 80 EFFICIENCY (%) Figure 3. Note that the charge pump design requires an input voltage of at least 1.7V to generate a regulated 3.3V output. Comparable inductor-based solutions require larger packages and more external components, making them unsuitable to applications where board space is at a premium, or too expensive for cost sensitive applications. 100 70 60 50 40 30 20 10 0 0.01 0.1 1 10 100 LOAD CURRENT (mA) 1k Figure 7. Efficiency versus load for the Li-ion to 5V converter Conclusion Many of today’s battery powered portable devices, such as MP3 players, medical instruments and digital cameras can benefit from the small size, simplicity and extended battery life offered by the LTC3525. Its tiny, low profile SC70 package and minimal external part count make it a viable, high performance alternative to less efficient charge pump designs. Its 400mA switch current and low output ripple allow it to replace more expensive fixed frequency converters in cost-sensitive applications. stuck low. It breaks a large capacitive bus into smaller pieces and allows I/O cards to be hot-swapped into and out of live systems. It logs faults, reports to the master, and works with the master to resolve faults efficiently. for the latest information on LTC products, visit www.linear.com Linear Technology Magazine • December 2005 DESIGN IDEAS Tiny DC/DC Buck Controller Provides High Efficiency and Low Ripple by Theo Phillips Introduction Circuit Description Figure 1 shows a typical application for the LTC3772. This circuit provides a regulated output of 2.5V from a typical input voltage of 5V, but it can also be powered from any input voltage between 2.75V and 9.8V (depending on the voltage rating of the P-channel power MOSFETs). This wide input range makes the LTC3772 suitable for a variety of input supplies, including 1- and 2-cell Li-Ion and 9V batteries, as well as 3.3V and 5V supply rails. The internal soft-start ramps the output voltage smoothly from 0V to its final value in 1ms (Figure 2). At low load currents (≤10% of IMAX), the LTC3772 enters Burst Mode operation. Compared with other power saving schemes, this variant of Burst Mode operation surrenders a modicum of efficiency to obtain very low output voltage ripple. Typically producing just 30mV for a typical application using ceramic output capacitors, the LTC3772 is ideal for noise-sensitive portable applications. Figure 3 illustrates inductor current and output voltage waveforms for Burst Mode operation. The LTC3772 uses the drain to source voltage (VDS) of the power Linear Technology Magazine • December 2005 680pF 20k VIN 2.75V TO 8V VIN ITH/RUN LTC3772 GND PGATE IPRG 82.5k 22µF 10V FDC638P L1 3.3µH VFB SW B220A 22pF 47µF 6.3V 174k VOUT 2.5V 2A L1: TOKO A916CY Figure 1. Typical application delivering 2.5V at 2A VOUT 1V/DIV VOUT 50mV/DIV ITH/RUN 1V/DIV IL 500mA/DIV IL 2A/DIV VIN = 5V VOUT = 2.5V LOAD = 2A 500µs/DIV Figure 2. The output voltage rises smoothly without requiring a soft-start capacitor as seen in this startup waveform for the converter in Figure 1. P-Channel MOSFET to sense the inductor current. The maximum load current that the converter can provide is determined by the RDS(ON) of the MOSFET, which is a function of the input supply voltage (which supplies the gate drive). The maximum load current can also be changed using the 100 2A APPLICATION 10µs/DIV VIN = 5V VOUT = 2.5V ILOAD = 12mA Figure 3. The LTC3772’s Burst Mode operation maintains light load efficiency while holding output voltage ripple to just 20mV in this application. current limit programming pin IPRG, which sets the peak current sense voltage across the MOSFET to one of three states; each voltage is associated with its own inductor current limit. With IPRG floating, the circuit of Figure 1 can reliably provide 2.5V at 2A from a 3.3V input supply. Efficiency for this circuit exceeds 93%, as shown in Figure 4. In drop out, the LTC3772 can operate at 100% duty 90 EFFICIENCY (%) To secure a foothold in today’s congested circuit boards, a power controller must deliver the most functionality in the smallest package. With a blend of popular features squeezed into a SOT23 or 3mm × 2mm DFN, the LTC3772 makes a power supply designer’s job easy. This versatile DC-DC controller supports a wide input voltage range, 2.5V to 9.8V, and maintains high efficiency over a variety of output current levels. Its 550kHz switching frequency trims solution size by permitting the use of small passive components. Its No RSENSE™ constant frequency architecture also eliminates the need for a sense resistor. 80 VOUT 100mV/DIV OFFSET = 2.5V 70 IL 2A/DIV 60 ILOAD 2A/DIV 1 10 100 1k LOAD CURRENT (mA) 10k Figure 4. Efficiency vs load current for the converter in Figure 1, with input of 3.3V 20µs/DIV Figure 5. Transient performance of the converter in Figure 1, with input of 5V 43 DESIGN IDEAS cycle, providing maximum operating life in battery-powered systems. Figure 1, using just one 47µF output capacitor. The response is quite fast, even though it involves a transition from Burst Mode operation to continuous conduction mode. OPTI-LOOP Compensation To meet stringent transient response requirements, some switching regulators use many large and expensive output capacitors to reduce the output voltage droop during a load step. The LTC3772, with OPTI-LOOP compensation, is stable for a wide variety of output capacitors, including tantalum, aluminum electrolytic, and ceramic capacitors. The ITH pin of the LTC3772 allows users to choose the proper component values to compensate the loop LTC4063 , continued from page 33 which can be made in various shapes including very thin cells, ideal for cell phones and other small handheld devices. Although the discharge characteristics and performance of the different types of Li-Ion cells vary, the charging characteristics are essentially the same. Rechargeable lithium battery technology is relatively new, and because of that, many improvements in future battery performance are almost guaranteed. Different materials, chemicals and construction will undoubtedly produce a battery that is ever closer to that perfect battery. The recommended charge voltage is a compromise between cell capacity, cell life and cell safety. Higher charge voltages increase the mAhr cell capacity, but shorten the cell lifetime. There are also upper limits that must be adhered to for safety reasons. The most common charge voltage is 4.2V±1% although future battery designs may have a slightly higher voltage. In applications that favor cycle life over cell capacity, a lower charge voltage greatly increases cycle life. Shallow rather than deep discharge cycles increase cycle life as well. The end of life for a Li-Ion battery is typically when its capacity drops to 80% of its rating. One lesser known fact about Li-Ion batteries is their aging characteristics. 44 Conclusion Figure 6. A typical LTC3772 application occupies just 1.5 square centimeters. so that the transient response can be optimized with the minimum number of output capacitors. Figure 4 shows a transient response for the circuit in Li-Ion batteries have a limited lifetime whether they are stored or in daily use. The permanent capacity loss, especially for lithium manganese chemistries, increases with charge level and temperature. For example, storing a battery at a 40% charge level at 25°C for a year could result in a permanent capacity loss of 4%, whereas if stored at a 100% charge level, the permanent capacity loss would be close to 20%. Stored at 100% charge level at 40°C could produce a permanent capacity loss up to 35% after one year. Of course, further improvements in Li-Ion battery technology will surely minimize aging Li-Ion batteries cannot absorb overcharge. Charge current must be completely stopped when the battery reaches full charge. Overcharge can cause internal lithium metal plating, which is a safety concern. Also, Li-Ion batteries should not be discharged below 2.5V to 3V ,depending on battery chemistry, as internal copper plating can form causing a short circuit. Battery Pack Protection: What Is It? Most manufacturers of Li-Ion batteries will not sell batteries unless they include built in battery pack protection circuitry for safety and to prolong battery life. The circuitry includes a FET switch in series with the battery For single-output designs with load currents as high as 5A from input voltages up to 9.8V, the LTC3772 delivers the most popular features of PFET controllers in a very small package. With small ancillary components and no sense resistor, the overall solution is unmatched where board space is at a premium. that turns off in the event of an over voltage, under voltage, over current and over temperature condition when either charging or discharging the battery. A prolonged overvoltage when charging can result in the battery overheating, bursting or even exploding. When discharging, the pack protection disconnects the battery if the battery voltage drops below a predetermined threshold level or if the battery current exceeds a preset limit. Without pack protection, Li-Ion batteries can easily be damaged or worse, can cause damage to other circuitry or bodily injury. Conclusion The LTC4063 Li-Ion battery charger provides the user with an excellent combination of packaging (3mm × 3mm DFN), high charge current (1A), tight float voltage (0.35%), low IDETECT current capability (5mA), choice of termination and an integrated 100mA LDO regulator. Two other chargers share similar charging characteristics but differ on features. The LTC4061 has no regulator but includes a NTC temperature qualification input, a USB current select input and an additional status output. The LTC4062 replaces the LDO regulator with a programmable comparator and reference and also includes a USB current select input. Linear Technology Magazine • December 2005 NEW DEVICE CAMEOS New Device Cameos Unprecedented Power Density from Breakthrough 10A DC/DC µModule A breakthrough DC/DC power converter combines the best features of two heretofore separate design approaches. It mixes the design simplicity of a power module with the power densities of a high performance IC to create a device that is unprecedented in ease-of-use, versatility and power density. The first of many µModules™ provides designers a complete 10A switching power supply in a tiny (15mm × 15mm) footprint, low profile (2.8mm) land grid array (LGA) package. The LTM®4600 is a synchronous switch mode DC/DC step-down regulator with built-in inductor, supporting power components and compensation circuitry. By simplifying power system development, this new high-density power supply reduces development time for a broad range of systems, including network routers, blade servers, cellular base stations, medical diagnostic equipment, test instrumentation and RAID systems. The LTM4600 accommodates a wide input voltage range of 4.5V to 28V. The high level of integration and synchronous current mode operation allows the LTM4600 to deliver superior transient response and up to 10A continuous current (14A peak) at up to 92% efficiency. It simplifies power supply design and construction, requiring only input and output bulk capacitors and a single resistor to set the output voltage within a range of 0.6V to 5V. The LTM4600 DC/DC µModule is a complete stand-alone surface-mount power supply that can be handled and assembled like a standard integrated circuit. Moreover, its low profile design permits the LTM4600 to be soldered onto the back side of a circuit board, freeing up valuable board space. The LTM4600 DC/DC µModules are self-protected against overvoltage Linear Technology Magazine • December 2005 and short circuit conditions. Its fast transient response minimizes required bulk output capacitance. Furthermore, two LTM4600s can be operated in parallel, increasing load current capability to 20A. The LTM4600 is offered in two versions: standard and high input voltage. The LTM4600EV operates from 4.5V to 20V, whereas the LTM4600HVEV has an operating voltage range from 4.5V to 28V. The LTM4600IV and LTM4600HVIV are tested and guaranteed to operate over the –40°C to 85°C temperature range. 16-bit, 130Msps ADC Delivers 100dBc SFDR for High Performance Receivers and Instrumentation The LTC2208 is a 130Msps, sampling 16-bit A/D converter designed for digitizing high frequency, wide dynamic range signals with input frequencies up to 700MHz. The input range of the ADC can be optimized with the PGA front end. The LTC2208 is perfect for demanding communications applications, with AC performance that includes 78dBFS Noise Floor and 100dB spurious free dynamic range (SFDR). Ultra low jitter of 70fsRMS allows undersampling of high input frequencies with excellent noise performance. Maximum DC specs include ±4LSB INL, ±1LSB DNL (no missing codes). The digital output can be either differential LVDS or single-ended CMOS. There are two format options for the CMOS outputs: a single bus running at the full data rate or demultiplexed buses running at half data rate. A separate output power supply allows the CMOS output swing to range from 0.5V to 3.6V. The ENC+ and ENC– inputs may be driven differentially or single-ended with a sine wave, PECL, LVDS, TTL or CMOS inputs. An optional clock duty cycle stabilizer allows high performance at full speed with a wide range of clock duty cycles. The LTC2208 packages an extensive feature set in a 9mm x 9mm QFN package delivering low power consumption at 1250mW without the need for heat sinking. Most importantly, both the power consumption and total solution size with integrated bypass capacitance are less than half that of the nearest competitor. The LTC2208 family includes speed grades of 130Msps, 105Msps, 80Msps, 65Msps, 40Msps, 25Msps and 10Msps all with superior SFDR and SNR performance. In addition to the 16-bit parts, 14-bit versions of this family will also be available. All devices are supported with demo boards for quick device evaluation. Tiny Controller Makes it Easy to Rapidly Charge Large Capacitors The LT3750 is a current-mode flyback controller optimized for charging large value capacitors to a predetermined target voltage. This target voltage is set by the turns ratio of the flyback transformer and just two resistors in a simple, low voltage network, so there is no need to connect components to the high voltage output. The charging current is set by an external sense resistor and is monitored on a cycleby-cycle basis. The device is compatible with a wide range of control circuitry, being equipped with a simple interface consisting of a CHARGE command input bit and an open drain DONE status flag. Both of these signals are compatible with most digital systems, yet are tolerant to voltages as high as 24V. The architecture balances a high degree of integration with the flexibility of leaving key parameters definable by the user. This leaves only a few issues to consider in order to complete the design: Input capacitor sizing, transformer design, and output diode selection. The LT3750 is available in a space saving, 10-lead MSOP package. 45 NEW DEVICE CAMEOS Performance of 50µA CMOS Amplifier Rivals Best Bipolar Op Amps with 0.7µV/°C Drift The LTC6078/LTC6079 are dual/ quad, low offset, low noise operational amplifiers with low power consumption and rail-to-rail input/output swing. Input offset voltage is trimmed to less than 25µV and the CMOS inputs draw less than 50pA of bias current. The low offset drift, excellent CMRR, and high voltage gain make it a good choice for precision signal conditioning. Each amp draws only 54µA current on a 3V supply. The micropower, rail-to-rail operation of the LTC6078/ LTC6079 is well suited for portable instruments and single supply applications. LTC2950/51, continued from page 4 ure 5 shows an actual ESD event. Note the arc onto the PB pin. The ESD strike fed directly onto the pin; there were no series resistors or parallel capacitors. This strike did not damage the pin, nor did it generate any leakage. LTC2950-1 and LTC2950-2 Versions The LTC2950-1 (high true EN) and LTC2950-2 (low true EN) differ only by the polarity of the EN/EN pin. Both versions allow the user to extend the amount of time that the PB must be held low in order to begin a valid power on/off sequence. An external capacitor placed on the ONT pin adds additional time to the turn-on time. An external capacitor placed on the LTC3454, continued from page 24 transconductance amplifier with sink only capability—takes control of the regulation loop and prevents VOUT runaway. The VOUT threshold at which this happens is approximately 5V. If the LED faults as a short circuit, the regulation loop continues to regulate the output current to its programmed current level. 46 The LTC6078/LTC6079 are specified on power supply voltages of 3V and 5V from –40°C to 125°C. The dual amplifier LTC6078 is available in 8-lead MSOP and 10-lead DFN packages. The quad amplifier LTC6079 is available in 16-lead SSOP and DFN packages. Dual and Quad, 1.8V, 13µA Precision Rail-to-Rail Op Amps The LT6001 and LT6002 are dual and quad precision rail-to-rail input and output operational amplifiers. Designed to maximize battery life in always-on applications, the devices operate on supplies down to 1.8V while drawing only 13µA quiescient current. The low supply current and low voltage operation is combined with precision OFFT pin adds additional time to the turn-off time. If no capacitor is placed on the ONT (OFFT) pin, then the turn on (off) duration is given by an internally fixed 32ms timer. The LTC2950 fixes the KILL turn off delay time (tKILL(OFF DELAY)) at 1024ms (the amount of time from interrupting the µP to turning off power). LTC2951-1 and LTC2951-2 Versions The LTC2951 fixes the turn on debounce time at 128ms. The turn off debounce time is the same as the LTC2950: 32ms internal plus the optional additional external when a capacitor is placed on the OFFT pin. The KILLT pin in the LTC2951-1 and LTC2951-2 provides extendable KILL specifications—for instance, input offset is guaranteed less than 500µV. The performance on 1.8V supplies is fully specified and guaranteed over temperature. A shutdown feature in the 10-lead dual version can be used to extend battery life by allowing the amplifiers to be switched off during periods of inactivity. The LT6001 is available in the 8-lead MSOP package and a 10-lead version with the shutdown feature in a tiny, dual fine pitch leadless package (DFN). The quad LT6002 is available in a 16lead SSOP package and a 16-lead DFN package. These devices are specified over the commercial and industrial temperature range. Authors can be contacted at (408) 432-1900 turn off timer, tKILL(OFF DELAY, ADDITIONAL), by connecting an optional external capacitor on the KILLT pin. The default power down delay time is 128ms, tKILL(OFF DELAY). Conclusion The LTC2950/LTC2951 is a family of micro-power (6µA), wide input voltage range (2.7V to 26.4V) push button controllers. The parts lower system cost and preserve battery life by integrating flexible push button timing, a high voltage LDO, and a simple µP interface that provides intelligent power up and power down. The device is available in space saving 8-lead 3mm × 2mm DFN and ThinSOT™ packages. Conclusion The LTC3454 adds to Linear Technology’s family of LED drivers. High efficiencies can be achieved over the entire Li-Ion range with a minimal number of external components. Additionally, it draws zero current when in shutdown, helping conserve battery life in hand held battery powered applications. The LTC3454 is available in a low profile small footprint 3mm × 3mm DFN package. for the latest information on LTC products, visit www.linear.com Information furnished herein by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits, as described herein, will not infringe on existing patent rights. Linear Technology Magazine • December 2005 DESIGN TOOLS DESIGN TOOLS CD-ROM Product Information The December 2005 CD-ROM contains product data sheets, application notes and Design Notes. Use your browser to view product categories and select products from parametric tables or simply choose products and documents from part number, application note or design note indexes. Linear Technology offers high-performance analog products across a broad product range. Current product information and design tools are available at www.linear.com. Our CD-ROM product selector tool, which is updated quarterly, and our most recent databook series can be obtained from your local Linear Sales office (see the back of this magazine) or requested from www.linear.com. www.linear.com Product information and application solutions are available at www.linear.com through powerful search tools, which yield weighted results from our data sheets, application notes, design notes, Linear Technology magazine issues and other LTC publications. The LTC website simplifies the product selection process by providing convenient search methods, complete application solutions and design simulation programs for power, filter, op amp and data converter applications. Search methods include a text search for a particular part number, keyword or phrase, or a powerful parametric search engine. After selecting a desired product category, engineers can specify and sort by key parameters and specifications that satisfy their design requirements. Purchase Products Online Credit Card Purchases—Purchase online direct from Linear Technology at www.linear.com using a credit card. Create a personalized account to check order history, shipment information and reorder products. Linear Express Distribution — Get the parts you need. Fast. Most devices are stocked for immediate delivery. Credit terms and low minimum orders make it easy to get you up and running. Place and track orders online. Apply today at www.linear.com or call (866) 546-3271. Applications Handbooks Linear Applications Handbook, Volume I — Almost a thousand pages of application ideas covered in depth by 40 Application Notes and 33 Design Notes. This catalog covers a broad range of real world linear circuitry. In addition to detailed, systems-oriented circuits, this handbook contains broad tutorial content together with liberal use of schematics and scope photography. A special feature in this edition includes a 22-page section on SPICE macromodels. Linear Applications Handbook, Volume II — Continues the stream of real world linear circuitry initiated by Volume I. Similar in scope to Volume I, this book covers Application Notes 40 through 54 and Design Notes 33 through 69. References and articles from non-LTC publications that we have found useful are also included. Linear Applications Handbook, Volume III — This 976-page handbook includes Application Notes 55 through 69 and Design Notes 70 through 144. Subjects include switching regulators, measurement and control circuits, filters, video designs, interface, data converters, power products, battery chargers and CCFL inverters. An extensive subject index references circuits in Linear data sheets, design notes, application notes and Linear Technology magazines. Linear Technology Magazine • December 2005 Brochures and observing the results. With FCAD, you can design lowpass, highpass, bandpass or notch filters with a variety of responses, including Butterworth, Bessel, Chebychev, elliptic and minimum Q elliptic, plus custom responses. Download at www.linear.com SPICE Macromodel Library — This library includes LTC op amp SPICE macromodels. The models can be used with any version of SPICE for analog circuit simulations. These models run on SwitcherCAD III/LTC SPICE. Power Management & Wireless Solutions for Handheld Products — The solutions in this product selection guide solve real-life problems for cell phones, digital cameras, PDAs and other portable devices, maximizing battery run time and saving space. Circuits are shown for LiIon battery chargers, battery managers, USB support, system power regulation, display drivers, white LED drivers, photoflash chargers, DC/DC converters and RF PA power supply and control. Noise Program — This PC program allows the user to calculate circuit noise using LTC op amps, determine the best LTC op amp for a low noise application, display the noise data for LTC op amps, calculate resistor noise and calculate noise using specs for any op amp. Automotive Electronic Solutions — This selection guide features high performance, high reliability solutions for a wide range of functions commonly used in today’s automobiles, including telematics, infotainment systems, body electronics, engine management, safety systems and GPS navigation systems. Amplifiers (Book 2 of 2) — • Operational Amplifiers • Instrumentation Amplifiers • Application Specific Amplifiers Industrial Signal Chain — This product selection guide highlights analog-to-digital converters, digital-to-analog converters, amplifiers, comparators, filters, voltage references, RMS-to-DC converters and silicon oscillators designed for demanding industrial applications. These precise, flexible and rugged devices feature parameters fully guaranteed over the –40°C to 85°C temperature range. Battery Charger Solutions — This guide identifies optimum charging solutions for single-cell batteries, multi-cell batteries and battery packs, regardless of chemistry. Linear offers a broad range of charger solutions, including linear chargers, linear chargers with regulators, pulse chargers, switchmode monolithic chargers, switchmode controller chargers, and switchmode smart battery chargers. Software SwitcherCAD™ III/LTC SPICE — LTC SwitcherCAD III is a fully functional SPICE simulator with enhancements and models to ease the simulation of switching regulators. This SPICE is a high performance circuit simulator and integrated waveform viewer, and also includes schematic capture. Our enhancements to SPICE result in much faster simulation of switching regulators than is possible with normal SPICE simulators. SwitcherCAD III includes SPICE, macromodels for 80% of LTC’s switching regulators and over 200 op amp models. It also includes models of resistors, transistors and MOSFETs. With this SPICE simulator, most switching regulator waveforms can be viewed in a few minutes on a high performance PC. Circuits using op amps and transistors can also be easily simulated. Download at www.linear.com FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aided design program for creating filters with Linear Technology’s filter ICs. FilterCAD is designed to help users without special expertise in filter design to design good filters with a minimum of effort. It can also help experienced filter designers achieve better results by playing “what if” with the configuration and values of various components Databooks Amplifiers (Book 1 of 2) — • Operational Amplifiers References, Filters, Comparators, Special Functions, RF & Wireless — • Voltage References • Special Functions • Monolithic Filters • RF & Wireless • Comparators • Optical Communications • Oscillators Monolithic Switching Regulators — • Micropower Switching Regulators • Continuous Switching Regulators Switching Regulator Controllers (Book 1 of 2) — • DC/DC Controllers Switching Regulator Controllers (Book 2 of 2) — • DC/DC Controllers • Digital Voltage Programmers • Off-Line AC/DC Controllers Linear Regulators, Charge Pumps, Battery Chargers — • Linear Regulators • Charge Pump DC/DC Converters • Battery Charging & Management Hot Swap Controllers, MOSFET Drivers, Special Power Functions — • Hot Swap Controllers • Power Switching & MOSFET Drivers • PCMCIA Power Controllers • CCFL Backlight Converters • Special Power Functions Data Converters (Book 1 of 2) — • Analog-to-Digital Converters Data Converters (Book 2 of 2) — • Analog-to-Digital Converters • Digital-to-Analog Converters • Switches & Multiplexers Interface, System Monitoring & Control — • Interface — RS232/562, RS485, Mixed Protocol, SMBus/I2C • System Monitoring & Control — Supervisors, Margining, Sequencing & Tracking Controllers 47 SALES OFFICES NORTH AMERICA GREATER BAY AREA Bay Area 720 Sycamore Dr. Milpitas, CA 95035 Phone: (408) 428-2050 FAX: (408) 432-6331 Sacramento Phone: (408) 432-6326 PACIFIC NORTHWEST Denver Phone: (303) 926-0002 Portland 6700 SW 105th Ave., Ste. 207 Beaverton, OR 97008 Phone: (503) 520-9930 FAX: (503) 520-9929 Salt Lake City Phone: (801) 731-8008 Seattle 2018 156th Ave. NE, Ste. 100 Bellevue, WA 98007 Phone: (425) 748-5010 FAX: (425) 748-5009 SOUTHWEST Los Angeles 21243 Ventura Blvd., Ste. 238 Woodland Hills, CA 91364 Phone: (818) 703-0835 FAX: (818) 703-0517 Orange County 15375 Barranca Pkwy., Ste. 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