ISL6527, ISL6527A ® Data Sheet November 18, 2008 Single Synchronous Buck Pulse-Width Modulation (PWM) Controller The ISL6527, ISL6527A make simple work out of implementing a complete control and protection scheme for a DC/DC step-down converter. Designed to drive N-Channel MOSFETs in a synchronous buck topology, the ISL6527, ISL6527A integrate the control, output adjustment, monitoring and protection functions into a single package. The ISL6527, ISL6527A provide simple, single feedback loop, voltage-mode control with fast transient response. The output voltage can be regulated to as low as the provided external reference. A fixed frequency oscillator reduces design complexity, while balancing typical application cost and efficiency. The error amplifier features a 15MHz gain-bandwidth product and 6V/µs slew rate, which enables high converter bandwidth for fast transient performance. The resulting PWM duty cycles range from 0% to 100%. Protection from overcurrent conditions is provided by monitoring the rDS(ON) of the upper MOSFET to inhibit PWM operation appropriately. This approach simplifies the implementation and improves efficiency by eliminating the need for a current sense resistor. FN9056.10 Features • Operates from 3.3V to 5V Input • Utilizes an External Reference • Drives N-Channel MOSFETs • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High Bandwidth Error Amplifier - Full 0% to 100% Duty Cycle • Lossless, Programmable Overcurrent Protection - Uses Upper MOSFET’s rDS(ON) • Converter Can Source and Sink Current • Small Converter Size - Internal Fixed Frequency Oscillator - ISL6527: 300kHz - ISL6527A: 600kHz • Internal Soft-Start • 14 Ld SOIC or 16 Ld 5x5 QFN • QFN package: - Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat No Leads - Package Outline - Near Chip-Scale Package Footprint, which Improves PCB Efficiency and has a Thinner Profile • Pb-Free (RoHS compliant) Applications • Power Supplies for Microprocessors - PCs - Embedded Controllers • Subsystem Power Supplies - PCI/AGP/GTL+ Busses - ACPI Power Control - DDR SDRAM Bus Termination Supply • Cable Modems, Set-Top Boxes, and DSL Modems • DSP and Core Communications Processor Supplies • Memory Supplies • Personal Computer Peripherals • Industrial Power Supplies • 3.3V Input DC/DC Regulators • Low Voltage Distributed Power Supplies 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2003-2008. All Rights Reserved All other trademarks mentioned are the property of their respective owners. ISL6527, ISL6527A Ordering Information PART NUMBER (Note) PART MARKING TEMP RANGE (°C) PACKAGE (Pb-Free) PKG DWG. # ISL6527ACBZ* 6527ACBZ 0 to +70 14 Ld SOIC M14.15 ISL6527ACRZ* ISL6527 ACRZ 0 to +70 16 Ld 5x5 QFN L16.5x5B ISL6527AIBZ* 6527AIBZ -40 to +85 14 Ld SOIC M14.15 ISL6527AIRZ* ISL6527 AIRZ -40 to +85 16 Ld 5x5 QFN L16.5x5B ISL6527CBZ* 6527CBZ 0 to +70 14 Ld SOIC M14.15 ISL6527CRZ* ISL6527 CRZ 0 to +70 16 Ld 5x5 QFN L16.5x5B ISL6527IBZ* 6527IBZ -40 to +85 14 Ld SOIC M14.15 ISL6527IRZ* ISL6527 IRZ -40 to +85 16 Ld 5x5 QFN L16.5x5B ISL6527EVAL1 ISL6527 SOIC Evaluation Board ISL6527EVAL2 ISL6527 QFN Evaluation Board ISL6527AEVAL1 ISL6527A SOIC Evaluation Board ISL6527AEVAL2 ISL6527A QFN Evaluation Board *Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. Pinouts ISL6527, ISL6527A (16 LD QFN) TOP VIEW 13 BOOT LGATE GND UGATE BOOT ISL6527, ISL6527A (14 LD SOIC) TOP VIEW 12 PHASE 16 15 14 13 14 UGATE GND 1 LGATE 2 CPVOUT 3 CT1 4 11 VCC CT2 5 10 CPGND OCSET/SD 6 9 REF_IN FB 7 8 COMP CPVOUT 1 12 PHASE CT1 2 11 VCC CT2 3 10 CPGND 2 5 6 7 8 FB COMP REF_IN 9 NC OCSET/SD 4 NC FN9056.10 November 18, 2008 ISL6527, ISL6527A Typical Application - 3.3V Input 3.3V VIN CIN CBULK VCC OCSET/SD CT1 ROCSET CPUMP CPVOUT CT2 ISL6527, ISL6527A DBOOT CDCPL CHF BOOT CPGND CBOOT UGATE GND Q1 PHASE VREF REF_IN LGATE COMP Q2 LOUT VOUT COUT FB CI RFB RF CF ROFFSET 3 FN9056.10 November 18, 2008 ISL6527, ISL6527A Typical Application - 5V Input +5V VIN CBULK VCC OCSET/SD CT1 ROCSET CPVOUT N/C ISL6527, ISL6527A CT2 DBOOT CIN CHF BOOT CPGND CBOOT UGATE GND Q1 PHASE VREF REF_IN LGATE COMP Q2 LOUT VOUT COUT FB CI RFB RF CF ROFFSET Block Diagram VCC CT1 CPVOUT CHARGE PUMP CT2 POWER-ON RESET (POR) CPGND BOOT + - OCSET/SD SOFT-START OC COMPARATOR 20μA UGATE ERROR AMP + - REF_IN PWM COMPARATOR + - INHIBIT PHASE GATE CONTROL LOGIC PWM LGATE FB OSCILLATOR COMP FIXED 300kHz or 600kHz GND 4 FN9056.10 November 18, 2008 ISL6527, ISL6527A Functional Pin Description UGATE 14 LD (SOIC) TOP VIEW 14 UGATE GND 1 13 BOOT LGATE 2 12 PHASE CPVOUT 3 CT1 4 11 VCC CT2 5 10 CPGND OCSET/SD 6 9 REF_IN FB 7 8 COMP Connect this pin to the upper MOSFET’s gate. This pin provides the PWM-controlled gate drive for the upper MOSFET. This pin is also monitored by the adaptive shoot-through protection circuitry to determine when the upper MOSFET has turned off. BOOT This pin provides ground referenced bias voltage to the upper MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive a logic-level N-channel MOSFET. LGATE Connect this pin to the lower MOSFET’s gate. This pin provides the PWM-controlled gate drive for the lower MOSFET. This pin is also monitored by the adaptive shoot-through protection circuitry to determine when the lower MOSFET has turned off. LGATE GND UGATE BOOT 16 LD 5X5 (QFN) TOP VIEW 16 15 14 13 CPVOUT 1 OCSET/SD 12 PHASE CT1 2 11 VCC CT2 3 10 CPGND 5 6 7 8 FB COMP REF_IN 9 NC OCSET/SD 4 NC VCC This pin provides the bias supply for the ISL6527, ISL6527A. Connect a well-decoupled 3.3V supply to this pin. COMP and FB Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET (VIN). ROCSET, an internal 20µA current source (IOCSET), and the upper MOSFET ON-resistance (rDS(ON)) set the converter overcurrent (OC) trip point according to Equation 1: I OCSET xR OCSET I PEAK = ------------------------------------------------r DS ( ON ) (EQ. 1) An overcurrent trip cycles the soft-start function. Pulling OCSET/SD to a voltage level below 0.8V disables the controller. Disabling the ISL6527, ISL6527A causes the oscillator to stop, the LGATE and UGATE outputs to be held low, and the soft-start circuitry to re-arm. COMP and FB are two of the three available external pins of the error amplifier. The FB pin is the inverting input of the internal error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the converter. CT1 and CT2 REF_IN This pin represents the output of the charge pump. The voltage at this pin is the bias voltage for the IC. Connect a decoupling capacitor from this pin to ground. The value of the decoupling capacitor should be at least 10x the value of the charge pump capacitor. This pin may be tied to the bootstrap circuit as the source for creating the BOOT voltage. This pin is the third available external pin of the error amplifier and represents the non-inverting input to the error amplifier. Connect the desired reference voltage to this pin. Voltage applied to this pin must not exceed 1.5V. GND This pin represents the signal and power ground for the IC. Tie this pin to the ground island/plane through the lowest impedance connection available. PHASE These pins are the connections for the external charge pump capacitor. A minimum of a 0.1µF ceramic capacitor is recommended for proper operation of the IC. CPVOUT CPGND This pin represents the signal and power ground for the charge pump. Tie this pin to the ground island/plane through the lowest impedance connection available. Connect this pin to the upper MOSFET’s source. This pin is used to monitor the voltage drop across the upper MOSFET for overcurrent protection. 5 FN9056.10 November 18, 2008 ISL6527, ISL6527A Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+7V Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V Absolute Error Amplifier Input, VEA+ . . . . . . . . . . . . . . . . . . . . +1.5V Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . +7.0V Input, Output or I/O Voltage . . . . . . . . . . . GND -0.3V to VCC +0.3V Thermal Resistance θJA (°C/W) θJC (°C/W) SOIC Package (Note 1) . . . . . . . . . . . . 67 N/A QFN Package (Notes 2, 3). . . . . . . . . . 35 5 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . +150°C Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Operating Conditions Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +3.3V ±10% Ambient Temperature Range . . . . . . . . . . . . . . . . . . .-40°C to +85°C Junction Temperature Range. . . . . . . . . . . . . . . . . .-40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 3. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Recommended Operating Conditions, unless otherwise noted VCC = 3.3V ±5% and TA = +25°C PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS 6.1 7.1 7.7 mA Commercial 4.25 4.30 4.42 V Industrial 4.10 4.30 4.50 V 0.3 0.6 0.9 V IC = ISL6527C, Commercial 275 300 325 kHz IC = ISL6527I, Industrial 250 300 340 kHz IC = ISL6527AC, Commercial 554 600 645 kHz IC = ISL6527AI, Industrial 524 600 650 kHz - 1.5 - VP-P - 5.1 - V - 2 - % (Note 4) - 88 - dB GBWP (Note 4) - 15 - MHz SR (Note 4) - 6 - V/µs VREF_IN (Note 4) - - 1.5 V Commercial; VREF_IN = 1.5V 6.2 - 7.3 ms Industrial; VREF_IN = 1.5V 6.2 - 7.6 ms VCC SUPPLY CURRENT Nominal Supply IBIAS POWER-ON RESET Rising CPVOUT POR Threshold POR CPVOUT POR Threshold Hysteresis OSCILLATOR Frequency fOSC ΔVOSC Ramp Amplitude CHARGE PUMP Nominal Charge Pump Output VCPVOUT VVCC = 3.3V, No Load Charge Pump Output Regulation ERROR AMPLIFIER DC Gain Gain-Bandwidth Product Slew Rate Non-Inverting Input Voltage SOFT-START Soft-start Slew Rate GATE DRIVERS Upper Gate Source Current IUGATE-SRC VBOOT - VPHASE = 5V, VUGATE = 4V - -1 - A Upper Gate Sink Current IUGATE-SNK - 1 - A Lower Gate Source Current ILGATE-SRC VVCC = 3.3V, VLGATE = 4V - -1 - A Lower Gate Sink Current ILGATE-SNK - 2 - A 6 FN9056.10 November 18, 2008 ISL6527, ISL6527A Electrical Specifications Recommended Operating Conditions, unless otherwise noted VCC = 3.3V ±5% and TA = +25°C (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS Commercial 18 20 22 µA Industrial 16 20 22 µA - - 0.8 V PROTECTION/DISABLE OCSET Current Source IOCSET Disable Threshold VOCSET/SD NOTE: 4. Limits should be considered typical and are not production tested. Functional Description . Initialization (1V/DIV) CPVOUT (5V) The ISL6527, ISL6527A automatically initialize upon receipt of power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the the output voltage of the Charge Pump. During POR, the Charge Pump operates on a free running oscillator. Once the POR level is reached, the Charge Pump oscillator is synched to the PWM oscillator. The POR function also initiates the soft-start operation after the Charge Pump Output Voltage exceeds its POR threshold. Soft-Start The POR function initiates the digital soft-start sequence. The PWM error amplifier reference is clamped to a level proportional to the soft-start voltage. As the soft-start voltage slews up, the PWM comparator generates PHASE pulses of increasing width that charge the output capacitor(s). This method provides a rapid and controlled output voltage rise. The soft-start sequence typically takes about 6.5ms when VREF_IN is 1.5V If VREF_IN is less that 1.5V, the soft-start rise time will be proportionally smaller as shown in Equation 2: V REFIN RiseTime = t SS × --------------------1.5V (EQ. 2) Figure 1 shows the soft-start sequence for a typical application. At T0, the +3.3V VCC voltage starts to ramp. At time t1, the Charge Pump begins operation and the +5V CPVOUT IC bias voltage starts to ramp up. Once the voltage on CPVOUT crosses the POR threshold at time t2, the output begins the soft-start sequence. The triangle waveform from the PWM oscillator is compared to the rising error amplifier output voltage. As the error amplifier voltage increases, the pulse-width on the UGATE pin increases to reach the steady-state duty cycle at time t3. 7 VCC (3.3V) VOUT (2.50V) 0V t0 t1 t2 t3 TIME FIGURE 1. SOFT-START INTERVAL Shoot-Through Protection A shoot-through condition occurs when both the upper MOSFET and lower MOSFET are turned on simultaneously, effectively shorting the input voltage to ground. To protect the regulator from a shoot-through condition, the ISL6527, ISL6527A incorporate specialized circuitry, which insures that the complementary MOSFETs are not ON simultaneously. The adaptive shoot-through protection utilized by the ISL6527, ISL6527A looks at the lower gate drive pin, LGATE, and the upper gate drive pin, UGATE, to determine whether a MOSFET is ON or OFF. If the voltage from UGATE or from LGATE to GND is less than 0.8V, then the respective MOSFET is defined as being OFF and the complementary MOSFET is turned ON. This method of shoot-through protection allows the regulator to sink or source current. Since the voltage of the lower MOSFET gate and the upper MOSFET gate are being measured to determine the state of the MOSFET, the designer is encouraged to consider the repercussions of introducing external components between the gate drivers and their respective MOSFET gates before actually implementing such measures. Doing so may interfere with the shoot-through protection. FN9056.10 November 18, 2008 ISL6527, ISL6527A Output Voltage Selection The output voltage can be programmed to any level between VIN and the supplied external reference. An external resistor divider is used to scale the output voltage relative to the reference voltage and feed it back to the inverting input of the error amplifier, see Figure 2. However, since the value of R1 affects the values of the rest of the compensation components, it is advisable to keep its value less than 5kΩ. R4 can be calculated based on Equation 3: R 1 × V REF R 4 = ---------------------------------------V OUT1 – V REF is quickly shutdown and the internal soft-start function begins producing soft-start ramps. The delay interval seen by the output is equivalent to three soft-start cycles. The fourth internal soft-start cycle initiates a normal soft-start ramp of the output, at time t1. The output is brought back into regulation by time t2, as long as the overcurrent event has cleared. VOUT (2.5V) (EQ. 3) If the output voltage desired is VREF, simply route the output back to the FB pin through R1, but do not populate R4. +3.3V 0V VIN VCC CPVOUT INTERNAL SOFT-START FUNCTION D1 BOOT C4 Q1 UGATE ISL6527, ISL6527A DELAY INTERVAL LOUT PHASE Q2 LGATE VOUT + COUT t1 t0 t2 TIME FB C1 COMP R1 C3 R3 R2 C2 R4 REF_IN VREF FIGURE 3. OVERCURRENT PROTECTION RESPONSE Had the cause of the overcurrent still been present after the delay interval, the overcurrent condition would be sensed and the regulator would be shut down again for another delay interval of three soft-start cycles. The resulting hiccup mode style of protection would continue to repeat indefinitely. FIGURE 2. OUTPUT VOLTAGE SELECTION The overcurrent function will trip at a peak inductor current (IPEAK) determined by Equation 4: Overcurrent Protection The overcurrent function protects the converter from a shorted output by using the upper MOSFET ON-resistance, rDS(ON), to monitor the current. This method enhances the converter’s efficiency and reduces cost by eliminating a current sensing resistor. The overcurrent function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) programs the overcurrent trip level (see Typical Application diagrams on page 3 and page 4). An internal 20µA (typical) current sink develops a voltage across ROCSET that is referenced to VIN. When the voltage across the upper MOSFET (also referenced to VIN) exceeds the voltage across ROCSET, the overcurrent function initiates a soft-start sequence. Figure 3 illustrates the protection feature responding to an overcurrent event. At time t0, an overcurrent condition is sensed across the upper MOSFET. As a result, the regulator 8 I OCSET x R OCSET I PEAK = ----------------------------------------------------r DS ( ON ) (EQ. 4) where IOCSET is the internal OCSET current source (20µA typical). The OC trip point varies mainly due to the MOSFET rDS(ON) variations. To avoid overcurrent tripping in the normal operating load range, find the ROCSET resistor from the Equation 4 with: 1. The maximum rDS(ON) at the highest junction temperature. 2. The minimum IOCSET from the “Specification Table” on page 7. ( ΔI ) 3. Determine IPEAK for I PEAK > I OUT ( MAX ) + --------2 , where ΔI is the output inductor ripple current. FN9056.10 November 18, 2008 ISL6527, ISL6527A For an equation for the ripple current see the section under component guidelines titled “Output Inductor Selection” on page 11. A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage. Current Sinking The ISL6527, ISL6527A incorporate a MOSFET shoot-through protection method, which allows a converter to sink current as well as source current. Care should be exercised when designing a converter with the ISL6527, ISL6527A when it is known that the converter may sink current. When the converter is sinking current, it is behaving as a boost converter that is regulating its input voltage. This means that the converter is boosting current into the input rail of the regulator. If there is nowhere for this current to go, such as to other distributed loads on the rail or through a voltage limiting protection device, the capacitance on this rail will absorb the current. This situation will allow the voltage level of the input rail to increase. If the voltage level of the rail is boosted to a level that exceeds the maximum voltage rating of any components attached to the input rail, then those components may experience an irreversible failure or experience stress that may shorten their lifespan. Ensuring that there is a path for the current to flow other than the capacitance on the rail will prevent this failure mode. External Reference The ISL6527, ISL6527A allow the designer to determine the reference voltage that is used. This allows the ISL6527, ISL6527A to be used in many specialized applications, such as the VTT termination voltage in a DDR Memory power supply, which must track the VDDQ voltage by 50%. Care must be taken to insure that this voltage does not exceed 1.5V. Application Guidelines Layout Considerations Layout is very important in high frequency switching converter design. With power devices switching efficiently at 300kHz or 600kHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device over-voltage stress. Careful component layout and printed circuit board design minimizes the voltage spikes in the converters. As an example, consider the turn-off transition of the PWM MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the lower MOSFET. Any parasitic inductance in the switched current path generates 9 a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in a DC/DC converter using the ISL6527, ISL6527A. The switching components are the most critical because they switch large amounts of energy, and therefore tend to generate large amounts of noise. Next, are the small signal components, which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 4 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the GATE pins to the MOSFET gates should be kept short and wide enough to easily handle the 1A of drive current. The switching components should be placed close to the ISL6527, ISL6527A first. Minimize the length of the connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper MOSFET and lower MOSFET and the load. The critical small signal components include any bypass capacitors, feedback components, and compensation components. Position the bypass capacitor, CBP, close to the VCC pin with a via directly to the ground plane. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors for both regulators should also be located as close as possible to the relevant FB pin with vias tied straight to the ground plane as required. Feedback Compensation Figure 5 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). FN9056.10 November 18, 2008 ISL6527, ISL6527A +3.3V VIN VIN DRIVER OSC ISL6527, ISL6527A PWM COMPARATOR VCC + DVOSC CVCC LO CPVOUT DRIVER VOUT PHASE CO ESR (PARASITIC) ZFB VE/A CBP CIN GND ZIN + D1 BOOT ERROR AMP REFERENCE CBOOT DETAILED COMPENSATION COMPONENTS Q1 UGATE LOUT ZFB VOUT PHASE VOUT C1 ZIN PHASE COUT C2 LOAD Q2 LGATE C3 R2 R3 R1 COMP COMP FB C2 + C1 R2 R1 ISL6527 FB R4 REFERENCE C3 R3 FIGURE 5. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN KEY ISLAND ON POWER PLANE LAYER these guidelines for locating the poles and zeros of the compensation network: ISLAND ON CIRCUIT PLANE LAYER 1. Pick gain (R2/R1) for desired converter bandwidth. VIA CONNECTION TO GROUND PLANE FIGURE 4. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage ΔVOSC . Modulator Break Frequency Equations 1 F LC = -----------------------------------------2π x L O x C O (EQ. 5) 1 F ESR = ------------------------------------------2π x ESR x C O (EQ. 6) The compensation network consists of the error amplifier (internal to the ISL6527, ISL6527A) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180°. Equations 7 through 10 relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 5. Use 10 2. Place first zero below filter’s double pole (~75% FLC). 3. Place second zero at filter’s double-pole. 4. Place first pole at the ESR zero. 5. Place second pole at half the switching frequency. 6. Check gain against error amplifier’s open-loop gain. 7. Estimate phase margin; repeat if necessary. Compensation Break Frequency Equations 1 F Z1 = ---------------------------------2π × R 2 × C 2 (EQ. 7) 1 F P1 = --------------------------------------------------------⎛ C 1 x C 2⎞ 2π x R 2 x ⎜ ----------------------⎟ ⎝ C1 + C2 ⎠ (EQ. 8) 1 F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3 (EQ. 9) 1 F P2 = -----------------------------------2π x R 3 x C 3 (EQ. 10) Figure 6 shows an asymptotic plot of the DC/DC converter’s gain vs frequency. The actual modulator gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 6. Using the above guidelines should give a compensation gain similar to the curve plotted. The open loop FN9056.10 November 18, 2008 ISL6527, ISL6527A error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The closed loop gain is constructed on the graph of Figure 6 by adding the modulator gain (in dB) to the compensation gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45°. Include worst-case component variations when determining phase margin. FZ1 FZ2 FP1 FP2 100 OPEN LOOP ERROR AMP GAIN ⎛ V IN ⎞ 20 log ⎜ ------------------⎟ ⎝ V OSC⎠ 80 GAIN (dB) 60 40 COMPENSATION GAIN 20 0 -20 -40 -60 R2 20 log ⎛ --------⎞ ⎝ R1⎠ MODULATOR GAIN 10 100 the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the Equivalent Series Inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Output Inductor Selection LOOP GAIN FLC 1k FESR 10k 100k 1M 10M FREQUENCY (Hz) FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Component Selection Guidelines ΔI = Charge Pump Capacitor Selection A capacitor across pins CT1 and CT2 is required to create the proper bias voltage for the ISL6527, ISL6527A when operating the IC from 3.3V. Selecting the proper capacitance value is important so that the bias current draw and the current required by the MOSFET gates do not overburden the capacitor. A conservative approach is presented in Equation 11: I BiasAndGate C PUMP = ------------------------------------ × 1.5 V CC × f s (EQ. 11) Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern digital ICs can produce high transient load slew rates. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by 11 The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by Equations 12 and 13: VIN - VOUT fs x L x ΔVOUT = ΔI x ESR VOUT VIN (EQ. 12) (EQ. 13) Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL6527, ISL6527A will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. Equations 14 and 15 give the FN9056.10 November 18, 2008 ISL6527, ISL6527A approximate response time interval for application and removal of a transient load: tRISE = tFALL = L x ITRAN VIN - VOUT (EQ. 14) L x ITRAN VOUT (EQ. 15) where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. The worst case response time can be either at the application or removal of load. Be sure to check both Equations 14 and 15 at the minimum and maximum output levels for the worst case response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2 . factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor. The switching losses seen when sourcing current will be different from the switching losses seen when sinking current. When sourcing current, the upper MOSFET realizes most of the switching losses. The lower switch realizes most of the switching losses when the converter is sinking current (see Equations 17 and 18). These equations assume linear voltage-current transitions and do not adequately model power loss due the reverse-recovery of the upper and lower MOSFET’s body diode. The gate-charge losses are dissipated by the ISL6527, ISL6527A and don't heat the MOSFETs. However, large gate-charge increases the switching interval, tSW which increases the MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25x greater than the maximum input voltage and a voltage rating of 1.5x is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. Losses while Sourcing current The maximum RMS current required by the regulator may be closely approximated through Equation 16: where: D is the duty cycle = VOUT/VIN, tSW is the combined switch ON- and OFF-time, and fS is the switching frequency. I RMS MAX = 2 V OUT ⎛ 2 1 ⎛ V IN – V OUT V OUT⎞ ⎞ ---------------- × ⎜ I OUT + ------ × ⎜ -------------------------------- × ----------------⎟ ⎟ V IN V IN ⎠ ⎠ L × fs 12 ⎝ MAX ⎝ (EQ. 16) For a through-hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. Some capacitor series available from reputable manufacturers are surge current tested. MOSFET Selection/Considerations The ISL6527, ISL6527A require two N-Channel power MOSFETs. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design 12 2 1 P UPPER = Io × r DS ( ON ) × D + --- ⋅ Io × V IN × t SW × f s 2 PLOWER = Io2 x rDS(ON) x (1 - D) (EQ. 17) Losses while Sinking current PUPPER = Io2 x rDS(ON) x D 2 1 P LOWER = Io × r DS ( ON ) × ( 1 – D ) + --- ⋅ Io × V IN × t SW × f s 2 (EQ. 18) Given the reduced available gate bias voltage (5V), logic-level or sub-logic-level transistors should be used for both N-MOSFETs. Caution should be exercised with devices exhibiting very low VGS(ON) characteristics. The shoot-through protection present aboard the ISL6527, ISL6527A may be circumvented by these MOSFETs if they have large parasitic impedances and/or capacitances that would inhibit the gate of the MOSFET from being discharged below its threshold level before the complementary MOSFET is turned on. Bootstrap Component Selection External bootstrap components, a diode and capacitor, are required to provide sufficient gate enhancement to the upper MOSFET. The internal MOSFET gate driver is supplied by the external bootstrap circuitry as shown in Figure 7. The boot capacitor, CBOOT, develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle, when DBOOT conducts, to a voltage of CPVOUT less the boot diode drop, VD, plus the voltage rise across QLOWER. FN9056.10 November 18, 2008 ISL6527, ISL6527A The minimum bootstrap capacitance can be calculated by rearranging Equation 19 and solving for CBOOT in Equation 20. CPVOUT DBOOT ISL6527, ISL6527A + VD - VIN Q GATE C BOOT ≥ ----------------------------------------------------V BOOT1 – V BOOT BOOT2 CBOOT UGATE QUPPER PHASE + NOTE: VG-S ≈ VCC -VD QLOWER LGATE NOTE: VG-S ≈ VCC GND FIGURE 7. UPPER GATE DRIVE BOOTSTRAP Just after the PWM switching cycle begins and the charge transfer from the bootstrap capacitor to the gate capacitance is complete, the voltage on the bootstrap capacitor is at its lowest point during the switching cycle. The charge lost on the bootstrap capacitor will be equal to the charge transferred to the equivalent gate-source capacitance of the upper MOSFET as shown in Equation 19: Q GATE = C BOOT × ( V BOOT1 – V BOOT2 ) (EQ. 19) where QGATE is the maximum total gate charge of the upper MOSFET, CBOOT is the bootstrap capacitance, VBOOT1 is the bootstrap voltage immediately before turn-on, and VBOOT2 is the bootstrap voltage immediately after turn-on. The bootstrap capacitor begins its refresh cycle when the gate drive begins to turn-off the upper MOSFET. A refresh cycle ends when the upper MOSFET is turned on again, which varies depending on the switching frequency and duty cycle. (EQ. 20) Typical gate charge values for MOSFETs considered in these types of applications range from 20 to 100nC. Since the voltage drop across QLOWER is negligible, VBOOT1 is simply VCPVOUT - VD. A schottky diode is recommended to minimize the voltage drop across the bootstrap capacitor during the on-time of the upper MOSFET. Initial calculations with VBOOT2 no less than 4V will quickly help narrow the bootstrap capacitor range. For example, consider an upper MOSFET is chosen with a maximum gate charge, Qg, of 100nC. Limiting the voltage drop across the bootstrap capacitor to 1V results in a value of no less than 0.1µF. The tolerance of the ceramic capacitor should also be considered when selecting the final bootstrap capacitance value. A fast recovery diode is recommended when selecting a bootstrap diode to reduce the impact of reverse recovery charge loss. Otherwise, the recovery charge, QRR, would have to be added to the gate charge of the MOSFET and taken into consideration when calculating the minimum bootstrap capacitance. ISL6527, ISL6527A DC/DC Converter Application Circuit Figure 8 shows an application circuit of a DC/DC Converter. Detailed information on the circuit, including a complete Bill-of-Materials and circuit board description, can be found in Application Note AN1021. http://www.intersil.com/data/an/an1021.pdf 13 FN9056.10 November 18, 2008 ISL6527, ISL6527A 3.3V C1 0.1µF GND 11 U1 TP1 4 VCC OCSET/SD CT1 C2 1000pF C3 6 R1 9.76kΩ TP3 C4 0.22µF CPVOUT 5 ISL6527, ISL6527A CT2 10 1 0.8V EXTERNAL REFERENCE BOOT 13 CPGND C7 UGATE GND REF_IN LGATE COMP 8 C5 10µF CERAMIC D1 PHASE 9 3 0.1μF 14 L1 12 2.5V @ 5A 2 C8,9 Q1 FB 7 C10 R3 33pF R2 C6 1µF C11 2.26kΩ R4 6.49kΩ 5600pF R5 C12 124Ω 8200pF GND Component Selection Notes: C3, C8, C9 - Each 150mF, Panasonic EEF-UE0J151R or Equivalent. D1 - 30mA Schottky Diode, MA732 or Equivalent L1 - 1µH Inductor, Panasonic P/N ETQ-P6F1ROSFA or Equivalent. Q1- Fairchild MOSFET; ITF86110DK8. FIGURE 8. 3.3V TO 2.5V 5A DC/DC CONVERTER 14 FN9056.10 November 18, 2008 ISL6527, ISL6527A Package Outline Drawing L16.5x5B 16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 2, 02/08 4X 2.4 5.00 12X 0.80 A B 13 6 PIN 1 INDEX AREA 6 PIN #1 INDEX AREA 16 12 5.00 1 3 . 10 ± 0 . 15 9 (4X) 4 0.15 5 8 TOP VIEW 0.10 M C A B +0.15 16X 0 . 60 -0.10 4 0.33 +0.07 / -0.05 BOTTOM VIEW SEE DETAIL "X" 0.10 C 1.00 MAX C BASE PLANE SEATING PLANE 0.08 C ( 4 . 6 TYP ) SIDE VIEW ( ( 12X 0 . 80 ) 3 . 10 ) C ( 16X 0 .33 ) 0 . 2 REF 5 ( 16 X 0 . 8 ) 0 . 00 MIN. 0 . 05 MAX. TYPICAL RECOMMENDED LAND PATTERN DETAIL "X" NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 15 FN9056.10 November 18, 2008 ISL6527, ISL6527A Small Outline Plastic Packages (SOIC) M14.15 (JEDEC MS-012-AB ISSUE C) N INDEX AREA H 0.25(0.010) M 14 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE B M E INCHES -B- 1 2 3 L SEATING PLANE -A- h x 45o A D -C- α e A1 B 0.25(0.010) M C A M SYMBOL MIN MAX MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - A1 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.3367 0.3444 8.55 8.75 3 E 0.1497 0.1574 3.80 4.00 4 e C 0.10(0.004) B S 0.050 BSC 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. 1.27 BSC - H 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 N NOTES: MILLIMETERS α 14 0o 14 8o 0o 7 8o Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 16 FN9056.10 November 18, 2008