NCV8870 D

NCV8870
Automotive Grade
Non-Synchronous Boost
Controller
The NCV8870 is an adjustable output non−synchronous boost
controller which drives an external N-channel MOSFET. The device
uses peak current mode control with internal slope compensation.
The IC incorporates an internal regulator that supplies charge to the
gate driver.
Protection features include internally-set soft-start, undervoltage
lockout, cycle-by-cycle current limiting, hiccup-mode short-circuit
protection and thermal shutdown.
Additional features include low quiescent current sleep mode and
externally-synchronizable switching frequency.
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SOIC−8
D SUFFIX
CASE 751
PIN CONNECTIONS
Features
•
•
•
•
•
•
•
•
•
•
•
•
Peak Current Mode Control with Internal Slope Compensation
1.2 V ±2% Reference voltage
Fixed Frequency Operation
Wide Input Voltage Range of 3.2 V to 40 Vdc, 45 V Load Dump
Input Undervoltage Lockout (UVLO)
Internal Soft-Start
Low Quiescent Current in Sleep Mode
Cycle-by-Cycle Current Limit Protection
Hiccup-Mode Overcurrent Protection (OCP)
Hiccup-Mode Short-Circuit Protection (SCP)
Thermal Shutdown (TSD)
This is a Pb-Free Device
EN/SYNC 1
8 VFB
ISNS 2
7 VC
GND 3
6 VIN
GDRV 4
5 VDRV
MARKING DIAGRAM
8
8870xx
ALYW
G
1
(Top View)
8870xx = Specific Device Code
xx = 00, 01
A
= Assembly Location
L
= Wafer Lot
Y
= Year
W
= Work Week
G
= Pb−Free Package
ORDERING INFORMATION
Device
Package
Shipping†
NCV887000D1R2G
SOIC−8
(Pb−Free)
2500 /
Tape & Reel
NCV887001D1R2G
SOIC−8
(Pb−Free)
2500 /
Tape & Reel
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
© Semiconductor Components Industries, LLC, 2015
December, 2015 − Rev. 9
1
Publication Order Number:
NCV8870/D
NCV8870
6
TEMP
VDRV
FAULT
LOGIC
EN/
5
Vg
VIN
Cg
L
CDRV
VDRV
D
Vo
CLK
EN/SYNC
OSC
1
SYNC
DRIVE
LOGIC
VC
PWM
SC
7
4
2
CL
CSA
3
+
Q
GDRV
ISNS
RF1
SCP
RC
CC
Co
RSNS
GND
8
Gm
VFB
RF2
SS
Vref
Figure 1. Simplified Block Diagram and Application Schematic
PACKAGE PIN DESCRIPTIONS
Pin No.
Pin Symbol
1
EN/SYNC
Function
2
ISNS
Current sense input. Connect this pin to the source of the external N-MOSFET, through a current-sense resistor to ground to sense the switching current for regulation and current limiting.
3
GND
Ground reference.
4
GDRV
Gate driver output. Connect to gate of the external N-MOSFET. A series resistance can be added
from GDRV to the gate to tailor EMC performance. An RGND = 15 kW (typical) GDRV−GND resistor is
strongly recommended.
5
VDRV
Driving voltage. Internally-regulated supply for driving the external N-MOSFET, sourced from VIN.
Bypass with a 1.0 mF ceramic capacitor to ground.
6
VIN
Input voltage. If bootstrapping operation is desired, connect a diode from the input supply to VIN,
in addition to a diode from the output voltage to VDRV and/or VIN.
7
VC
Output of the voltage error amplifier. An external compensator network from VC to GND is used to
stabilize the converter.
8
VFB
Output voltage feedback. A resistor from the output voltage to VFB with another resistor from VFB to
GND creates a voltage divider for regulation and programming of the output voltage.
Enable and synchronization input. The falling edge synchronizes the internal oscillator. The part is
disabled into sleep mode when this pin is brought low for longer than the enable time-out period.
ABSOLUTE MAXIMUM RATINGS (Voltages are with respect to GND, unless otherwise indicated)
Rating
Value
Unit
−0.3 to 40
V
Peak Transient Voltage (Load Dump on VIN)
45
V
Dc Supply Voltage (VDRV, GDRV)
12
V
−0.3 to 6
V
−0.3 to 3.6
V
Dc Voltage (EN/SYNC)
−0.3 to 6
V
Dc Voltage Stress (VIN − VDRV)*
−0.7 to 40
V
Operating Junction Temperature
−40 to 150
°C
Storage Temperature Range
−65 to 150
°C
Peak Reflow Soldering Temperature: Pb−Free, 60 to 150 seconds at 217°C
265 peak
°C
Dc Supply Voltage (VIN)
Peak Transient Voltage (VFB)
Dc Voltage (VC, VFB, ISNS)
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
*An external diode from the input to the VIN pin is required if bootstrapping VDRV and VIN off of the output voltage.
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2
NCV8870
PACKAGE CAPABILITIES
Characteristic
ESD Capability (All Pins)
Human Body Model
Machine Model
Value
Unit
≥ 2.0
≥ 200
kV
V
1
−
100
°C/W
Moisture Sensitivity Level
Package Thermal Resistance
Junction-to-Ambient, RqJA (Note 1)
1. 1 in2, 1 oz copper area used for heatsinking.
Device Variations
The NCV8870 features several variants to better fit a
multitude of applications. The table below shows the typical
values of parameters for the parts that are currently
available.
TYPICAL VALUES
Part No.
Dmax
fs
tss
Sa
Vcl
Isrc
Isink
VDRV
SCE
NCV887000
93%
50 kHz
26 ms
15 mV/ms
400 mV
800 mA
600 mA
10.5 V
Y
NCV887001
93%
100 kHz
13 ms
33 mV/ms
400 mV
800 mA
600 mA
10.5 V
Y
DEFINITIONS
Symbol
Characteristic
Dmax
Symbol
Characteristic
Symbol
Characteristic
Maximum Duty Cycle
fs
Switching Frequency
tss
Soft-Start Time
Sa
Slope Compensating Ramp
Vcl
Current Limit Trip Voltage
Isrc
Gate Drive Sourcing Current
Isink
Gate Drive Sinking Current
VDRV
Drive Voltage
SCE
Short Circuit Enable
ELECTRICAL CHARACTERISTICS
(−40°C < TJ < 150°C, 3.2 V < VIN < 40 V, unless otherwise specified) Min/Max values are guaranteed by test, design or statistical correlation.
Characteristic
Symbol
Conditions
Min
Typ
Max
Unit
GENERAL
Quiescent Current, Sleep Mode
Iq,sleep
VIN = 13.2 V, EN = 0, TJ = 25°C
−
2.0
−
mA
Quiescent Current, Sleep Mode
Iq,sleep
VIN = 13.2 V, EN = 0, −40°C < TJ < 125°C
−
2.0
6.0
mA
Quiescent Current, No switching
Iq,off
Into VIN pin, EN = 1, No Switching
−
1.5
2.5
mA
Quiescent Current, Switching,
Normal Operation
Iq,on
Into VIN pin, EN = 1, Switching
−
3.0
6.0
mA
200
250
300
ns
OSCILLATOR
Minimum Pulse Width
ton,min
Maximum Duty Cycle
Dmax
NCV887000
NCV887001
91
91
93
93
95
95
%
Switching Frequency
fs
NCV887000
NCV887001
45
90
50
100
55
110
kHz
Soft-Start Time
tss
From start of switching with VFB = 0 until
reference voltage = VREF
NCV887000
NCV887001
21
10.5
26
13
31
15.5
From EN → 1 until start of switching with
VFB = 0
−
720
840
NCV887000
NCV887001
12
28
15
33
18
38
Soft-Start Delay
Slope Compensating Ramp
tss,dly
Sa
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3
ms
ms
mV/ms
NCV8870
ELECTRICAL CHARACTERISTICS (continued)
(−40°C < TJ < 150°C, 3.2 V < VIN < 40 V, unless otherwise specified) Min/Max values are guaranteed by test, design or statistical correlation.
Characteristic
Symbol
Conditions
Min
Typ
Max
Unit
−
5.0
10
mA
2.0
−
5.0
V
ENABLE/SYNCHRONIZATION
EN/SYNC Pull-down Current
IEN/SYNC
EN/SYNC Input High Voltage
Vs,ih
EN/SYNC Input Low Voltage
Vs,il
EN/SYNC Time-out Ratio
%ten
SYNC Minimum Frequency Ratio
SYNC Maximum Frequency
%fsync,min
VEN/SYNC = 5 V
0
−
800
mV
From SYNC falling edge, to oscillator control (EN high) or shutdown (EN low), Percent of typical switching period
−
−
350
%
Percent of fs
−
−
80
%
fsync,max
Synchronization Delay
ts,dly
Synchronization Duty Cycle
Dsync
1.1
−
−
MHz
−
50
100
ns
25
−
75
%
Input-to-output gain at dc, ISNS ≤ 1 V
0.9
1.0
1.1
V/V
2.5
−
−
MHz
−
30
50
mA
360
360
400
400
440
440
−
80
125
ns
125
150
175
%
−
80
125
ns
0.8
1.2
1.5
mS
From SYNC falling edge to GDRV falling
edge
CURRENT SENSE AMPLIFIER
Low-frequency Gain
Acsa
Bandwidth
BWcsa
Gain of Acsa − 3 dB
ISNS Input Bias Current
Isns,bias
Out of ISNS pin
Current Limit Threshold Voltage
Current Limit,
Response Time
Vcl
tcl
Overcurrent Protection,
Threshold Voltage
%Vocp
Overcurrent Protection,
Response Time
tocp
Voltage on ISNS pin
NCV887000
NCV887001
CL tripped until GDRV falling edge,
VISNS = Vcl(typ) + 60 mV
Percent of Vcl
From overcurrent event, Until switching
stops, VISNS = VOCP + 40 mV
mV
VOLTAGE ERROR OPERATIONAL TRANSCONDUCTANCE AMPLIFIER
Transconductance
gm,vea
VEA Output Resistance
Ro,vea
VFB Input Bias Current
Ivfb,bias
VFB – Vref = ±20 mV
2.0
−
−
MW
−
0.5
2.0
mA
Vref
1.176
1.200
1.224
V
VEA Maximum Output Voltage
Vc,max
2.5
−
−
V
VEA Minimum Output Voltage
Vc,min
−
−
0.3
V
VEA Sourcing Current
Isrc,vea
VEA output current, Vc = 2.0 V
80
100
−
mA
VEA Sinking Current
Isnk,vea
VEA output current, Vc = 0.7 V
80
100
−
mA
VDRV ≥ 6 V, VDRV − VGDRV = 2 V
NCV887000
NCV887001
600
600
800
800
−
−
VGDRV ≥ 2 V
NCV887000
NCV887001
500
500
600
600
−
−
VIN − VDRV, IvDRV = 10 mA
−
0.2
0.35
V
VIN − VDRV = 1 V
10
15
−
mA
V
Reference Voltage
Current out of VFB pin
GATE DRIVER
Sourcing Current
Sinking Current
Driving Voltage Dropout
Isrc
Isink
Vdrv,do
mA
mA
Driving Voltage Source Current
Idrv
Backdrive Diode Voltage Drop
Vd,bd
VDRV − VIN, Id,bd = 5 mA
−
−
0.7
Driving Voltage
VDRV
IVDRV = 0.1 − 25 mA
NCV887000
NCV887001
10
10
10.5
10.5
11
11
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4
V
NCV8870
ELECTRICAL CHARACTERISTICS (continued)
(−40°C < TJ < 150°C, 3.2 V < VIN < 40 V, unless otherwise specified) Min/Max values are guaranteed by test, design or statistical correlation.
Characteristic
Symbol
Conditions
Min
Typ
Max
Unit
UVLO
Undervoltage Lock-out,
Threshold Voltage
Vuvlo
VIN falling
3.0
3.1
3.2
V
Undervoltage Lock-out,
Hysteresis
Vuvlo,hys
VIN rising
50
125
200
mV
Startup Blanking Period
%tscp,dly
From start of soft-start, Percent of tss
100
120
150
%
Hiccup-mode Period
%thcp,dly
From shutdown to start of soft-start,
Percent of tss
65
80
95
%
VFB as percent of Vref
60
67
75
%
tscp
From VFB < Vscp to stop switching
−
35
100
ns
Thermal Shutdown Threshold
Tsd
TJ rising
160
170
180
°C
Thermal Shutdown Hysteresis
Tsd,hys
TJ falling
10
15
20
°C
Thermal Shutdown Delay
tsd,dly
From TJ > Tsd to stop switching
−
−
100
ns
SHORT CIRCUIT PROTECTION
Short Circuit Threshold Voltage
Short Circuit Delay
%Vscp
THERMAL SHUTDOWN
Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product
performance may not be indicated by the Electrical Characteristics if operated under different conditions.
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5
NCV8870
TYPICAL PERFORMANCE CHARACTERISTICS
6
7
Iq,sleep, SLEEP CURRENT (mA)
Iq,sleep, SLEEP CURRENT (mA)
TJ = 25°C
6
5
4
3
2
1
0
0
10
20
30
VIN, INPUT VOLTAGE (V)
VIN = 13.2 V
5
4
3
2
1
0
−50
40
Figure 2. Sleep Current vs. Input Voltage
100
150
200
252
VIN = 13.2 V
3.35
ton,min MINIMUM ON TIME (ns)
Iq,on, QUIESCENTCURRENT (mA)
50
Figure 3. Sleep Current vs. Temperature
3.40
fs = 100 kHz
3.30
3.25
3.20
3.15
3.10
3.05
−50
fs = 50 kHz
0
50
100
150
TJ, JUNCTION TEMPERATURE (°C)
250
248
246
244
242
240
238
236
−50
200
0
50
100
150
200
TJ, JUNCTION TEMPERATURE (°C)
Figure 4. Quiescent Current vs. Temperature
Figure 5. Minimum On Time vs. Temperature
1.010
Vref, REFERENCE VOLTAGE (V)
NORMALIZED CURRENT LIMIT (25°C)
1.205
1.005
1.000
0.995
0.990
−40
0
TJ, JUNCTION TEMPERATURE (°C)
10
60
110
TJ, JUNCTION TEMPERATURE (°C)
1.203
1.201
1.199
1.197
1.195
−40
160
10
60
110
TJ, JUNCTION TEMPERATURE (°C)
Figure 6. Normalized Current Limit vs.
Temperature
Figure 7. Reference Voltage vs. Temperature
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6
160
NCV8870
TYPICAL PERFORMANCE CHARACTERISTICS
8.0
TJ = 25°C
6
Ienable, PULLDOWN CURRENT (mA)
Ienable, PULLDOWN CURRENT (mA)
7
5
4
3
2
1
0
0
1
2
3
4
Venable, VOLTAGE (V)
5
7.5
7.0
6.5
6.0
5.5
5.0
−40
6
10
60
110
160
TJ, JUNCTION TEMPERATURE (°C)
Figure 8. Enable Pulldown Current vs. Voltage
Figure 9. Enable Pulldown Current vs.
Temperature
THEORY OF OPERATION
VIN
L
VOUT
Oscillator
S
GDRV
Q
Gate
Drive
PWM Comparator
−
R
CO
RL
+
+
+
Voltage Error
ISNS
−
CSA
VFB
−
+
VEA
NCV8870
Compensation
Figure 10. Current Mode Control Schematic
Current Mode Control
found in voltage mode controllers. The second benefit
comes from inherent pulse-by-pulse current limiting by
merely clamping the peak switching current. Finally, since
current mode commands an output current rather than
voltage, the filter offers only a single pole to the feedback
loop. This allows for a simpler compensation.
The NCV8870 also includes a slope compensation
scheme in which a fixed ramp generated by the oscillator is
added to the current ramp. A proper slope rate is provided to
improve circuit stability without sacrificing the advantages
of current mode control.
The NCV8870 incorporates a current mode control
scheme, in which the PWM ramp signal is derived from the
power switch current. This ramp signal is compared to the
output of the error amplifier to control the on-time of the
power switch. The oscillator is used as a fixed-frequency
clock to ensure a constant operational frequency. The
resulting control scheme features several advantages over
conventional voltage mode control. First, derived directly
from the inductor, the ramp signal responds immediately to
line voltage changes. This eliminates the delay caused by the
output filter and the error amplifier, which is commonly
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7
NCV8870
Current Limit
and disabling the part. The falling edge of the square wave
corresponds to the start of the switching cycle. If device is
disabled, it must be disabled for 7 clock cycles before being
re-enabled.
The NCV8870 features two current limit protections,
peak current mode and over current latch off. When the
current sense amplifier detects a voltage above the peak
current limit between ISNS and GND after the current limit
leading edge blanking time, the peak current limit causes the
power switch to turn off for the remainder of the cycle. Set
the current limit with a resistor from ISNS to GND, with
R = VCL / Ilimit.
If the voltage across the current sense resistor exceeds the
over current threshold voltage the device enters over current
hiccup mode. The device will remain off for the hiccup time
and then go through the soft−start procedure.
UVLO
Input Undervoltage Lockout (UVLO) is provided to
ensure that unexpected behavior does not occur when VIN
is too low to support the internal rails and power the
controller. The IC will start up when enabled and VIN
surpasses the UVLO threshold plus the UVLO hysteresis
and will shut down when VIN drops below the UVLO
threshold or the part is disabled.
Short Circuit Protection
Internal Soft-Start
If the short circuit enable bit is set (SCE = Y) the device
will attempt to protect the power MOSFET from damage.
When the output voltage falls below the short circuit trip
voltage, after the initial short circuit blanking time, the
device enters short circuit latch off. The device will remain
off for the hiccup time and then go through the soft-start.
To insure moderate inrush current and reduce output
overshoot, the NCV8870 features a soft start which charges a
capacitor with a fixed current to ramp up the reference voltage.
This fixed current is based on the switching frequency, so
that if the NCV8870 is synchronized to twice the default
switching frequency the soft start will last half as long.
EN/SYNC
VDRV
The Enable/Synchronization pin has three modes. When
a dc logic high (CMOS/TTL compatible) voltage is applied
to this pin the NCV8870 operates at the programmed
frequency. When a dc logic low voltage is applied to this pin
the NCV8870 enters a low quiescent current sleep mode.
When a square wave of at least %fsync,min of the free running
switching frequency is applied to this pin, the switcher
operates at the same frequency as the square wave. If the
signal is slower than this, it will be interpreted as enabling
An internal regulator provides the drive voltage for the
gate driver. Bypass with a ceramic capacitor to ground to
ensure fast turn on times. The capacitor should be between
0.1 mF and 1 mF, depending on switching speed and charge
requirements of the external MOSFET.
GDRV
An RGND = 15 kW (typical) GDRV−GND resistor is
strongly recommended.
APPLICATION INFORMATION
Design Methodology
VOUT: output voltage [V]
IOUT(max): maximum output current [A]
ICL: desired typical cycle-by-cycle current limit [A]
This section details an overview of the component selection
process for the NCV8870 in continuous conduction mode
boost. It is intended to assist with the design process but does
not remove all engineering design work. Many of the
equations make heavy use of the small ripple approximation.
This process entails the following steps:
1. Define Operational Parameters
2. Select Current Sense Resistor
3. Select Output Inductor
4. Select Output Capacitors
5. Select Input Capacitors
6. Select Feedback Resistors
7. Select Compensator Components
8. Select MOSFET(s)
9. Select Diode
10. Determine Feedback Loop Compensation Network
From this the ideal minimum and maximum duty cycles
can be calculated as follows:
D min + 1 *
D max + 1 *
V IN(max)
V OUT
V IN(min)
V OUT
Both duty cycles will actually be higher due to power loss
in the conversion. The exact duty cycles will depend on
conduction and switching losses. If the maximum input
voltage is higher than the output voltage, the minimum duty
cycle will be negative. This is because a boost converter
cannot have an output lower than the input. In situations
where the input is higher than the output, the output will
follow the input, minus the diode drop of the output diode
and the converter will not attempt to switch.
If the calculated Dmax is higher the Dmax of the NCV8870,
the conversion will not be possible. It is important for a boost
1. Define Operational Parameters
Before beginning the design, define the operating
parameters of the application. These include:
VIN(min): minimum input voltage [V]
VIN(max): maximum input voltage [V]
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NCV8870
4. Select Output Capacitors
converter to have a restricted Dmax, because while the ideal
conversion ration of a boost converter goes up to infinity as
D approaches 1, a real converter’s conversion ratio starts to
decrease as losses overtake the increased power transfer. If
the converter is in this range it will not be able to regulate
properly.
If the following equation is not satisfied, the device will
skip pulses at high VIN:
The output capacitors smooth the output voltage and
reduce the overshoot and undershoot associated with line
transients. The steady state output ripple associated with the
output capacitors can be calculated as follows:
V OUT(ripple) +
DI OUT(max)
fC OUT
D min
w t on(min)
fs
I Cout(RMS) + I OUT
2. Select Current Sense Resistor
Current sensing for peak current mode control and current
limit relies on the MOSFET current signal, which is
measured with a ground referenced amplifier. The easiest
method of generating this signal is to use a current sense
resistor from the source of the MOSFET to device ground.
The sense resistor should be selected as follows:
V IN(min)D
2fL
Ǔ
R ESR
Ǹ
D WC
D
) WC
12
DȀ WC
ǒ
DȀ WC
L
R
OUT
T
SW
Ǔ
2
The input capacitor reduces voltage ripple on the input to
the module associated with the ac component of the input
current.
V CL
I CL
I Cin(RMS) +
V IN(min) 2 D WC
Lf sV OUT2 Ǹ3
6. Select Feedback Resistors
The feedback resistors form a resistor divider from the
output of the converter to ground, with a tap to the feedback
pin. During regulation, the divided voltage will equal Vref.
The lower feedback resistor can be chosen, and the upper
feedback resistor value is calculated as follows:
The output inductor controls the current ripple that occurs
over a switching period. A high current ripple will result in
excessive power loss and ripple current requirements. A low
current ripple will result in a poor control signal and a slow
current slew rate in case of load steps. A good starting point
for peak to peak ripple is around 20−40% of the inductor
current at the maximum load at the worst case VIN, but
operation should be verified empirically. The worst case VIN
is half of VOUT, or whatever VIN is closest to half of VOUT.
After choosing a peak current ripple value, calculate the
inductor value as follows:
R upper + R lower
ǒV out * V refǓ
V ref
The total feedback resistance (Rupper + Rlower) should be in
the range of 1 kW – 100 kW.
7. Select Compensator Components
Current Mode control method employed by the NCV8870
allows the use of a simple, Type II compensation to optimize
the dynamic response according to system requirements.
V IN(WC) D WC
DI L,max f s
Where: VIN(WC): VIN value as close as possible to
half of VOUT [V]
DWC: duty cycle at VIN(WC)
DIL,max: maximum peak to peak ripple [A]
The maximum average inductor current can be calculated
as follows:
8. Select MOSFET(s)
In order to ensure the gate drive voltage does not drop out
the MOSFET(s) chosen must not violate the following
inequality:
Q g(total) v
V OUTI OUT(max)
V IN(min)h
I drv
fs
Where: Qg(total): Total Gate Charge of MOSFET(s) [C]
Idrv: Drive voltage current [A]
fs: Switching Frequency [Hz]
The maximum RMS Current can be calculated as follows:
The Peak Inductor current can be calculated as follows:
I L,peak + I L,avg )
1*D
)
5. Select Input Capacitors
3. Select Output Inductor
I L,AVG +
I OUT(max)
The use of parallel ceramic bypass capacitors is strongly
encouraged to help with the transient response.
Where: RS: sense resistor [W]
VCL: current limit threshold voltage [V]
ICL: desire current limit [A]
L+
ǒ
The capacitors need to survive an RMS ripple current as
follows:
Where: fs: switching frequency [Hz]
ton(min): minimum on time [s]
RS +
)
DI L,max
2
I Q(max) + I out
Where: IL,peak: Peak inductor current value [A]
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9
ǸD
DȀ
NCV8870
boost inductor, MOSFET, current sensing and boost diode
losses also influence the gain and compensation
expressions. The OTA has an ESD protection structure
(RESD ≈ 502 W, data not provided in the datasheet) located
on the die between the OTA output and the IC package
compensation pin (VC). The information from the OTA
PWM feedback control signal (VCTRL) may differ from the
IC-VC signal if R2 is of similar order of magnitude as RESD .
The compensation and gain expressions which follow take
influence from the OTA output impedance elements into
account.
Type-I compensation is not possible due to the presence
of RESD . The Figures 11 and 12 compensation networks
correspond to a Type-II network in series with RESD .
The resulting control-output transfer function is an accurate
mathematical model of the IC in a boost converter topology.
The model does have limitations and a more accurate SPICE
model should be considered for a more detailed analysis:
• The attenuating effect of large value ceramic capacitors
in parallel with output electrolytic capacitor ESR is not
considered in the equations.
• The CCM Boost control-output transfer function
includes operating efficiency as a correction factor to
improve modeling accuracy under low input voltage
and high output current operating conditions where
operating losses becomes significant.
The maximum voltage across the MOSFET will be the
maximum output voltage, which is the higher of the
maximum input voltage and the regulated output voltaged:
V Q(max) + V OUT(max)
9. Select Diode
The output diode rectifies the output current. The average
current through diode will be equal to the output current:
I D(avg) + I OUT(max)
Additionally, the diode must block voltage equal to the
higher of the output voltage and the maximum input voltage:
V D(max) + V OUT(max)
The maximum power dissipation in the diode can be
calculated as follows:
P D + V f (max) I OUT(max)
Where: Pd: Power dissipation in the diode [W]
Vf(max): Maximum forward voltage of the diode [V]
10. Determine Feedback Loop Compensation Network
The purpose of a compensation network is to stabilize the
dynamic response of the converter. By optimizing the
compensation network, stable regulation response is
achieved for input line and load transients.
Compensator design involves the placement of poles and
zeros in the closed loop transfer function. Losses from the
L
rL
VIN
Vd
VOUT
rCf
COUT
Rds(on)
VC
GDRV
R2
RESD
C2
C1
ISNS
VCTRL
OTA
Ri
R0
VFB
GND
R1
Rlow
Figure 11. NCV8870 Boost Converter OTA and Compensation
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10
ROUT
NCV8870
Vd
VIN
VOUT
1:N
rCf
Lp
COUT
Rds(on)
ROUT
VC
GDRV
R2
RESD
C2
ISNS
C1
VCTRL
OTA
R0
Ri
VREF
VFB
GND
R1
Rlow
Figure 12. NCV8870 Flyback Converter OTA and Compensation
Necessary equations for describing the modulator gain
(Vctrl-to-Vout gain) Hctrl_output (f) are described next. Boost
continuous conduction mode (CCM) and discontinuous
conduction mode (DCM) transfer function expressions are
summarized in Table 1. Flyback CCM and DCM transfer
function expressions are summarized in Table 2.
The following equations may be used to select compensation
components R2 , C1 , C2 for Figures 11 & 12 power supply.
Required input design parameters for analysis are:
Vd = Output diode Vf (V)
VIN = Power supply input voltage (V)
N = Ns /Np (Flyback transformer turns ratio)
Ri = Current sense resistor (W)
RDS(on) = MOSFET RDS(on) (W)
(Rsw_eq = RDS(on) + Ri for the boost continuous conduction
mode (CCM) expressions)
COUT = Bulk output capacitor value (F)
rCF = Bulk output capacitor ESR (W)
ROUT = Equivalent resistance of output load (W)
Pout = Output Power (W)
L = Boost inductor value or flyback transformer primary
side inductance (H)
rL = Boost inductor ESR (W)
Ts = 1/fs , where fs = clock frequency (Hz)
R1 and Rlow = Feedback resistor divider values used to set the
output voltage (W)
VOUT = Device specific output voltage (defined by R1 and
Rlow values) (V)
R0 = OTA output resistance = 3 MW
Sa = IC slope compensation (e.g. 33 mV/ms for NCV887001)
gm = OTA transconductance = 1.2 mS
D = Controller duty ratio
D’ = 1 − D
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11
NCV8870
Table 1. BOOST CCM AND DCM TRANSFER FUNCTION EXPRESSIONS
CCM
Duty Ratio (D)
ȡ
ȧ
ȧ
ȧ
Ȣ
-V OUT
DCM
ƪ
2R OUTV dV IN* R sw_eq)R OUT
Ǹ ǒ
ǒ
V IN
V OUT
*2
V OUT 2
Ǔ
ǒ
VOUT/VIN DC
Conversion Ratio
(M)
Ǹ2tLM(M * 1)
L
Where: t L +
R OUTT s
Ǔ
1
1
1*D
2
ǒ Ǹ
1)
1)
2D 2
tL
Ǔ
V IN * I Laveǒr L ) R sw_eqǓ
Inductor On-slope
(Sn ), V/s
Ri
L
Where average inductor current:
Compensation
Ramp (mc )
(1 * D)
L
Low Frequency
Modulator Pole
(wp1 )
2
V IN
I Lave +
P out
L
Sa
1)
Sn
Sn
1
1
r CFC OUT
r CFC OUT
ǒ
r CFR OUT
R OUT *
r CF ) R OUT
Ǔ
*
rL
R OUT
L
M 2L
Ts
2
)
mc
R OUT
LM 3
1
R CFC OUT
C OUT
High Frequency
Modulator Pole
(wp2 )
2F SW
p
Sampling Double
Pole (wn )
2M * 1
@
M*1
ǒ Ǔ
−
1
1*M
2
D
−
Ts
Sampling Quality
Coefficient (Qp )
Ri
V INh
Sa
1)
Cout ESR Zero
(wz1 )
1
−
p(m c(1 * D) * 0.5)
Fm
1
ǒ
R OUTT s 1
Sa
2M )
)
2
Sn
LM 2
Hd
Control-Output
Transfer Function
(Hctrl_output (f))
ȣ
ȧ
ȧ
ȧ
Ȥ
R OUTV IN 2)2R sw_eqV INV OUT*4V dR sw_eqV IN
R OUT
)R sw_eq 2V OUT 2
-4R sw_eqV OUT 2*4r LV dV IN*4r LV OUT 2
2R OUT V OUT 2 ) V dV IN
Right-Half-Plane
Zero (wz2 )
Ǔƫ
1
Ǔ
S nm cT s
hR OUT
2V OUT
Ri
D
F mH d
ǒ
2pf
1 )jw
p1
Ǔǒ
z2
1)j
2pf 2
2pf
) ǒj w Ǔ
n
w nQ p
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12
M*1
2M * 1
ǒ1 ) j w2pfǓǒ1 * j w2pfǓ
ǒ1 ) j w2pfǓǒ1 * j w2pfǓ
z1
@
Ǔ
F mH d
z1
ǒ
2pf
1 )jw
p1
Ǔǒ
z2
2pf
1 ) jw
p2
Ǔ
NCV8870
Table 2. FLYBACK CCM AND DCM TRANSFER FUNCTION EXPRESSIONS
CCM
DCM
Duty ratio (D)
V OUT
V OUT
V OUT ) NV IN
VOUT/VIN DC Conversion Ratio (M)
Compensation Ramp (mc )
Ǹ2 @ tL
V IN
V IN
Cout ESR Zero (wz1 )
Right-Half-Plane Zero (wz2 )
Ri
Lp
Sa
1)
1
r CFC OUT
(1 * D) R OUT
R OUT
DL pN 2
N 2L
@
p
1
M(M ) 1)
Ǔ
2
R OUTC OUT
R OUTC OUT
ǒ Ǔ
wp2
−
1
ǒ
2F SW
V IN
R iN
ǒ1 ) j w2pfǓǒ1 * j w2pfǓ
1
1)M
2pf
1 ) jw
p1
F mH d
Ǔ
Ǹ
1
2t L
ǒ1 ) j w2pfǓǒ1 * j w2pfǓ
z2
z1
ǒ
2
S nm cT s
R OUT
F mH d
1
D
1
Ǔ
Sa
t L 1 ) 2 S n ) 2M ) 1
Control-output Transfer Function (Hctrl_output (f))
Sn
1
ǒ
Hd
Ri
r CFC OUT
Sa
DȀ 3
t L 1 ) 2 S n ) 1 ) D
DȀ 2
T sR OUT
Sa
1)
Sn
2
Fm
N 2L p
N@D
1*D
Lp
Modulator Pole (wp1 )
Where: t L +
N@D
Inductor On-slope (Sn ), V/s
Ǹ2t L
NV IN
z1
ǒ
2pf
1 )jw
p1
Ǔǒ
z2
2pf
1 ) jw
p2
Ǔ
component value adjustments may become necessary when
R2 ≤ ~10·Resd as a result of approximations for determining
components R2 , C1 , C2 .
Once the desired cross-over frequency (fc ) gain
adjustment and necessary phase boost are determined from
the Hctrl_output (f) gain and phase plots, the Table 3 equations
may be used. It should be noted that minor compensation
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13
NCV8870
Table 3. OTA COMPENSATION TRANSFER FUNCTION AND COMPENSATION VALUES
Desired OTA Gain at Cross-over Frequency fc (G)
desired_G fc_gain_db
20
10
ǒq
Desired Phase Boost at Cross-over Frequency fc (boost)
Ǔ 180°
* 90° Ǔ
p
ǒ
( )
margin * arg H ctrl_output fc
p
180°
w p1e
Select OTA Compensation Zero to Coincide
with Modulator Pole at fp1 (fz )
2p
f zf c ) f c 2 tan(boost)
Resulting OTA High Frequency Pole Placement (fp )
f c * f z tan(boost)
Compensation Resistor R2
V OUT
f p * f z 1.2g m
f pG
Ǹ
1)
Ǹ
1)
Compensation Capacitor C1
ǒǓ
2
fc
fp
ǒǓ
fz
fp
1
2pf zR 2
Compensation Capacitor C2
1
2pf pG
OTA DC Gain (G0_OTA )
@
R lowg m
R low ) R 1
R low
R low ) R 1
ȳ
R R C
1*
1*4
ȧ
ȧ
2 R R C
ǒR ) R Ǔ C ȴ
Ȳ
ǒ
Ǔ
ȳ
R R C
1 R )R ȱ
1)
1*4
ȧ
ȧ
2 R R C
ǒR ) R Ǔ C ȴ
Ȳ
ǒ
Ǔȱ
ȳ
R ǒR ) R ǓC
1 R )R )R
1
*
1
*
4
ȧ
ȧ
2 R ǒR ) R ǓC
ǒR ) R ) R Ǔ C ȴ
Ȳ
ǒ
Ǔȱ
ȳ
R ǒR ) R ǓC
1 R )R )R
1)
1*4
ȧ
ȧ
2 R ǒR ) R ǓC
ǒR ) R ) R Ǔ C ȴ
Ȳ
ǒ1 ) j w Ǔ ǒ1 ) j w2pf Ǔ
Low Frequency Zero (wz1e )
1
ǒR2 ) ResdǓȱ
@ gm @ R0
2 esd 2
High Frequency Zero (wz2e )
2
0
2
High Frequency Pole (wp2e )
2
0
0
2
OTA Transfer Function (GOTA (f))
2
0
2
esd
esd
2 esd 2
2
2
1
esd
Ǹ
Ǹ
esd
esd
2 esd 2
2
2
1
esd
Ǹ
esd
2 esd 2
Low Frequency Pole (wp1e )
Ǹ
2
2
0
0
2
2
0
esd
2
2
1
esd
0
esd
2
esd
2
2
1
2pf
z2e
z1e
-G 0_OTA
The open-loop-response in closed-loop form to verify the
gain/phase margins may be obtained from the following
expression.
ǒ
2pf
1 ) jw
p1e
Ǔǒ
2pf
1 )jw
p2e
Ǔ
device. Simply connect the voltage you would like to boost
to the inductor and connect the stable voltage to the VIN pin
of the device. In boost configuration, the output of the
converter can be used to power the device. In some cases it
may be desirable to connect 2 sources to VIN pin, which can
be accomplished simply by connecting each of the sources
through a diode to the VIN pin.
T(f) + G OTA(f) H ctrl_output(f)
Low Voltage Operation
If the input voltage drops below the UVLO or MOSFET
threshold voltage, another voltage may be used to power the
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14
NCV8870
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AK
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
K
−Y−
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
M
D
0.25 (0.010)
M
Z Y
S
X
J
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and the
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at www.onsemi.com/site/pdf/Patent−Marking.pdf. SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation
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NCV8870/D