NCP1607 Cost Effective Power Factor Controller The NCP1607 is an active power factor controller specifically designed for use as a pre−converter in ac−dc adapters, electronic ballasts, and other medium power off line converters (typically up to 250 W). It utilizes Critical Conduction Mode (CRM) to ensure unity power factor across a wide range of input voltages and power levels. The NCP1607 minimizes the number of external components. The integration of comprehensive safety protection features makes it an excellent choice for designing robust PFC stages. It is available in a SOIC−8 package. www.onsemi.com • • • • • • • “Unity” Power Factor No Need for Input Voltage Sensing Latching PWM for Cycle by Cycle On Time Control (Voltage Mode) High Precision Voltage Reference (±1.6% over the Temperature Range) Very Low Startup Current Consumption (≤ 40 mA) Low Typical Operating Current (2.1 mA) Source 500 mA / Sink 800 mA Totem Pole Gate Driver Undervoltage Lockout with Hysteresis Pin to Pin Compatible with Industry Standards This is a Pb−Free Device This Device uses Halogen−Free Molding Compound 1607B ALYW G 1 A L Y W G = Assembly Location = Wafer Lot = Year = Work Week = Pb−Free Package PIN CONNECTION FB Control Ct CS VCC DRV GND ZCD (Top View) ORDERING INFORMATION Safety Features • • • • 8 1 SO−8 D SUFFIX CASE 751 General Features • • • • MARKING DIAGRAMS 8 Programmable Overvoltage Protection Open Feedback Loop Protection Accurate and Programmable On Time Control Accurate Overcurrent Detector Device Package Shipping† NCP1607BDR2G SOIC−8 (Pb−Free) 2500 / Tape & Reel †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. Typical Applications • AC−DC Adapters, TVs, Monitors • Off Line Appliances Requiring Power Factor Correction • Electronic Light Ballast LBOOST VOUT DBOOST LOAD (Ballast, SMPS, etc.) RZCD AC Line EMI Filter + CIN ROUT1 NCP1607 1 CCOMP FB 2 Control DRV 3 ROUT2 VCC 4 CT Ct GND CS ZCD VCC 8 + CBULK 7 6 5 RS Figure 1. Typical Application © Semiconductor Components Industries, LLC, 2015 July, 2015 − Rev. 2 1 Publication Order Number: NCP1607/D NCP1607 VCC Shutdown POK VOUT + ROUT1 VUVP FB ROUT2 (Enable EA) E/A − + ESD + DBOOST + CBULK VCC UVP − + VREF Dynamic OVP VDD Static OVP ESD Static OVP is triggered when clamp is activated. VEAH Clamp LBOOST POK VDD CT PWM ICHARGE Add VEAL Offset − + ESD S Q DRV + − LEB CS uVDD Fault VEAL Clamp VCONTROL CT VDDGD VDD Reg IEAsink ESD + R Q OCP VCC VCS(limit) RS ZCD RZCD + VCL(POS) Clamp VZCDL − + R Q R Q + − DRV S Q S Q VZCDH + VCL(NEG) Active Clamp UVLO Demag + − VDD + AC IN VDD Isink>Iovp Enable Control UVLO Measure RFB CCOMP + − VDDGD Off Timer uVDD S Q Reset Shutdown GND R Q VSDL uVDD S Q R Q *All SR Latches are Reset Dominant Figure 2. Block Diagram www.onsemi.com 2 POK NCP1607 PIN FUNCTION DESCRIPTION Pin Name Function 1 FB The FB pin is the inverting input of the internal error amplifier. An external resistor divider scales the output voltage to the internal reference voltage to maintain regulation. The feedback information is also used for the programmable overvoltage and undervoltage protections. The controller is disabled when this pin is below the undervoltage protection threshold, VUVP, typically 0.3 V. 2 Control The Control pin is the output of the internal error amplifier. A compensation network is placed between the Control and FB pins to set the loop bandwidth. A low enough bandwidth is needed to obtain a high power factor ratio and a low THD. 3 Ct The Ct pin sources a current to charge an external timing capacitor. The circuit controls the power switch on time by comparing the Ct voltage to an internal voltage derived from the regulation block. The Ct pin discharges the external timing capacitor at the end of the switching cycle. 4 CS The CS pin limits the cycle−by−cycle current through the power switch. When the CS voltage exceeds the internal threshold, the MOSFET driver turns off. The sense resistor that connects to the CS pin programs the maximum switch current. 5 ZCD The voltage of an auxiliary winding is applied to this pin to detect when the inductor is demagnetized for critical conduction mode operation. The controller is disabled when this pin is grounded. 6 GND Analog ground. 7 DRV Integrated MOSFET driver capable of driving a high gate charge power MOSFET. 8 VCC The VCC pin is the positive supply of the controller. The controller is enabled when VCC exceeds VCC(on) and remains enabled until VCC decreases below VCC(off). MAXIMUM RATINGS Symbol Value Unit Supply Voltage Rating VCC −0.3 to 20 V Supply Current ICC ±20 mA VDRV −0.3 to 20 V IDRV(sink) 800 mA IDRV(source) 500 mA VFB −0.3 to 10 V DRV Voltage DRV Sink Current DRV Source Current FB Voltage IFB ±10 mA Control Voltage VCONTROL −0.3 to 10 V Control Current ICONTROL −2 to 10 mA Ct Voltage VCt −0.3 to 6 V Ct Current ICt ±10 mA CS Voltage VCS −0.3 to 6 V CS Current ICS ±10 mA ZCD Voltage VZCD −0.3 to 10 V ZCD Current IZCD ±10 mA PD(SO) RqJA(SO) 450 178 mW °C/W FB Current Power Dissipation and Thermal Characteristics D suffix, Plastic Package, Case 751 Maximum Power Dissipation @ TA = 70°C Thermal Resistance Junction−to−Air Operating Junction Temperature Range Maximum Junction Temperature Storage Temperature Range Lead Temperature (Soldering, 10 s) TJ −40 to 125 °C TJ(MAX) 150 °C TSTG −65 to 150 °C TL 300 °C Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality should not be assumed, damage may occur and reliability may be affected. 1. This device series contains ESD protection and exceeds the following tests: Pins 1 − 8: Human Body Model 2000 V per JEDEC Standard JESD22−A114E, Charged Device Model 1000 V per JEDEC Standard JESD22−C101E. 2. This device contains latch−up protection and exceeds ±100 mA per JEDEC Standard JESD78. www.onsemi.com 3 NCP1607 ELECTRICAL CHARACTERISTICS (For typical values, TJ = 25°C. For min/max values, TJ = −40°C to +125°C, unless otherwise specified, VCC = 12 V, VFB = 2.4 V, VCS = 0 V, VCONTROL = open, VZCD = open, CDRV = 1 nF, CT = 1 nF) Characteristics Symbol Min Typ Max 11.0 10.9 11.8 11.8 13.0 13.1 8.7 8.5 9.5 9.5 10.3 10.5 Unit VCC UNDERVOLTAGE LOCKOUT SECTION VCC Startup Threshold (Undervoltage Lockout Threshold, Vcc rising) −25°C < TJ < +125°C −40°C < TJ < +125°C VCC(on) V VCC Disable Voltage after Turn On (Undervoltage Lockout Threshold, VCC falling) −25°C < TJ < +125°C −40°C < TJ < +125°C VCC(off) Undervoltage Lockout Hysteresis HUVLO 2.2 2.5 2.8 V ICC(startup) − 23.5 40 mA ICC consumption after turn on at No Load, 70 kHz switching ICC1 − 1.4 2.0 mA ICC consumption after turn on at 70 kHz switching ICC2 − 2.17 3.0 mA ICC(fault) − 1.2 1.6 mA VREF 2.475 2.465 2.460 2.50 2.50 2.50 2.525 2.535 2.540 V VREF(line) −2.0 − 2.0 mV 8.0 −2.0 17 −6.0 − − V DEVICE CONSUMPTION ICC consumption during startup: 0 V < VCC < VCC(on) − 200 mV ICC consumption after turn on at no switching (such as during OVP fault, UVP fault, or grounding ZCD) REGULATION BLOCK (ERROR AMPLIFIER) Voltage Reference TJ = 25 °C −25°C < TJ < +125°C −40°C < TJ < +125°C VREF Line Regulation from VCC(on) + 200 mV < VCC < 20 V, TJ = 25°C Error Amplifier Current Capability: (Note 3) Sink (VControl = 4 V, VFB = 2.6 V): Source (VControl = 4 V, VFB = 2.4 V): IEA Error Amplifier Open Loop DC Gain (Note 4) GOL − 80 − dB Unity Gain Bandwidth (Note 4) BW − 1.0 − MHz FB Bias Current (VFB = 2.5 V) IFB 0.25 0.53 1.25 mA FB Pull Down Resistor (VFB = 2.5 V) RFB 2.0 4.7 10 MW ICONTROL −1.0 − 1.0 mA Control Pin Bias Current (FB = 0 V and VCONTROL = 4.0 V) mA VCONTROL (IEASOURCE = 0.5 mA, VFB = 2.4 V) VEAH 4.9 5.3 5.7 V VCONTROL (IEASINK = 0.5 mA, VFB = 2.6 V) VEAL 1.85 2.1 2.4 V VEA(diff) 3.0 3.2 3.4 V VEA(diff) = VEAH − VEAL CURRENT SENSE BLOCK VCS(limit) 0.45 0.5 0.55 V Leading Edge Blanking Duration Overcurrent Voltage Threshold tLEB 150 256 350 ns Overcurrent Voltage Propagation Delay tCS 40 100 170 ns CS Bias Current (VCS = 2 V) ICS −1.0 − 1.0 mA Zero Current Detection Threshold (VZCD rising) VZCDH 1.9 2.1 2.3 V Zero Current Detection Threshold (VZCD falling) VZCDL 1.45 1.6 1.75 V VZCD(HYS) 300 500 800 mV IZCD −2.0 − +2.0 mA Upper Clamp Voltage (IZCD = 2.5 mA) VCL(POS) 5.0 5.7 6.5 V Current Capability of the Positive Clamp at VZCD = VCL(POS) + 200 mV: ICL(POS) 5.0 8.5 − mA Negative Active Clamp Voltage (IZCD = −2.5 mA) VCL(NEG) 0.45 0.6 0.75 V ZERO CURRENT DETECTION VZCDH − VZCDL Maximum ZCD bias Current (VZCD = 5 V) 3. Parameter values are valid for transient conditions only. 4. Parameter characterized and guaranteed by design, but not tested in production. www.onsemi.com 4 NCP1607 ELECTRICAL CHARACTERISTICS (For typical values, TJ = 25°C. For min/max values, TJ = −40°C to +125°C, unless otherwise specified, VCC = 12 V, VFB = 2.4 V, VCS = 0 V, VCONTROL = open, VZCD = open, CDRV = 1 nF, CT = 1 nF) Characteristics Symbol Current Capability of the Negative Active Clamp: in normal mode (VZCD = 300 mV) in shutdown mode (VZCD = 100 mV) Min Typ Max Unit 2.5 35 3.7 70 5.0 100 mA mA VSDL 150 205 250 mV ICL(NEG) Shutdown Threshold (VZCD falling) Enable Threshold (VZCD rising) VSDH − 290 350 mV VSD(HYS) − 85 − mV Zero Current Detection Propagation Delay tZCD − 100 170 ns Minimum Detectable ZCD Pulse Width tSYNC − 70 − ns Drive off Restart Timer tSTART 75 179 300 ms ICHARGE 243 235 270 270 297 297 mA tCT(discharge) − − 100 ns VCTMAX 2.9 2.9 3.2 3.2 3.3 3.4 V tPWM − 142 220 ns 9.0 8.7 10.5 − 11.8 12.1 IOVP(HYS) − 8.5 − mA Static OVP Threshold Voltage VOVP − VEAL + 100 mV − V Undervoltage Protection (UVP) Threshold Voltage VUVP 0.25 0.302 0.4 V Gate Drive Resistance: ROH @ ISOURCE = 100 mA ROL @ ISINK = 100 mA ROH ROL − − 12 6.0 18 10 Drive voltage rise time from 10% VCC to 90% VCC trise − 30 80 ns tfall − 25 70 ns VOUT(start) − − 0.2 V Shutdown Comparator Hysteresis RAMP CONTROL Ct Charge Current (VCT = 0 V) −25°C < TJ < +125°C −40°C < TJ < +125°C Time to discharge a 1 nF Ct capacitor from VCT = 3.4 V to 100 mV. Maximum Ct level before DRV switches off −25°C < TJ < +125°C −40°C < TJ < +125°C PWM Propagation Delay OVER AND UNDERVOLTAGE PROTECTION Dynamic Overvoltage Protection (OVP) Triggering Current: TJ = 25°C TJ = −40°C to +125°C mA IOVP Hysteresis of the dynamic OVP current before the OVP latch is released GATE DRIVE SECTION W Drive voltage fall time from 90% VCC to 10% VCC Driver output voltage at VCC = VCC(on) − 200 mV and Isink = 10 mA 3. Parameter values are valid for transient conditions only. 4. Parameter characterized and guaranteed by design, but not tested in production. Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product performance may not be indicated by the Electrical Characteristics if operated under different conditions. www.onsemi.com 5 NCP1607 274 14 272 12 270 10 Ct = 1 nF ton, ON TIME (ms) 268 266 264 4 2 260 −50 0 −25 0 25 50 75 100 125 150 0 1 2 3 4 5 6 TEMPERATURE (°C) VCONTROL (V) Figure 3. Ct Charge Current vs. Temperature Figure 4. On Time vs. VCONTROL Level tPWM, PWM PROPAGATION DELAY (ns) VCTMAX, MAXIMUM Ct LEVEL (V) 6 262 3.30 3.25 3.20 3.15 3.10 3.05 3.00 −50 −25 0 25 50 75 100 125 150 170 160 150 140 130 −50 −25 0 25 50 75 100 125 TEMPERATURE (°C) TEMPERATURE (°C) Figure 5. Maximum Ct Level vs. Temperature Figure 6. PWM Propagation Delay vs. Temperature 2.505 2.500 2.495 2.490 2.485 2.480 2.475 2.470 −50 −25 0 25 50 75 100 125 150 80 160 GAIN 60 120 PHASE 40 80 20 40 0 0 −20 10 100 1k 10k 100k 1M −40 10M TEMPERATURE (°C) FREQUENCY (Hz) Figure 7. Reference Voltage vs. Temperature Figure 8. Error Amplifier Open Loop Gain and Phase www.onsemi.com 6 150 200 100 GOL, OPEN LOOP GAIN (dB) VREF, REFERENCE VOLTAGE (V) 8 PHASE (°) ICHARGE, Ct CHARGE CURRENT (mA) TYPICAL CHARACTERISTICS NCP1607 TYPICAL CHARACTERISTICS 7 11 IOVP 10 9 IOVP(HYS) 8 7 −50 −25 0 25 50 75 100 6 5 4 3 2 1 0 −50 125 150 −25 0 50 75 100 125 150 TEMPERATURE (°C) Figure 9. Dynamic OVP Triggering Current vs. Temperature Figure 10. Feedback Resistor vs. Temperature 2.30 2.25 2.20 2.15 2.10 2.05 2.00 −50 −25 0 25 50 75 100 125 26 24 22 20 18 16 14 −50 150 −25 0 TEMPERATURE (°C) 50 75 100 125 150 Figure 12. Startup Current vs. Temperature 13 tSTART, RESTART TIMER (ms) 200 VCC(on) 12 11 10 VCC(off) 9 8 −50 25 TEMPERATURE (°C) Figure 11. Switching Supply Current vs. Temperature VCC, SUPPLY VOLTAGE THRESHOLD (V) 25 TEMPERATURE (°C) ICC(startup), STARTUP CURRENT (mA) ICC2, SWITCHING SUPPLY CURRENT (mA) RFB, FEEDBACK RESISTOR (MW) IOVP, DYNAMIC OVP TRIGGERING CURRENT (mA) 12 −25 0 25 50 75 100 125 150 190 180 170 160 −50 TEMPERATURE (°C) −25 0 25 50 75 100 125 TEMPERATURE (°C) Figure 13. Supply Voltage Thresholds vs. Temperature Figure 14. Restart Timer vs. Temperature www.onsemi.com 7 150 NCP1607 18 280 16 tLEB, LEB DURATION (ns) ISOURCE = 100 mA 14 ROH 12 10 ISINK = 100 mA 8 ROL 6 4 270 260 250 2 0 −50 −25 0 25 50 75 100 125 240 −50 150 0 25 50 75 100 125 TEMPERATURE (°C) TEMPERATURE (°C) Figure 16. LEB Duration vs. Temperature VUVP, UVP THRESHOLD VOLTAGE (V) 0.520 0.515 0.510 0.505 0.500 0.495 0.490 0.485 0.480 −50 −25 Figure 15. Gate Drive Resistance vs. Temperature −25 0 25 50 75 100 125 150 0.315 0.310 0.305 0.300 0.295 0.290 0.285 0.280 −50 −25 0 25 TEMPERATURE (°C) 75 100 125 Figure 18. Undervoltage Protection Threshold Voltage vs. Temperature 0.35 VSDH 0.25 VSDL 0.20 0.15 −50 50 TEMPERATURE (°C) 0.30 −25 0 25 50 75 100 125 TEMPERATURE (°C) Figure 19. Shutdown Thresholds vs. Temperature www.onsemi.com 8 150 0.320 Figure 17. Overcurrent Threshold Voltage vs. Temperature VSDH/SDL, SHUTDOWN THRESHOLD (V) VCS(limit), OVERCURRENT THRESHOLD VOLTAGE (V) ROH/OL, GATE DRIVE RESISTANCE (W) TYPICAL CHARACTERISTICS 150 150 NCP1607 Introduction The NCP1607 is a voltage mode power factor correction (PFC) controller designed to drive cost effective pre−converters to meet input line harmonic regulations. This controller operates in critical conduction mode (CRM) for optimal performance in applications up to 250 W. Its voltage mode scheme enables it to obtain unity power factor without the need for a line sensing network. The output voltage is accurately controlled by a high precision error amplifier. The controller also implements a comprehensive array of safety features for robust designs. The key features of the NCP1607 are as follows: • Constant on time (Voltage Mode) CRM operation. High power factor ratios are easily obtained without the need for input voltage sensing. This allows for optimal standby power consumption. • Accurate and Programmable On Time Limitation. The NCP1607 uses an accurate current source and an external capacitor to generate the on time. • High Precision Voltage Reference. The error amplifier reference voltage is guaranteed at 2.5 V ±1.6% over process, temperature, and voltage supply levels. This results in very accurate output voltages. • Very Low Startup Current Consumption. The circuit consumption is reduced to a minimum (< 40 mA) during the startup phase, allowing fast, low loss, charging of VCC. The architecture of the NCP1607 gives a controlled undervoltage lockout level and provides ample VCC hysteresis during startup. • Powerful Output Driver. A Source 500 mA / Sink 800 mA totem pole gate driver is used to provide rapid turn on and turn off times. This allows for improved efficiencies and the ability to drive higher power MOSFETs. Additionally, a combination of active and passive circuitry is used to ensure that the driver output voltage does not float high while VCC is below its turn on level. • Programmable Overvoltage Protection (OVP). The adjustable OVP feature protects the PFC stage against excessive output overshoots that could damage the application. These events can typically occur during the startup phase or when the load is abruptly removed. • Protection against Open Feedback Loop (Undervoltage Protection). Undervoltage protection (UVP) disables the PFC stage when the output voltage is excessively low. This also protects the circuit in case of a failure in the feedback network: if no voltage is applied to FB because of a poor connection or if the FB pin is floating, UVP is activated shutting down the converter. • Overcurrent Limitation. The peak current is accurately limited on a pulse by pulse basis. The level is adjustable by modifying the current sense resistor. An • integrated LEB filter reduces the chance of noise prematurely triggering the overcurrent limit. Shutdown Features. The PFC pre−converter is placed in a shutdown mode by grounding the FB pin or the ZCD pin. During this mode, the ICC current consumption is reduced and the error amplifier is disabled. Application information Most electronic ballasts and switching power supplies use a diode bridge rectifier and a bulk storage capacitor to produce a dc voltage from the utility ac line (Figure 20). This DC voltage is then processed by additional circuitry to drive the desired output. Rectifiers AC Line Converter + Bulk Storage Capacitor Load Figure 20. Typical Circuit without PFC This simple rectifying circuit draws power from the line when the instantaneous ac voltage exceeds the capacitor voltage. Since this occurs near the line voltage peak, the resulting current draw is non sinusoidal and contains a very high harmonic content. This results in a poor power factor (typically < 0.6) and consequently, the apparent input power is much higher than the real power delivered to the load. Additionally, if multiple devices are tied to the same input line, the effect is magnified and a “line sag” effect can be produced (see Figure 21). Vpk Rectified DC 0 Line Sag AC Line Voltage 0 AC Line Current Figure 21. Typical Line Waveforms without PFC Increasingly, government regulations and utility requirements necessitate control over the line current harmonic content. To meet this need, power factor correction is implemented with either a passive or active circuit. Passive circuits usually contain a combination of large capacitors, inductors, and rectifiers that operate at the ac line frequency. Active circuits incorporate some form of a high frequency switching converter that regulates the www.onsemi.com 9 NCP1607 input current to stay in phase with the input voltage. These circuits operate at a higher frequency and so they are smaller, lighter in weight, and more efficient than a passive circuit. With proper control of an active PFC stage, almost any complex load can be made to appear in phase with the ac line, thus significantly reducing the harmonic current PFC Preconverter Rectifiers AC Line content. Because of these advantages, active PFC circuits have become the most popular way to meet harmonic content requirements. Generally, they consist of inserting a PFC pre−regulator between the rectifier bridge and the bulk capacitor (Figure 22). + High Frequency Bypass Capacitor Converter Bulk Storage Capacitor + NCP1607 Load Figure 22. Active PFC Pre−Converter with the NCP1607 The boost (or step up) converter is the most popular topology for active power factor correction. With the proper control, it produces a constant voltage while drawing a sinusoidal current from the line. For medium power (<300 W) applications, critical conduction mode (also called borderline conduction mode) is the preferred control method. Critical conduction mode (CRM) occurs at the boundary between discontinuous conduction mode Diode Bridge (DCM) and continuous conduction mode (CCM). In CRM, the next driver on time is initiated when the boost inductor current reaches zero. CRM operation is an ideal choice for medium power PFC boost stages because it combines the lower peak currents of CCM operation with the zero current switching of DCM operation. The operation and waveforms in a PFC boost converter are illustrated in Figure 23. Diode Bridge IL + VIN IL + L VIN Vdrain L + + + IN VOUT IN Vdrain − − The power switch is ON The power switch is OFF With the power switch voltage being about zero, the input voltage is applied across the coil. The coil current linearly increases with a (VIN/L) slope. The coil current flows through the diode. The coil voltage is (VOUT − VIN) and the coil current linearly decays with a (VOUT − VIN)/L slope. Coil Current (VOUT − VIN)/L VIN/L IL(pk) Critical Conduction Mode: Next current cycle starts as soon as the core is reset. Vdrain VOUT VIN If next cycle does not start then Vdrain rings towards VIN Figure 23. Schematic and Waveforms of an Ideal CRM Boost Converter www.onsemi.com 10 NCP1607 When the switch is closed, the inductor current increases linearly to its peak value. When the switch opens, the inductor current linearly decreases to zero. At this point, the drain voltage of the switch (Vd) is essentially floating and begins to drop. If the next switching cycle does not start, then the voltage will ring with a dampened frequency around Vin. A simple derivation of equations (such as found in AND8123), leads to the result that good power factor correction in CRM operation is achieved when the on time is constant across an ac cycle and is equal to: ton + 2 @ P OUT @ L IL(pk) IL(t) IIN(pk) MOSFET (eq. 1) h @ Vac 2 VIN(t) VIN(pk) IIN(t) ON OFF Figure 24. Inductor Waveform During CRM Operation A simple plot of this switching over an ac line cycle is illustrated in Figure 24. The off time varies based on the instantaneous line voltage, but the on time is kept constant. This naturally causes the peak inductor current (IL(pk)) to follow the ac line voltage. The NCP1607 represents an ideal method to implement this constant on time CRM control in a cost effective and robust solution. The device incorporates an accurate regulation circuit, a low power startup circuit, and advanced protection features. ERROR AMPLIFIER REGULATION The NCP1607 is configured to regulate the boost output voltage based on its built in error amplifier (EA). The error amplifier ’s negative terminal is pinned out to FB, the positive terminal is tied to a 2.5 V ± 1.6% reference, and the output is pinned out to Control (Figure 25). VOUT ROUT1 PWM BLOCK EA FB − + + RFB ton(MAX) VREF ROUT2 Slope + CCOMP VCONTROL Control Ct I CHARGE ton tPWM VEAL VEAH VCONTROL Figure 25. Error Amplifier and On Time Regulation Circuits A resistor divider from the boost output to the input of the EA sets the FB level. If the output voltage is too low, then the FB level will drop and the EA will cause the control voltage to increase. This increases the on time of the driver, which increases the power delivered and brings the output back into regulation. Alternatively, if the output voltage (and hence FB voltage) is too high, then the control level decreases and the driver on times are shortened. In this way, the circuit regulates the output voltage (VOUT) so that the VOUT portion that is applied to FB through the resistor divider ROUT1 and ROUT2 is equal to the internal reference (2.5 V). The output voltage is set using Equation 2: VOUT + V REF @ ǒ R OUT1 ) R EQ R EQ Ǔ (eq. 2) Where REQ is the parallel combination of ROUT2 and RFB. REQ is calculated using Equation 3: REQ + www.onsemi.com 11 R OUT2 @ R FB R OUT2 ) R FB (eq. 3) NCP1607 A compensation network is placed between the FB and Control pins to reduce the speed at which the EA responds to changes in the boost output. This is necessary due to the nature of an active PFC circuit. The PFC stage absorbs a sinusoidal current from a sinusoidal line voltage. Hence, the converter provides the load with a power that matches the average demand only. Therefore, the output capacitor must “absorb” the difference between the delivered power and the power consumed by the load. This means that when the power fed to the load is lower than the demand, the output capacitor discharges to compensate for the lack of power. Alternatively, when the supplied power is higher than that absorbed by the load, the output capacitor charges to store the excess energy. The situation is depicted in Figure 26. Iac Vac PIN POUT VOUT Figure 26. Output Voltage Ripple for a Constant Output Power As a consequence, the output voltage exhibits a ripple at a frequency of either 100 Hz (for 50 Hz mains such as in Europe) or 120 Hz (for 60 Hz mains in the USA). This ripple must not be taken into account by the regulation loop because the error amplifier’s output voltage must be kept constant over a given ac line cycle for a proper shaping of the line current. Due to this constraint, the regulation bandwidth is typically set below 20 Hz. For a simple type 1 compensation network, only a capacitor is placed between FB and Control (see Figure 1). In this configuration, the capacitor necessary to attenuate the bulk voltage ripple is given by: VDD ICHARGE Ct + PWM − + ton DRV VEAL VCt G 10 20 CCOMP + 4 @ p fline @ ROUT1 VCONTROL Control VCt(off) VCONTROL − VEAL (eq. 4) where G is the attenuation level in dB (commonly 60 dB) ton ON TIME SEQUENCE Since the NCP1607 is designed to control a CRM boost converter, its switching pattern must accommodate constant on times and variable off times. The Controller generates the on time via an external capacitor connected to pin 3 (Ct). A current source charges this capacitor to a level determined by the Control pin voltage. Specifically, Ct is charged to VCONTROL minus the VEAL offset (2.1 V typical). Once this level is exceeded, the drive is turned off (Figure 27). DRV Figure 27. On Time Generation Since VCONTROL varies with the RMS line level and output load, this naturally satisfies equation 1. And if the values of compensation components are sufficient to filter www.onsemi.com 12 NCP1607 DRV out the bulk voltage ripple, then this on time is truly constant over the ac line cycle. Note that the maximum on time of the controller occurs when VCONTROL is at its maximum. Therefore, the Ct capacitor must be sized to ensure that the required on time can be delivered at full power and the lowest input voltage condition. The maximum on time is given by: ton(MAX) + Ct @ VCTMAX I CHARGE VOUT Drain (eq. 5) VZCD(off) (eq. 6) VZCD(on) VCL(POS) VZCDH VZCDL Combining this equation with equation 1, gives: Ct w 2 @ P OUT @ L @ I CHARGE h @ Vac 2 @ V CTMAX where VCTMAX = 2.9 V (min) ICHARGE = 297 mA (max) Winding ZCD VCL(NEG) OFF TIME SEQUENCE While the on time is constant across the ac cycle, the off time in CRM operation varies with the instantaneous input voltage. The NCP1607 determines the correct off time by sensing the inductor voltage. When the inductor current drops to zero, the drain voltage (“Vdrain” in Figure 23) is essentially floating and naturally begins to drop. If the switch is turned on at this moment, then CRM operation will be achieved. To measure this high voltage directly on the inductor is generally not economical or practical. Rather, a smaller winding is taken off of the boost inductor. This winding, called the zero current detector (ZCD) winding, gives a scaled version of the inductor output and is more useful to the controller. Figure 28. Voltage Waveforms for Zero Current Detection Figure 28 gives typical operating waveforms with the ZCD winding. When the drive is on, a negative voltage appears on the ZCD winding. And when the drive is off, a positive voltage appears. When the inductor current drops to zero, then the ZCD voltage falls and starts to ring around zero volts. The NCP1607 detects this falling edge and starts the next driver on time. To ensure that a ZCD event has truly occurred, the NCP1607’s logic (Figure 29) waits for the ZCD pin voltage to rise above VZCDH (2.1 V typical) and then fall below VZCDL (1.6 V typical). In this way, CRM operation is easily achieved. NB Vin NZCD + − + VDD RSENSE S VZCDH + − VCL(NEG) Active Clamp DRIVE VZCDL + ZCD RZCD + VCL(POS) Clamp − + Shutdown VSDL Figure 29. Implementation of the ZCD Winding www.onsemi.com 13 Q Reset Dominant Latch R Q Demag NCP1607 To prevent negative voltages on the ZCD pin, the pin is internally clamped to VCL(NEG) (600 mV typical) when the ZCD winding is negative. Similarly, the ZCD pin is clamped to VCL(POS) (5.7 V typical), when the voltage rises too high. Because of these clamps, a resistor (RZCD in Figure 29) is necessary to limit the current from the ZCD winding to the ZCD pin. At startup, there is no energy in the ZCD winding and therefore no voltage signal to activate the ZCD comparators. This means that the driver could never turn on. Therefore, to enable the PFC stage to startup under these conditions, an internal watchdog timer is integrated into the controller. This timer turns the drive on if the driver has been off for more than 180 ms (typical). This feature is deactivated during a fault mode (OVP, UVP, or Shutdown), and reactivated when the fault is removed. level, the internal references and logic of the NCP1607 turn on. The controller has an undervoltage lockout (UVLO) feature which keeps the part active until VCC drops below VCC(off) (9.5 V typical). This hysteresis allows ample time for the auxiliary winding to take over and supply the necessary power to VCC (Figure 30). VCC(on) VCC VCC(off) Figure 30. Typical VCC Startup Waveform When the PFC pre−converter is loaded by a switch mode power supply (SMPS), then it is often preferable to have the SMPS controller startup first. The SMPS can then supply the NCP1607 VCC directly. Advanced controllers, such as the NCP1230 or NCP1381, can control when to turn on the PFC stage (see Figure 31) leading to optimal system performance. This setup also eliminates the startup resistors and therefore improves the no load power dissipation of the system. STARTUP Generally, a resistor connected between the ac input and VCC (pin 8) charges the VCC capacitor to the VCC(on) level (12 V typical). Because of the very low consumption of the NCP1607 during this stage (< 40 mA), most of the current goes directly to charging up the VCC capacitor. This provides faster startup times and reduced standby power dissipation. When the VCC voltage exceeds the VCC(on) DBOOST + CBULK PFC_VCC 8 2 3 4 NCP1607 1 1 8 7 2 7 6 3 6 5 4 5 + VCC + + + NCP1230 Figure 31. NCP1607 Supplied by a Downstream SMPS Controller (NCP1230) QUICK START and SOFT START At startup, the error amplifier is enabled and Control is pulled up to VEAL (2.1 V typical). This is the lowest level of control voltage which produces output drives. This feature, called “quick start,” eliminates the delay at startup associated with charging the compensation network to its minimum level. This also produces a natural “soft−start” mode where the controller’s power ramps up from zero to the required power (see Figure 32). www.onsemi.com 14 NCP1607 VCC(on) OUTPUT DRIVER The NCP1607 includes a powerful output driver capable of peak currents of Source 500 mA / Sink 800 mA. This enables the controller to efficiently drive power MOSFETs for medium power (up to 300 W) applications. Additionally, the driver stage is equipped with both passive and active pull down clamps (Figure 33). The clamps are active when VCC is off and force the driver output to well below the threshold voltage of a power MOSFET. VCC VCC(off) IM VREF FB Control VEAL Natural Soft Start VOUT Figure 32. Startup Timing Diagram Showing the Natural Soft Start of the Control Pin VCC + − VDD UVLO UVLO DRV IN DRV VDDGD + VDDREG uVDD GND Figure 33. Output Driver Stage and Pull Down Clamps Overvoltage Protection and disables the driver until the output voltage returns to nominal levels. This keeps the output voltage within an acceptable range. The limit is adjustable so that the overvoltage level can be optimally set. The level must not be so low that it is triggered by the 100 or 120 Hz ripple of the output voltage, but it must be low enough so as not to require a larger voltage rating of the output capacitor. Figure 34 depicts the operation of the OVP circuitry. The low bandwidth of the feedback network makes active PFC stages very slow systems. One consequence of this is the risk of huge overshoots in abrupt transient phases (startup, load steps, etc.). For reliable operation, it is critical that some form of overvoltage protection (OVP) effectively prevents the output voltage from rising too high. The NCP1607 detects these excessive VOUT levels www.onsemi.com 15 NCP1607 VOUT UVP − + + IROUT1 ROUT1 VUVP IRFB (Enable EA) E/A FB − + + RFB IROUT2 ROUT2 Dynamic OVP ICONTROL > Iovp VREF Measure ICONTROL Fault VDD CCOMP VEAL Static OVP Clamp Static OVP is triggered when clamp is activated. Enable VCONTROL Control ICONTROL VEAH Clamp Figure 34. OVP and UVP Circuit Blocks When the output voltage is in steady state equilibrium, ROUT1 and ROUT2 regulate the FB voltage to VREF. During this equilibrium state, no current flows through the compensation capacitor (CCOMP shown in Figure 34). These facts allow the following equations to be derived: • The ROUT1 current is: V * V REF IROUT1 + OUT R OUT1 IROUT1 + V REF R EQ + I ROUT2 ) I FB V OUT * V REF R OUT1 * V REF R EQ IControl + I ROUT1 * I EQ + (eq. 7) V REF R EQ ROUT1 (eq. 11) V OUT ) DVOUT * VREF ROUT1 * VREF REQ (eq. 12) The combination of Equations 2 and 12 yield a simple expression of the current sunk by the error amplifier: (eq. 8) ICONTROL + DV OUT R OUT1 The current absorbed by pin 2 (IControl) is proportional to the output voltage excess. The circuit senses this current and disables the drive (pin 7) when IControl exceeds IOVP (10.4 mA typical). The OVP threshold is calculated using Equation 13. (eq. 9) Under stable conditions, Equations 7 through 9 are true. Conversely, when VOUT is not at the target voltage, the output of the error amplifier sinks or sources the current necessary to maintain VREF on pin 1. In the case of an overvoltage condition: • The error amplifier maintains VREF on pin 1, and the REQ current remains the same as the steady state value: IEQ + V OUT ) DVOUT * VREF where DVOUT is the output voltage excess. • And since no current flows through CCOMP, IROUT1 + ROUT1 + • The error amplifier sinks: • The REQ current is: IEQ + V OUT(OVP) * V REF VOUT(OVP) + V OUT ) R OUT1 @ I OVP (eq. 13) The OVP limit is set by adjusting ROUT1. ROUT1 is calculated using Equation 14. ROUT1 + V OUT(OVP) * V OUT (eq. 14) IOVP For example, if 440 V is the maximum output voltage and 400 V is the target output voltage, then ROUT1 is calculated using Equation 14. (eq. 10) • The ROUT1 current is increased and is calculated using ROUT1 + 440 * 400 + 3.846 MW 10.4m Equation 11: If ROUT1 is selected as 4 MW,, then VOUT(OVP) = 442 V. www.onsemi.com 16 NCP1607 STATIC OVERVOLTAGE PROTECTION However, if the FB pin voltage increases and exceeds the UVP level, then the controller will start the application up normally. If the OVP condition lasts for a long time, it may happen that the error amplifier output reaches its minimum level (i.e. Control = VEAL). It would then not be able to sink any current and maintain the OVP fault. Therefore, to avoid any discontinuity in the OVP disabling effect, the circuit incorporates a comparator which detects when the lower level of the error amplifier is reached. This event, called “static OVP”, disables the output drives. Once the OVP event is over, and the output voltage has dropped to normal, then Control rises above the lower limit and the driver is re−enabled (Figure 35). VCC(on) VCC VCC(off) VOUT VOUT FB 2.5 V VUVP VOUT VEAH UVP Fault is “Removed” Control VEAL DRV UVP Wait VEAH VCONTROL UVP Wait UVP VEAL Figure 36. The NCP1607’s Startup Sequence with and without a UVP Fault IOVPH IOVPL ICONTROL The voltage on the output which exits a UVP fault is given by: Dynamic OVP VOUT(UVP) + R OUT1 ) R EQ @ V UVP R EQ (eq. 15) If ROUT1 = 4 MW and REQ = 25.16 kW, then the VOUT UVP threshold is 48 V. This corresponds to an input voltage of approximately 34 Vac. Static OVP Figure 35. OVP Timing Diagram Open Feedback Loop Protection NCP1607 Undervoltage Protection (UVP) The NCP1607 features comprehensive protection against open feedback loop conditions by including OVP, UVP, and Floating Pin Protection (FPP). Figure 37 illustrates three conditions in which the feedback loop is open. The corresponding number below describes each condition shown in Figure 37. 1. UVP Protection: The connection from resistor ROUT1 to the FB pin is open. ROUT2 pulls down the FB pin to ground. The UVP comparator detects a UVP fault and the drive is disabled. 2. OVP Protection: The connection from resistor ROUT2 to the FB pin is open. ROUT1 pulls up the FB pin to the output voltage. The ESD diode clamps the FB voltage to 10 V and ROUT1 limits the current into the FB pin. The VEAL clamp detects a static OVP fault and the drive is disabled. 3. FPP Protection: The FB pin is floating. The internal pulldown resistor RFB pulls down the FB voltage below the UVP threshold. The UVP comparator detects a UVP fault and the drive is disabled. When the PFC stage is plugged in, the output voltage is forced to roughly equate the peak line voltage. The NCP1607 detects an undervoltage fault when this output voltage is unusually low, such that the feedback voltage is below VUVP (300 mV typical). In an UVP fault, the drive output and error amplifier (EA) are disabled. The latter is done so that the EA does not source a current which would increase the FB voltage and prevent the UVP event from being accurately detected. The UVP feature helps to protect the application if something is wrong with the power path to the bulk capacitor (i.e. the capacitor cannot charge up) or if the controller cannot sense the bulk voltage (i.e. the feedback loop is open). Furthermore, the NCP1607 incorporates a novel startup sequence which ensures that undervoltage conditions are always detected at startup. It accomplishes this by waiting approximately 180 ms after VCC reaches VCC(on) before enabling the error amplifier (Figure 36). During this wait time, it looks to see if the feedback (FB) voltage is greater than the UVP threshold. If not, then the controller enters a UVP fault and leaves the error amplifier disabled. www.onsemi.com 17 NCP1607 UVP and OVP protect the system from low bulk voltages and rapid operating point changes respectively, while the FPP protects the system against floating feedback pin conditions. If FPP is not implemented and a manufacturing error causes the feedback pin to float, then the feedback voltage is dependent on the coupling within the system and the surrounding environment. The coupled feedback voltage may be within the regulation limits (i.e. above the UVP threshold, but below VREF) and cause the controller to deliver excessive power. The result is that the output voltage rises until a component fails due to the voltage stress. The tradeoff for including FPP is that the value of RFB causes an error in the output voltage. The output voltage including the error caused by RFB (VOUT) is calculated using Equation 16: VOUT + V OUT ) R OUT1 @ V REF The error caused by RFB is compensated by adjusting ROUT2. The parallel combination of RFB and ROUT2 form an equivalent resistor REQ that is calculated using Equation 17. REQ + R OUT1 @ REQ + 4 M @ ROUT2 + 2.5 (eq. 18) R FB * R EQ 25.16 k @ 4.7 M 4.7 M * 25.16 k + 25.29 kW The compensated output voltage is calculated using Equation 19. (eq. 16) VOUT + VREF @ + 402 V ǒ Ǔ ROUT1 ) ROUT2 VOUT + 2.5 @ VOUT 2.5 + 25.16 kW 400 * 2.5 R EQ @ R FB ROUT2 + RFB 4.7 M (eq. 17) V OUT * V REF REQ is used to calculate ROUT2. Using the values from the OVP calculation, the output voltage including the error caused by RFB is equal to: VOUT + 400 ) 4 M @ V REF R OUT2 ǒ 4 M ) 25.29 k 25.29 k Ǔ ) ROUT1 @ )4 M@ VREF (eq. 19) RFB 2.5 4.7 M + 400 V UVP + + ROUT1 VUVP Condition 1 E/A Condition 3 FB RFB Condition 2 (Enable EA) + Dynamic OVP ICONTROL > Iovp + Measure VREF ROUT2 ICONTROL VDD CCOMP Enable VCONTROL Control Fault VEAL Static OVP Clamp Static OVP is triggered when clamp is activated VEAH Clamp ICONTROL Figure 37. Open Feedback Loop Protection Overcurrent Protection (OCP) An internal LEB filter (Figure 38) reduces the likelihood of switching noise falsely triggering the OCP limit. This filter blanks out the first 250 ns (typical) of the current sense signal. If additional filtering is necessary, a small RC filter can be added between RSENSE and the CS pin. A dedicated pin on the NCP1607 senses the peak current and limits the driver on time if this current exceeds VCS(limit). This level is 0.5 V (typical). Therefore, the maximum peak current can be adjusted by changing RSENSE according to: Ipeak + V CS(limit) RS (eq. 20) www.onsemi.com 18 NCP1607 SHUTDOWN MODE The NCP1607 allows for two methods to place the controller into a standby mode of operation. The FB pin can be pulled below the UVP level (300 mV typical) or the ZCD pin can be pulled below the VSDL level (200 mV typical). If the FB pin is used for shutdown (Figure 39(a)), care must be taken to ensure that no significant leakage current exists on the shutdown circuitry. This could impact the output voltage regulation. If the ZCD pin is used for shutdown (Figure 39(b)), then any parasitic capacitance created by the shutdown circuitry will add to the delay in detecting the zero inductor current event. DRV CS + RS OCP + − LEB VCS(limit) optional Figure 38. OCP Circuitry with Optional External RC Filter LBOOST VOUT ROUT1 NCP1607 NCP1607 CCOMP Shutdown ROUT2 1 FB VCC 8 1 2 Control DRV 7 2 Control DRV 7 3 Ct GND 6 3 Ct GND 6 4 CS ZCD 5 4 CS ZCD 5 FB VCC 8 RZCD Shutdown Figure 39(a) Figure 39(b) Figure 39. Shutting Down the PFC Stage by Pulling FB to GND (A) or Pulling ZCD to GND (B) To activate the shutdown feature on ZCD, the internal clamp must first be overcome. This clamp will draw a maximum of ICL(NEG) (5.0 mA maximum) before releasing and allowing the ZCD pin voltage to drop low enough to shutdown the part (Figure 40). After shutdown, the comparator includes approximately 90 mV of hysteresis to ensure noise free operation. A small current source (70 mA typical) is also activated to pull the unit out of the shutdown condition when the external pull down is released. 5 mA ~70 mA IZCD Shutdown Controller Disabled Controller Enabled VSDL VSDH VCL(NEG) ~1 V Figure 40. Shutdown Comparator and Current Draw to Overcome Negative Clamp www.onsemi.com 19 NCP1607 Application Information The electronic design tool allows the user to easily determine most of the system parameters of a boost pre−converter. The demonstration board is a boost pre−converter that delivers 100 W at 400 V. The circuit schematic is shown in Figure 41. The pre−converter design is described in Application Note AND8353/D. ON Semiconductor provides an electronic design tool, a demonstration board and an application note to facilitate the design of the NCP1607 and reduce development cycle time. All the tools can be downloaded or ordered at www.onsemi.com. RSTART1 RSTART2 LBOOST DBOOST J3 NTC t BRIDGE F1 R1 D1 C3 RCTUP1 L1 L2 J2 C1 RO1A + DAUX CVCC DVCC RZCD C2 RO1B RCTUP2 J1 CIN CBUL- U1 NCP1607 CCOMP CCOMP1RCOMP2 1 2 FB K VCC 8 Control DRV 7 3 Ct GND 6 4 CS ZCD 5 RCT RCS CT2 CT1 CCS CVCC DDRV 2 Q1 RDRV ROUT2B RS3 CZCD Figure 41. Application Board Circuit Schematic www.onsemi.com 20 RS2 RS1 ROUT2A + NCP1607 BOOST DESIGN EQUATIONS Components are identified in Figure 1 RMS Input Current Maximum Inductor Peak Current Ipk(MAX) + Inductor Value 2 @ Ǹ2 @ P OUT h @ Vac LL 2 @ Vac 2 @ Lv ton(MAX) + Boost Turns to ZCD Turns Ratio Resistor from ZCD winding to the ZCD pin (pin 5) Boost Output Voltage Vac 2 @ h @ 2 @ L @ P OUT Ct w ǒ 1* The maximum on time occurs at the lowest line voltage and maximum output power. The off time is greatest at the peak of the ac line voltage and approaches zero at the ac line zero crossings. Theta (q) represents the angle of the ac line voltage. Ǔ V * Vac HL @ Ǹ2 NB : N ZCD v OUT V ZCDH RZCD w Vac HL @ Ǹ2 I CL(NEG) @ (N B : N ZCD) VOUT + V REF @ RZCD must be large enough so that the shutdown comparator is not inadvertently activated. R OUT2 ) R FB V OUT(OVP) * V OUT IOVP is given in the NCP1607 specification table. IOVP V REF V OUT * V REF R EQ ) R FB R FB * R EQ VOUT(UVP) + V UVP @ R OUT1 ) R EQ VUVP is given in the NCP1607 specification table. R EQ POUT C BULK @ 2 @ p @ fline @ VOUT IL(RMS) + ID(RMS)MAX + 4 @ 3 Where VacHL is the maximum line input voltage. The turns ratio must be low enough so as to trigger the ZCD comparators at high line. R OUT2 @ R FB ROUT1 + Vripple(pk−pk) + ICHARGE and VCTMAX are given in the NCP1607 specification table. R OUT1 ) R EQ R EQ VOUT(OVP) + V OUT ) ǒI OVP @ R OUT1Ǔ Bulk Cap Ripple Boost Diode RMS Current fSW(MIN) is the minimum desired switching frequency. The maximum L must be calculated at low line and high line. 2 @ P OUT @ L @ I CHARGE h @ Vac 2 @ V CTMAX ROUT2 + Inductor RMS Current Ǔ Vac @ |sin q| @ Ǹ2 V OUT REQ + R OUT1 @ Minimum output voltage necessary to exit undervoltage protection (UVP) * Vac V OUT *1 Vac@Ťsin(q)Ť@Ǹ2 REQ + Maximum VOUT voltage prior to OVP activation and the necessary ROUT1 and ROUT2. OUT Ǹ2 ton toff + Pin 3 Capacitor V 2 @ L @ P OUT h @ Vac LL 2 Off Time fSW + ǒ Where VacLL is the minimum line input voltage. Ipk(MAX) occurs at the lowest line voltage. VOUT @ Vac @ I pk(MAX) @ fSW(min) Maximum On Time Frequency h (the efficiency of only the Boost PFC stage) is generally in the range of 90 − 95% POUT h @ Vac Iac + 2 @ P OUT Ǹ3 @ Vac @ h LL Ǹ2 @pǸ2 @ P OUT h @ ǸVac LL @ VOUT www.onsemi.com 21 Use fline = 47 Hz for worst case at universal lines. The ripple must not exceed the OVP level for VOUT. NCP1607 BOOST DESIGN EQUATIONS Components are identified in Figure 1 MOSFET RMS Current Pout IM(RMS)MAX + 2 @ @ Ǹ3 h @ Vac LL MOSFET Sense Resistor RS + Ǹ ǒ 1* Ǔ 8 @ Ǹ2 @ Vac LL 3 p @ V out V CS(limit) I pk(MAX) VCS(limit) is given in the NCP1607 specification table. PRS + I M(RMS) 2 @ RS Bulk Capacitor RMS Current Type 1 CCOMP IC(RMS) + Ǹ 32 @ Ǹ2 @ P OUT 2 * (ILOAD(RMS)) 2 9 @ p @ Vac LL @ VOUT @ h2 CCOMP + 10 Gń20 4 @ p @ f line @ ROUT1 www.onsemi.com 22 G is the desired attenuation in decibels (dB). Typically it is 60 dB. NCP1607 PACKAGE DIMENSIONS SOIC−8 NB CASE 751−07 ISSUE AJ NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSION A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. 6. 751−01 THRU 751−06 ARE OBSOLETE. NEW STANDARD IS 751−07. −X− A 8 5 S B 0.25 (0.010) M Y M 1 4 K −Y− G C N DIM A B C D G H J K M N S X 45 _ SEATING PLANE −Z− 0.10 (0.004) H M D 0.25 (0.010) M Z Y S X J S MILLIMETERS MIN MAX 4.80 5.00 3.80 4.00 1.35 1.75 0.33 0.51 1.27 BSC 0.10 0.25 0.19 0.25 0.40 1.27 0 _ 8 _ 0.25 0.50 5.80 6.20 INCHES MIN MAX 0.189 0.197 0.150 0.157 0.053 0.069 0.013 0.020 0.050 BSC 0.004 0.010 0.007 0.010 0.016 0.050 0 _ 8 _ 0.010 0.020 0.228 0.244 SOLDERING FOOTPRINT* 1.52 0.060 7.0 0.275 4.0 0.155 0.6 0.024 1.270 0.050 SCALE 6:1 mm Ǔ ǒinches *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. ON Semiconductor and the are registered trademarks of Semiconductor Components Industries, LLC (SCILLC) or its subsidiaries in the United States and/or other countries. SCILLC owns the rights to a number of patents, trademarks, copyrights, trade secrets, and other intellectual property. A listing of SCILLC’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent− Marking.pdf. SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner. PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colorado 80217 USA Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada Fax: 303−675−2176 or 800−344−3867 Toll Free USA/Canada Email: [email protected] N. American Technical Support: 800−282−9855 Toll Free USA/Canada Europe, Middle East and Africa Technical Support: Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81−3−5817−1050 www.onsemi.com 23 ON Semiconductor Website: www.onsemi.com Order Literature: http://www.onsemi.com/orderlit For additional information, please contact your local Sales Representative NCP1607/D