ISL8104 ® Data Sheet February 13, 2006 FN9257.0 Synchronous Buck Pulse-Width Modulator (PWM) Controller Features The ISL8104 is a high performance synchronous controller for demanding DC/DC converter applications. It provides overcurrent fault protection and is designed to safely start-up into prebiased output loads. • Excellent Output Voltage Regulation - 0.597V Internal Reference - ±1% Over the Commercial Temperature Range - ±1.5% Over the Industrial Temperature Range The output voltage of the converter can be precisely regulated to as low as 0.597V, with a maximum tolerance of ±1% over the commercial temperature range, and ±1.5% over the industrial temperature range. • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Operates from an +8V ±5% to +14V ±10% Input The ISL8104 provides simple, single feedback loop, voltagemode control with fast transient response. It includes a triangle-wave oscillator that is adjustable from below 50kHz to over 1.5MHz. Full (0% to 100%) PWM duty cycle support is provided. • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Ratio - Leading and Falling Edge Modulation • Small Converter Size - Constant Frequency Operation - Oscillator Programmable from 50kHz to Over 1.5MHz The error amplifier features a 15MHz gain-bandwidth product and 6V/µs slew rate which enables high converter bandwidth for fast transient performance. • 14V High Speed MOSFET Gate Drivers - 2.0A Source/3A Sink at 14V Low Side Gate Drive - 1.25A Source/2A Sink at 14V High Side Gate Drive - Drives Two N-Channel MOSFETs The ISL8104's overcurrent protection monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. • Overcurrent Fault Monitor - High-Side MOSFET’s rDS(ON) Sensing - Reduced ~120ns Blanking Time Pinouts • Converter can Source and Sink Current ISL8104 (14 LD NARROW SOIC AND 16 LD QFN) TOP VIEW RT 1 OCSET 2 SS 3 • Soft-Start Done and an External Reference Pin for Tracking Applications are Available in the QFN Package • Pin Compatible with ISL6522 and ISL6535 14 VCC • Supports Start-Up into Prebiased Loads 13 PVCC • Pb-Free Plus Anneal Available (RoHS Compliant) 12 LGATE Applications COMP 4 11 PGND FB 5 10 BOOT • Test & Measurement Instruments EN 6 9 UGATE • Routers Switches GND 7 8 PHASE • Medical Instrumentation • Industrial Applications SS SSDONE OCSET RT VCC • Telecom/Datacom Applications 16 15 14 13 1 Ordering Information PART # (Note) 12 PVCC ISL8104CBZ 8104CBZ ISL8104IBZ COMP 2 11 LGATE FB 3 10 PGND 6 1 7 8 UGATE 5 PHASE 9 GND 4 REFIN EN PART TEMP. MARKING RANGE (°C) ISL8104CRZ 8104CRZ ISL8104IRZ BOOT 8104IBZ 8104IRZ PACKAGE (Pb-free) PKG. DWG. # 0 to 70 14 Ld SOIC M14.15 -40 to 85 14 Ld SOIC M14.15 0 to 70 16 Ld 4x4 QFN L16.4x4 -40 to 85 16 Ld 4x4 QFN L16.4x4 Add “-T” suffix for tape and reel. NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2006. All Rights Reserved All other trademarks mentioned are the property of their respective owners. Block Diagram EN SS VCC OCSET INTERNAL REGULATOR 30µA 6µA 200µA POWER-ON RESET (POR) 2 REFERENCE VREF = 0.597 V BOOT REFIN (QFN ONLY) SOFT-START AND FAULT LOGIC SOURCE OCP UGATE EA PHASE PWM PVCC COMP OSCILLATOR LGATE GND PGND SSDONE (QFN ONLY) RT ISL8104 FB GATE CONTROL LOGIC FN9257.0 February 13, 2006 ISL8104 Simplified Power System Diagram ROCSET +8V to +14V +1.2V to +14VIN Q1 Cvcc LOUT VOUT ISL8104 COUT RFS Q2 CSS R1 R2 Typical Application +14VIN LIN RFILTER CHFIN CBIN DBOOT CF2 CF1 VCC PVCC BOOT ROCSET OCSET SSDONE (QFN ONLY) COCSET Q1 UGATE REFIN (QFN ONLY) CBOOT LOUT VOUT PHASE Q2 LGATE EN CHFOUT CBOUT PGND RRT RT SS ISL8104 COMP CSS C2 C1 C3 R3 R2 FB R1 GND 3 RO FN9257.0 February 13, 2006 ISL8104 Absolute Maximum Ratings Thermal Information Supply Voltage, VPVCC,VVCC . . . . . . . . . . . . . . GND - 0.3V to +16V Enable Voltage, VEN . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +16V Soft-start Done Voltage, VSSDONE . . . . . . . . . . GND - 0.3V to +16V Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +36V Phase Voltage, VPHASE . . . . . . . . . VBOOT - 16V to VBOOT + 0.3V All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5.0V Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W) SOIC Package (Note 1) . . . . . . . . . . . . 95 N/A QFN Package (Note 2). . . . . . . . . . . . . 47 8.5 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150°C Maximum Storage Temperature Range . . . . . . . . . . . -65°C to 150°C Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300°C (SOIC - Lead tips only) Operating Conditions ESD Ratings Supply Voltage, VVCC . . . . . . . . . . . . . . . . .+8V ±5% to +14V ±10% Supply Voltage, VPVCC . . . . . . . . . . . . . . . .+8V ±5% to +14V ±10% Boot to Phase Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . <VPVCC Ambient Temperature Range, ISL8104C . . . . . . . . . . . . 0°C to 70°C Ambient Temperature Range, ISL8104I. . . . . . . . . . . .-40°C to 85°C ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. 3. Parameters designated by GBD are "Guaranteed by Design." Electrical Specifications Recommended Operating Conditions, unless otherwise noted specifications in bold are valid for process, temperature, and line operating conditions. PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS VCC SUPPLY CURRENT Shutdown Supply VCC Shutdown Supply VPVCC IVCC SS/EN = 0V 3.5 6.1 8.5 mA IPVCC SS/EN = 0V 0.30 0.5 0.75 mA POWER-ON RESET VCC/VPVCC Rising Threshold 6.55 7.10 7.55 V VCC/VPVCC Hysteresis 170 250 500 mV OCSET Rising Threshold 0.70 0.73 0.75 V OCSET Hysteresis 180 200 220 mV Enable - Rising Threshold 1.4 1.5 1.60 V Enable - Hysteresis 175 250 325 mV 175 200 220 kHz OSCILLATOR Trim Test Frequency RRT = OPEN VVCC = 12 - ±15 - % 1.7 1.9 2.15 VP-P RL = 10kΩ, CL= 100pF - GBD - 88 - dB GBWP RL = 10kΩ, CL= 100pF - GBD - 15 - MHz SR RL = 10kΩ, CL= 100pF - GBD - 6 - V/µs µA 8kΩ < RRT to GND < 200kΩ - GBD Total Variation ∆VOSC Ramp Amplitude RRT = OPEN ERROR AMPLIFIER DC Gain Gain-Bandwidth Product Slew Rate PROTECTION OCSET Current OCSET Current IOCSET TJ = 0°C to 70°C 180 200 220 IOCSET TJ = -40°C to 85°C 176 200 224 µA - ±10 - mV 22 30 38 µA OCPOFFSET OCSET= 1.5V to 15.4V - GBD OCSET Measurement Offset Soft-start Current ISS 4 FN9257.0 February 13, 2006 ISL8104 Electrical Specifications Recommended Operating Conditions, unless otherwise noted specifications in bold are valid for process, temperature, and line operating conditions. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS TJ = 0°C to 70°C 0.591 0.597 0.603 V TJ = -40°C to 85°C 0.588 0.597 0.606 V TJ = 0°C to 70°C -1.0 - 1.0 % TJ = -40°C to 85°C -1.5 - 1.5 % -4 -6 -8 µA 2.10 - 3.50 V -3 - 3 mV - 1.25 - A - 2.0 - Ω REFERENCE Reference Voltage System Accuracy REFIN Current Source (QFN Only) REFIN Threshold (QFN Only) REFIN Offset (QFN Only) GATE DRIVERS VBOOT - VPHASE = 14V, 3nF Load - GBD Upper Drive Source Current IU_SOURCE Upper Drive Source Impedance RU_SOURCE 90mA Source Current IU_SINK VBOOT - VPHASE = 14V, 3nF Load - GBD - 2 - A Upper Drive Sink Impedance RU_SINK 90mA Source Current - 1.3 - Ω Lower Drive Source Current IL_SOURCE VPVCC = 14V, 3nF Load - GBD - 2 - A Lower Drive Source Impedance RL_SOURCE 90mA Source Current - 1.3 - Ω Upper Drive Sink Current Lower Drive Sink Current IL_SINK VPVCC = 14V, 3nF Load - GBD - 3 - A Lower Drive Sink Impedance RL_SINK 90mA Source Current - 0.94 - Ω 0.30 V SSDONE (QFN ONLY) SSDONE Low Output Voltage ISSDONE = 2mA Typical Performance Curves 80 70 RRT PULLUP TO +14V 60 IPVCC+VCC (mA) RESISTANCE (kΩ) 1000 100 RRT PULLDOWN TO GND CGATE = 3300pF 50 CGATE = 1000pF 40 30 20 10 CGATE = 10pF 10 10 100 SWITCHING FREQUENCY (kHz) 1000 FIGURE 1. RRT RESISTANCE vs FREQUENCY 0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (kHz) FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY Functional Pin Description (SOIC/QFN) RT (Pin 1/14) This pin provides oscillator switching frequency adjustment. By placing a resistor (RRT) from this pin to GND, the switching frequency is set from between 200kHz and 1.5MHz according to the following equation: 6500 R RT [ kΩ ] ≈ ------------------------------------------------------- – 1.3kΩ F s [ kHz ] – 200 [ kHz ] 5 (RRT to GND) Alternately ISL8104’s switching frequency can be lowered from 200kHz to 50kHz by connecting the RT pin with a resistor to VCC according to the following equation: 55000 R RT [ kΩ ] ≈ ------------------------------------------------------- + 70kΩ 200 [ kHz ] – F s [ kHz ] (RRT to VCC) FN9257.0 February 13, 2006 ISL8104 OCSET (Pin 2/15) VCC (Pin 14/13) The current limit is programmed by connecting this pin with a resistor and capacitor to the drain of the high side MOSEFT. A 200µA current source develops a voltage across the resistor which is then compared with the voltage developed across the high side MOSFET. A blanking period of 120ns is provided for noise immunity. Provide an 8V to 14V bias supply for the chip to this pin. The pin should be bypassed with a capacitor to GND. SS (Pin 3/1) Connect a capacitor from this pin to ground. This capacitor, along with an internal 30µA current source, sets the soft-start interval of the converter. REFIN (QFN ONLY Pin 5) Upon enable if REFIN is less than 2.2V, the external reference pin is used as the control reference instead of the internal 0.597V reference. An internal 6µA pull up to 5V is provided for disabling this functionality. SSDONE (QFN ONLY Pin 16) Provides an open drain signal at the end of soft-start. COMP (Pin 4/2) and FB (Pin 5/3) Functional Description COMP and FB are the available external pins of the error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the converter. Initialization EN (Pin 6/4) This pin is a TTL compatible input. Pull this pin below 0.8V to disable the converter. In shutdown the soft-start pin is discharged and the UGATE and LGATE pins are held low. GND (Pin 7/6) Signal ground for the IC. All voltage levels are measured with respect to this pin. The ISL8104 automatically initializes upon receipt of power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the bias voltage at the VCC pin and the driver input on the PVCC pin. When the voltages at VCC and PVCC exceed their rising POR thresholds, a 30µA current source driving the SS pin is enabled. Upon the SS pin exceeding 1V, the ISL8104 begins ramping the non-inverting input of the error amplifier from GND to the System Reference. During initialization the MOSFET drivers pull UGATE to PHASE and LGATE to PGND. Soft-Start PHASE (Pin 8/7) This pin connects to the source of the high side MOSFET and the drain of the low side MOSFET. This pin represents the return path for the high side gate driver. During normal switching, this pin is used for high side current sensing. UGATE (Pin 9/8) Connect UGATE to the upper MOSFET gate. This pin provides the gate drive for the upper MOSFET. During soft-start, an internal 30µA current source charges the external capacitor (CSS) on the SS pin up to ~4V. If the ISL8104 is utilizing the internal reference, then as the SS pin’s voltage ramps from 1V to 3V, the soft-start function scales the reference input (positive terminal of error amp) from GND to VREF (0.597V nominal). If the ISL8104 is utilizing an VEN BOOT (Pin 10/9) VOUT This pin provides bias to the upper MOSFET driver. A bootstrap circuit may be used to create a BOOT voltage suitable to drive a standard N-Channel MOSFET. VSS PGND (Pin 11/10) This is the power ground connection. Tie the lower MOSFET source and board ground to this pin. LGATE (Pin 12/11) Connect LGATE to the lower MOSFET gate. This pin provides the gate drive for the lower MOSFET. tSS PVCC (Pin 13/12) Provide an 8V to 14V bias supply for the lower gate drive to this pin. This pin should be bypassed with a capacitor to PGND. 6 FIGURE 3. TYPICAL SOFT-START INTERVAL externally supplied reference, when the voltage on the SS pin reaches 1V, the internal reference input (into of the error amp) ramps from GND to the externally supplied reference at the same rate as the voltage on the SS pin. Figure 3 shows a FN9257.0 February 13, 2006 ISL8104 typical soft-start interval. The rise time of the output voltage is, therefore, dependent upon the value of the soft-start capacitor, CSS. If the internal reference is used, then the softstart capacitance value can be calculated through: 30µA ⋅ t SS C SS = ---------------------------2V If an external reference is used, then the soft-start capacitance can be calculated through: 30µA ⋅ t SS C SS = ---------------------------V REFEXT Prebiased Load Start-up Drivers are held in tri-state (UG pulled to Phase, LG pulled to PGND) at the beginning of a soft-start cycle until two PWM pulses are detected. The low side MOSFET is turned on first to provide for charging of the bootstrap capacitor. This method of driver activation provides support for start-up into prebiased loads by not activating the drivers until the control loop has entered its linear region, thereby substantially reducing output transients that would otherwise occur had the drivers been activated at the beginning of the soft-start cycle. A 120ns blanking period is used to reduce the current sampling error due to leading-edge switching noise. An additional simultaneous 120ns low pass filter is used to further reduce measurement error due to noise. OCP faults cause the regulator to disable (upper and lower drives disabled, SSDONE pulled low, soft-start capacitor discharged) itself for a fixed period of time, after which a normal soft-start sequence is initiated. If the voltage on the SS pin is already at 4V and an OCP is detected, a 30µA current sink is immediately applied to the SS pin. If an OCP is detected during soft-start, the 30µA current sink will not be applied until the voltage on the SS pin has reached 4V. This current sink discharges the CSS capacitor in a linear fashion. Once the voltage on the SS pin has reached approximately 0V, the normal soft-start sequence is initiated. If the fault is still present on the subsequent restart, the ISL8104 will repeat this process in a hiccup mode. Figure 4 shows a typical reaction to a repeated overcurrent condition that places the regulator in a hiccup mode. If the regulator is repeatedly tripping overcurrent, the hiccup period can be approximated by the following formula: 8V ⋅ C SS T HICCUP = ----------------------30µA SSDONE VSSDONE Soft-start done is only available in the 16 Lead QFN packaging option of the ISL8104. When the soft-start pin reaches 4V, an open drain signal is provided to support sequencing requirements. SSDONE is deasserted by disabling of the part, including pulling SS low, and by POR and OCP events. Oscillator VSS IOCP The oscillator is a triangular waveform, providing for leading and falling edge modulation. The peak to peak of the ramp amplitude is set at 1.9V and varies as a function of frequency. At 50kHz the peak to peak amplitude is approximately 1.8V while at 1.5MHz it is approximately 2.2V. In the event the regulator operates at 100% duty cycle for 64 clock cycles an automatic boot cap refresh circuit will activate turning on LG for approximately 1/2 of a clock cycle. ILOAD THICCUP FIGURE 4. TYPICAL OVERCURRENT PROTECTION Overcurrent Protection The OCP function is enabled with the drivers at start-up. OCP is implemented via a resistor (ROCSET) and a capacitor (COCSET) connecting the OCSET pin and the drain of the high side MOSEFT. An internal 200mA current source develops a voltage across ROCSET which is then compared with the voltage developed across the high side MOSFET at turn on as measured at the PHASE pin. When the voltage drop across the MOSFET exceeds the voltage drop across the resistor, a sourcing OCP event occurs. COCSET is placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input bus. 7 The OCP trip point varies mainly due to MOSFET rDS(ON) variations and layout noise concerns. To avoid overcurrent tripping in the normal operating load range, find the ROCSET resistor from the following equations with: 1. The maximum rDS(ON) at the highest junction temperature; 2. The minimum IOCSET from the specification table; FN9257.0 February 13, 2006 ISL8104 Determine the overcurrent trip point greater than the maximum output continuous current at maximum inductor ripple current. Simple OCP Equation I OC_SOURCE • r DS ( ON ) R OCSET = --------------------------------------------------------------200µA Detailed OCP Equation ∆I I + ----- • r OC_SOURCE 2 DS ( ON ) R OCSET = ---------------------------------------------------------------------------------I HSOC • N U N U = NUMBER OF HIGH SIDE MOSFETs V IN - V OUT V OUT ∆I = --------------------------------- • ---------------F SW • L OUT V IN F SW = Regulator Switching Frequency traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible using ground plane construction or single point grounding. A multi-layer printed circuit board is recommended. Figure 5 shows the critical components of the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. High Speed MOSFET Gate Driver +14V VCC CBP_PVCC PVCC CBP_VCC ISL8104 VIN CIN UGATE Reference Input BOOT The REFIN pin allows the user to bypass the internal 0.597V reference with an external reference. If REFIN is NOT above ~2.2V, the external reference pin is used as the control reference instead of the internal 0.597V reference. When not using the external reference option the REFIN pin should be left floating. An internal 6mA pull-up keeps this REFIN pin above 2.2V in this situation. Internal Reference and System Accuracy The Internal Reference is set to 0.597V. The total DC system accuracy of the system is to be within 1.0% over commercial temperature range and 1.5% over the industrial temperature range. System Accuracy includes Error Amplifier offset, and Reference Error. The use of REFIN may add up to 3mV of offset error into the system (as the Error Amplifier offset is trimmed out via the internal System reference.) Application Guidelines Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit 8 Q1 CIN LOUT VOUT PHASE COUT LGATE LOAD The integrated driver has the same drive capability and feature as the Intersil’s 12V gate driver, ISL6612. The PWM tri-state feature helps prevent a negative transient on the output voltage when the output is being shut down. This eliminates the Schottky diode that is used in some systems for protecting the microprocessor from reversed-outputvoltage damage. See the ISL6612 datasheet for specification parameters that are not defined in the current ISL8104 electrical specifications table. Q2 SS GND PGND CSS KEY TRACE SIZED FOR 3A PEAK CURRENT SHORT TRACE, MINIMUM IMPEDANCE ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER VIA CONNECTION TO GROUND PLANE FIGURE 5. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS FN9257.0 February 13, 2006 ISL8104 Locate the ISL8104 within 2 to 3 inches of the MOSFETs, Q1 and Q2 (1 inch or less for 500kHz or higher operation). The circuit traces for the MOSFETs’ gate and source connections from the ISL8104 must be sized to handle up to 3A peak current. Minimize any leakage current paths on the SS pin and locate the capacitor, Css close to the SS pin as the internal current source is only 30µA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT pin and the phase node. C2 R2 COMP C3 R3 C1 - E/A + FB R1 VREF GND Compensating the Converter The ISL8104 Single-phase converter is a voltage-mode controller. This section highlights the design consideration for a voltage-mode controller requiring external compensation. To address a broad range of applications, a type-3 feedback network is recommended (see Figure 6). VOUT OSCILLATOR VIN PWM CIRCUIT C2 R2 C1 R3 UGATE COMP HALF-BRIDGE DRIVE FB C3 VOSC R1 ISL8104 ISL8104 Figure 7 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage is regulated to the reference voltage level. The error amplifier output is compared with the oscillator triangle wave to provide a pulse-width modulated wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter. The output filter capacitor bank’s equivalent series resistance is represented by the series resistor ESR. The modulator transfer function is the small-signal transfer function of VOUT /VCOMP. This function is dominated by a DC gain and shaped by the output filter, with a double pole break frequency at FLC and a zero at FCE . For the purpose of this analysis, L and DCR represent the output inductance and its DCR, while C and ESR represents the total output capacitance and its equivalent series resistance. 1 F LC = --------------------------2π ⋅ L ⋅ C 1 F CE = --------------------------------2π ⋅ C ⋅ ESR DCR PHASE C ESR LGATE VOUT FIGURE 6. COMPENSATION CONFIGURATION FOR THE ISL8104 CIRCUIT L EXTERNAL CIRCUIT FIGURE 7. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN The compensation network consists of the error amplifier (internal to the ISL8104) and the external R1-R3, C1-C3 components. The goal of the compensation network is to provide a closed loop transfer function with high 0dB crossing frequency (F0; typically 0.1 to 0.3 of FSW) and adequate phase margin (better than 45 degrees). Phase margin is the difference between the closed loop phase at F0dB and 180°. The equations that follow relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figures 6 and 7. Use the following guidelines for locating the poles and zeros of the compensation network: 1. Select a value for R1 (1kΩ to 10kΩ, typically). Calculate value for R2 for desired converter bandwidth (F0). If setting the output voltage to be equal to the reference set voltage as shown in Figure 7, the design procedure can be followed as presented. As the ISL8104 supports 100% V OSC ⋅ R 1 ⋅ F 0 R 2 = --------------------------------------------D MAX ⋅ V IN ⋅ F LC duty cycle, DMAX equals 1. The ISL8104 uses a fixed ramp amplitude (VOSC) of 1.9V, the above equation simplifies to: 1.9 ⋅ R 1 ⋅ F 0 R 2 = -----------------------------V IN ⋅ F LC 9 FN9257.0 February 13, 2006 ISL8104 2. Calculate C1 such that FZ1 is placed at a fraction of the FLC, at 0.1 to 0.75 of FLC (to adjust, change the 0.5 factor below to the desired number). The higher the quality factor of the output filter and/or the higher the ratio FCE/FLC, the lower the FZ1 frequency (to maximize phase boost at FLC). 1 C 1 = ----------------------------------------------2π ⋅ R 2 ⋅ 0.5 ⋅ F LC compensation gain at FP2 against the capabilities of the error amplifier. The closed loop gain, GCL, is constructed on the log-log graph of Figure 8 by adding the modulator gain, GMOD (in dB), to the feedback compensation gain, GFB (in dB). This is equivalent to multiplying the modulator transfer function and the compensation transfer function and then plotting the resulting gain. 3. Calculate C2 such that FP1 is placed at FCE. FZ1 FZ2 FP1 FP2 GAIN C1 C 2 = ------------------------------------------------------2π ⋅ R 2 ⋅ C 1 ⋅ F CE – 1 R2 20 log -------- R1 R1 R 3 = --------------------F SW ------------ – 1 F LC D V MAX ⋅ IN 20 log ---------------------------------V OSC 0 GFB GCL LOG 4. Calculate R3 such that FZ2 is placed at FLC. Calculate C3 such that FP2 is placed below FSW (typically, 0.3 to 1.0 times FSW). FSW represents the switching frequency of the regulator. Change the numerical factor (0.7) below to reflect desired placement of this pole. Placement of FP2 lower in frequency helps reduce the gain of the compensation network at high frequency, in turn reducing the HF ripple component at the COMP pin and minimizing resultant duty cycle jitter. MODULATOR GAIN COMPENSATION GAIN CLOSED LOOP GAIN OPEN LOOP E/A GAIN GMOD LOG FLC FCE F0 FREQUENCY FIGURE 8. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN 1 C 3 = ------------------------------------------------2π ⋅ R 3 ⋅ 0.7 ⋅ F SW It is recommended that a mathematical model be used to plot the loop response. Check the loop gain against the error amplifier’s open-loop gain. Verify phase margin results and adjust as necessary. The following equations describe the frequency response of the modulator (GMOD), feedback compensation (GFB) and closed-loop response (GCL): D MAX ⋅ V IN 1 + s ( f ) ⋅ ESR ⋅ C G MOD ( f ) = ------------------------------- ⋅ ----------------------------------------------------------------------------------------------------------2 V OSC 1 + s ( f ) ⋅ ( ESR + DCR ) ⋅ C + s ( f ) ⋅ L ⋅ C 1 + s ( f ) ⋅ R2 ⋅ C1 G FB ( f ) = ---------------------------------------------------- ⋅ s ( f ) ⋅ R1 ⋅ ( C1 + C2 ) Component Selection Guidelines 1 + s ( f ) ⋅ ( R1 + R3 ) ⋅ C3 ----------------------------------------------------------------------------------------------------------------------- C1 ⋅ C2 -------------------( 1 + s ( f ) ⋅ R3 ⋅ C3 ) ⋅ 1 + s ( f ) ⋅ R2 ⋅ C 1 + C 2 G CL ( f ) = G MOD ( f ) ⋅ G FB ( f ) where, s ( f ) = 2π ⋅ f ⋅ j COMPENSATION BREAK FREQUENCY EQUATIONS 1 F Z1 = ------------------------------2π ⋅ R 2 ⋅ C 1 1 F P1 = --------------------------------------------C1 ⋅ C2 2π ⋅ R 2 ⋅ --------------------C1 + C2 1 F Z2 = ------------------------------------------------2π ⋅ ( R 1 + R 3 ) ⋅ C 3 1 F P2 = ------------------------------2π ⋅ R 3 ⋅ C 3 Figure 8 shows an asymptotic plot of the DC/DC converter’s gain vs. frequency. The actual Modulator Gain has a high gain peak dependent on the quality factor (Q) of the output filter, which is not shown. Using the above guidelines should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the 10 A stable control loop has a gain crossing with close to a -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. The mathematical model presented makes a number of approximations and is generally not accurate at frequencies approaching or exceeding half the switching frequency. When designing compensation networks, select target crossover frequencies in the range of 10% to 30% of the switching frequency, FSW. Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance FN9257.0 February 13, 2006 ISL8104 components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor's ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: V IN - V OUT V OUT ∆I = -------------------------------- • ---------------Fs x L V IN ∆VOUT= ∆I x ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL8104 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: L O × I TRAN T RISE = ------------------------------V IN – V OUT L O × I TRAN T FALL = -----------------------------V OUT where: ITRAN is the transient load current step, TRISE is the response time to the application of load, and TFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the 11 application or removal of load and dependent upon the output voltage setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2. The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select a bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage, a voltage rating of 1.5 times greater is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The ISL8104 requires at least 2 N-Channel power MOSFETs. These should be selected based upon rDS(ON), gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. At a 300kHz switching frequency, the conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor (see the following equations). Only the upper MOSFET exhibits switching losses, since the schottky rectifier clamps the switching node before the synchronous rectifier turns on. These equations assume linear voltage-current transitions and do not adequately model power loss due the reverserecovery of the lower MOSFETs body diode. The gate-charge losses are dissipated by the ISL8104 and don't heat the MOSFETs. However, large gate-charge increases FN9257.0 February 13, 2006 ISL8104 PUPPER = IO2 x rDS(ON) x D + 1 Io x VIN x TSW x Fs 2 +14V DBOOT PLOWER = IO2 x rDS(ON) x (1 - D) + where: D is the duty cycle = VO / VIN, TSW is the switching interval, and Fs is the switching frequency. ISL8104 VD +1.2V TO +14V - BOOT CBOOT Q1 UGATE the switching interval, TSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. Standard-gate MOSFETs are normally recommended for use with the ISL8104. However, logic-level gate MOSFETs can be used under special circumstances. The input voltage, upper gate drive level, and the MOSFETs absolute gate-tosource voltage rating determine whether logic-level MOSFETs are appropriate. NOTE: VG-S ≈ VCC - VD PHASE PVCC +14V Q2 LGATE + D2 NOTE: VG-S ≈ PVCC PGND GND FIGURE 9. UPPER GATE DRIVE - BOOTSTRAP OPTION +14V Figure 9 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from +14V. The boot capacitor, CBOOT develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle to a voltage of +12V less the boot diode drop (VD) when the lower MOSFET, Q2 turns on. A MOSFET can only be used for Q1 if the MOSFETs absolute gate-to-source voltage rating exceeds the maximum voltage applied to +14V. For Q2, a logic-level MOSFET can be used if its absolute gate-to-source voltage rating also exceeds the maximum voltage applied to +14V. Figure 10 shows the upper gate drive supplied by a direct connection to +14V. This option should only be used in converter systems where the main input voltage is +5VDC or less. The peak upper gate-to-source voltage is approximately +14V less the input supply. For +5V main power and +14VDC for the bias, the gate-to-source voltage of Q1 is 9V. A logic-level MOSFET is a good choice for Q1 and a logic-level MOSFET can be used for Q2 if its absolute gate-to-source voltage rating exceeds the maximum voltage applied to PVCC. This method reduces the number of required external components, but does not provide for immunity to phase node ringing during turn on and may result in lower system efficiency. 12 +5V OR LESS ISL8104 BOOT Q1 UGATE PVCC + LGATE PGND NOTE: VG-S ≈ VCC - 5V +14V Q2 D2 NOTE: VG-S ≈ PVCC GND FIGURE 10. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION Schottky Selection Rectifier D2 is a clamp that catches the negative inductor swing during the dead time between turning off the lower MOSFET and turning on the upper MOSFET. The diode must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is acceptable to omit the diode and let the body diode of the lower MOSFET clamp the negative inductor swing, but efficiency could slightly decrease as a result. The diode's rated reverse breakdown voltage must be greater than the maximum input voltage. FN9257.0 February 13, 2006 ISL8104 Small Outline Plastic Packages (SOIC) M14.15 (JEDEC MS-012-AB ISSUE C) N INDEX AREA H 0.25(0.010) M 14 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE B M E INCHES -B- 1 2 3 L SEATING PLANE -A- h x 45o A D -C- α e A1 B 0.25(0.010) M C A M SYMBOL MIN MAX MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - A1 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.3367 0.3444 8.55 8.75 3 E 0.1497 0.1574 3.80 4.00 4 e C 0.10(0.004) B S 0.050 BSC 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. 1.27 BSC - H 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 N NOTES: MILLIMETERS α 14 0o 14 8o 0o 7 8o Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. 13 FN9257.0 February 13, 2006 ISL8104 Quad Flat No-Lead Plastic Package (QFN) Micro Lead Frame Plastic Package (MLFP) L16.4x4 16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE (COMPLIANT TO JEDEC MO-220-VGGC ISSUE C) MILLIMETERS SYMBOL MIN NOMINAL MAX NOTES A 0.80 0.90 1.00 - A1 - - 0.05 - A2 - - 1.00 A3 b 0.23 D 0.28 9 0.35 5, 8 4.00 BSC D1 D2 9 0.20 REF - 3.75 BSC 1.95 2.10 9 2.25 7, 8 E 4.00 BSC - E1 3.75 BSC 9 E2 1.95 e 2.10 2.25 7, 8 0.65 BSC - k 0.25 - - - L 0.50 0.60 0.75 8 L1 - - 0.15 10 N 16 2 Nd 4 3 Ne 4 3 P - - 0.60 9 θ - - 12 9 Rev. 5 5/04 NOTES: 1. Dimensioning and tolerancing conform to ASME Y14.5-1994. 2. N is the number of terminals. 3. Nd and Ne refer to the number of terminals on each D and E. 4. All dimensions are in millimeters. Angles are in degrees. 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 7. Dimensions D2 and E2 are for the exposed pads which provide improved electrical and thermal performance. 8. Nominal dimensions are provided to assist with PCB Land Pattern Design efforts, see Intersil Technical Brief TB389. 9. Features and dimensions A2, A3, D1, E1, P & θ are present when Anvil singulation method is used and not present for saw singulation. 10. Depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (L1) maybe present. L minus L1 to be equal to or greater than 0.3mm. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 14 FN9257.0 February 13, 2006