MIC26950 12A Hyper Speed ControlTM Synchronous DC-DC Buck Regulator SuperSwitcher IITM General Description Features The Micrel MIC26950 is a constant-frequency, synchronous buck regulator featuring a unique digitally modified adaptive ON-time control architecture. The MIC26950 operates over an input supply range of 4.5V to 26V and provides a regulated output at up to 12A of output current. The output voltage is adjustable down to 0.8V with a typical accuracy of ±1%, and the device operates at a switching frequency of 300kHz. Micrel’s Hyper Speed ControlTM architecture allows for ultrafast transient response while reducing the output capacitance and also makes (High VIN)/(Low VOUT) operation possible. This digitally modified adaptive tON ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. The MIC26950 offers a full suite of protection features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, fold-back current limit, “hiccup” mode shortcircuit protection and thermal shutdown. All support documentation can be found on Micrel’s web site at: www.micrel.com. • • • • • • • • • • • • • • Hyper Speed ControlTM architecture enables - High delta V operation (VIN = 26V and VOUT = 0.8V) - Small output capacitance 4.5V to 26V input voltage Adjustable output from 0.8V to 5.5V (±1% accuracy) Any CapacitorTM Stable - Zero-ESR to high-ESR output capacitance 12A output current capability 300kHz switching frequency Internal compensation Up to 95% efficiency 6ms internal soft-start Foldback current-limit and “hiccup” mode short-circuit protection Thermal shutdown Supports safe start-up into a pre-biased load –40°C to +125°C junction temperature range 28-pin 5mm X 6mm MLF® package Applications • Distributed power systems • Communications/networking infrastructure • Set-top box, gateways and routers • Printers, scanners, graphic cards and video cards ____________________________________________________________________________________________________________ Typical Application Efficiency (VIN = 12V) vs. Output Current 100 EFFICIENCY (%) 95 5.0V 3.3V 90 2.5V 85 1.5V 80 1.0V 75 70 65 60 0 3 6 9 12 15 OUTPUT CURRENT (A) Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc. MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com July 2011 M9999-070111-C Micrel, Inc. MIC26950 Ordering Information Part Number Voltage Switching Frequency Junction Temperature Range Package Lead Finish MIC26950YJL Adjustable 300kHz –40°C to +125°C 28-pin 5mm × 6mm MLF® Pb-Free Pin Configuration 28-Pin 5mm X 6mm MLF® (YJL) Pin Description Pin Number Pin Name 13,14,15, 16,17,18, PVIN 19 Pin Function High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from 4.5V to 26V. Input capacitors between PVIN and the power ground (PGND) are required. 24 EN Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating = enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically 0.8mA). 25 FB Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 26 SGND Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. 27 VDD VDD Bias (Input): Power to the internal reference and control sections of the MIC26950. The VDD operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin-to-PGND is recommended for clean operation. PGND Power Ground. PGND is the ground path for the MIC26950 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (SGND) loop. 2, 5, 6, 7, 8, 21 22 July 2011 CS Current Sense (Input): High current output driver return. The CS pin connects directly to the switch node. Due to the high speed switching on this pin, the CS pin should be routed away from sensitive nodes. CS pin also senses the current by monitoring the voltage across the low-side internal MOSFET during OFF-time. 2 M9999-070111-C Micrel, Inc. MIC26950 Pin Description (Continued) Pin Number Pin Name 20 BST Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected between the BST pin and the SW pin. 4, 9, 10, 11, 12 SW Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET drain. 23 VIN Power Supply Voltage (Input): Requires bypass capacitor to SGND. 1, 3, 28 NC No Connect. July 2011 Pin Function 3 M9999-070111-C Micrel, Inc. MIC26950 Absolute Maximum Ratings(1,2) Operating Ratings(3) PVIN to PGND................................................ −0.3V to +28V VIN to PGND ....................................................−0.3V to PVIN VDD to PGND ................................................... −0.3V to +6V VSW, VCS to PGND .............................. −0.3V to (PVIN +0.3V) VBST to VSW ........................................................ −0.3V to 6V VBST to PGND .................................................. −0.3V to 34V VEN to PGND ...................................... −0.3V to (VDD + 0.3V) VFB to PGND....................................... −0.3V to (VDD + 0.3V) PGND to SGND ........................................... −0.3V to +0.3V Junction Temperature .............................................. +150°C Storage Temperature (TS).........................−65°C to +150°C Lead Temperature (soldering, 10sec)........................ 260°C Supply Voltage (PVIN, VIN)............................... 4.5V to 26V Output Voltage Range (VOUT)........................... 0.8V to 5.5V Bias Voltage (VDD)............................................ 4.5V to 5.5V Enable Input (VEN) ................................................. 0V to VDD Junction Temperature (TJ) ........................ −40°C to +125°C Maximum Power Dissipation......................................Note 4 Package Thermal Resistance(4) 5mm x 6mm MLF®(θJA) .....................................36°C/W Electrical Characteristics(5) PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units 26 V Power Supply Input Input Voltage Range (VIN, PVIN) 4.5 VDD Bias Voltage Operating Bias Voltage (VDD) Undervoltage Lockout Trip Level VDD Rising 4.5 5 5.5 V 2.4 2.7 3.2 V UVLO Hysteresis Quiescent Supply Current Shutdown Supply Current 50 mV VFB = 1.5V 1.4 3 VDD = VBST = 5.5V, VIN = 26V 0.7 2 SW = unconnected, VEN = 0V mA mA Reference Feedback Reference Voltage 0°C ≤TJ ≤ 85°C (±1.0%) 0.792 0.8 0.808 −40°C ≤TJ ≤ 125°C (±1.5%) 0.788 0.8 0.812 V Load Regulation IOUT = 0A to 12A 0.2 % Line Regulation VIN = (VOUT + 3.0V) to 26V 0.1 % FB Bias Current VFB = 0.8V 5 nA 0.85 V Enable Control EN Logic Level High 4.5V < VDD < 5.5V EN Logic Level Low 4.5V < VDD < 5.5V EN Bias Current VEN = 0V 1.2 0.78 50 0.4 V µA Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF. 3. The device is not guaranteed to function outside operating range. 4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.” 5. Specification for packaged product only. July 2011 4 M9999-070111-C Micrel, Inc. MIC26950 Electrical Characteristics(5) PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units 225 300 375 kHz Oscillator Switching Frequency (6) Maximum Duty Cycle (7) Minimum Duty Cycle VFB = 0V 87 % VFB > 0.8V 0 % 360 ns 6 ms 27 A Minimum Off-Time Soft-Start Soft-Start Time Short-Circuit Protection Current-Limit Threshold VFB = 0.8V Short-Circuit Current VFB = 0V 8 A Top-MOSFET RDS (ON) ISW = 1A 17 mΩ Bottom-MOSFET RDS (ON) ISW = 1A 6 mΩ SW Leakage Current PVIN = 26V, VSW = 26V, VEN = 0V, VBST = 31.5 V 60 µA VIN Leakage Current PVIN = 26V, VSW = 0V, VEN = 0V, VBST = 31.5V 25 µA 13.2 Internal FETs Thermal Protection Over-temperature Shutdown TJ Rising Over-temperature Shutdown Hysteresis 155 °C 10 °C Notes: 6. Measured in test mode. 7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns. July 2011 5 M9999-070111-C Micrel, Inc. MIC26950 Typical Characteristics VIN Shutdown Current vs. Input Voltage VIN Operating Supply Current vs. Input Voltage 8.0 6.0 4.0 VOUT = 1.2V VDD = 5V SWITCHING IOUT = 0A 2.0 16 12 8 V DD = 5V 4 V EN = 0V 4 10 16 22 8 V OUT = 1.2V 4 10 16 22 28 4 10 16 22 INPUT VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V) Feedback Voltage vs. Input Voltage Total Regulation vs. Input Voltage Current Limit vs. Input Voltage 0.804 0.802 0.800 0.798 0.796 VOUT = 1.2V VDD = 5V 0.794 25 0.8% CURRENT LIMIT (A) TOTAL REGULATION (%) 0.806 0.6% 0.4% V OUT = 1.2V 0.2% V DD = 5V 16 22 15 10 5 VOUT = 1.2V VDD = 5V 0.0% 10 20 IOUT = 0A to 12A IOUT = 0A 0.792 28 INPUT VOLTAGE (V) 28 30 1.0% 0.808 4 12 0 4 28 16 V DD = 5V SWITCHING 0 0.0 FEEDBACK VOLTAGE (V) 20 SUPPLY CURRENT (mA) 20 SHUTDOWN CURRENT (µA) SUPPLY CURRENT (mA) 10.0 VDD Operating Supply Current vs. Input Voltage 0 4 10 16 22 INPUT VOLTAGE (V) 28 4 8 12 16 20 24 28 INPUT VOLTAGE (V) Switching Frequency vs. Input Voltage SWITCHING FREQUENCY (kHz) 390 VOUT = 1.2V VDD = 5V 345 IOUT = 0A 300 255 210 4 10 16 22 28 INPUT VOLTAGE (V) July 2011 6 M9999-070111-C Micrel, Inc. MIC26950 Typical Characteristics (Continued) VDD Shutdown Current vs. Temperature VDD Operating Supply Current vs. Temperature 10.0 1 2.8 0.8 2.7 VDD UVLO Threshold vs. Temperature 6.0 4.0 VIN = 12V VOUT = 1.2V VDD = 5V 2.0 IOUT = 0A SWITCHING 0.0 0.6 0.4 VIN = 12V IOUT = 0A VDD = 5V 0.2 -20 10 40 70 100 130 -50 -20 TEMPERATURE (°C) VIN Operating Supply Current vs. Temperature 40 70 100 6.0 VIN = 12V 4.0 VOUT = 1.2V VDD = 5V IOUT = 0A SWITCHING 2.0 -20 10 40 70 100 -50 6.0 4.0 VIN = 12V VDD = 5V 2.0 IOUT = 0A -20 10 40 70 100 130 0.798 0.796 0.794 0.792 -20 10 40 70 100 130 15 VIN = 12V 10 VOUT = 1.2V VDD = 5V -20 10 40 70 Line Regulation vs. Temperature 0.5% VIN = 12V 0.8% VOUT = 1.2V VDD = 5V IOUT = 0A to 12A 0.6% 0.4% 0.2% 130 20 TEMPERATURE (°C) 0.0% -50 100 25 -50 LINE REGULATION (%) IOUT = 0A 0.800 130 0 -50 LOAD REGULATION (%) 0.802 100 30 5 1.0% VOUT = 1.2V 70 35 Load Regulation vs. Temperature VDD = 5V 40 40 Feedback Voltage vs. Temperature 0.804 10 Current Limit vs. Temperature TEMPERATURE (°C) VIN = 12V -20 VIN Shutdown Current vs. Temperature TEMPERATURE (°C) 0.806 VIN = 12V TEMPERATURE (°C) 8.0 130 0.808 2.5 TEMPERATURE (°C) 0.0 -50 Falling 2.3 130 CURRENT LIMIT (A) 8.0 0.0 FEEDBACK VOLTAGE (V) 10 10.0 SUPPLY CURRENT (µA) SUPPLY CURRENT (mA) 10.0 2.6 2.4 VEN = 0V 0 -50 VIN =6V to 26V 0.4% VOUT = 1.2V VDD = 5V 0.3% 0.2% 0.1% 0.0% -50 -20 TEMPERATURE (°C) 10 40 70 100 130 TEMPERATURE (°C) -50 -20 10 40 70 100 130 TEMPERATURE (°C) EN Bias Current vs. Temperature Switching Frequency vs. Temperature 100 345 VIN = 12V 330 EN BIAS CURRENT (µA) SWITCHING FREQUENCY (kHz) VDD THRESHOLD (V) SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) Rising 8.0 VOUT = 1.2V VDD = 5V 315 IOUT = 0A 300 285 270 80 60 40 VIN = 12V 20 VOUT = 1.2V VDD = 5V 255 0 -50 -20 10 40 70 TEMPERATURE (°C) July 2011 100 130 -50 -20 10 40 70 100 130 TEMPERATURE (°C) 7 M9999-070111-C Micrel, Inc. MIC26950 Typical Characteristics (Continued) 12VIN 85 18V IN 24VIN 80 75 70 VOUT = 1.2V 65 0 3 6 0.806 0.804 0.802 0.800 0.798 0.796 VIN = 12V VOUT = 1.2V 0.794 VDD = 5V 60 9 VDD = 5V 0 3 OUTPUT CURRENT (A) V OUT = 1.2V V DD = 5V 0.3% 0.2% 0.1% 0.0% 3 6 9 300 V IN = 12V 270 V OUT = 1.2V V DD = 5V 240 3 6 9 4.4 4.2 3.8 0 40 VIN = 12V 20 VOUT = 1.2V 60 40 3 6 9 EFFICIENCY (%) 1.2V 1.0V 0.9V 0.8V 90 85 9 12 OUTPUT CURRENT (A) July 2011 15 9 12 95 1.2V 1.0V 0.9V 0.8V 80 6 Efficiency (VIN = 24) vs. Output Current 70 6 3 OUTPUT CURRENT (A) 5.0V 3.3V 2.5V 1.8V 1.5V 75 70 3 VOUT = 1.2V 0 12 95 0 VIN = 24V 20 VDD = 5V Efficiency (VIN = 12V) vs. Output Current 100 3.3V 2.5V 1.8V 1.5V 15 80 OUTPUT CURRENT (A) 95 12 0 0 Efficiency (VIN = VDD = 5V) vs. Output Current 75 9 Die Temperature* (VIN = 24V) vs. Output Current 60 12 80 6 V DD = 5V 9 85 3 OUTPUT CURRENT (A) 80 OUTPUT CURRENT (A) 90 TA 25ºC 85ºC 125ºC 4 0 100 VFB < 0.8V 100 VDD = 5V 6 VIN = 5V 3.4 12 DIE TEMPERATURE (°C) VIN = 5V VOUT = 1.2V 12 3.6 210 0 DIE TEMPERATURE (°C) 40 9 VDD = 5V Die Temperature* (VIN = 12V) vs. Output Current 60 6 4.6 OUTPUT CURRENT (A) 80 3 3 Output Voltage (VIN = 5V) vs. Output Current 5 330 12 0 EFFICIENCY (%) 0 100 0 V DD = 5V 4.8 Die Temperature* (VIN = 5V) vs. Output Current 20 V IN = 12V V OUT = 1.2V -0.4% OUTPUT CURRENT (A) 360 OUTPUT CURRENT (A) 100 -0.3% 12 OUTPUT VOLTAGE (V) SWITCHING FREQUENCY (kHz) LINE REGULATION (%) 9 390 V IN = 6V to 26V 0 -0.2% Switching Frequency vs. Output Current 0.5% 0.4% -0.1% OUTPUT CURRENT (A) Line Regulation vs. Output Current DIE TEMPERATURE (°C) 6 0.0% -0.5% 0.792 12 Feedback Voltage (%) vs. Output Current 0.1% 5.0V 3.3V 2.5V 90 EFFICIENCY (%) EFFICIENCY (%) FEEDBACK VOLTAGE (V) 6VIN 90 Feedback Voltage vs. Output Current 0.808 FEEDBACK VOLTAGE (%) Efficiency vs. Output Current 95 85 1.8V 1.5V 80 1.2V 1.0V 0.9V 0.8V 75 70 0 3 6 9 12 OUTPUT CURRENT (A) 8 15 0 3 6 9 12 15 OUTPUT CURRENT (A) M9999-070111-C Micrel, Inc. OUTPUT CURRENT (A) 16 0.8V 12 1.5V 8 4 Thermal Derating* vs. Ambient Temperature 20 VIN = 5V 16 1.8V 12 3.3V 8 4 VIN = 5V VOUT = 0.8, 1.2, 1.5V -25 0 25 50 75 100 125 -50 OUTPUT CURRENT (A) OUTPUT CURRENT (A) 2.5V 12 4 5 5V VIN = 12V VIN = 12V 0 25 50 75 100 125 -50 -25 0 25 50 75 100 125 AMBIENT TEMPERATURE (°C) Thermal Derating* vs. Ambient Temperature 20 16 8 -25 AMBIENT TEMPERATURE (°C) Thermal Derating* vs. Ambient Temperature 1.8V 10 0 AMBIENT TEMPERATURE (°C) 20 0.8V 15 VOUT = 0.8, 1.2, 1.8V 0 -50 20 VOUT = 1.8, 2.5, 3.3V 0 Thermal Derating* vs. Ambient Temperature 25 OUTPUT CURRENT (A) Thermal Derating* vs. Ambient Temperature 20 OUTPUT CURRENT (A) MIC26950 16 0.8V 12 2.5V 8 4 VOUT = 2.5, 3.3, 5V VIN = 24V VOUT = 0.8, 1.2, 2.5V 0 0 -50 -25 0 25 50 75 100 AMBIENT TEMPERATURE (°C) 125 -50 -25 0 25 50 75 100 125 AMBIENT TEMPERATURE (°C) Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26950 case mounted on a 5 square inch 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. July 2011 9 M9999-070111-C Micrel, Inc. MIC26950 Functional Characteristics July 2011 10 M9999-070111-C Micrel, Inc. MIC26950 Functional Characteristics (Continued) July 2011 11 M9999-070111-C Micrel, Inc. MIC26950 Functional Characteristics (Continued) July 2011 12 M9999-070111-C Micrel, Inc. MIC26950 Functional Diagram Figure 1. MIC26950 Block Diagram July 2011 13 M9999-070111-C Micrel, Inc. MIC26950 It is not recommended to use MIC26950 with a OFF-time close to tOFF(min) during steady-state operation. Also, as VOUT increases, the internal ripple injection will increase and reduce the line regulation performance. Therefore, the maximum output voltage of the MIC26950 should be limited to 5.5V. Please refer to “Setting Output Voltage” subsection in “Application Information” for more details. The actual ON-time and resulting switching frequency will vary with the part-to-part variation in the rise and fall times of the internal MOSFETs, the output load current, and variations in the VDD voltage. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications, such as 26V to 1.0V. The minimum tON measured on the MIC26950 evaluation board is about 184ns. During load transients, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop operation, will be analyzed both the steady-state and load transient scenarios. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the feedback voltage. Figure 2 shows the MIC26950 control loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. Functional Description The MIC26950 is an adaptive ON-time synchronous step-down DC-DC regulator. It is designed to operate over a wide input voltage range from 4.5V to 26V and provides a regulated output voltage at up to 12A of output current. A digitally modified adaptive ON-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. Over-current protection is implemented without the use of an external sense resistor. The device includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. Theory of Operation Figure 1 illustrates the block diagram for the control loop of the MIC26950. The output voltage is sensed by the MIC26950 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low gain transconductance (gm) amplifier. If the feedback voltage decreases and the output of the gm amplifier is below 0.8V, then the error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the “FIXED tON ESTIMATION” circuitry: t ON(estimated) = VOUT VIN × 300kHz (1) where VOUT is the output voltage and VIN is the power stage input voltage. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 360ns, the MIC26950 control logic will apply the tOFF(min) instead. tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. The maximum duty cycle is obtained from the 360ns tOFF(min): D max = t S − t OFF(min) tS = 1− 360ns tS (2) Figure 2. MIC26950 Control Loop Timing where tS = 1/300kHz = 3.33μs. July 2011 14 M9999-070111-C Micrel, Inc. MIC26950 Figure 3 shows the operation of the MIC26950 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This will cause the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time tOFF(min) is generated to charge CBST since the feedback voltage is still below VREF. Then, the next ON-time period is triggered due to the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC26950 converter. Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC26950 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with a 9.7mV step. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function behavior correctly. Current Limit The MIC26950 uses the RDS(ON) of the internal low-side power MOSFET to sense over-current conditions. This method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. In each switching cycle of the MIC26950 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. If the inductor current is greater than 27A, then the MIC26950 turns off the highside MOSFET and a soft-start sequence is triggered. This mode of operation is called “hiccup mode” and its purpose is to protect the downstream load in case of a hard short. The current limit threshold has a fold back characteristic related to the feedback voltage. As shown in Figure 4. Figure 3. MIC26950 Load Transient Response Unlike true current-mode control, the MIC26950 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. The MIC26950 control loop has the advantage of eliminating the need for slope compensation. In order to meet the stability requirements, the MIC26950 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV. If a low ESR output capacitor is selected, the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to “Ripple Injection” subsection in “Application Information” for more details about the ripple injection technique. July 2011 Current Limit Threshold vs. Feedback Voltage CURRENT LIMIT THRESHOLD (A) 30.0 25.0 20.0 15.0 10.0 5.0 0.0 0.0 0.2 0.4 0.6 0.8 1.0 FEEDBACK VOLTAGE (V) Figure 4. MIC26950 Current Limiting Circuit 15 M9999-070111-C Micrel, Inc. MIC26950 MOSFET Gate Drive The Block Diagram of Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. ΔBST = 10mA x 3.33μs/0.1μF = 333mV. July 2011 When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the VDD supply voltage. The nominal low-side gate drive voltage is VDD and the nominal high-side gate drive voltage is approximately VDD – VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. 16 M9999-070111-C Micrel, Inc. MIC26950 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC26950 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by Equation 8: Application Information Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the Equation 4: L= 2 PINDUCTOR(Cu) = IL(RMS) × RWINDING VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × 20% × IOUT(max) (4) The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. where: fSW = switching frequency, 300kHz 20% = ratio of AC ripple current to DC output current VIN(max) = maximum power stage input voltage The peak-to-peak inductor current ripple is: ΔIL(pp) = VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × L PWINDING(Ht) = RWINDING(20°C) × 1 + 0.0042 × (TH – T20°C)) (5) Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. (6) The RMS inductor current is used to calculate the I2R losses in the inductor. 2 IL(RMS) = IOUT(max) + July 2011 ΔIL(PP) 12 (9) where: TH = temperature of wire under full load T20°C = ambient temperature RWINDING(20°C) = room temperature winding resistance (usually specified by the manufacturer) The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) =IOUT(max) + 0.5 × ΔIL(pp) (8) 2 (7) 17 M9999-070111-C Micrel, Inc. MIC26950 The maximum value of ESR is calculated: ESR COUT ≤ ΔVOUT(pp) Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend upon the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: (10) ΔIL(PP) where: ΔVOUT(pp) = peak-to-peak output voltage ripple ΔIL(PP) = peak-to-peak inductor current ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated in Equation 11: 2 ΔVOUT(pp) ΔVIN = IL(pk) × ESRCIN ΔIL(PP) ⎞ ⎛ 2 ⎟ + ΔIL(PP) × ESR C = ⎜⎜ OUT ⎟ C f 8 × × OUT SW ⎠ ⎝ (11) ( ) The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: where: D = duty cycle COUT = output capacitance value fSW = switching frequency As described in the “Theory of Operation” subsection in “Functional Description”, the MIC26950 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the “Ripple Injection” subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 12: ICOUT (RMS) = ΔIL(PP) ICIN(RMS) ≈ IOUT(max) × D × (1 − D) (15) The power dissipated in the input capacitor is: PDISS(CIN) = ICIN(RMS)2 × ESRCIN (16) Ripple Injection The VFB ripple required for proper operation of the MIC26950 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as a 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator can’t sense it, the MIC26950 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1) Enough ripple at the feedback voltage due to the large ESR of the output capacitors. (12) 12 (14) The power dissipated in the output capacitor is: 2 PDISS(COUT ) = ICOUT (RMS) × ESR COUT July 2011 (13) 18 M9999-070111-C Micrel, Inc. MIC26950 As shown in Figure 5a, the converter is stable without any ripple injection. The feedback voltage ripple is: ΔVFB(pp) = R2 × ESR COUT × ΔIL (pp) R1 + R2 (17) where ΔIL(pp) is the peak-to-peak value of the inductor current ripple. 2) Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 5b. The typical Cff value is between 1nF and 100nF. With the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: ΔVFB(pp) ≈ ESR × ΔIL (pp) Figure 5c. Invisible Ripple at FB In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 5c. The injected ripple is: (18) ΔVFB(pp) = VIN × K div × D × (1 - D) × 1 fSW × τ (19) 3) Virtually no ripple at the FB pin voltage due to the very low ESR of the output capacitors. K div = R1//R2 R inj + R1//R2 where VIN = Power stage input voltage D = duty cycle fSW = switching frequency τ = (R1//R2//Rinj) × Cff In equations (19) and (20), it is assumed that the time constant associated with Cff must be much greater than the switching period: Figure 5a. Enough Ripple at FB 1 T = << 1 fSW × τ τ (21) If the voltage divider resistors R1 and R2 are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. Figure 5b. Inadequate Ripple at FB July 2011 (20) 19 M9999-070111-C Micrel, Inc. MIC26950 Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 22: K div = ΔVFB(pp) VIN × fSW × τ D × (1 − D) In addition to the external ripple injection added at the FB pin, internal ripple injection is added at the inverting input of the comparator inside the MIC26950, as shown in Figure 7. The inverting input voltage VINJ is clamped to 1.2V. As VOUT is increased, the swing of VINJ will be clamped. The clamped VINJ reduces the line regulation because it is reflected back as a DC error on the FB terminal. Therefore, the maximum output voltage of the MIC26950 should be limited to 5.5V to avoid this line regulation problem. (22) Then the value of Rinj is obtained as: R inj = (R1//R2) × ( 1 K div − 1) (23) Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. Setting Output Voltage The MIC26950 requires two resistors to set the output voltage as shown in Figure 6. Figure 7. Internal Ripple Injection Thermal Measurements Measuring the IC’s case temperature is recommended to ensure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. Figure 6. Voltage-Divider Configuration The output voltage is determined by the equation: VOUT = VFB × (1 + R1 ) R2 (24) where, VFB = 0.8V. A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 = July 2011 VFB × R1 VOUT − VFB (25) 20 M9999-070111-C Micrel, Inc. MIC26950 Inductor PCB Layout Guidelines Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC26950 converter. IC • The signal ground pin (SGND) must be connected directly to the ground planes. Do not route the SGND pin to the PGND Pad on the top layer. • Place the IC close to the point-of-load (POL). • Use fat traces to route the input and output power lines. • Signal and power grounds should be kept separate and connected at only one location. Place the input capacitor next. • Place the input capacitors on the same side of the board and as close to the IC as possible. • Keep both the VIN Pin and PGND connections short. • Place several vias to the ground plane close to the input capacitor ground terminal. • Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. • Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. • If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. • In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. July 2011 • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (SW) away from the feedback (FB) pin. • The CS pin should be connected directly to the SW pin to accurate sense the voltage across the lowside MOSFET. To minimize noise, place a ground plane underneath the inductor. Output Capacitor • Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. • Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. RC Snubber • Place the RC snubber on the same side of the board and as close to the SW pin as possible. Input Capacitor • Keep the inductor connection to the switch node (SW) short. • The 2.2µF ceramic capacitor, which is connected to the VDD terminal, must be located right at the IC. The VDD terminal is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the VDD and PGND pins. • • 21 M9999-070111-C Micrel, Inc. MIC26950 Evaluation Board Schematic Figure 8. Schematic of MIC26950 Evaluation Board (J13, R13, R15 are for testing purposes) July 2011 22 M9999-070111-C Micrel, Inc. MIC26950 Bill of Materials Item C1 Part Number B41125A7227M C2, C3 C4, C5 C6, C7, C10 C8,C9 C11 C12 C13 EPCOS AVX Murata(3) Murata(3) GRM188R71H104KA93D TDK 0805ZC225MAT2A AVX(2) Murata(3) GRM21BR71A225KA01L TDK 06035C102KAT2A AVX(2) Murata(3) GRM188R71H102KA01D 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 3 2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 2 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1 560µF OSCON Capacitor, 6.3V 1 Small Signal Schottky Diode 1 5.6V Zener Diode 1 2.2µH Inductor, 15A Saturation Current 1 (4) C1608X7R1H102K TDK 06035C223KAZ2A AVX(2) GRM188R71H223K Murata(3) (4) C1608X7R1H223K TDK 12106D107MAT2A AVX(2) Murata(3) GRM32ER60J107ME20L CMDZ5L6 HCF1305-2R2-R FCX619 CRCW06034R75FKEA R3, R4 2 (4) C2012X7R1A225K SANYO(5) Diodes Inc CRCW08051R21FKEA CRCW060310K0FKEA (6) Vishay(7) SD103AWS R2, R16 4.7µF Ceramic Capacitor, X7R, Size 1210, 50V (4) C1608X7R1H104K SD103AWS-7 R1 1 AVX(2) 06035C104KAT2A 6SEPC560MX Q1 220µF Aluminum Capacitor, SMD, 35V Open C15 L1 Qty. (2) GRM32ER71H475KA88L Open D2 Description (1) 12105C475KAZ2A C14 D1 Manufacturer Central Semi(8) (9) Cooper Bussmann (6) ZETEX 50V NPN Transistor 1 (7) 4.75Ω Resistor, Size 0603, 1% 1 (7) 1.21Ω Resistor, Size 0805, 1% 2 (7) 10kΩ Resistor, Size 0603, 1% 2 Vishay Dale Vishay Dale Vishay Dale Notes: 1. EPCOS: www.epcos.com. 2. AVX: www.avx.com. 3. Murata: www.murata.com. 4. TDK: www.tdk.com. 5. SANYO: www.sanyo.com. 6. Diode Inc.: www.diodes.com. 7. Vishay: www.vishay.com. 8. Central Semi: www.centralsemi.com. 9. Cooper Bussmann: www.cooperbussmann.com. 10. Micrel, Inc.: www.micrel.com. July 2011 23 M9999-070111-C Micrel, Inc. MIC26950 Bill of Materials (Continued) Item R5 Part Number CRCW060380K6FKEA Manufacturer Description Qty. (7) 80.6kΩ Resistor, Size 0603, 1% 1 (7) Vishay Dale R6 CRCW060340K2FKEA Vishay Dale 40.2kΩ Resistor, Size 0603, 1% 1 R7 CRCW060320K0FKEA Vishay Dale(7) 20kΩ Resistor, Size 0603, 1% 1 CRCW060311K5FKEA (7) 11.5kΩ Resistor, Size 0603, 1% 1 (7) 8.06kΩ Resistor, Size 0603, 1% 1 (7) 4.75kΩ Resistor, Size 0603, 1% 1 (7) 3.24kΩ Resistor, Size 0603, 1% 1 (7) 1.91kΩ Resistor, Size 0603, 1% 1 (7) 0Ω Resistor, Size 0603, 5% 1 (7) 5.23kΩ Resistor, Size 0603, 1% 1 (7) 49.9Ω Resistor, Size 0603, 1% 1 26V/12A Synchronous Buck DC-DC Regulator 1 R8 R9 R10 R11 R12 R13 R14 R15 U1 July 2011 CRCW06038K06FKEA CRCW06034K75FKEA CRCW06033K24FKEA CRCW06031K91FKEA CRCW06030000FKEA CRCW06035K23FKEA CRCW060349R9FKEA MIC26950YJL Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc. (10) 24 M9999-070111-C Micrel, Inc. MIC26950 PCB Layout Figure 9. MIC26950 Evaluation Board Top Layer Figure 10. MIC26950 Evaluation Board Mid-Layer 1 (Ground Plane) July 2011 25 M9999-070111-C Micrel, Inc. MIC26950 PCB Layout (Continued) Figure 11. MIC26950 Evaluation Board Mid-Layer 2 Figure 12. MIC26950 Evaluation Board Bottom Layer July 2011 26 M9999-070111-C Micrel, Inc. MIC26950 Recommended Land Pattern July 2011 27 M9999-070111-C Micrel, Inc. MIC26950 Package Information 28-Pin 5mm x 6mm MLF® (YJL) MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2010 Micrel, Incorporated. July 2011 28 M9999-070111-C