MIC2176-1/-2/-3 Wide Input Voltage, Synchronous Buck Controllers Featuring Adaptive On-Time Control Hyper Speed Control™ Family General Description Features The Micrel MIC2176-1/-2/-3 is a family of constant-frequency, synchronous buck controllers featuring a unique digitally modified adaptive ON-time control architecture. The MIC2176 family operates over an input supply range of 4.5V to 75V and can be used to supply up to 15A of output current. The output voltage is adjustable down to 0.8V with a guaranteed accuracy of ±1%, and the device operates at a constant switching frequency of 100kHz, 200kHz, and 300kHz. Micrel’s Hyper Speed ControlTM architecture allows for ultrafast transient response while reducing the output capacitance and also makes (High VIN)/(Low VOUT) operation possible. This digitally modified adaptive tON ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. The MIC2176 offers a full suite of protection features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, fold-back current limit, “hiccup” mode shortcircuit protection and thermal shutdown. All support documentation can be found on Micrel’s web site at: www.micrel.com. • • • • • • • • • • • • Hyper Speed ControlTM architecture enables - High delta V operation (VIN = 75V and VOUT = 1.2V) - Small output capacitance 4.5V to 75V input voltage Output down to 0.8V with ±1% accuracy Any CapacitorTM Stable - Zero-ESR to high-ESR output capacitance 100kHz/200kHz/300kHz switching frequency Internal compensation 6ms Internal soft-start Foldback current limit and “hiccup” mode short-circuit protection Thermal shutdown Supports safe start-up into a pre-biased output –40°C to +125°C junction temperature range Available in 10-pin MSOP package Applications • Distributed power systems • Networking/Telecom Infrastructure • Printers, scanners, graphic cards and video cards ___________________________________________________________________________________________________________ Typical Application Efficiency vs. Output Current 95 EFFICIENCY (%) 90 85 80 28VIN 48VIN 75 70 60VIN 65 60 MIC2176-2 VOUT = 3.3V 55 50 VDD = 5V LINEAR 45 40 0 1 2 3 4 5 OUTPUT CURRENT (A) MIC2176-2 Adjustable Output 200KHz Buck Converter Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc. MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com November 2010 M9999-111710-A Micrel, Inc. MIC2176 Ordering Information Output Voltage Switching Frequency Junction Temperature Range Package Lead Finish MIC2176-1YMM Adjustable 100kHz –40°C to +125°C 10-pin MSOP Pb-Free MIC2176-2YMM Adjustable 200kHz –40°C to +125°C 10-pin MSOP Pb-Free MIC2176-3YMM Adjustable 300kHz –40°C to +125°C 10-pin MSOP Pb-Free Part Number Pin Configuration 10-Pin MSOP (MM) Pin Description Pin Number Pin Name 1 HSD 2 EN Enable (Input): A logic level control of the output. The EN pin is CMOS compatible. Logic high or floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is reduced (typically 0.7mA). Do not connect the EN pin to the HSD pin. 3 FB Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 4 GND Signal ground. GND is the ground path for the device bias voltage VDD and the control circuitry. The loop for the signal ground should be separate from the power ground (PGND) loop. 5 VDD VDD Bias (Input): Power to the internal reference and control sections of the MIC2176. The VDD operating voltage range is from 4.5V to 5.5V. A 1µF ceramic capacitor from the VDD pin to the PGND pin must be placed next to the IC. 6 DL 7 PGND Power Ground. PGND is the ground path for the buck converter power stage. The PGND pin connects to the sources of low-side N-Channel external MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (GND) loop. DH High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving voltage is floating on the switch node voltage (VSW). Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFETs can slow down the turn-on and turn-off time of the MOSFETs. 8 November 2010 Pin Function High-Side MOSFET Drain Connection (Input): The HSD pin in the input of the adaptive On-time control circuitry. A 0.1uF ceramic capacitor between the HSD pin and the power ground (PGND) is required and must be place as close as possible to the IC. Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving voltage swings from ground to VDD. 2 M9999-111710-A Micrel, Inc. MIC2176 Pin Description (Continued) Pin Number Pin Name 9 10 November 2010 Pin Function SW Switch Node and Current-Sense input (Input): High current output driver return. The SW pin connects directly to the switch node. Due to the high-speed switching on this pin, the SW pin should be routed away from sensitive nodes. The SW pin also senses the current by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to the SW pin using a Kelvin connection. BST Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected between the BST pin and the SW pin. Adding a small resistor in series with the BST pin can slow down the turn-on time of high-side N-Channel MOSFETs. 3 M9999-111710-A Micrel, Inc. MIC2176 Absolute Maximum Ratings(1, 2) Operating Ratings(3) VHSD to PGND................................................ −0.3V to +76V VDD to PGND ................................................... −0.3V to +6V VSW to PGND......................................−0.3V to (VHSD +0.3V) VBST to VSW ........................................................ −0.3V to 6V VBST to PGND .................................................. −0.3V to 82V VEN to PGND ...................................... −0.3V to (VDD + 0.3V) VFB to PGND....................................... −0.3V to (VDD + 0.3V) PGND to GND .............................................. −0.3V to +0.3V Junction Temperature .............................................. +150°C Storage Temperature (TS).........................−65°C to +150°C Lead Temperature (soldering, 10sec)........................ 260°C Supply Voltage (VHSD) ....................................... 4.5V to 75V Bias Voltage (VDD)............................................ 4.5V to 5.5V Enable Input (VEN) ................................................. 0V to VDD Junction Temperature (TJ) ........................ −40°C to +125°C Junction Thermal Resistance MSOP (θJA) ..................................................130.5°C/W Continuous Power Dissipation (derate 5.6mW/°C above 70°C) (TA = 70°C) ........................................................421mW ESD (Human Body Mode) .......................................... 1.5kV Electrical Characteristics(4) VIN = VHSD = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units 75 V Power Supply Input HSD Voltage Range (VHSD)(5) 4.5 VDD Bias Voltage Operating Bias Voltage (VDD) Undervoltage Lockout Trip Level VDD Rising 4.5 5 5.5 V 3.2 3.85 4.45 V UVLO Hysteresis 370 mV Quiescent Supply Current VFB = 1.5V 1.4 3 mA Shutdown Supply Current SW = unconnected, VEN = 0V 0.7 2 mA Reference Feedback Reference Voltage FB Bias Current 0°C ≤ TJ ≤ 85°C (±1.0%) 0.792 0.8 0.808 -40°C ≤ TJ ≤ 125°C (±1.5%) 0.788 0.8 0.812 5 500 nA 0.78 0.4 V 50 100 µA VFB = 0.8V V Enable Control EN Logic Level High 4.5V < VDD < 5.5V EN Logic Level Low 4.5V < VDD < 5.5V EN Bias Current VEN = 0V 1.2 0.85 V Oscillator Switching Frequency (6) Maximum Duty Cycle (7) Minimum Duty Cycle MIC2176-1 75 100 125 MIC2176-2 150 200 250 MIC2176-3 225 300 375 MIC2176-1, VFB = 0V, HSD=4V, VO = 3.3V 96 MIC2176-2, VFB = 0V, HSD=4V, VO = 3.3V 93 MIC2176-3, VFB = 0V, HSD=4V, VO = 3.3V 89 VFB > 0.8V 0 kHz % % Minimum Off-Time 360 ns Minimum On-Time 60 ns November 2010 4 M9999-111710-A Micrel, Inc. MIC2176 Electrical Characteristics(4) (Continued) VIN = VHSD = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C. Parameter Condition Min. Typ. Max. Units Soft Start Soft-Start time 6 ms Short Circuit Protection Current-Limit Threshold VFB = 0.8V 103 130 162 mV Short-Circuit Threshold VFB = 0V 19 48 77 mV 0.1 V FET Drivers DH, DL Output Low Voltage ISINK = 10mA DH, DL Output High Voltage ISOURCE = 10mA VDD - 0.1V V Or VBST - 0.1V DH On-Resistance, High State 2.1 3.3 Ω DH On-Resistance, Low State 1.8 3.3 Ω DL On-Resistance, High State 1.8 3.3 Ω DL On-Resistance, Low State 1.2 2.3 Ω SW Leakage Current VSW = 48V, VDD = 5V, VBST = 53V 55 µA HSD Leakage Current VSW = 48V, VDD = 5V, VBST = 53V 55 µA Thermal Protection Over-Temperature Shutdown TJ Rising Over-Temperature Shutdown Hysteresis 160 °C 25 °C Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF. 3. The device is not guaranteed to function outside operating range. 4. Specification for packaged product only. 5. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have enough low voltage VTH. 6. Measured in test mode. 7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns. November 2010 5 M9999-111710-A Micrel, Inc. MIC2176 Typical Characteristics VIN Shutdown Current vs. Input Voltage VIN Operating Supply Current vs. Input Voltage 40 25 20 15 MIC2176-2 VOUT = 3.3V 10 IOUT = 0A 5 VDD = 5V SWITCHING 10 35 SUPPLY CURRENT (mA) SHUTDOWN CURRENT (µA) 30 25 20 15 10 VDD = 5V VEN = 0V 5 0 0 0 10 20 30 40 50 60 0 70 10 20 30 40 50 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Feedback Voltage vs. Input Voltage Total Regulation vs. Input Voltage 60 TOTAL REGULATION (%) 0.806 0.804 0.802 0.800 0.798 0.796 VOUT = 3.3V VDD = 5V 0.794 6 MIC2176-2 VOUT = 3.3V 4 IOUT = 0A VDD = 5V SWITCHING 2 0 10 20 30 40 20 VOUT = 3.3V 0.8% VDD = 5V IOUT = 0A to 5A 0.6% 0.4% 0.2% 15 10 5 VOUT = 1.2V 0 0.0% 20 30 40 50 70 VDD = 5V 0.792 10 60 Current Limit vs. Input Voltage IOUT = 0A 0 50 INPUT VOLTAGE (V) 1.0% 0.808 8 0 70 CURRENT LIMIT (A) SUPPLY CURRENT (mA) 30 FEEDBACK VOLTAGE (V) VDD Operating Supply Current vs. Input Voltage 60 70 INPUT VOLTAGE (V) 0 10 20 30 40 50 INPUT VOLTAGE (V) 60 70 0 10 20 30 40 50 60 70 INPUT VOLTAGE (V) Switching Frequency vs. Input Voltage SWITCHING FREQUENCY (kHz) 240 MIC2176-2 V OUT = 3.3V 220 IOUT = 0A V DD = 5V 200 180 160 0 10 20 30 40 50 60 70 INPUT VOLTAGE (V) November 2010 6 M9999-111710-A Micrel, Inc. MIC2176 Typical Characteristics (Continued) VDD Shutdown Current vs. Temperature VDD Operating Supply Current vs. Temperature 10 VDD UVLO Threshold vs. Temperature 4.2 1 6 MIC2176-2 VIN = 48V 4 VOUT = 3.3V IOUT = 0A 2 VDD = 5V SWITCHING 0 -20 10 40 70 100 0.6 0.4 VIN = 48V IOUT = 0A VDD = 5V 0.2 130 VEN = 0V 3.8 3.6 3.4 Falling V IN = 48V 3.2 -20 10 40 70 100 130 -50 -20 10 40 70 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) VIN Operating Supply Current vs. Temperature VIN Shutdown Current vs. Temperature Current Limit vs. Temperature 18 16 MIC2176-2 V IN = 48V 14 V OUT = 3.3V IOUT = 0A V DD = 5V SWITCHING 12 25 CURRENT LIMIT (A) SUPPLY CURRENT (µA) 20 20 15 10 VIN = 48V VDD = 5V 5 -20 10 40 70 100 -20 10 TEMPERATURE (°C) 40 70 100 -50 130 V OUT = 3.3V IOUT = 0A V DD = 5V 0.800 0.798 0.796 0.794 0.792 0.8% VIN = 48V VDD = 5V IOUT = 0A to 5A 0.4% 0.2% -20 10 40 70 100 130 70 0.8% VIN = 6V to 70V VOUT = 3.3V IOUT = 0A 0.6% VDD = 5V 0.4% 0.2% 0.0% 0.0% -0.2% -0.2% -50 40 1.0% VOUT = 3.3V 0.6% 10 Line Regulation vs. Temperature LINE REGULATION (%) LOAD REGULATION (%) V IN = 48V 0.802 -20 TEMPERATURE (°C) 1.0% 0.804 VOUT = 3.3V VDD = 5V Load Regulation vs. Temperature Feedback Voltage vs. Temperature 0.806 130 VIN = 48V 5 TEMPERATURE (°C) 0.808 100 10 0 -50 130 130 15 IOUT = 0A 0 -50 100 20 30 10 FEEBACK VOLTAGE (V) 4.0 3.0 -50 22 SUPPLY CURRENT (mA) 0.8 0 -50 -50 -20 TEMPERATURE (°C) 10 40 70 100 130 -50 -20 10 40 70 100 130 TEMPERATURE (°C) TEMPERATURE (°C) EN Bias Current vs. Temperature Switching Frequency vs. Temperature 100 240 MIC2176-1 VIN = 48V 220 EN BIAS CURRENT (µA) SWITCHING FREQUENCY (kHz) VDD THRESHOLD (V) SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) Rising 8 VOUT = 3.3V IOUT = 0A VDD = 5V 200 180 80 60 40 VIN = 48V 20 VEN = 0V VDD = 5V 160 0 -50 -20 10 40 70 TEMPERATURE (°C) November 2010 100 130 -50 -20 10 40 70 100 130 TEMPERATURE (°C) 7 M9999-111710-A Micrel, Inc. MIC2176 Typical Characteristics (Continued) Feedback Voltage vs. Output Current Efficiency vs. Output Current 0.806 28VIN 80 48VIN 75 70 60VIN 65 60 MIC2176-2 VOUT = 3.3V 55 50 0.802 0.800 0.798 VIN = 48V 0.796 VOUT = 3.3V 0.794 VDD = 5V LINEAR 45 0.804 0 1 2 3 4 0 5 1 SWITCHING FREQUENCY (kHz) LINE REGULATION (%) VIN = 6V to 70V VOUT = 3.3V VDD = 5V 0.4% 0.2% 0.0% -0.2% 1 2 3 4 3 4 3.278 0 5 1 MIC2176-2 VIN = 48V VOUT = 3.3V 180 VDD = 5V 1 2 3 4 5 CASE TEMPERATURE (°C) 40 MIC2176-2 V IN = 48V V OUT = 3.3V 5 40 MIC2176-2 VIN = 28V 20 VOUT = 3.3V VDD = 5V 0 1 2 3 4 5 OUTPUT CURRENT (A) OUTPUT CURRENT (A) 60 4 0 160 100 3 IC Case Temperature* (VIN = 28V) vs. Output Current 200 0 80 2 OUTPUT CURRENT (A) 220 5 IC Case Temperature* (VIN = 48V) vs. Output Current CASE TEMPERATURE (°C) 3.289 60 OUTPUT CURRENT (A) 20 2 240 1.0% 0 3.300 Switching Frequency vs. Output Current Line Regulation vs. Output Current 0.6% VDD = 5V 3.311 OUTPUT CURRENT (A) OUTPUT CURRENT (A) 0.8% VOUT = 3.3V 3.267 0.792 40 VIN = 48V 3.322 VDD = 5V CASE TEMPERATURE (°C) EFFICIENCY (%) 85 3.333 OUTPUT VOLTAGE (V) 0.808 90 FEEDBACK VOLTAGE (V) 95 Output Voltage vs. Output Current IC Case Temperature* (VIN = 60V) vs. Output Current 80 60 40 MIC2176-2 VIN = 60V 20 VOUT = 3.3V VDD = 5V V DD = 5V 0 0 0 1 2 3 OUTPUT CURRENT (A) 4 5 0 1 2 3 4 5 OUTPUT CURRENT (A) Case Temperature* : The temperature measurement was taken at the hottest point on the MIC2176 case mounted on a 5 square inch PCB, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. November 2010 8 M9999-111710-A Micrel, Inc. MIC2176 Typical Characteristics (Continued) 95 85 80 75 80 70 75 70 65 60 55 50 MIC2176-2 VDD = 5V LINEAR 65 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 85 EFFICIENCY (%) EFFICIENCY (%) 90 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V 90 Efficiency (VIN = 48V) vs. Output Current MIC2176-2 VDD = 5V LINEAR 45 60 EFFICIENCY (%) Efficiency (VIN = 28V) vs. Output Current 40 0 1 2 3 4 5 OUTPUT CURRENT (A) November 2010 6 7 0 1 2 3 4 5 OUTPUT CURRENT (A) 9 6 7 Efficiency (VIN = 60V) vs. Output Current 90 85 80 75 70 65 60 55 50 45 40 35 30 5.0V 3.3V 2.5V 1.8V 1.5V 1.2V 1.0V 0.9V 0.8V MIC2176-2 VDD = 5V LINEAR 0 1 2 3 4 5 6 7 OUTPUT CURRENT (A) M9999-111710-A Micrel, Inc. MIC2176 Functional Characteristics November 2010 10 M9999-111710-A Micrel, Inc. MIC2176 Functional Characteristics (Continued) November 2010 11 M9999-111710-A Micrel, Inc. MIC2176 Functional Characteristics (Continued) November 2010 12 M9999-111710-A Micrel, Inc. MIC2176 Functional Diagram Figure 1. MIC2176 Functional Diagram November 2010 13 M9999-111710-A Micrel, Inc. MIC2176 The maximum duty cycle is obtained from the 360ns tOFF(min): Functional Description The MIC2176 is an adaptive on-time synchronous buck controller family built for high-input voltage and low output voltage applications. It is designed to operate over a wide input voltage range from, 4.5V to 75V and the output is adjustable with an external resistive divider. A digitally modified adaptive on-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. Over-current protection is implemented by sensing low-side MOSFET’s RDS(ON). The device features internal softstart, enable, UVLO, and thermal shutdown. D max = VOUT VIN × fSW tS = 1− 360ns tS (2) where tS = 1/fSW. It is not recommended to use MIC2176 with a OFF-time close to tOFF(min) during steady-state operation. The adpative ON-time control scheme results in a constant switching frequency in the MIC2176. The actual ON-time and resulting switching frequency will vary with the different rising and falling times of the external MOSFETs. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications, such as 48V to 1.0V. The minimum tON measured on the MIC2176 evaluation board is about 60ns. During load transients, the switching frequency is changed due to the varying OFF-time. To illustrate the control loop operation, we will analyze both the steady-state and load transient scenarios. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the feedback voltage. Figure 2 shows the MIC2176 control loop timing during steady-state operation. During steady-state, the gm amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple plus injected voltage ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. Theory of Operation Figure 1 illustrates the block diagram of the MIC2176. The output voltage is sensed by the MIC2176 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low-gain transconductance (gm) amplifier. If the feedback voltage decreases and the amplifier output is below 0.8V, then the error comparator will trigger the control logic and generate an ON-time period. The ONtime period length is predetermined by the “Fixed tON Estimator” circuitry: tON(estimated) = t S − t OFF(min) (1) where VOUT is the output voltage, VIN is the power stage input voltage, and fSW is the switching frequency (100kHz for MIC2176-1, 200kHz for MIC2176-2, and 300kHz for MIC2176-3). At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gm amplifier is below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 360ns, the MIC2176 control logic will apply the tOFF(min) instead. tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. Figure 2. MIC2176 Control Loop Timing November 2010 14 M9999-111710-A Micrel, Inc. MIC2176 Figure 3 shows the operation of the MIC2176 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This will cause the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time tOFF(min) is generated to charge CBST since the feedback voltage is still below VREF. Then, the next ON-time period is triggered due to the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC2176 converter. Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. The MIC2176 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with 9.7mV steps. Therefore, the output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. VDD must be powered up at the same time or after VIN to make the soft-start function correctly. Current Limit The MIC2176 uses the RDS(ON) of the low-side power MOSFET to sense over-current conditions. This method will avoid adding cost, board space and power losses taken by discrete current sense resistors. The low-side MOSFET is used because it displays much lower parasitic oscillations during switching than the high-side MOSFET. In each switching cycle of the MIC2176 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. The sensed voltage is compared with a current-limit threshold voltage VCL after a blanking time of 150ns. If the sensed voltage is over VCL, which is 133mV typical at 0.8V VFB, then the MIC2176 turns off the high-side and low-side MOSFETs and a soft-start sequence is triggered. This mode of operation is called “hiccup mode” and its purpose is to protect the downstream load in case of a hard short. The current limit threshold VCL has a foldback characteristic related to the FB voltage. Please refer to the “Typical Characteristics” for the curve of current limit threshold vs. FB voltage percentage. The circuit in Figure 4 illustrates the MIC2176 current limiting circuit. Figure 3. MIC2176 Load Transient Response Unlike true current-mode control, the MIC2176 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor current ripple if the ESR of the output capacitor is large enough. The MIC2176 control loop has the advantage of eliminating the need for slope compensation. In order to meet the stability requirements, the MIC2176 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended feedback voltage ripple is 20mV~100mV. If a low ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. In these cases, ripple injection is required to ensure proper operation. Please refer to “Ripple Injection” subsection in Application Information for more details about the ripple injection technique. Figure 4. MIC2176 Current Limiting Circuit November 2010 15 M9999-111710-A Micrel, Inc. MIC2176 Using the typical VCL value of 130mV, the current-limit value is roughly estimated as: ICL ≈ 130mV R DS(ON) MOSFET Gate Drive The MIC2176 high-side drive circuit is designed to switch an N-Channel MOSFET. Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode D1 is reverse biased and CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. ΔBST = 10mA x 3.33μs/0.1μF = 333mV. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor RG, which is in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. The drive voltage is derived from the VDD supply voltage. The nominal low-side gate drive voltage is VDD and the nominal high-side gate drive voltage is approximately VDD – VDIODE, where VDIODE is the voltage drop across D1. An approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. (3) For designs where the current ripple is significant compared to the load current IOUT, or for low duty cycle operation, calculating the current limit ICL should take into account that one is sensing the peak inductor current and that there is a blanking delay of approximately 150ns. ICL = 130mV VOUT × TDLY ΔIL(pp) + − RDS(ON) L 2 V × (1− D) ΔIL(pp) = OUT f SW × L (4) (5) where VOUT = The output voltage TDLY = Current-limit blanking time, 150ns typical ΔIL(pp) = Inductor current ripple peak-to-peak value D = Duty Cycle fSW = Switching frequency The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to ICL in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect SW pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON). November 2010 16 M9999-111710-A Micrel, Inc. MIC2176 accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET: Application Information MOSFET Selection The MIC2176 controller works from power stage input voltages of 4.5V to 73V and has an external 4.5V to 5.5V VIN to provide power to turn the external N-Channel power MOSFETs for the high- and low-side switches. For applications where VDD < 5V, it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VDD > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. There are different criteria for choosing the high-side and low-side MOSFETs. These differences are more significant at lower duty cycles. In such an application, the high-side MOSFET is required to switch as quickly as possible to minimize transition losses, whereas the low-side MOSFET can switch slower, but must handle larger RMS currents. When the duty cycle approaches 50%, the current carrying capability of the high-side MOSFET starts to become critical. It is important to note that the on-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current limit. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2176 gate-drive circuit. At 300kHz switching frequency, the gate charge can be a significant source of power dissipation in the MIC2176. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is: IG[high-side] (avg) = Q G × f SW IG[low -side] (avg) = C ISS × VGS × f SW Since the current from the gate drive comes from the VDD, the power dissipated in the MIC2176 due to gate drive is: PGATEDRIVE = VDD × (IG[high- side] (avg) + IG[low -side] (avg)) (8) A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON) × QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2176. Also, the RDS(ON) of the low-side MOSFET will determine the current-limit value. Please refer to “Current Limit” subsection is Functional Description for more details. Parameters that are important to MOSFET switch selection are: • Voltage rating • On-resistance • Total gate charge The voltage ratings for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VHSD. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitic elements. The power dissipated in the MOSFETs is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses during the period of time when the MOSFETs turn on and off (PAC). (6) PSW = PCONDUCTION + PAC where: IG[high-side](avg) = Average high-side MOSFET gate current QG = Total gate charge for the high-side MOSFET taken from the manufacturer’s data sheet for VGS = VDD. fSW = Switching Frequency (9) PCONDUCTION = ISW(RMS) 2 × RDS(ON) (10) PAC = PAC(off ) + PAC(on) (11) where: RDS(ON) = On-resistance of the MOSFET switch D = Duty Cycle = VOUT / VHSD The low-side MOSFET is turned on and off at VDS = 0 because an internal body diode or external freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more November 2010 (7) 17 M9999-111710-A Micrel, Inc. MIC2176 The inductance value is calculated by Equation 14: Making the assumption that the turn-on and turn-off transition times are equal; the transition times can be approximated by: L= tT = C ISS × VIN + C OSS × VHSD IG where: fSW = Switching frequency, 300kHz 20% = Ratio of AC ripple current to DC output current VIN(max) = Maximum power stage input voltage The peak-to-peak inductor current ripple is: The total high-side MOSFET switching loss is: ΔIL(pp) = (13) where: tT = Switching transition time VD = Body diode drop (0.5V) fSW = Switching Frequency VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × L (15) The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(pk) =IOUT(max) + 0.5 × ΔIL(pp) The high-side MOSFET switching losses increase with the switching frequency and the input voltage VHSD. The low-side MOSFET switching losses are negligible and can be ignored for these calculations. (16) The RMS inductor current is used to calculate the I2R losses in the inductor. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. November 2010 (14) (12) where: CISS and COSS are measured at VDS = 0 IG = Gate-drive current PAC = (VHSD + VD ) × IPK × t T × f SW VOUT × (VIN(max) − VOUT ) VIN(max) × fsw × 20% × IOUT(max) 2 IL(RMS) = IOUT(max) + ΔIL(PP) 12 2 (17) Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2176 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. 18 M9999-111710-A Micrel, Inc. MIC2176 where: D = duty cycle COUT = output capacitance value fsw = switching frequency Copper loss in the inductor is calculated by Equation 18: PINDUCTOR(Cu) = IL(RMS)2 × RWINDING (18) The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C)) As described in the “Theory of Operation” subsection in Functional Description, the MIC2176 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the “Ripple Injection” subsection for more details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 22: (19) where: TH = temperature of wire under full load T20°C = ambient temperature RWINDING(20°C) = room temperature winding resistance (usually specified by the manufacturer) Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated: ICOUT (RMS) = ΔIL(PP) (22) 12 The power dissipated in the output capacitor is: 2 ESR COUT ≤ PDISS(COUT ) = ICOUT (RMS) × ESR COUT ΔVOUT(pp) (20) ΔIL(PP) Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: where: ΔVOUT(pp) = peak-to-peak output voltage ripple ΔIL(PP) = peak-to-peak inductor current ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated in Equation 21: 2 ΔVOUT(pp) ΔIL(PP) ⎞ ⎛ 2 ⎟ + ΔIL(PP) × ESR C = ⎜⎜ OUT ⎟ ⎝ C OUT × f SW × 8 ⎠ (21) November 2010 ( (23) ) ΔVIN = IL(pk) × ESRCIN 19 (24) M9999-111710-A Micrel, Inc. MIC2176 less than 20mV. If the feedback voltage ripple is so small that the gm amplifier and error comparator can’t sense it, then the MIC2176 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1. Enough ripple at the feedback voltage due to the large ESR of the output capacitors. As shown in Figure 6a, the converter is stable without any ripple injection. The feedback voltage ripple is: The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN(RMS) ≈ IOUT(max) × D × (1 − D) (25) The power dissipated in the input capacitor is: PDISS(CIN) = ICIN(RMS)2 × ESRCIN (26) Voltage Setting Components The MIC2176 requires two resistors to set the output voltage as shown in Figure 5: ΔVFB(pp) = ΔVFB(pp) ≈ ESR × ΔIL (pp) The output voltage is determined by the equation: R1 ) R2 (29) where ΔIL(pp) is the peak-to-peak value of the inductor current ripple. 2. Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 6b. The typical Cff value is between 1nF and 100nF. With the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: Figure 5. Voltage-Divider Configuration VOUT = VFB × (1 + R2 × ESR COUT × ΔIL (pp) R1 + R2 (30) 3. Virtually no ripple at the FB pin voltage due to the very-low ESR of the output capacitors: (27) where, VFB = 0.8V. A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 = VFB × R1 VOUT − VFB (28) Figure 6a. Enough Ripple at FB Ripple Injection The VFB ripple required for proper operation of the MIC2176 gm amplifier and error comparator is 20mV to 100mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as a 1V, the output voltage ripple is only 10mV to 20mV, and the feedback voltage ripple is November 2010 20 M9999-111710-A Micrel, Inc. MIC2176 If the voltage divider resistors R1 and R2 are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. Step 2. Select Rinj according to the expected feedback voltage ripple using Equation 24: Figure 6b. Inadequate Ripple at FB K div = ΔVFB(pp) VIN × fSW × τ D × (1 − D) (34) Then the value of Rinj is obtained as: Figure 6c. Invisible Ripple at FB R inj = (R1//R2) × ( In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 6c. The injected ripple is: ΔVFB(pp) = VIN × K div × D × (1 - D) × K div = R1//R2 R inj + R1//R2 1 fSW × τ 1 K div − 1) (35) Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. (31) (32) where: VIN = Power stage input voltage D = Duty cycle fSW = Switching frequency τ = (R1//R2//Rinj) × Cff In Equations 21 and 22, it is assumed that the time constant associated with Cff must be much greater than the switching period: 1 T = << 1 fSW × τ τ November 2010 (33) 21 M9999-111710-A Micrel, Inc. MIC2176 Inductor PCB Layout Guidelines Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2176 converter. IC • The signal ground pin (GND) must be connected directly to the ground planes. Do not route the GND pin to the PGND pin on the top layer. • Place the IC close to the point of load (POL). • Use fat traces to route the input and output power lines. • Signal and power grounds should be kept separate and connected at only one location. Keep the inductor connection to the switch node (SW) short. • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (SW) away from the feedback (FB) pin. • The SW pin should be connected directly to the drain of the low-side MOSFET to accurate sense the voltage across the low-side MOSFET. • To minimize noise, place a ground plane underneath the inductor. Output Capacitor The 1µF ceramic capacitor, which is connected to the VDD pin, must be located right at the IC. The VDD pin is very noise sensitive and placement of the capacitor is very critical. Use wide traces to connect to the VDD and PGND pins. • • • Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. • Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. Input Capacitor MOSFETs • Place the input capacitor next. • • Place the input capacitors on the same side of the board and as close to the MOSFETs as possible. • Place several vias to the ground plane close to the input capacitor ground terminal. Low-side MOSFET gate drive trace (DL pin to MOSFET gate pin) must be short and routed over a ground plane. The ground plane should be the connection between the MOSFET source and PGND. • • Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Chose a low-side MOSFET with a high CGS/CGD ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. • Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. • Do not put a resistor between the Low-side MOSFET gate drive output and the gate. • Use a 4.5V VGS rated MOSFET. Its higher gate threshold voltage is more immune to glitches than a 2.5V or 3.3V rated MOSFET. MOSFETs that are rated for operation at less than 4.5V VGS should not be used. • If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. • In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. RC Snubber • Place the RC snubber on the same side of the board and as close to the SW pin as possible. November 2010 22 M9999-111710-A Micrel, Inc. MIC2176 Evaluation Board Schematic Figure 7. Schematic of MIC2176 Evaluation Board (J1, J9, J12, R12, and R13 are for testing purposes) November 2010 23 M9999-111710-A Micrel, Inc. MIC2176 Bill of Materials Item C1 Part Number B41125A9336M Manufacturer (1) EPCOS Description Qty 33µF Aluminum Capacitor, SMD, 100V (2) 1 C2, C3 GRM32ER72A225K Murata 2.2µF/100V Ceramic Capacitor, X7R, Size 1210 2 C4 6SEPC470M Sanyo(3) 470µF/6.3V OSCON Capacitor 1 C5, C15 GRM32ER60J104KA94D Murata(2) 100µF/6.3V Ceramic Capacitor, X7R, Size 1210 2 GRM188R71H104KA94L Murata (2) 0.1µF/6.3V Ceramic Capacitor, X7R, Size 0603 3 Murata (2) 1µF/6.3V Ceramic Capacitor, X7R, Size 0603 1 (2) 0.1µF/100V Ceramic Capacitor, X7R, Size 0603 2 (2) 1nF/100V Cermiac Capacitor, X7R, Size 0603 1 Murata (2) 10nF/50V Ceramic Capacitor, X7R, Size 0603 1 Murata (2) 4.7µF/6.3V Ceramic Capacitor, X5R, Size 1206 1 100V Small Signal Schottky Diode, SOD123 1 5.6V Zener Diode, SOD323 1 C6, C7, C16 C8 GRM188R70J105KA01D C9, C10 C11 C12 C14 D1 GRM188R72A104KA35D GRM188R72A102KA01D GRM188R71H103K GRM31CR60J475KA01L BAT46W D2 CMDZ5L6 L1 HCL1305-4R0-R Q1 SIR432DP Q2 SIR804DP Q3 FCX493 R1, R3 R2 CRCW060310K0FKEA CRCW08051R21FKEA R4 CRCW060380K6FKEA R5 CRCW060340K2FKEA R6 CRCW060320K0FKEA Murata Murata Diodes, Inc. (4) (5) Central Semi Cooper Bussmann(6) 4.0µH Inductor, 10A RMS Current 1 (7) MOSFET, N-CH, Power SO-8 1 (7) MOSFET, N-CH, Power SO-8 1 Vishay Vishay (4) ZETEX 100V NPN Transistor, SOT89 1 (7) 10kΩ Resistor, Size 0603, 1% 2 (7) 1.21Ω Resistor, Size 0805, 5% 1 (7) 80.6kΩ Resistor, Size 0603, 1% 1 (7) 40.2kΩ Resistor, Size 0603, 1% 1 (7) 20kΩ Resistor, Size 0603, 1% 1 Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Notes: 1. EPCOS: www.epcos.com. 2. Murata: www.murata.com. 3. Sanyo: www.sanyo.com. 4. Diodes Inc.: www.diodes.com. 5. Central Semi: www.centralsemi.com. 6. Cooper Bussmann: www.cooperbussmann.com. 7. Vishay: www.vishay.com. November 2010 24 M9999-111710-A Micrel, Inc. MIC2176 Bill of Materials (Continued) Item R7 Part Number CRCW060311K5FKEA Manufacturer Description Qty (7) 11.5kΩ Resistor, Size 0603, 1% 1 (7) Vishay Dale R8 CRCW06038K06FKEA Vishay Dale 8.06kΩ Resistor, Size 0603, 1% 1 R9 CRCW06034K75FKEA Vishay Dale(7) 4.75kΩ Resistor, Size 0603, 1% 1 CRCW06033K24FKEA (7) 3.24kΩ Resistor, Size 0603, 1% 1 (7) 1.91kΩ Resistor, Size 0603, 1% 1 (7) 49.9Ω Resistor, Size 0603, 1% 1 (7) 0Ω Resistor, Size 0603, 5% 2 (7) 9.7kΩ Resistor, Size 0805, 5% 1 (8) 75V Synchronous Buck DC-DC Regulator 1 R10 R11 R12 R13, R21 R14 U1 CRCW06031K91FKEA CRCW060349K24FKEA CRCW06030000FKEA CRCW08059K7FKEA MIC2176-2YMM Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel. Inc. Notes: 8. Micrel, Inc.: www.micrel.com. November 2010 25 M9999-111710-A Micrel, Inc. MIC2176 PCB Layout Figure 8. MIC2176 Evaluation Board Top Layer November 2010 26 M9999-111710-A Micrel, Inc. MIC2176 PCB Layout (Continued) Figure 9. MIC2176 Evaluation Board Mid-Layer 1 (Ground Plane) November 2010 27 M9999-111710-A Micrel, Inc. MIC2176 PCB Layout (Continued) Figure 10. MIC2176 Evaluation Board Mid-Layer 2 November 2010 28 M9999-111710-A Micrel, Inc. MIC2176 PCB Layout (Continued) Figure 11. MIC2176 Evaluation Board Bottom Layer November 2010 29 M9999-111710-A Micrel, Inc. MIC2176 Recommended Land Pattern November 2010 30 M9999-111710-A Micrel, Inc. MIC2176 Package Information 10-Pin MSOP (MM) MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2010 Micrel, Incorporated. November 2010 31 M9999-111710-A