MIC28500

MIC28500
75V/4A Hyper Speed Control™
Synchronous DC-DC Buck Regulator
SuperSwitcher II™
General Description
Features
The Micrel MIC28500 is an adjustable frequency,
synchronous buck regulator featuring a unique adaptive ontime control architecture. The MIC28500 operates over an
input supply range of 30V to 75V and provides a regulated
output of up to 4A of output current. The output voltage is
adjustable down to 0.8V with a guaranteed accuracy of ±1%.
•
Micrel’s Hyper Speed Control™ architecture allows for ultrafast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This adaptive tON ripple control architecture combines the
advantages of fixed-frequency operation and fast transient
response in a single device.
The MIC28500 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup” mode shortcircuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
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•
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•
•
•
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Hyper Speed Control™ architecture enables
-High Delta V operation (VIN = 75V and VOUT = 0.8V)
-Small output capacitance
30V to 75V voltage input
Adjustable output down to 0.8V
±1% FB accuracy
Any Capacitor™ Stable
- Zero-ESR to high-ESR output capacitors
4A output current capability, up to 90% efficiency
100kHz to 500kHz switching frequency
Internal compensation
Foldback current-limit and “hiccup” mode short-circuit
protection
Thermal shutdown
Supports safe startup into a pre-biased load
–40°C to +125°C junction temperature range
28-pin 5mm × 6mm MLF® package
Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
___________________________________________________________________________________________________________
Typical Application
Efficiency (VIN = 48V)
vs. Output Current
100
90
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.8V
EFFICIENCY (%)
80
70
60
fSW = 250kHz
50
40
30
20
10
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
June 2011
M9999-060311-B
Micrel, Inc.
MIC28500
Ordering Information
Part Number
Voltage
Switching Frequency
Junction Temperature Range
Package
Lead Finish
MIC28500YJL
Adjustable
Adjustable
−40°C to +125°C
28-pin 5mm × 6mm MLF®
Pb-Free
Pin Configuration
28-Pin 5mm × 6mm MLF® (YJL)
Pin Description
Pin
Number
Pin Name
Pin Function
PVIN
High-Side internal N-channel MOSFET Drain Connection (Input): The PVIN operating voltage range
is from 30V to 75V. Input capacitors between the PVIN pins and the power ground (PGND) are
required and keep the connection short. Enabling the device below 30V VIN and under maximum
loading could heat up the device beyond safe operating conditions.
24
EN
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA). Do not pull the EN pin above the VDD supply. Enabling the device below
30V VIN and under maximum loading could heat up the device beyond safe operating conditions.
25
FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
26
SGND
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to
the PGND Pad on the top layer, see PCB layout guidelines for details.
VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC28500. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin to the
PGND pin must be placed next to the IC. VDD must be powered up at the same time or after PVIN to
make the soft-start function correctly.
PGND
Power Ground. PGND is the ground path for the MIC28500 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The loop for the power ground should be
as small as possible and separate from the signal ground (SGND) loop.
13, 14, 15,
16, 17, 18,
19
27
2, 5, 6, 7,
8, 21
22
June 2011
CS
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high-speed switching on this pin, the CS pin should be routed away from sensitive
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.
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Micrel, Inc.
MIC28500
Pin Description (Continued)
Pin
Number
Pin Name
20
BST
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected
between the BST pin and the SW pin.
4, 9, 10,
11, 12
SW
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain.
23
FS
Frequency Setting Pin.
28
PVDD
1, 3
NC
June 2011
Pin Function
Power Supply for gate driver of bottom MOSFET.
No Connect.
3
M9999-060311-B
Micrel, Inc.
MIC28500
Absolute Maximum Ratings(1, 2)
Operating Ratings(3)
PVIN to PGND................................................ −0.3V to +76V
FS to PGND ....................................................−0.3V to PVIN
PVDD, VDD to PGND ......................................... −0.3V to +6V
VSW, VCS to PGND .............................. −0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 82V
VEN to PGND ...................................... −0.3V to (VDD + 0.3V)
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Supply Voltage (PVIN) ........................................ 30V to 75V
Bias Voltage (PVDD, VDD).................................. 4.5V to 5.5V
Enable Input (VEN) ................................................. 0V to VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF® (θJA) ....................................36°C/W
Electrical Characteristics(5)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range ( PVIN)
30
75
V
FS Voltage Range
2
75
V
VDD Bias Voltage
Operating Bias Voltage (VDD)
Under-Voltage Lockout Trip Level
VDD Rising
4.5
5
5.5
V
3.2
3.85
4.45
V
UVLO Hysteresis
Quiescent Supply Current (IVDD)
Shutdown Supply Current (IVDD)
380
mV
VFB = 1.5V
1.4
3
mA
VDD = VBST = 5.5V, VIN = 48V
0.7
2
mA
SW = unconnected, VEN = 0V
Reference
Feedback Reference Voltage
0°C ≤ TJ ≤ 85°C (±1.0%)
0.792
0.8
0.808
−40°C ≤ TJ ≤ 125°C (±1.5%)
0.788
0.8
0.812
V
Load Regulation
IOUT = 0A to 4A
0.04
%
Line Regulation
PVIN = 30 to 75V
0.1
%
FB Bias Current
VFB = 0.8V
-0.5
0.005
EN Logic Level High
4.5V < VDD < 5.5V
1.2
0.85
EN Logic Level Low
4.5V < VDD < 5.5V
EN Bias Current
VEN = 0V
0.5
µA
Enable Control
V
0.78
0.4
V
50
100
µA
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.
June 2011
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MIC28500
Electrical Characteristics(5) (Continued)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
VFS=PVIN
375
500
625
kHz
Oscillator
Switching Frequency (6)
Maximum Duty Cycle
(7)
Minimum Duty Cycle
VFB = 0V, VFS=PVIN
82
%
VFB > 0.8V
0
%
360
ns
6
ms
Minimum Off-time
Soft-Start
Soft-Start time
Short Circuit Protection
Current-Limit Threshold
Short-Circuit Current
VFB = 0.8V, TJ = 25°C
5.5
7
9
A
VFB = 0.8V, TJ = 125°C
4.2
5.5
8.5
A
2
3.6
5.2
A
VFB = 0V
Internal FETs
Top-MOSFET RDS (ON)
ISW = 1A
175
mΩ
Bottom-MOSFET RDS (ON)
ISW = 1A
31
mΩ
SW Leakage Current
PVIN = 36V, VSW = 36V, VEN = 0V, VBST = 41V
55
µA
PVIN Leakage Current
PVIN = 36V, VSW = 0V, VEN = 0V, VBST = 41V
55
µA
Thermal Protection
Over-Temperature Shutdown
TJ Rising
Over-Temperature Shutdown
Hysteresis
160
°C
25
°C
Notes:
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
June 2011
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Micrel, Inc.
MIC28500
Typical Characteristics
IOUT = 0A
16
VDD = 5V
fSW = 250kHz
12
8
4
0
4
VDD = 5V
VEN = 0V
3
2
1
35
40
45
50
55
60
65
70
75
30
35
40
INPUT VOLTAGE (V)
8
fSW = 250kHz
6
4
2
45
50
55
60
65
70
30
75
35
40
IOUT = 0A
fSW = 250kHz
0.800
0.796
60
65
70
75
VOUT = 3.3V
VOUT = 3.3V
0.6%
VDD = 5V
0.4%
IOUT = 0A to 4A
0.2%
fSW = 250kHz
CURRENT LIMIT (A)
TOTAL REGULATION (%)
VDD = 5V
0.8%
55
15
1.0%
VOUT = 3.3V
50
Current Limit
vs. Input Voltage
Total Regulation
vs. Input Voltage
0.808
45
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
0.804
VOUT = 3.3V
VDD = 5V
0
0
30
FEEDBACK VOLTAGE (V)
VDD SUPPLY CURRENT (mA)
VOUT = 3.3V
VDD Operating Supply Current
vs. Input Voltage
10
5
SHUTDOWN CURRENT (mA)
20
SUPPLY CURRENT (mA)
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
0.0%
-0.2%
-0.4%
VDD = 5V
12
fSW = 250kHz
9
6
3
-0.6%
-0.8%
0.792
0
-1.0%
30
35
40
45
50
55
60
65
70
75
30
35 40 45 50
INPUT VOLTAGE (V)
30
60 65 70 75
VDD = 5V
IOUT = 1A
R18=100k Ω
R19=100k Ω
100
SHUTDOWN CURRENT (mA)
SUPPLY CURRENT (mA)
VOUT = 3.3V
VOUT = 3.3V
10
VDD = 5V
IOUT = 0A
8
fSW = 250kHz
6
4
2
35
40
45
50
55
60
65
70
75
55
60
65
70
75
VIN = 48V
IOUT = 0A
0.8
VDD = 5V
VEN = 0V
0.6
0.4
0.2
0.0
0
30
50
1.0
VIN = 48V
250
45
VDD Shutdown Current
vs. Temperature
12
150
40
INPUT VOLTAGE (V)
VDD Operating Supply Current
vs. Temperature
300
200
35
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
SWITCHING FREQUENCY (kHz)
55
-50
-25
0
25
50
75
TEMPERATURE (°C)
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
INPUT VOLTAGE (V)
June 2011
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Micrel, Inc.
MIC28500
Typical Characteristics (Continued)
4.2
VIN Shutdown Current
vs. Temperature
20
3.8
3.7
3.6
3.5
Falling
16
-25
0
25
50
75
100
VDD = 5V
IOUT = 0A
fSW = 250kHz
12
8
4
3.3
-50
SHUTDOWN CURRENT (mA)
Rising
3.4
VOUT = 3.3V
SUPPLY CURRENT (mA)
4.0
3.9
5
VIN = 48V
VIN = 48V
IOUT = 0A
4.1
VDD THRESHOLD (V)
VIN Operating Supply Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
V IN = 48V
V EN = 0V
2
1
0
-50
TEMPERATURE (°C)
IOUT = 0A
3
0
125
V DD = 5V
4
-25
0
25
50
75
100
-50
125
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
Current Limit
vs. Temperature
Feedback Voltage
vs. Temperature
15
0.808
V OUT = 3.3V
12
V DD = 5V
fSW = 250kHz
9
6
3
VIN = 48V
VIN = 48V
LOAD REGULATION (%)
FEEBACK VOLTAGE (V)
V IN = 48V
CURRENT LIMIT (A)
Load Regulation
vs. Temperature
0.4%
VOUT = 3.3V
VDD = 5V
0.804
IOUT = 0A
0.800
0.796
0.3%
VOUT = 3.3V
0.2%
IOUT = 0A to 4A
VDD = 5V
fSW = 250kHz
0.1%
0.0%
-0.1%
-0.2%
-0.3%
0
-25
0
25
50
75
100
125
-50
50
75
100
Line Regulation
vs. Temperature
Switching Frequency
vs. Output Current
300
SWITCHING FREQUENCY (kHz)
0.3%
VDD = 5V
0.2%
IOUT = 0A
0.1%
0.0%
-0.1%
-0.2%
-0.3%
0
25
50
75
TEMPERATURE (°C)
June 2011
25
TEMPERATURE (°C)
VIN = 30V to
75V
VOUT = 3.3V
-25
0
TEMPERATURE (°C)
0.4%
-50
-25
100
125
-25
125
0
25
50
75
100 125
TEMPERATURE (°C)
EN Bias Current
vs. Temperature
100
VIN = 48V
-40C
EN BIAS CURRENT (µA)
-50
LINE REGULATION (%)
-50
0.792
250
25C
125C
200
VIN = 48V
VOUT = 3.3V
VDD = 5V
150
VDD = 5V
80
VEN = 0V
60
40
20
0
-50
100
0
1
2
3
OUTPUT CURRENT (A)
7
4
-25
0
25
50
75
100
125
TEMPERATURE (°C)
M9999-060311-B
Micrel, Inc.
MIC28500
Typical Characteristics (Continued)
Efficiency
vs. Output Current
Enable Threshold
vs. Temperature
90
VDD = 5V
85
EFFICIENCY (%)
0.9
Rising
0.8
30VIN
FEEDBACK VOLTAGE (V)
VIN = 48V
ENABLE THRESHOLD (V)
0.808
95
1.0
0.7
Falling
80
48VIN
75
75VIN
VOUT = 3.3V
70
VDD = 5V
65
fSW = 250kHz
60
0.6
Feedback Voltage
vs. Output Current
VIN = 48V
VOUT = 3.3V
VDD = 5V
0.804
fSW = 250kHz
0.800
0.796
55
0.792
50
0.5
0
-50
-25
0
25
50
75
100
125
1
2
3
OUTPUT CURRENT (A)
TEMPERATURE (°C)
4
0
1
2
3
4
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current
0.3%
LINE REGULATION (%)
VIN = 30V to 75V
0.2%
VOUT = 3.3V
0.1%
fSW = 250kHz
VDD = 5V
0.0%
-0.1%
-0.2%
-0.3%
0
1
2
3
4
OUTPUT CURRENT (A)
June 2011
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MIC28500
Typical Characteristics (Continued)
80
DIE TEMPERATURE (°C)
DIE TEMPERATURE (°C)
80
60
40
VIN = 30V
VOUT = 3.3V
20
Die Temperature* (VIN = 48V)
vs. Output Current
VDD = 5V
fSW = 250kHz
60
40
VIN = 48V
VOUT = 3.3V
20
Die Temperature* (VIN = 75V)
vs. Output Current
100
DIE TEMPERATURE (°C)
Die Temperature* (VIN = 30V)
vs. Output Current
80
60
40
VIN = 75V
VOUT = 3.3V
20
VDD = 5V
VDD = 5V
fSW = 250kHz
fSW = 250kHz
0
0
1
2
3
4
0
0
0
OUTPUT CURRENT (A)
1
2
3
0
4
1
OUTPUT CURRENT (A)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
70
60
VIN = 30V
fSW = 250kHz
50
70
60
50
VIN = 48V
40
40
90
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
80
EFFICIENCY (%)
80
fSW = 250kHz
0
1
2
3
4
5
50
20
0
1
2
3
4
5
10
6
0
VOUT = 2.5V
1
Maximum Ambient Temperature (°C)
100
5
6
VIN = 48V
fSW =250kHz
L=10uH
Tj_MAX =125°C
3
VOUT = 1.2V
2
VOUT = 0.8V
1
0
0
85
4
4
Load Current (A)
Load Current (A)
VOUT = 5V
2
70
3
VIN = 48V
fSW =250kHz
L=10uH
Tj_MAX =125°C
55
2
Thermal Derating
4
40
1
OUTPUT CURRENT (A)
Thermal Derating
VIN = 48V
VOUT = 3.3V
fSW = 250kHz
OUTPUT CURRENT (A)
4
3
VIN = 75V
40
20
Thermal Derating
Load Current (A)
60
30
OUTPUT CURRENT (A)
25
70
30
6
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
80
10
30
4
100
100
90
90
3
Efficiency (VIN = 75V)
vs. Output Current
EFFICIENCY (%)
100
EFFICIENCY (%)
Efficiency (VIN = 48V)
vs. Output Current
Efficiency (VIN = 30V)
vs. Output Current
2
OUTPUT CURRENT (A)
25
40
55
70
85
Maximum Ambient Temperature (°C)
100
VOUT = 12V
3
fSW=250kHz
L=33μH
Tj_MAX =125°C
2
1
0
25
40
55
70
85
100
Maximum Ambient Temperature (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC28500 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz. finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
June 2011
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MIC28500
Functional Characteristics
June 2011
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MIC28500
Functional Characteristics (Continued)
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MIC28500
Functional Characteristics (Continued)
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MIC28500
Functional Diagram
Figure 1. MIC28500 Block Diagram
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MIC28500
where tS = 1/fSW. It is not recommended to use
MIC28500 with a OFF-time close to tOFF(min) during
steady-state operation..
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 75V to 1.0V. The minimum tON
measured on the MIC28500 evaluation board is about
184ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
Figure 2 shows the MIC28500 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ONtime is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Functional Description
The MIC28500 is an adaptive ON-time synchronous
step-down DC-DC regulator. It is designed to operate
over a wide input voltage range from, 30V to 75V, and
provides a regulated output voltage at up to 4A of output
current. A digitally modified adaptive ON-time control
scheme is employed in to obtain a constant switching
frequency and to simplify the control compensation.
Over current protection is implemented without the use
of an external sense resistor. The device includes an
internal soft-start function which reduces the power
supply input surge current at start-up by controlling the
output voltage rise time.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC28500. The output voltage is sensed by the
MIC28500 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
t ON(estimated) =
VOUT
VIN × fSW
Eq. 1
where VOUT is the output voltage and VIN is the power
stage input voltage and fSW is the switching frequency.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, then the MIC28500 control logic will apply the
tOFF(min) instead. The minimum tOFF(min) period is required
to maintain enough energy in the boost capacitor (CBST)
to drive the high-side MOSFET. The maximum duty
cycle is obtained from the 360ns tOFF(min):
Dmax =
June 2011
t S − t OFF(min)
tS
= 1−
360ns
tS
Figure 2. MIC28500 Control Loop Timing
Eq. 2
14
M9999-060311-B
Micrel, Inc.
MIC28500
Figure 3 shows the operation of the MIC28500 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC28500 converter.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC28500 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC28500 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC28500 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the peak inductor current
is greater than 7A, then the MIC28500 turns off the highside MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The current-limit threshold has a foldback
characteristic related to the feedback voltage, as shown
in Figure 4.
Figure 3. MIC28500 Load Transient Response
Unlike true current-mode control, the MIC28500 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC28500 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
June 2011
Current Limit Threshold
vs. Feedback Voltage
CURRENT LIMIT THRESHOLD
(A)
12
VIN = 48V
10
8
6
4
2
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC28500 Current Limit Foldback Characteristic
15
M9999-060311-B
Micrel, Inc.
MIC28500
Internal MOSFET Gate Drive
Figure 1 (Block Diagram) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA x 3.33μs/0.1μF = 333mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
The drive voltage is derived from the PVDD supply
voltage. The nominal low-side gate drive voltage is PVDD
and the nominal high-side gate drive voltage is
approximately PVDD – VDIODE, where VDIODE is the voltage
drop across D1. An approximate 30ns delay between the
high-side and low-side driver transitions is used to
prevent current from simultaneously flowing unimpeded
through both MOSFETs.
June 2011
16
M9999-060311-B
Micrel, Inc.
MIC28500
programmed to either lower end or higher end, the
design needs optimization.
Application Information
Setting the Switching Frequency
The MIC28500 is an adjustable-frequency, synchronous
buck regulator featuring a unique digitally modified
adaptive on-time control architecture. The switching
frequency can be adjusted between 100kHz and 500kHz
by changing the resistor divider connected network
consisting of R18 and R19.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by Equation 3:
Figure 5. Switching Frequency Adjustment
L=
The following formula gives the estimated switching
frequency:
R19
Eq. 2
fSW _ ADJ = fO ×
R18 + R19
Eq. 3
VIN(max) × fsw × 20% × IOUT(max)
where:
fSW = switching frequency, 300kHz
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
Where fO = Switching Frequency when R18 is 100k and
R19 being open, fO should be typically 500kHz. For more
precise setting, it is recommended to use the following
graph.
Switching Frequency
ΔIL(pp) =
500
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × L
Eq. 4
R18 = 100k, IOUT =1A
450
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
VIN = 48V
400
SW FREQ (kHz)
VOUT × (VIN(max) − VOUT )
350
300
VIN = 75V
250
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
200
Eq. 5
150
100
The RMS inductor current is used to calculate the I2R
losses in the inductor.
50
0
10.00
100.00
1000.00
10000.00
R19 (k Ohm)
2
IL(RMS) = IOUT(max) +
ΔIL(PP)
12
2
Eq. 6
Figure 6. Switching Frequency vs. R19
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
The evaluation board design is optimized for a switching
frequency of 250kHz. If the switching frequency is
June 2011
17
M9999-060311-B
Micrel, Inc.
MIC28500
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
high frequency operation of the MIC28500 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 7:
PINDUCTOR(Cu) = IL(RMS)2 × RWINDING
2
ΔIL(PP)
⎛
⎞
2
⎟ + ΔIL(PP) × ESR C
ΔVOUT(pp) = ⎜⎜
OUT
⎟
×
×
C
f
8
⎝ OUT SW
⎠
Eq. 10
As described in the “Theory of Operation” subsection in
Functional Description, the MIC28500 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be 20%
greater for aluminum electrolytic or OS-CON. The output
capacitor RMS current is calculated in Equation 11:
Eq. 7
PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 8
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
ICOUT (RMS) =
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are ceramic, low-ESR aluminum electrolytic, OSCON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
ΔVOUT(pp)
ΔIL(PP)
ΔIL(PP)
Eq. 11
12
The power dissipated in the output capacitor is:
2
PDISS(COUT ) = ICOUT (RMS) × ESR COUT
Eq. 12
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
Eq. 9
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
June 2011
)
where:
COUT = output capacitance value
fSW = switching frequency
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature:
ESR COUT ≤
(
18
M9999-060311-B
Micrel, Inc.
MIC28500
ΔVIN = IL(pk) × CESR
ΔVFB(pp) ≈ ESR × ΔIL (pp)
Eq. 13
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
ICIN(RMS) ≈ IOUT(max) × D × (1 − D)
Eq. 17
Eq. 14
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)2 × CESR
Eq. 15
Figure 7a. Enough Ripple at FB
Ripple Injection
The VFB ripple required for proper operation of the
MIC28500 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC28500 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
As shown in Figure 7a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
ΔVFB(pp) =
R2
× ESR COUT × ΔIL (pp)
R1 + R2
Figure 7b. Inadequate Ripple at FB
Eq. 16
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 7b. The typical Cff value is between 1nF and
22nF.
With the feedforward capacitor, the feedback voltage
ripple is very close to the output voltage ripple:
June 2011
Figure 7c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 7c.
19
M9999-060311-B
Micrel, Inc.
MIC28500
The injected ripple is:
ΔVFB(pp) = VIN × K div × D × (1 - D) ×
K div =
R1//R2
R inj + R1//R2
1
fSW × τ
Setting Output Voltage
The MIC28500 requires two resistors to set the output
voltage as shown in Figure 8.
Eq. 18
Eq. 19
where
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
Figure 8. Voltage-Divider Configuration
The output voltage is determined by Equation 23:
In Equations 18 and 19, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
fSW × τ
=
T
τ
<< 1
R1 ⎞
⎛
VO = VFB × ⎜1 +
⎟
⎝ R2 ⎠
where, VFB = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
Eq. 20
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 22nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
22nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 21:
K div =
ΔVFB(pp)
VIN
×
fSW × τ
D × (1 − D)
R2 =
1
− 1)
K div
Eq. 24
Thermal Measurements
Measuring the IC’s case temperature is recommended to
ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
Eq. 21
Eq. 22
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
June 2011
VFB × R1
VOUT − VFB
The inverting input voltage VINJ is clamped to 1.2V. As
the injected ripple increases, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected back as a DC error on the FB
terminal.
Then the value of Rinj is obtained as:
R inj = (R1//R2) × (
Eq. 23
20
M9999-060311-B
Micrel, Inc.
MIC28500
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
External VIN Limiter Circuit
The external VIN limiter circuit can be implemented either
on EN pin or VDD pin. Only one of these VIN limiter
circuits is required. The external VIN limiter circuit limits
the minimum input to 30V. If the minimum input in
certain applications is more than 30V then neither of
these limiter circuits is needed. Enabling the device
below 30V VIN and under maximum loading could heat
up the device beyond safe operating conditions. The
following figures show the external VIN limiter circuit on
EN and VDD pins Figure 9A and Figure 9B respectively.
The VIN limiter on EN consists of D5, R22, D6, R23 and
VIN limiter on the VDD pin along with VDD supply
regulator consists of D4, R14, D2, and Q1.
Figure 9B: VIN Limiter On VDD pin
Figure 9A: VIN Limiter On EN pin
June 2011
21
M9999-060311-B
Micrel, Inc.
MIC28500
placed in parallel with the input capacitor.
PCB Layout Guidelines
•
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths. Thickness of the copper planes
is also important in terms of dissipating heat. The 2oz
copper thickness is adequate from thermal point of view
and also thick copper plain helps in terms of noise
immunity. Keep in mind thinner planes can be easily
penetrated by noise
•
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
Inductor
The following guidelines should be followed to insure
proper operation of the MIC28500 converter.
IC
•
The 2.2µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
•
The signal ground pin (SGND) must be connected
directly to the ground planes. Do not route the
SGND pin to the PGND Pad on the top layer.
•
Place the IC close to the point of load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
•
Keep the inductor connection to the switch node
(SW) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the lowside MOSFET.
•
To minimize noise, place a ground plane underneath
the inductor.
•
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
Input Capacitor
•
Place the input capacitor next to the power pins.
•
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
Keep both the PVIN pin and PGND connections
short.
•
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
June 2011
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
RC Snubber
• Place the RC snubber on either side of the board
and as close to the SW pin as possible.
22
M9999-060311-B
Micrel, Inc.
MIC28500
Evaluation Board Schematic
Figure 10. Schematic of MIC28500 Evaluation Board
(J9, J10, J11, R13, R15 are for testing purposes)
June 2011
23
M9999-060311-B
Micrel, Inc.
MIC28500
Bill of Materials
Item
C1
C2, C3
C13
C6
Part Number
EEU-FC2A101B
GRM32ER72A225KA35L
C8, C9
GRM32ER60J107ME20L
Murata
12106D107MAT2A
AVX(4)
06035C104KAT2A
AVX
GRM188R71H104KA93D
GRM188R72A104KA35D
AVX
AVX
TDK
06035C472KAT2A
AVX
Open
Open
C14, C15
Open
Murata
TDK
BAT46W-TP
MCC(5)
BAT46W-7-F
Diodes Inc. (6)
MMXZ5232B-TP
CMDZ5L6
1
2.2µF Ceramic Capacitor, X7R, Size 1210, 100V
2
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1
0.1µF Ceramic Capacitor, X7R, Size 0603, 100V
1
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V
2
1nF Ceramic Capacitor, X7R, Size 0603, 100V
1
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
1
1µF Ceramic Capacitor, X7R, Size 0805, 10V
1
Small Signal Schottky Diode
1
5.6V Zener Diode
1
Murata
C1608X7R2A472K
C2012X7R1A105K
100µF Aluminum Capacitor, SMD, 100V
Murata
06031C102KAT2A
GRM21BR71A105KA01L
Qty.
TDK
TDK
C7
D2
Murata
C1608X7R2A102K
C4, C5
D1
Murata
0805ZC225MAT2A
GRM21BR71A225KA01L
Description
TDK
TDK
GRM188R71H472KA01D
C16
Murata
C1608X7S2A104K
GRM188R72A102KA01D
C12
Murata
(1)
(2)
TDK(3)
C2012X7R1A225K
C11
Panasonic
C3225X7R2A225KT5
C1608X7R1H104K
C10
Manufacturer
MCC
(7)
Central Semi
1
D5
CMDZ24L-MIC
Central Semi
24V Zener
1
D6
CMDZ3L6-MIC
Central Semi
3.6V Zener
1
D4
Open
L1
DR125-100-R
Cooper Bussmann(8) 10µH Inductor, 5.35A RMS, 7A Saturation Current
Diodes Inc/ZETEX
1
100V NPN Transistor
1
Q1
FCX493
R1
CRCW06034R75FKEA
Vishay Dale
4.75Ω Resistor, Size 0603, 1%
1
R2, R16
CRCW08051R21FKEA
Vishay Dale
1.21Ω Resistor, Size 0805, 1%
2
R3
CRCW060319K6FKEA
Vishay Dale
19.6kΩ Resistor, Size 0603, 1%
1
R4
CRCW060310K0FKEA
Vishay Dale
10kΩ Resistor, Size 0603, 1%
1
R5
CRCW060380K6FKEA
Vishay Dale
80.6kΩ Resistor, Size 0603, 1%
1
R6
CRCW060340K2FKEA
Vishay Dale
40.2kΩ Resistor, Size 0603, 1%
1
June 2011
24
M9999-060311-B
Micrel, Inc.
MIC28500
Bill of Materials (Continued)
Item
Part Number
Manufacturer
Description
Qty.
R7
CRCW060320K0FKEA
Vishay Dale
20kΩ Resistor, Size 0603, 1%
R8
CRCW060311K5FKEA
Vishay Dale
11.5kΩ Resistor, Size 0603, 1%
1
R9
CRCW06038K06FKEA
Vishay Dale
8.06kΩ Resistor, Size 0603, 1%
1
R10, R23
CRCW06034K75FKEA
Vishay Dale
4.75kΩ Resistor, Size 0603, 1%
1
R11
CRCW06033K24FKEA
Vishay Dale
3.24kΩ Resistor, Size 0603, 1%
1
R12
CRCW06031K91FKEA
Vishay Dale
1.91kΩ Resistor, Size 0603, 1%
1
R13, R24
CRCW06030000Z0EAHP
Vishay Dale
0Ω Resistor, Size 0603
2
R14
CRCW080510K0JNEA
Vishay Dale
10kΩ Resistor, Size 0805, 1%
1
R15
CRCW060349R9FKEA
Vishay Dale
49.9Ω Resistor, Size 0603, 1%
1
R17 (OPEN) CRCW0603715RFKEA
Vishay Dale
715Ω Resistor, Size 0603, 1%
R18, R19
CRCW0603100KFKEAHP
Vishay Dale
100kΩ Resistor, Size 0603, 1%
2
R20
1
CRCW06032R00FKEA
Vishay Dale
2Ω Resistor, Size 0603, 1%
1
R21 (OPEN) CRCW060333K2FKEA
Vishay Dale
33.2kΩ Resistor, Size 0603, 1%
1
R22
Vishay Dale
36.5kΩ Resistor, Size 0603, 1%
1
75V/4A Synchronous Buck DC-DC Regulator
1
U1
CRCW060336K5FKEA
MIC28500YJL
Micrel. Inc.
(9)
Notes:
1.
Panasonic: www.panasonic.com.
2.
Murata: www.murata.com.
3.
TDK: www.tdk.com.
4.
AVX: www.avx.com.
5.
MCC: www.mccsemi.com.
6.
Diode Inc.: www.diodes.com.
7.
Central Semi: www.centralsemi.com.
8.
Cooper: www.cooperbussman.com.
9.
Micrel, Inc.: www.micrel.com.
June 2011
25
M9999-060311-B
Micrel, Inc.
MIC28500
PCB Layout
Figure 11. MIC28500 Evaluation Board Top Layer
Figure 12. MIC28500 Evaluation Board Mid-Layer 1 (Ground Plane)
June 2011
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M9999-060311-B
Micrel, Inc.
MIC28500
PCB Layout (Continued)
Figure 13. MIC28500 Evaluation Board Mid-Layer 2
Figure 14. MIC28500 Evaluation Board Bottom Layer
June 2011
27
M9999-060311-B
Micrel, Inc.
MIC28500
Recommended Land Pattern
June 2011
28
M9999-060311-B
Micrel, Inc.
MIC28500
Package Information
28-Lead 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2010 Micrel, Incorporated.
June 2011
29
M9999-060311-B