AN1680/D Design Considerations for Clamping Networks for Very High Voltage Monolithic Off-line PWM Controllers http://onsemi.com Prepared by: Christophe BASSO MOTOROLA SPS BP-1029, Le Mirail 31023 Toulouse France email: [email protected]; Tel.: 33 5 61 19 90 12 APPLICATION NOTE INTRODUCTION these devices and how to predict its efficiency in the application. In the large family of Switch-Mode Power Supply (SMPS) components, the recently introduced high-voltage monolithic switchers start to play an important role. First of all because they provide an easy mean to instantaneously build an efficient off-line supply but also because their internal structure offers everything a designer needs: internal clock, pulse-by-pulse limitation, Leading Edge Blanking (LEB) etc. However, the internal MOSFET exhibits a low-energy capability body-diode which no longer protects the device against accidental avalanche. This element thus needs an adequate protection network against the electromagnetic leakage energy. This paper details what network is best adapted to the protection of The Leakage Inductance Figure 1. shows a transformer wound across a standard magnetic material. The primary side made of Np turns creates the necessary force F which gives birth to two components: fm who links both windings, but also fl1 which does not couple to the secondary and corresponds to a leakage path through the air. Thanks to fm, a current Io circulates in the secondary, but this current also gives birth to another leakage flux fl2 whose polarity is opposite of that of fm. It is important to note that fm produces Io while fl2 is a consequence of it. Ip Io fl1 fl2 Np Ns Vexcitation Vo fm Magnetic material with a linear reluctance ℜm Figure 1. A Two-winding Transformer Showing the Leakage Paths As one can see from the picture, φl1 and φl2 close through the air. As any (magnetic) medium, the air is affected by a Reluctance ℜ, or its inverse, the permeance P. These permeances create in the primary and secondary two leakage inductance with a value of: Lleak = N2 ⋅ Pair , with N © Semiconductor Components Industries, LLC, 2000 September, 2000 − Rev. 0 being the primary or secondary turns. As an effect, these parasitic leakage elements degrade the energy transfer between the primary and secondary (ies). In a FLYBACK converter, the presence of the leakage element will a) generate a voltage spike at turn-off and b) divert a portion 1 Publication Order Number: AN1680/D AN1680/D primary and leakage coils. When the ON periods stops, the MOSFET opens and interrupts its current. Since no current discontinuities can take place in an inductor, both magnetic fields collapse and the voltage across the inductances reverses in an attempt to keep the amps-turn constant: Lp energy is thus coupled to the secondary and gives birth to the output current charge. Since Lleakage cannot find a circulating path, it pulls-up 2 the drain voltage until Dclamp starts to conduct and protects the switcher at a maximum theoretical level of: Vmains + Vclamp = 650 V. Figure 2. (b) shows the results of an INTUSOFT’s IsSpice4 (San Pedro, CA) simulations. When all the leakage energy is released, a short parasitic oscillation takes place involving Lleakage and all the parasitic capacitive elements present in the circuit (transformer’s primary capacitance, MOSFET’s Coss etc.) of the primary current into the clamping network. Point a) implies the use of protection network to prevent a lethal drain voltage excursion while point b) explains the root of a degraded open-loop gain: the peak current needs to be higher than theoretically calculated to deliver the full rated output power. The Principle of a Protection Network The goal of the clamping network is to prevent the drain voltage to exceed a given limit. For instance, in the new ON Semiconductor MC3337X family, the maximum voltage shall stay within 700 V. A worse case arises when the mains is at its highest level, e.g., 285 VAC in a universal mains application. To prevent the drain from reaching this value, Figure 2. (a) shows how a perfect network would work: when the MOSFET closes, the current builds-up in both RATIO_POW = −0.183 RATIO_AUX = −0.142 MC33374 Vclamp 250 V R6 3.6 C1 50 mF C2 1 mF C4 100 n R7 75 k 13 1 Rp 100 m 17 Aux 16 D1 1N4146 R8 15 k C5 10 mF Lleak 4 mH MOS GND VDCmains 330 V CTL VCC + FB 2 L3 5 mF UP TI Lp 250 mH + 3 Dclamp Int 4 7 MC 3337X X2 MBR20100 10 C3 150 mF Int 6 R12 270 Aux 12 C6 10 nF 14 MOC8101 15 UP TI 16 Figure 2. (a) A Simple FLYBACK Configuration Implementing a Clamping Network http://onsemi.com 2 Rload 5 Vout AN1680/D 700 500 VDS CLIPPING EFFECT X = 212.29U Y = 650.84 Fosc + 300 1 2@p@ ǸLleak @ Clump 100 −100 210.20U 211.20U 212.20U 213.20U 214.20U Figure 2. (b) The Drain is Safely Clipped Below 700 V at High Mains Diverting the Primary Current where the secondary diode catches-up with the primary current. The slope of the decreasing primary current is simply Figure 2. (a) is interesting because it helps understanding how the reflected secondary voltage resets the leakage energy and how much of primary current this leakage inductance “steals away” by diverting it into the clamp. Everything is detailed on Figure 2. (c) graph. When the MOSFET turns-off, a reset voltage is applied to the leakage inductance. This reset voltage depends on the clipping voltage but also on the FLYBACK’s. The higher this level, the faster the leakage energy drops to zero and authorizes the secondary current to take place. The time Δt needed to complete the energy transfer is easily defined by: Dt + N @ (Vout ) Vf sec) , Lp but this equation can also be written as: N @ (Vout ) Vfsec) Ip * Ipx . + Dt Lp Replacing Δt and solving for Ipx gives: Ipx +1* Ip Lp @ L @ Ip leak , V * (Vout ) Vf sec) @ N clamp ǒ L leak . V clamp *1 (V )Vf)@N out This last equation gives you the effective percentage of primary current stolen by the leakage inductance. Applying Figure 2. numerical values gives: Ipx = 98.4% of Ip. Since Ip grows up to 2.73 A, then the theoretical peak secondary current establishes at: 0.984 ⋅ 2.73 ⋅ 12.5 = 33.58 A. Figure 2. (d) validates the calculation. where Ip is the final primary current, N the transformer ratio secondary to primary, Vfsec the secondary diode forward drop and Lleak the primary leakage inductance. Estimating the percentage of diverted current tells you the real peak current you will actually put in the primary to deliver the rated power. Figure 2. (c)’s Ipx point shows Iprimary Ip Ipx Vin/Lp Vo @ N Lp ILleak Iprimary ILleak Isecondary ton Ǔ Vclamp * Vo @ N L leak Dt Figure 2. (c) Waveforms at Turn-off: the Leakage Coil Prevents an Immediate Transfer http://onsemi.com 3 AN1680/D 30 x = 202.37U y = 33.600 VCC + Vz Isec. peak VO.N Ip = 2.73 A 20 10 Vdrain-source Ileakage Iprimary 0 Vout @ turn-off = 10 V 200.60U 201.79U Ip 202.98U 204.17U 205.37U Izener diode Figure 2. (d) IsSpice4 Simulation of Figure 2’s Circuit Dt Protecting the MOSFET with an Active Element The easiest way to clamp at a known level is to replace the null-impedance Vclamp source by a zener diode or a transient suppressor. Figures 4. (a) and 4. (b) detail the connections and their associated waveforms. Since we have two diodes in series, we have to care about both dissipated powers. Diodes can be modeled by a voltage source V (which equals Vzener or Vforward) in series with a dynamic resistance Rd. The total average conducted power dissipated is therefore: Figure 3. (b) Clipping Waveforms with a Zener Diode To account for the Rd term, MOTOROLA specifies a clamping factor FC which gives the real peak zener voltage at a given peak current: VZ(Peak) = VZ(Nom) . FC. From this formula, we can write: Vz(Nom) + Rd ⋅ Iz(peak) = Fc ⋅ Vz(Nom). Solving for Rd gives: 2 (Fc * 1) @ V z(Nom) , Rd + P PK(Nom) Pavg = V ⋅ Iavg + Rd ⋅ I2rms. For the zener diode, the first part is easily deducted from Figure 3. (b): with PPK(Nom) being the maximum peak power accepted by the zener diode or the transient suppressor. From Figure 3(b), let’s now calculate the RMS and average values of the zener current: V @ Ip Dt Pavg1 + z @ 2 T or, once solving with Δt: I zener(t) + Ip @ Dt * t, Dt Vz @ Ip 2 @ L @F leak , P avg1 + 2 @ (Vz * (Vout ) Vfsec) @ N) to obtain the RMS value simply solve: Ǹŕ with F the switching frequency and Vz the nominal zener level. 8 5 Ǹ3Dt@ T, T is the switching period. Pavg2 is thus: Vout P avg2 + Cout Lp 4 I 2zener(t).dt or I zenerRMS + Ip @ 0 1 Dclamp 1 T D1 N:1 DCrail Dt Rd @ Ip 2 @ F @ Dt. 3 The IzenerAVG value, which affects the conduction losses of the diode in series with the zener is evaluated by: 2 R1 I zenerAVG + Ip @ Dt @ F . 2 As we previously wrote, the diode conduction losses are expressed the same way as the zener’s, except that Lleak Rd + dVf dId extracted from the Vf versus Id curve in the diode data-sheet at Id = Ip. To summarize we have: 7 Ip 2 @ L @ F @ (V z ) 0.66 @ Rd @ Ip) leak 2 @ (V z * N @ (V out ) Vf sec)) Ip 2 @ L @ F @ (V ) 0.66 @ Rd @ Ip) leak Ip f P + cond_diode 2 @ (V z * N @ (V out ) Vf sec)) P Drv Figure 3. (a) Clipping with a Zener is Possible Solution cond_zener http://onsemi.com 4 + AN1680/D a 180 V zener diode, then the series diode starts to clip at 605 V. However, because the injection time into the low-doped N-region takes about 50 ns, the dynamic resistance of the MUR160 gradually drops to its nominal value, accordingly generating an overshoot upon the drain. Measurements have to be carried upon the final board to confirm the safety of the final drain level. The final clipping level will be affected by two components: the zener clamping factor FC but also the time the series diode takes to react. If we select a fast MOTOROLA MUR160, the data-sheet specifies a turn-on time of 50 ns. In presence of drain voltage rising with a 1.5 kV/μs slope, the diode will start to conduct at VmainsDC + Vz . FC. If we take the highest mains level of 275VAC and RATIO_AUX = −0.156 RATIO_AUX = −0.156 Rclamp 100 k RC CLAMPING NETWORK 14 R7 15 k 13 9 D4 1N4148 LI1 58 m 16 10 MC33363 Vaux R5 110 k 13 6 11 7 10 8 9 Vclamp + Vin (Passive Network) BVDSS (MOSFET Alone) Aux Vdrain 12 Rload 9.8 6 1 3 5 C10 470 mF 8 Lp 3.56 M 1 4 Vout 17 2 12 X1 MC33363 VDCmains 320 Aux D1 1N5821 R1 3.5 Dclamp 1N4937 + 4 10 Cclamp 1 nF test Overvoltage Protect. Vout.N R8 4.7 k Ip 11 20 C2 10 mF C4 7 100 nF R6 24 k 16 R10 921 C6 C5 820 p 100 nF Dt Figure 4. (a) An RC Network Clamps the Drain Voltage at Turn-off 0 Figure 4. (b) Turn-off Waveforms with a MOSFET Clipping the Peak When to Use a Zener or a Transient Suppressor? A Passive RC Network to Clamp the Drain If the above solution provides a stable clamping level rather independent from peak current variations, the cost of those zener elements can degrade the overall price of your SMPS. The alternative lies in implementing a passive RC network as the one depicted in Figure 4. . If we assume we do not have any external clipping network, we can calculate the amount of energy ET dissipated in the transistor every time it opens, assuming it would safely avalanche the voltage (Figure 4. (b)). As we said, the leakage tries to keep the current circulating at its level (Ip, when the transistor opens) during Δt and pushes the drain voltage up to BVDSS. Ip(t) can be expressed by: There are little technology differences behind a standard zener diode and a transient. However, the die area is far bigger for a transient suppressor than that of zener. A 5 W zener diode like the 1N5388B will accept 180 W peak power if it lasts less than 8.3 ms. If the peak current in the worse case (e.g., when the PWM circuit maximum current limit works) multiplied by the nominal zener voltage exceeds these 180 W, then the diode will be destroyed when the supply experiences overloads. A transient suppressor like the P6KE200 still dissipates 5 W of continuous power but is able to accept surges up to 600 W @ 1 ms. If the peak power is really high, then turn to a 1.5KE200 which accepts up to 1.5 kW @ 1 ms. Ip(t) + Ip @ Dt * t. Dt http://onsemi.com 5 AN1680/D You calculate the energy by integrating over time the cross-over area between current and voltage: It is also sometimes interesting to know the clamp voltage set by known RC elements. In that case, simply apply: Dt E + T ŕ Id(t) @ VDS(t) @ dt + 12 @ Ip @ BVDSS @ Dt 0 V + 1 @ Vout @ N ) 1 @ clamp 2 2 Ǹ Vout . If we now introduce the term Δt previously calculated, we get: P +1@ V @ Ip @ Dt @ F clamp clamp 2 . Δt has already been defined, but this time, BVDSS is replaced by Vclamp. Once introduced in the previous equation, we obtain the power dissipated in the clipping network: V clamp + 1 @ Ip 2 @ L @F@ clamp leak 2 V * (Vout ) Vfsec)N clamp Since this power will mainly be dissipated by the resistor Rclamp at steady-state, we can write the following equality: 2 V V clamp clamp + 1 @ Ip 2 @ L @F leak 2 R V * (Vout ) Vf sec)N clamp clamp . By solving for Rclamp we calculate its value for a given level of clamping voltage: clamp + 2@V @ (V * (Vout ) Vf sec) @ N) clamp clamp L @ Ip 2 @ F leak It is important to minimize the ripple level Vripple superimposed on Vclamp. A capacitor Cclamp will fulfill this function. If we agree that the amount of charge Q will equally split between Rclamp and Cclamp at turn-off we can write: Vripple . Cclamp = IRclamp. T. Knowing that IRclamp = Vclamp / Rclamp and then solving for Cclamp it comes: C clamp + V ripple V clamp @F@R clamp In steady-state operation, by neglecting the RMS current circulating in Rclamp, the RMS current which flows in the capacitor is the same as in the fast series diode: I C + Ip @ RMS clamp clamp @L leak @ Ip 2 @ F If RC networks are more attractive than zeners when talking about price, they suffer from a poor behavior when the peak current varies. As a matter of fact, the clamping level will always be calculated at the highest primary current. The highest primary current depends, of course, on the internal current limit (at the highest TJ) but also from the turn-off response time due to the over-current comparator propagation delay. For the MC3337X family, this value is typically 280 ns, while the maximum current limit increases by roughly 3.5% at the maximum operating temperature. As an example, let us take a primary inductance of 290 μH. With this value in mind, a high mains of 285 VAC imposes a slope of 1.38 A/μs when the MOSFET closes. The maximum current limit of the MC33374 is set at 3.7 + 3.5% = 3.83 A. With the previous slope and a total delay of 280 ns, the switch will close when the current finally reaches: 3.83 A + 0.28 μs ⋅ 1.38 = 4.21 A. To have an idea of the peak current dependency of the RC network’s clamp level, we built a 13 W power supply supposed to operate on wide mains. Despite a 400 mA nominal peak current, we used an MC33373 to operate without an heatsink. MC33373 authorizes peak currents up to 3 A. The RC network was specified to clamp at 240 V @ Ip nominal and gave values of 11 kW and 100 nF. With a 118 μH leakage inductance and a 13.8 turn ratio, the normal operating clamping voltage was measured at 238 V. However, as one can see from Figure 5. , the clamping level grows-up by adding successive voltage steps, corresponding to every switch turn-off. At power-on, Vclamp and Vout are both at zero. The internal error amplifier pushes the duty-cycle toward 70% but each cycle is fortunately truncated when Idrain exceeds the internal limitation. Vclamp starts to grow and keeps rising until Vout nominal is reached and forces the PWM modulator to brake: Vclamp then diminishes to establish at the calculated 240 V. If for some reasons maximum Ipeak conditions would stay longer than expected (e.g., Vout could not reach it nominal value), Vclamp would continue its grow, no longer protecting the internal MOSFET. To avoid this condition, Figure 5. (b)’s plot depicting Vclamp = f (Ip) will help the designer to track worse case conditions and react by either lowering Rclamp or simply selecting a member of the MC3337X family exhibiting a maximum peak current of 400−500 mA (e.g., MC33369). Another solution is to use an adequate zener diode whose clamping level will be less sensitive to peak current variations. This result depicts the average power the transistor would be the seat of, if no mean were implemented to re-route the energy spike elsewhere. By wiring an RC network from drain to VDCmains via a fast diode, we will prevent the drain voltage to rise above VDCmains + Vclamp, the clipping voltage we want to impose (Figure 4. (b)). Let us first consider that the voltage across the RC network is constant (ripple is low compared to average voltage) and equals Vclamp. The average power dissipated at turn-off in the clamp is: R @ N2 ) 2 @ R Watch-out for the Current Dependency! BV DSS . P + 1 @ Ip 2 @ L @F@ T leak 2 * (Vout ) Vf sec) @ N BV DSS P 2 Ǹ3Dt@ T http://onsemi.com 6 AN1680/D V clamp (Ip) 1000 500 0 Figure 5. (a) Turn-on Sequence Showing Vclamp Running Away to Large Values Selecting the Right Active Components 0 1 2 3 Figure 5. (b) This Plot Shows How Vclamp Moves with Ip To illustrate this point, Figure 6. shows a little spike superimposed upon the drain voltage at turn-off due to this diode turn-on time. This shot is directly taken from Figure 4. circuit. Once again, this spike should never exceeds BVDSS. The series diode should be fast enough to clamp as soon as the drain voltage exceeds Vclamp + VDCrail. As you can understand, the switching time should be selected accordingly with the drain voltage rising slope, dVDS / dt. Figure 6. The Diode Turn-on Time Allows VDS to Continue its Rise Before it Actually Clamps The zener clamping level must be selected to be between 40 to 80 volts above the reflected output voltage when the supply is heavily loaded. The given formulae will help you determining the average losses of the zener but also its maximum peak power (e.g., during power-on). http://onsemi.com 7 AN1680/D The following ON Semiconductor references can be used as active clipping elements: Reference Nominal Voltage (V) Average Power (W) Maximum Peak Power 1N5953B 150 1.5 98 W @ 1 ms 1N5955B 180 1.5 98 W @ 1 ms 1N5383B 150 5 180 W @ 8.3 ms 1N5386B 180 5 180 W @ 8.3 ms 1N5388B 200 5 180 W @ 8.3 ms P6KE150A 150 5 600 W @ 1 ms P6KE180A 180 5 600 W @ 1 ms P6KE200A 200 5 600 W @ 1 ms 1.5KE150A 150 5 1.5 kW @ 1 ms 1.5KE180A 180 5 1.5 kW @ 1 ms 1.5KE200A 200 5 1.5 kW @ 1 ms X = Full Scale Span Another benefit of using a zener diode is to limit the inverse voltage applied upon the fast series diode during turn-on at VRRM = VDCmains. In the presence of an RC clamping network, the clamping level unfortunately adds to the input voltage and forces the adoption of a diode with better VRRM parameter: VRRM = VDCmains + Vclamp. Depending on the dVDS/dt slope and maximum reverse voltage conditions, the following MOTOROLA references can be used: Reference VRRM Ton (typical) IF max MUR160 600 V 50 ns 3A MUR100E 1000 V 25 ns 3A 1N4937 600 V 200 ns 1A MSR860* 600 V 100 ns 8A MSRB860-1* 600 V 100 ns 8A * soft recovery diodes Parameter Evolutions with Temperature variations with the ambient temperature, they must not jeopardize the SMPS operation. The below lines give an idea on the way these parameters move with temperature: Once protected, the equipment must survive in its operating environment. That is to say, despite the component’s key specs (e.g., Tfr for the series diode) Conclusion MC3337X’s MOSFET BVDSS: positive temperature coefficient MC3337X’s maximum current limit: +1.03% at TJ = 125°C MC3337X’s oscillator frequency: 115 kHz at max TJ Zener breakdown voltage: positive temperature coefficient Diode reverse recovery time: positive temperature coefficient Data-sheets can be downloaded http://mot-sps.com/books/current.html In FLYBACK converters, leakage inductance always stresses the switching elements. An efficient active or passive protection network is, therefore, mandatory to ensure the SMPS will survive in any operating conditions. This paper details some possibilities on how to implement the adequate level of protection. at: Spice models of diodes and MC3337X: http://mot-sps.com/models/html/models.html http://onsemi.com 8 AN1680/D Notes http://onsemi.com 9 AN1680/D Notes http://onsemi.com 10 AN1680/D Notes http://onsemi.com 11 AN1680/D ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. 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