Implementing the SOFA in AC/DC Power Supplies

AND8043/D
Implementing the SOFA in
AC/DC Power Supplies
Prepared by: Christophe Basso
ON Semiconductor
e–mail: [email protected]
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APPLICATION NOTE
INTRODUCTION
The novel ON Semiconductor NCP1205 represents
a real breakthrough in Switch Mode Power Supply
(SMPS) controllers. By combining free–running operation
and a Voltage Controlled Oscillator (VCO), the Soft
Oscillating Free–running chArger or SOFA, allows
true Quasi–Resonant (QR) or valley switching operation
and soft frequency foldback when the output power
demand diminishes. Implementing ON Semiconductor
state–of–the–art Very High–Voltage Integrated Circuitry
(VHVIC), the SOFA will naturally find a place in designs
where soft–switching operation and ease of implementation
are premium, as the following features emphasize:
Full Quasi–Resonant Operation: By detecting the end
of the transformer core demagnetization to initiate a new
cycle, the SOFA ensures drain–source valley switching or
QR operation. Furthermore, thanks a to comprehensive
logic circuitry, the device jumps between the valleys as
the built–in VCO starts to decrease the switching
frequency. As a result, Electromagnetic–Interference
(EMI) are reduced and turn–on losses are virtually null.
Voltage–Controlled Oscillator: An internal VCO takes
over as soon as the free–running frequency hits a
maximum user adjustable value. As the output power
demand further diminishes, the switching frequency is
naturally reduced to ensure a better efficiency at light
loads.
Low Standby–Power: If SMPS naturally exhibit a good
efficiency at nominal load, they begin to be less efficient
when the output power demand vanishes. By smoothly
reducing the number of switching cycles per second, the
SOFA drastically reduces the power wasted during light
load conditions. In no–load conditions, the SOFA allows
the total standby power to easily reach and exceed next
International Energy Agency (IEA) recommendations.
Short–Circuit Protection: By permanently monitoring
the feedback line activity, the IC is able to detect the
presence of a short–circuit, immediately reducing the
output power for a total system protection. Once the short
 Semiconductor Components Industries, LLC, 2002
January, 2001 – Rev. 0
1
has disappeared, the controller resumes and goes back to
normal operation. For given applications (e.g. constant
output power supplies), you can easily disconnect this
protective feature.
Over–Voltage Protection: By continuously checking its
own Vcc rail, the SOFA can safely go into permanent
latch–off phase when the operating voltage exceeds 36 V.
In Forward winding applications, this options lets you
also protect the design against transient mains over
voltages. For application where an adjustment is
necessary, the SO–16 versions pins out the dedicated
comparator input to let you select the protection level of
your choice.
Large Supply Range: Battery charger applications
require that the controller can still control the output
current when the output voltage is close to zero (e.g. a
discharged battery). This is called Constant–Current/
Constant–Voltage (CC–CV) operation. To allow the
controller self–supply when the output voltage
disappears, one needs to wire the auxiliary winding in the
Forward mode. However, most of today’s primary side
controllers have difficulty to cope with a Forward
auxiliary winding operated on a universal mains because
of the large voltage dynamics it implies. Fortunately, by
authorizing 7.0 V through 36 V operation, the SOFA eases
the designer task on the self–supply side.
Low Output Ripple in Standby: Some loads are
sensitive to the ripple present on the output. This is the
case for Li–Ion batteries where a clean voltage is required
to ensure the longest service. Standard hysteretic
controllers produce unacceptable output ripple. By
smoothly reducing the operating frequency, the SOFA
generates a lower ripple when entering the standby mode.
No Acoustic Noise While Operating: Instead of
reducing the switching frequency at high peak currents,
the SOFA waits until the peak current demand falls below
a fixed 1/3rd of the peak maximum limit. As a result,
frequency reduction takes place without having a singing
transformer … You can thus select cheap magnetic
components free of noise problems.
Publication Order Number:
AND8043/D
AND8043/D
External MOSFET Connection: By leaving the
external MOSFET external to the IC, you can select
avalanche proof devices which, in certain cases (e.g. low
output powers), let you work without an active clamping
network. Also, by controlling the MOSFET gate signal
flow, you have an option to slow down the device
commutation, therefore reducing the amount of
ElectroMagnetic Interference (EMI).
SPICE Model: A dedicated model that lets you run
transient cycle–by–cycle simulations is available to
verify your theoretical design. Ready–to–use templates
can be downloaded in OrCAD’s PSpice, INTUSOFT’s
IsSpice and Spectrum–Software’s µCap from ON
Semiconductor web site, NCP1205 related section.
Vf the diode forward drop. During this time, the primary
current decreases with a slope imposed by the reflected
voltage
N (Vout Vf)
(eq. 2). Figure 2 zooms on the
Lp
primary current, showing how it moves over one switching
cycle.
When the primary current reaches zero, the transformer
core is fully demagnetized: we are in Discontinuous
Conduction Mode (DCM). The primary inductance Lp
together with all the surrounding capacitive elements Clump
create an LC filter. When the secondary diode stops
conducting at Ip = 0, the drain branch is left floating since the
MOSFET is already open. As a result, a natural oscillation
takes place exhibiting the following frequency value:
Fring What is Valley Switching?
Figure 1 depicts a typical FLYBACK converter
drain–source waveform. During the ON time, Vds is close to
zero, the power switch being closed. The input voltage is
applied on the primary inductance Lp and the current ramps
1
2 · · Lp · Clump
(eq. 3) . As any sinusoidal
signal, there are peaks and valleys. When you restart the
switch in the valley, all the parasitic capacitor are at the
lowest possible level and the capacitive losses
1 · Clump · Vds · Fsw (eq. 4) are kept small: the MOSFET
2
V
up with a slope of in (eq. 1). When the controller dictates
is no longer the seat of turn–on losses and the usual turn–on
parasitics are removed. Unfortunately, the perfidious
leakage inductance degrades the beauty of Figure 1 and 2 by
also ringing at the switch opening. The frequency is much
higher than the natural frequency since Lleak is smaller than
Lp. Care must be taken to not cheat the SOFA
demagnetization block by this spurious signal.
Lp
the switch opening (because of the Pulse Width Modulator
or the peak current limit has been reached), the drain–source
quickly rises and the energy transfer between primary and
secondary takes place: the secondary diode conducts and the
output voltage is also seen on the primary, over Lp. This
plateau is equal to Vin + N . (Vout + Vf) where N is the
secondary to primary turn ratio, Vout the output voltage and
Leakage
Inductance
Ipeak
Core
Demagnetized
S = N. (Vout + Vf) / Lp
S = Vin / Lp
N. (Vout + Vf)
Vin
ON
ON
OFF
Ip = 0
0
OFF
Valleys
Figure 1. A Typical FLYBACK Drain–Source
Waveform
Figure 2. The Primary Current Ramps Up and
Down to Zero in DCM
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AND8043/D
Observing the Core Flux
The core flux can easily be observed through an auxiliary winding wired as Figure 3 proposes:
Vaux
6
Vout
1
20.0
10.0
0
–10.0
Gnd
–N. Vin
Vin
–20.0
20.0
Flyback Operation
N. Vin
10.0
3
Flyback Operation
65 mV
0
–10.0
(a)
–20.0
(b)
Figure 3. An auxiliary winding lets you observe the flux image in the transformer’s core
depending on the way you wire it: Flyback or Forward
Thanks to the coupling between the windings, the
auxiliary section will deliver a voltage image of the core’s
toff d
(eq. 5).
flux through the following formula: Vaux N ·
dt
Now, you can wire the winding either in Flyback (as the
power winding) or in Forward. The observed signals look
the same but the polarity is different as Figure 3b depicts.
Please note that both signals are centered around the ground.
We will see later on why battery chargers need Forward
operation for a proper behavior. Figure 3b shows how a
comparator featuring a reference DC level above ground can
easily toggle up and down as soon as DCM occurs. The
internal logic circuitry can then shape this signal to restart
the power switch.
Lp
inDC
· Ip
(Vout Vf)
(eq. 7)
· Ip
1 1
ton toff Ip Lp VinDC Np (V
out Vf)
Ns
1
(eq. 8)
Fsw
P
1
Pin out
2 · Lp · Ip? · Fsw
from eq. 9, Ip out
2Lp PFsw
(eq. 9)
(eq. 10)
Plugging (eq. 8) definition into the above equation (eq. 10):
What is Free–Running Operation or SOPS Mode?
Self–Oscillating Power Supply (SOPS) or free–run
operation portrays a power supply whose power switch is
activated immediately further to the transformer’s core
demagnetization occurrence. The controller thus no longer
needs an internal clock since external elements such as Lp,
Vout, Ip etc. rule the operation. If a small delay is added
further to the demagnetization comparator trip point, then
the switch can be restarted right in the middle of the
sinusoidal valley (See Figure 1). Let’s see how we can
describe the SOPS operation through the following
equations:
ton V
Lp
Np
Ns
Ip ⋅
(eq. 11)
2Pout
1
Ip V1 N(V 1 V
in
out
f)
Solving for Ip
Ip 2 Pout N (Vout Vf) Vin
(eq. 12)
Vin N (Vout Vf)
Now extracting Fsw from eq. 10 and plugging it into eq. 8
leads to the switching frequency expression versus the
output power:
Fsw (eq. 6)
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2 Pout
N(VoutVf)Vin
2
Lp 2 Pout ((V (N(V
in
outVf))))
(eq. 13)
AND8043/D
Feeding a math processor with eq. 13 and watching Fsw
in the Y–axis while moving Pout shows how the switching
frequency changes with power:
CCM
300
L > LC
IL
Not 0 at
I P turn ON
250
kHz
200
0
L = LC
ON
150
W
DCM
OFF
IL(avg)
Fmax
100
L > LC
0 before
turn ON
50
BCM
VCO
0
D/Fs
0
0
2
4
6
8
10
Dead Time
Time
Figure 5. Depending on the primary current at
turn–on, you can define various operating modes
Figure 4. Switching Frequency Evolution
with the Delivered Power
(Lp = 6.0 mH, Vin = 120 VDC, N = 6.5)
When the output power demands decreases, the natural
free–running switching frequency raises (Figure 4). As a
natural result, switching losses also increase and degrade the
SMPS efficiency. To overcome this problem, the maximum
switching frequency of the NCP1205 is clamped to typically
125 kHz when a 1.0 nF capacitor is connected as the timing
capacitor. As soon as the free running mode (also called
Borderline Control Mode, BCM) reaches this clamp value,
the internal VCO takes over and starts to decrease the
switching frequency: we are in Variable Frequency Mode
(VFM). Please note that during this transition phase, the
peak current is not fixed but is still decreasing because the
output power demand does. This ensures a very smooth
transition between free–run and VFM modes. However, the
frequency decrease is subordinate to the appearance of a
valley as we have stated before: if the VCO dictates a switch
restart, the logic latches the information until a valley is
detected. Also, if the VCO timer would ask for a restart, the
logic circuit will wait until the next valley level before
giving the final green light. As a result, in VFM, discrete
leaps between valleys occur and introduce a natural jitter:
during these leaps, the peak current also toggle between
different values to satisfy eq. 9 at every switching cycle. At
a given state, the peak current reaches a minimum ceil
(typically 250 mV/Rsense, eq. 14), and cannot go further
down: the switching frequency continues its decrease down
to a possible extremely low frequency. During normal
free–running operation and VFM, the controller always
ensures single drain–source valley switching. As soon as the
VCO dictates a smaller switching frequency, the controller
waits for the occurrence of the next valley before restarting
the power switching.
As one can see, the switching frequency can grow in
definitively if no clamping mean is provided. In the case of
the SOFA, a VCO will smoothly reduce the switching
frequency down to a few kilo–hertz as soon as the user
selectable maximum frequency is hit.
Theory of Operation
As we have seen before, the SOFA combines free–running
operation with minimum drain–source switching (so–called
valley switching or quasi–resonant operation), which
naturally reduces the peak current stress as well as the final
EMI content at the switch closing. At nominal output power,
the circuit implements a traditional current–mode SMPS
whose peak current set point is given by the feedback signal.
However, rather than keeping the switching frequency
constant, each cycle is initiated by the end of the transformer
core demagnetization. The system therefore operates at the
boundary between Discontinuous Conduction Mode
(DCM) and Continuous Conduction Mode (CCM) while the
switching frequency evolves with the operating conditions.
Figure 5 details this terminology.
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AND8043/D
force a frequency decrease. However, since the internal
logic has no other options than accepting valley transition
only, a second valley switching takes place. As the power
further decreases, the population of second valleys
increases. The peak current jumps between Ip1 and Ip2 to
cope with the discrete frequency transitions within the
cycles: one valley gives Fsw1 and Ip1 adjusts to satisfy
eq. 9, while two valleys give a lower frequency and Ip2
rises during this event, always to satisfy eq. 9.
At a certain moment where Pout has further diminished,
there are only two–valley switching events and a stable
condition takes place. Ip has no other option than going
down, until it is frozen at the very minimum ceil: 250 mV/
Rsense.
The VCO now controls the switching frequency with
fixed current pulses. Fswitching diminishes while other
valley jumps take place. Figure 6 shows a typical
frequency versus power graph for a low power charger
(Pnominal = 5.0 W).
To adjust the transmitted power, the PWM controller can
play on the switching frequency or the peak current set point.
To refine the control and thus offering a truly smooth
transition between VCO and VFM operation, the NCP1205
offers the ability to play on both parameters either altogether
or on an individual basis. In order to clarify the device
behavior, we can distinguish the following operating phases:
The load is at its nominal value, maximum output power.
The SMPS operates in borderline conduction mode
(BCM) and the switching frequency is imposed by the
external elements (Vin, Lp, Ip, Vout). The MOSFET is
turned on at the minimum drain–source level. The peak
current is controlled via the feedback voltage (VFB)
which undergoes an internal inversion: if VFB goes up →
Ip goes down and vice versa.
The load starts to decrease and, as a natural consequence,
the free–running frequency increases until it hits the
maximum level externally set by the timing capacitor. At
this time, the VCO takes over the loop control and tries to
120
Max F switching
SWITCHING FREQUENCY (kHz)
Permanent 2nd valley lp moving down
100
Appearance of the 2nd
valley switching, Vfb = 3 V
(VCO starts)
80
60
Peak current is frozen
here, Vfb = 3.08 V,
3rd Valley appears
40
2nd/no valley
jumping lp jumping
between lp1 and
lp2: delta = 13%
Multiple valleys
appearing until timeout
of 4 µs becomes active
20
Permanent valley
switching with lp
varying
0
0
1
2
3
4
5
6
OUTPUT POWER (W)
Figure 6. This Graph Shows how the Frequency and Current Move
until the Current is Frozen
220 mA
Figure 7 depicts how the waves were captured to plot
Figure 6 graph. The output current (Vout = constant) was
reduced and the drain–source signal was observed then
memorized.
210 mA
200 mA
180 mA
140 mA
Figure 7. Multiple pulses start to occur further to the
transition into VCO control
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AND8043/D
The Decaying Sine Wave
The drain–source wave has a sinusoidal shape
exponentially amortized over time. The decay is dependent
upon the ohmic losses of the equivalent Lp . Clump
composing the ringing circuit. As a matter of fact, the case
quickly arises that no more valleys are available to trigger
the internal SOFA circuitry when the VCO has lowered the
frequency. To circumvent this problem, the device
implements a timeout (timo) which is reset at each valley
occurrence. If the next valley does not appear within a time
window of 4.0 µs, then the timeout gives the authorization
to restart but the SOFA waits for the VCO clock appearance
before actually driving the power MOSFET. Figure 8
portrays different states showing how the pulses evolve with
power and how the timo automatically initiates a new cycle
at the end of the decay period:
Advantages of the Method
By implementing the aforementioned control scheme, the
SOFA brings the following advantages:
1. Discontinuous only operation: in DCM, the
FLYBACK is a first order system (at low frequencies)
and thus naturally eases the feedback loop
compensation.
2. A low–cost secondary rectifier can be used thanks to
soft blocking conditions (Ip goes down to zero and the
diode stops conducting, unlike in CCM where the
power switch turn–on event forces the diode to brutally
stop conducting).
3. Valley switching ensures minimum switching losses
brought by Coss and all the parasitic capacitances.
4. By folding back the switching frequency, you turn the
system into Pulse Duration Modulation. This method
prevents from generating uncontrolled output ripple as
with hysteretic controllers.
5. By letting you control the peak current value at which
the frequency goes down, you ensure that this level is
low enough to avoid transformer acoustic noise
generation even at audible frequencies.
Feedback, Current Set Point and Error Circuitry
The feedback voltage (FB pin) undergoes an internal
inversion before imposing the final current set point. This is
shown on Figure 9. As you can see, the error amplifier output
expression is: Verr 10 3 · VFB using the resistor values
as given on the schematic. The error signal is then divided
by three (to offer a sufficient dynamic on the FB pin) and is
classically clamped to 1.0 V. Now, if you carefully look at
Figure 9, you notice the presence of a diode and 250 mV
reference. The presence of these two elements prevents the
peak current to go below 250 mV/Rsense, Rsense being the
external power sense resistor. As a result, the peak current
can only move between 1.0 V/Rsense and 250 mV/Rsense.
When the 250 mV clamp is activated, it means that the VCO
has already taken the lead of the SMPS. This is depicted by
Figure 10 sketch where you can see the current set point
evolution.
Figure 8. Typical operating waveforms for different
output powers down to no–load operation
(bottom curve)
150 k
1
FB
0.3
50 k
Gain
–
+
2
7
3
+
1.0 V
Vref1
2.5 V
4
+
–
+
5
Current
Setpoint
3V
VCO Freqency
is fixed at Fmax
9
+
250 mV
BCM Mode
Peak current
can change
VCO Freqency
can decrease
Error Flag
1V
0.75 V
Peak Current is Frozen
Vref2
1.5 V
Figure 9. This component arrangement lets you monitor
the (+) signal and detect when it leaves its limits
Figure 10. The current is frozen below a given limit
and cannot go further down
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AND8043/D
Frequency Management
As we have detailed, the power switch restart is initiated
when two information are present: the core demagnetization
has occurred and the internal timer validates it. The VCO
simplified circuitry is shown on Figure 12. In free–run
operation, the timer capacitor is charged at the highest rate
(≈ 350 µA) and is transparent to the circuit: it always gives
the green light before the core’s demagnetization signal
happens. The capacitor timer signal thus ramps up to 5.0 V
and stays there until it is discharged by the driver signal.
During free–run operation, the error voltage controlling the
350 µA current source (in fact that is a voltage–controlled
current source) is clamped to 1.0 V. As soon as the internal
feedback voltage (the error amplifier output) goes below
1.0 V, then the charging current obeys the law Icharge =
Verror x 350 µA, down from 350 µA (Verror = 1.0 V) to
nearly zero (Verror ≈ 0). The switching frequency at a given
Ct is thus calculated knowing the capacitor swing (500 mV
to 3.0 V) and the maximum charging current of 350 µA:
Actually, the SOFA features an internal error amplifier
solely used to detect an over current problem (OCP). The
internal error amplifier is used to create a virtual ground
permanently biased at 2.5 V, an internal reference level. By
monitoring this virtual ground further called V(–), we have
the possibility to confirm the good behavior of the loop. If
by any mean the loop is broken (shorted optocoupler, open
LED etc.) or the regulation cannot be reached (true output
short–circuit), the OPAMP network is adjusted in order to no
longer be able to ensure the 2.5 V virtual point V(–). If V(–)
passes down the 1.5 V level (e.g. output shorted) for a time
longer than 128 ms, then the pulses are stopped for
8 x 128 ms. The controller enters a kind of burst mode with
bunch of pulses lasting 128 ms (typical with Ct = 1.0 nF) and
repeating every 8 x 128 ms. If the loop is restored within the
8 x 128 ms period, then the pulses are back again on the
output drive (synchronized with UVLOH). Figure 11 depicts
a scope shot of the drain–source pulses in presence of a short:
t Ct ·
V or 130 kHz @ Ct 1.0 nF (eq. 15),
350 A
including a total 500 ns discharge time. In practice,
accounting for the internal element precisions, a 110 kHz is
typically given in the data–sheet. Figure 13 gives the typical
frequency versus the feedback pin level shape for the SOFA.
The frequency starts to decrease when Vfb is around 3.1 V.
Figure 11. In presence of a short, the SOFA
enters a controlled burst mode to keep the
power dissipation within safe limits.
Error Voltage
SWITCHING FREQUENCY (kHz)
120
Green Light
6
2
1 350 µA
+
5V
Ct
3
5
7
+
–
+
3 V/0.5 V
Driver
100
80
60
40
20
0
3.05
3.1
3.15
3.2
3.25
3.3
3.35
FEEDBACK VOLTAGE (V)
Figure 12. This picture details the oscillator section
Figure 13. The VCO versus feedback
voltage transfer function (Ct = 1.0 nF)
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3.4
AND8043/D
Fault Management
As we saw before, the SOFA hosts a dedicated timing
circuitry only activated during the presence of a
short–circuit. As soon as the 2.5 V virtual ground is lost
(Figure 9), Ct is routed to a fixed charge/discharge current
generator which produces an internal 1.0 kHz frequency
(Ct = 1.0 nF). This 1.0 kHz is internally divided to produce
a 128 ms timeout. If the short circuit lasts more than 128 ms,
then the output pulses are permanently stopped for a
complete 8 x 128 ms or 1 second (Ct = 1.0 nF). During this
time, Vcc goes up and down. Once this period is elapsed, the
SOFA resumes and attempts to restart. If it cannot (because
the short is still there), then a new 128 ms cycle takes place.
Please note that Ct being routed to the timeout generator, the
SOFA enters the free–un operation in error mode.
In VCO mode (also called VFM), the charging current
diminishes and Ct reaching 3.0 V gives the VCO green light.
However, the power switch will effectively be driven high
only when both Ct and demagnetization end both give the
green light. This is important to respect the multiple wave
jumps. Figure 14 and 15 detail Ct signals at two different
points: free–run and VFM.
Vdrain
Ct/demag
are ok
Ct ok but no demag
Design Rule
VCt
The power supply start–up period (when
you plug the system into the wall until it
actually regulates) shall last less than
128 ms (if Ct = 1.0 nF) otherwise a
short–circuit like situation will occur and
the supply will fail to crank. Also, please
note that the fault timer circuitry is not
synchronized with the Vcc level. As a
result, the 128 ms burst can only appear
on the drive output for a going down Vcc
(15 to 8.0) and not during a going up Vcc
(8.0 to 15) where the pulses are
automatically invalidated (as with a
startup sequence). Because of this lack of
synchronization, the 128 ms burst can
sometimes be truncated if it starts in the
Vcc ramp–up portion. This is not a
problem in the application:
0V
45.0 U
55.0 U
65.0 U
Ct is 2 V/div
75.0 U
85.0 U
Figure 14. Free–run operation: Ct is okay to restart
but the demagnetization signal is not yet arrived . . .
Ct ok but no demag
Ct/demag are ok
VCt
0V
UVLOH
Ct is 2 V/div
335 U
345 U
355 U
365 U
UVLOH
375 U
Figure 15. Here, we are in VFM and Ct rules the
operation. Sometimes Ct says okay but, again, the
demagnetization has not arrived: next wave is fine.
Design Rule
The SOFA operates in free–run until the
frequency increases (Pout diminishes)
and the VCO starts to clamp then folds the
frequency back. Always keep a 20%
difference between the maximum
frequency imposed by the VCO (through
Ct) and the frequency occurring at
maximum power in free–run operation
(eq. 16). If this condition is not respected,
there are chances that the regulation is lost
at the highest output level. As an example,
if the maximum switching frequency is
selected to be 90 kHz, then the maximum
output power shall be reached at a
frequency less or equal than 70 kHz.
128 ms
< 128 ms
Figure 16. The burst can be truncated when it
occurs before UVLOH
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AND8043/D
Demagnetization Circuit and Auxiliary Winding
The demagnetization portion plays an important role in
the SOFA application. Its role consists in monitoring the
transformer core activity and restart the power switch once
the core reset has occurred. This function is accomplished
by checking the level on a dedicated auxiliary winding as
portrayed by Figure 19:
Startup Section
The SOFA includes a 3.0 mA startup current source
directly connected to the mains. It is capable to permanently
sustain up to 500 V without a problem. During the startup
sequence, the 3.0 mA charges the Vcc capacitor. Once
UVLOH has been reached, the error timer circuitry is
triggered because the supply output voltage does not
regulate Vout. After startup, providing the self–supply is
present, the current source is turned off, dissipating very few
mW. Figure 17 portrays a typical startup sequence:
HV
Vout
Rvalley
Np
Demag
Nsec
Rlimit
Vin
Naux
Ct starts to
count 128 ms
VCC, 5 V/div
VCt, 2 V/div
Figure 19. Implementing the demagnetization with
the auxiliary supply wired in Forward mode
Figure 17. A typical startup sequence showing Ct
activity at UVLOH
2k
Demag
1
2
To Internal Circuitry
∆I, Thermal
Coefficient
0
Figure 20. The front ESD zener diode exhibits a
parasitic capacitance of roughly 10 pF
Rvalley limits the maximum current flowing through the
SOFA demagnetization pin (≈ 3.0 mA is a good value) but
also forms a delay network when combined with the pin
input capacitance (10 pF typical at low voltages thanks to
ESD protection network, Figure 20). Once the power supply
prototype has been built, a few tweak on Rvalley will let you
find the right value to restart right in the minimum drain
valley (Figure 21):
1 mA/div
Figure 18. A typical startup shape illustrating
the thermal coefficient on the current source
(1.0 mA/div)
As Figure 18 depicts, a thermal coefficient is naturally
present on the high–voltage startup current source. Because
the die heats up, the current set point is modified to slowly
reduce the current delivered to the Vcc capacitor.
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400
Since we are operating the winding in Forward, it behaves
as a low impedance voltage source. A resistor is thus
required to limit the maximum peak current flowing through
the Vcc capacitor but also through the auxiliary diode
(typically a 1N4148 can do the job).
The SOFA has been designed to accept a demagnetization
polarity corresponding to an auxiliary winding operated in
the Forward mode (dot position on Figure 19 and Figure 3b
signals). The Aux Vcc node will thus be at a level equal to:
Right in the valley:
Freeze Rvalley
300
200
100
J
0
372
376
380
(U)
400
384
388
Vcc VHV · Naux . This is true when the duty–cycle is
Np
large enough (e.g. at a nominal load level). When the load
goes down, the duty–cycle also diminishes until the peak
current is frozen (≈ 250 mV/Rsense). The Vcc is therefore
affected by a ripple depending on the amount of charge you
bring in (during the ON time) and the amount of charge
delivered to the load (the SOFA IC). You thus need to check
that the Vcc stays at a minimum of 8.0 V at any load and
mains conditions but also remains lower than the maximum
allowable Vcc to not shut down the IC (highest mains and
maximum output current). We will tackle this point in the
design example.
In some applications, where you need to implement a
precise overvoltage protection, an auxiliary self–supply
configured in Flyback is mandatory to offer a primary image
of what is going on the secondary. To still allow the
demagnetization detection, Figure 22 proposes a solution
built around two extra diodes and one resistor.
To Soon:
Increase Rvalley
300
200
100
L
0
390
394
398
402
406
(U)
400
To Late:
Derease Rvalley
300
200
100
L
0
340
344
348
352
356
(U)
Figure 21. A tweak on Rvalley allows to restart
right in the middle of the wave
HV
Vout
Np
Daux
Nsec
Vin
Naux
Rvalley
Gnd
2 x 1N4148
Vdem
Figure 22. A simple component arrangement allows a detection in Flyback mode
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AND8043/D
During the ON time, the Demag point goes high but clamp
to one Vf. The auxiliary diode being blocked, Rvalley closes
the path to ground and fixes the output impedance. You can
thus directly drive the demagnetization pin from this point,
Rvalley inserting the same delay as on Figure 19. At the
switch opening, the auxiliary voltage also reverses and the
other diode ensures a Demag level of –Vf. When DCM
occurs, the Demag line crosses the 65 mV threshold and
reactivates the SOFA circuitry.
The Perfidious Leakage Inductance…
As usual, the leakage inductance will bother the designer
but this time not in the sense of a lethal kick but rather
because of its high frequency nature. The SOFA
demagnetization circuitry features a threshold of typically
65 mV which is very sensitive. If the leakage inductance is
high and the reflected voltage low, e.g. during power up,
then there are chances that the leakage inductance ringing is
internally interpreted as a demagnetization signal and the
SOFA will restart the switch: you enter CCM and the
switching frequency becomes very high, implying a
possible MOSFET over dissipation. Figure 23 shows how
the problem takes place:
Figure 24. Multiple high frequency restart occurs
because of the leakage
At the very beginning, just when UVLOH is crossed by the
Vcc line, the first pulse takes place. Because Vout is down to
zero, you only reflect (N x Vf) which can be pretty low,
especially with a Schottky diode. As a result, the leakage
ringing can be so strong (the peak current set point is pushed
into its upper stops) that it crosses the Vin reference level (or
the ground on the auxiliary winding, Figure 23) and triggers
the demagnetization comparator. In normal operation, it
could not append because the internal timer clamps the
maximum switching frequency to Fsw max = 110 kHz.
Unfortunately, at start–up, the timer is routed to the fault
detection circuitry and does not plays its usual clamping role
and total free–run is authorized. Pulses appear at a very high
recurrent frequency until Vout starts to be significant and
elevates the leakage inductance floor. At this time, the
demagnetization Zero Crossing Detector (ZCD) is no longer
bothered by this default. Figure 24 shows the true behavior
of an application board without any damping network. The
drain (lower trace) is close to the ground and the leakage
ringing restarts the SOFA controller. The switching
frequency is very high…
A possible cure consists in further integrating the ZCD
signal with an additional capacitor wired between the
feedback pin and ground. This option adds another delay and
the true valley switching can be lost. Another option takes
the form of an R–C damper wired on the transformer
primary. Trials will be necessary to find a balance between
the damping effect and a reasonable efficiency degradation.
Potential
Problem
N. (Vout + Vf)
This value is
close to zero
at start–up
150.8U
152.8U
154.8U
156.8U
158.8U
Figure 23. Potential problems can arise from an
energetic leakage effect (Spice simulation)
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AND8043/D
VCC
The Driving Stage
As we have seen, the SOFA sustains operating voltages up
to 35 V. Obviously, the driving level delivered by the circuit
cannot be that high, otherwise the power MOSFET
gate–source oxide would immediately be destroyed. The
SOFA features a patented non–dissipative driving stage
which clamps the Vgs to 12 V but offers a boost at low Vcc
levels. Because the driver behaves like a voltage source
(unlike a true current source with bipolar output stages), the
rising edge of the signal can be the seat of a short ringing
wave which occurs at the Miller transition. This pulse
occurrence is normal and gets higher as the driver clamps
more heavily. It does not have any effect the supply
operation:
6
5V
5
10 V
4
10 V
3
10 V
2k
Latched
OVP
2
+ 7
–
1
8
18 k
+
2.5 V
Figure 26. The OVP is made with four 10 V zener
diodes connected in series
Miller Plateau
VZ
OVP
Pin
VDS
Voltage to be
monitored
Cnoise
VGS
Figure 27. A possible OVP implementation with the
DIP14 version
Figure 25. A small glitch occurs at the
Miller transition
A Design Example
Now that we went through the description of the internal
SOFA blocks, we will detail how to calculate the main
components and particularly the transformer. The
specifications are the following and correspond to a generic
power supply application:
The SOFA Vcc operating range being quite large, always
check that the selected power MOSFET accepts Vgs down
to 6.0 V. A logic threshold device is recommended in case
of extended operation close to UVLOLOW (6.5 V min).
The decaying plateau on the top of the Vgs signal is due
to the internal bootstrap capacitor that slowly discharges
during the transition.
Vin: 90 to 250 VAC
VinDC min = 120 V
VinDC max = 350 V
Pout = 10 W
Vout = 6.5 V → IoutDC = 1.53 A, Rload = 4.2 Ω
Target efficiency = 80%
Overvoltage Protection
The SOFA offers two types of overvoltage protection
(OVP): one through the Vcc pin and another one via a
dedicated pin only available on the DIP14 package.
Figure 26 depicts the internal implementation which
reveals the presence of several zener diodes connected in
series. The DIP14 offers a direct access to the comparator
input for a different OVP level selection. The typical
threshold level is stated at 2.5 V. With a 20 kΩ input
impedance, we recommend the use of a zener diode as
proposed by Figure 27. A capacitor can be added in very
noisy environments. Final OVP level is thus Vz + 2.5 V.
For EMI reasons, we want to keep Fmax below 90 kHz.
The maximum power in free–run shall thus take place at
90–20% = 70 kHz. Now, as eq. 12 shows, we need several
elements to compute our primary peak current defined by:
Ip 2 · Pout ·
N · (Vout Vf) Vin
. Worse case appears
· Vin · N · (Vout Vf)
at nominal output power and lowest input line level (120
VDC): Ip is the highest and Fswitching the lowest.
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AND8043/D
N PIV Vout (eq. 20). If we select Np:Ns = 1: 0.08, then
VinDC max
Turn Ratio Calculation
The turn ratio N fixes the level of Ip but also the type of
secondary diode you will select. It actually affects several
parameters:
The drain plateau voltage during the OFF time: the
lowest plateau gives room for the leakage inductance
spike before reaching the MOSFET’s BVdss:
Vplateau PIV = 28 V at 350 VDC input voltage which is okay with the
selected diode. The plateau voltage at the drain will establish
around 430 VDC: it leaves up to 170 V for the leakage spike
with a 600 V MOSFET transistor.
The average diode Idavg current is the converter’s DC
output current which is 1.5 A, in line with our 3.0 A
MBRS340T3.
Np
· (Vout Vf) VinDC max (eq. 17).
Ns
The secondary Peak Inverse Voltage (PIV) is linked to the
turn ratio and the regulated output voltage by:
Primary Inductance
Plugging N into eq. 12 gives a peak current of 482 mA. We
should now select the primary inductance to cope with our
70 kHz requirement. A simple spreadsheet plotting
PIV Ns · VinDC max Vout (eq. 18). If you lower the
Np
plateau voltage, you will increase the reverse voltage the
secondary diode must sustain.
The amp–turns equation Np . Ip Ns . Is should
satisfy the average output current demand with
Ioutavg Ip · toff · Fsw · 2·
Np
Ns
Y 2 · Pout while displaying incremental values for Lp
· Lp · Ip2
in the X–axis will offer a possible selection for the primary
inductance:
(eq. 19). The α parameter
120
illustrates the energy diverted by the leakage inductance
at the switch opening (take 0.95 for low leakage designs).
With these numbers in mind, you can tweak the turn ratio
according to the MOSFET BVdss and the diode. Below are
given ON Semiconductor references for Schottky diodes:
kHz
100
80
L = 1.55 mH
Reference
VRRM (V)
Io (A)
Case
MBRM120LT3
20
1
PowerMite
MBRM130LT3
30
1
PowerMite
MBRA130LT3
30
1
SMA
MBRA140LT3
40
1
SMA
MBRS120LT3
20
1
SMB
MBRS130LT3
30
1
SMB
MBRS140LT3
40
1
SMB
MBRS190T3
90
1
SMB
MBRS1100T3
100
1
SMB
MBRS320T3
20
3
SMC
MBRS330T3
30
3
SMC
MBRS340T3
40
3
SMC
MBRS360T3
60
3
SMC
1N5817
20
1
Axial
1N5818
30
1
Axial
1N5819
40
1
Axial
mH
60
40
20
0
0
2
4
6
8
10
Figure 28. A graph plotting Fsw versus
Lp offers a fast selection process
A 70 kHz operating frequency is obtained with a primary
inductance of
2 · Pout
= 1.55 mH. In the lower
· 70 · 103 · Ip?
SOFA limit (0.9 V), we need to be able to develop our peak
current target of 480 mA. The minimum shunt value is
therefore: 0.9/0.48 = 1.87 Ω. Let’s pick–up an 1.8 Ω
normalized value. This shunt being affected by a ±5.0%
tolerance, the new worse case reveals a maximum peak of
1.1/1.71 = 643 mA. Furthermore, we need to account for the
maximum SOFA propagation delay (250 ns) which,
together with the primary slope, affects the final value.
Worse case arises at the highest line: 350 · 250 n 56 mA.
Please see brochure BR1487/D for thermal and package details.
1.55 m
The
If we select an MBRS340T3 (VRRM = 40 V), then the
PIV should be selected around 35 V at high line:
total
maximum
peak
643 mA 56 mA 699 mA.
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current
can
thus
be:
AND8043/D
Idrain RMS = Iprimary RMS = 182 mA
Idiode secondary RMS = 2.53 A, Iavg = Iout
Fswitching = 64 kHz
The basic transformer specs are therefore:
Lp = 1.55 mH
Np : Ns = 1 : 0.08
Maximum peak current = 691 mA
The above values let you evaluate the MOSFET
conduction losses (Pcond = Rdson @ Tj = 100°C x I_drain
RMS), the secondary diode conduction losses (Pcond = Vf x
Iavg + Rd x I_diode RMS). The primary/secondary RMS
currents will determine the wire size when designing your
transformer.
To verify our design, we can plug our numbers into a
SPICE simulator using the transient SOFA model. The
below schematic exemplifies the INTUSOFT’s IsSpice4
model where the components have been assigned with the
calculated values:
Rprim
0.5
Vin
Iout
X1
XFMR–AUX
RATIO _POW = –0.08
RATIO_AUX = 0.08
D2
1N5822
Vout
+
Vout
19
Y13
∆
R6
49 k
C6
10 nF
1
18
Resr1
60 m
24
Lprim
1.55 mH
Icoil
D1
1N4148
2
D3
MUR160
Vin
120
VCC
Caux
22 µF
IC = 14.99
+
IHV
R7
56
13
Dem
+
Rload
4.2
Cout1
470 µF
IC = 6.5
42
Lleak
30 µH
Vdrain
VCC
+
ICC
VCC
Vout
+
Idrain
Rled
470
VCC
11
DRV
R4
47 k
Dem
21
5
FB
Demag VCt
8
2
7
3
6
4
17
Ct
1 nF
1
SOFA
X2
NCP1201_8
X1
MTD1N60E
23
14
3
8
5
X7
MOC8101
Vsense
Vfb
Rp
10 k
R2
1.8
FB
C3
1 nF
Figure 29. A Typical SPICE Simulation Testing the Design Validity
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Rfb
4.7 k
15
BV = 6 V
AND8043/D
The following curves give you the most important wave shapes and let you deduct the other design parameters:
8.00
250
ID(D2) AMPS
Vdrain (VOLTS)
350
150
50.0
6.00
4.00
2.00
2.0
–50.0
1.058
1.062
1.066
1.070
0
1.074
Vdrain, TIME IN SECS (M)
ID(D2)
600
Idrain, AMPS (M)
Vout (VOLTS)
7.05
6.95
6.85
6.75
400
200
0
6.65
–200
1.01
1.03
1.05
1.07
1.09
Idrain
Vout, TIME IN SECS (M)
600
Icoil, AMPS (M)
Drv (VOLTS)
14.0
10.0
6.00
2.00
–2.00
400
200
0
–200
1.01
1.03
1.05
1.07
1.09
1.01
Drv, TIME IN SECS (M)
1.03
1.05
1.07
1.09
Icoil, TIME IN SECS (M)
Figure 30. Various Simulation Results Revealing Important Values
Protecting the Power MOSFET
If the leakage inductance is kept low, an avalanche rugged
MOSFET such as the MTD1N60E can withstand accidental
avalanche energy, e.g. during a high–voltage spike
superimposed over the mains, without the help of a clamping
network. However, if this leakage path permanently forces
a drain–source voltage above the MOSFET BVdss (e.g.
600 V), a clamping network is mandatory and must be built
around a passive RC network or a Transient Voltage
Suppressor (TVS). Figure 31 depicts the phenomenon while
the below lines details the calculation steps:
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AND8043/D
RC network or TVS
BV dss
HV rail
HV rail
You need
1
4
3
5
TVS
Ultra–fast
Figure 31. Care must be taken to ensure a safe operation of the MOSFET
1. RCD Network
Our 10 W power supply imposes Rclamp = 49 kΩ/1.0 W
Cclamp = 10 nF D = MUR160 from ON Semiconductor
The RCD network will permanently impose a fixed
clamping level that will oppose to the leakage voltage. As a
result, the drain will be clamp to VHVrail + Vclamp. You
normally select a clamping level between 40 to 80 volts
above the reflected output voltage when the supply is
heavily loaded.
To calculate the component values, the following
formulae will help you:
Rclamp 2. Transient Voltage Suppressor
Despite the low–cost offered by the above RC solution,
the clamping level unfortunately varies with the peak
current. If you need a very precise clamping level, you must
implement a zener diode or a TVS. There are little
technology differences behind a standard zener diode and a
TVS. However, the die area is far bigger for a transient
suppressor than that of zener. A 5.0 W zener diode like the
1N5388B will accept 180 W peak power if it lasts less than
8.3 ms. If the peak current in the worse case (e.g. when the
PWM circuit maximum current limit works) multiplied by
the nominal zener voltage exceeds these 180 W, then the
diode will be destroyed when the supply experiences
overloads. A transient suppressor like the P6KE200 still
dissipates 5.0 W of continuous power but is able to accept
surges up to 600 W @ 1.0 ms. If the peak power is really
high, then turn to a 1.5KE200 which accepts up to 1.5 kW
@ 1.0 ms.
Select the zener or TVS clamping level between 40 to 80
volts above the reflected output voltage when the supply is
heavily loaded.
2 · Vclamp · (Vclamp (Vout Vf sec) · N)
Lleak · Ip? · Fsw
(eq. 21)
Cclamp Vclamp
Vripple · Fsw · Rclamp
(eq. 22)
The power dissipated by Rclamp can also be expressed by:
PRclamp 1 · Lleak · Ip? · Fsw ·
2
Vclamp
(Vout Vf sec) · N
Vclamp
–1
(Vout Vf sec) · N
(eq. 23)
with:
Vclamp: the desired clamping level
Ip: the maximum peak current (e.g. during overload)
Vout + Vf: the regulated output voltage level + the
secondary diode voltage drop
Lleak: the primary leakage inductance
N: the Ns:Np conversion ratio
Fsw: the switching frequency
Vripple: the clamping ripple, could be around 20 V
3. Snubber Network
Another option lies in implementing a snubber network
which will damp the leakage oscillations but also provide
more capacitance at the MOSFET’s turn–off. The peak
voltage at which the leakage forces the drain is calculated
by: Vmax Ip ·
Lleak (eq. 24) where Clump represents
Clump
the total parasitic capacitance seen at the MOSFET opening.
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AND8043/D
frequency: Z 2 · · Fring · Lleak (eq. 25). Wiring a
resistor R whose value equals Z should solve the problem
but at the expense of a large power dissipation. To lower the
dissipated heat, you can wire a capacitor C in series with R:
Depending on the output power, you can either wire a simple
capacitor across the MOSFET or an R–C network, as shown
by Figure 32.
4
5
Csnubb
C
Csnubb
can directly wire this capacitor between the MOSFET drain
and ground (not between drain–source to avoid substrate
injection). Unfortunately, you discharge this capacitor in the
MOSFET every time it turns on . . . Further tweaking is thus
necessary to tune the dissipated power versus standby
power.
3
Rsnubb
2
1
RS
1
(eq. 26). If the output power is low, you
· Fring · R
RS
ON Semiconductor Protection Devices
SMPS protection clearly needs fast switching
components to ensure a reliable operation in the event of
dangerous transients. ON Semiconductor portfolio offers a
comprehensive list of semiconductors dedicated to
protection: fast diodes, zeners, TVS etc. Below is a small list
of typical component you can select to protect the MOSFET
in your application. The complete list of TVS devices can be
found at the following URL: www.onsemi.com or by
ordering the selection guide TVSPROMO1299/D by
sending an email to: [email protected]:
Figure 32. If the output power is low, you can wire a
simple capacitor MOSFET’s drain and ground
To calculate the values of this RC snubber, you need to
measure the ringing frequency imposed by the leakage
inductance and all the stray capacitances. Make sure you use
a low capacitance probe, otherwise you might affect the
observed frequency. Once you have it, calculate the
impedance of the leakage inductance at the ringing
Clipping Elements
Reference
Nominal Voltage (V)
Average Power (W)
Maximum Peak Power
1N5953B
150
1.5
98 W @ 1.0 ms
1N5955B
180
1.5
98 W @ 1.0 ms
1N5383B
150
5.0
180 W @ 8.3 ms
1N5386B
180
5.0
180 W @ 8.3 ms
1N5388B
200
5.0
180 W @ 8.3 ms
P6KE150A
150
5.0
600 W @ 1.0 ms
P6KE180A
180
5.0
600 W @ 1.0 ms
P6KE200A
200
5.0
600 W @ 1.0 ms
1.5KE150A
150
5.0
1.5 kW @ 1.0 ms
1.5KE180A
180
5.0
1.5 kW @ 1.0 ms
1.5KE200A
200
5.0
1.5 kW @ 1.0 ms
VRRM
Ton (Typical)
IF Max
Fast Diodes
Reference
MUR160
600 V
50 ns
3.0 A
MUR1100E
1000 V
25 ns
3.0 A
1N4937
600 V
200 ns
1.0 A
MSR860*
600 V
100 ns
8.0 A
MSRB860–1*
600 V
100 ns
8.0 A
*Soft Recovery Diodes
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AND8043/D
Auxiliary Supply
As Figure 29 shows, we are using the auxiliary winding in
a Forward mode. Further to what we have discussed,
the auxiliary voltage will then be fixed by the primary/
auxiliary turn ratio through the following formula:
8.60
6.60
Vcc
Vcc VHV · Naux . We have to ensure full operation at low
Np
4.60
line (min Vcc = 8.0 V) and be sure to not trigger the internal
SOFA OVP at high line (unless this becomes a safety
recommendation against accidental high line levels). With
line varying between 120 V and 350 VDC, the minimum
turn ratio is 8/120 = 0.066. If we include a design safety
margin of 15%, we end up with a turn ratio of 0.08. At high
line, the level will rise up to: 350 x 0.08 = 28 V which is ok.
However, in standby, the switching frequency will largely
diminish while the forward pulse become narrower. As a
result, the Vcc capacitor refresh rate could be too low to
ensure a correct level of self–supply at the lowest mains
condition. We need to verify this point through calculation
and simulation. At no–load we have:
2.60
600 M
39.0
97.0
155
213
271
231.5
232.5
(M)
9.00
8.90
Ipeak = Ip max/3.0 or 250 mV/Rsense = 138 mA. Including
the 250 ns propagation delay, Ip = 157 mA
Vout = 6.5 V
VFB ≈ 3.0 V over 4.7 kΩ → optocoupler collector current
≈ 640 µA
Isecondary bias ≈ 640 µA with a Current Transfer Ratio
(CTR) of 100%
Pout = 6.5 V x 640 µA = 4.0 mW.
8.80
8.70
Zoom
On VCC
8.60
228.5
229.5
230.5
(M)
The efficiency being rather low at these levels of power,
experience shows that an input power around 40 mW
including the MOSFET losses, ESR losses etc. is more
realistic. Applying eq. 9, we can obtain an estimation of the
Figure 34. We have been obliged to raise the
turn ratio in order to ensure an adequate Vcc
level around 8.0 V
switching frequency in standby: Pin · 2 2.0 kHz. With a
The simulation revealed a slightly low Vcc level at
120 VDC input line. Raising the auxiliary turn ratio from
0.08 up to 0.095 fixes the problem. In high line conditions,
Vcc will go up to 350 x 0.095 = 33.3 V, lower than SOFA trip
point. In case of over–voltage, the 36 V (min) SOFA trip
point will stop the operation at a 267 VAC line level.
Lp · Ip?
120 VDC (low line) input level, the power MOSFET closes
during: 1.55 m · 0.157 2.0 s. We can now feed our
120
simulator with the calculated values: pulses of 0.08 x 120 =
9.6 V, lasting 2.0 s and recurring with a 2.0 kHz frequency.
Figure 33 shows the proposed simulation setup where the
SOFA consumption is given by the 1.2 mA current source:
Final transformer specifications:
Lp = 1.55 mH
Np : Nspower = 1 : 0.08
Np : Nsaux = 1 : 0.095
Maximum primary peak current = 691 mA
Primary RMS current = 182 mA
Secondary RMS current = 2.53 A
+
Figure 33. A simple simulation setup lets you
evaluate the auxiliary Vcc level you will reach
in standby
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AND8043/D
Notes
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AND8043/D
ON Semiconductor and
are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes
without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular
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including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or
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