Data Sheet No. PD94705 IR3080 XPHASETM VRD10 CONTROL IC WITH VCCVID & OVERTEMP DETECT DESCRIPTION The IR3080 Control IC combined with an IR XPhaseTM Phase IC provides a full featured and flexible way to implement a complete VRD 10 power solution. The “Control” IC provides overall system control and interfaces with any number of “Phase ICs” which each drive and monitor a single phase of a multiphase converter. The XPhaseTM architecture results in a power supply that is smaller, less expensive, and easier to design while providing higher efficiency than conventional approaches. The IR3080 is intended for desktop applications and includes the VCCVID and VRHOT functions required for proper system operation. FEATURES • • • • • • • • • • • • • • • • • 6 bit VR10 compatible VID with 0.5% overall system accuracy 1 to X phases operation with matching phase ICs On-chip 700Ω VID Pull-up resistors with VID pull-up voltage input Programmable Dynamic VID Slew Rate No Discharge of output capacitors during Dynamic VID step-down (can be disabled) +/-300mV Differential Remote Sense Programmable 150kHz to 1MHz oscillator Programmable VID Offset and Load Line output impedance Programmable Softstart Programmable Hiccup Over-Current Protection with Delay to prevent false triggering Simplified Powergood provides indication of proper operation and avoids false triggering Operates from 12V input with 9.1V Under-Voltage Lockout 6.8V/5mA Bias Regulator provides System Reference Voltage 1.2V @ 150mA VCCVID Linear Regulator with Full Protection VCCVID Powergood with Programmable delay gates converter operation Programmable Converter Over-Temperature Detection and Output Small thermally enhanced 32L MLPQ package 25 LGND 26 27 HOTSET 28 VRHOT SS/DEL 29 30 PWRGD RMPOUT EAOUT FB VDRP 24 23 22 21 20 19 18 VDAC 17 16 15 14 ROSC OCSET TRM4 VID3 TRM3 IIN VOSNS- VID2 9 Page 1 of 41 VIDPGD VID1 VID4 8 VID0 13 7 VID5 12 6 BBFB IR3080 CONTROL IC TRM2 5 VBIAS VIDPWR TRM1 4 VCC VCCVID 11 3 VIDFB VIDDEL 32 2 10 1 31 PACKAGE PIN OUT 9/30/04 IR3080 ORDERING INFORAMATION Device Order Quantity IR3080MTR 3000 per reel * IR3080M 100 piece strips * Samples only ABSOLUTE MAXIMUM RATINGS Operating Junction Temperature……………..150oC Storage Temperature Range………………….-65oC to 150oC ESD Rating……………………………………..HBM Class 1C JEDEC standard PIN # 1 2 3 4-9 10, 11, 13,14 12 15 16 17 18 PIN NAME VIDFB VCCVID VIDPWR VID0-5 TRM1-4 VMAX 20V 20V 20V 20V Do Not Connect VMIN -0.3V -0.5V -0.3V -0.3V Do Not Connect ISOURCE 1mA 400mA 1mA 10mA Do Not Connect ISINK 1mA 50mA 400mA 10mA Do Not Connect VOSNSROSC VDAC OCSET IIN 0.5V 20V 20V 20V 20V -0.5V -0.5V -0.3V -0.3V -0.3V 10mA 1mA 1mA 1mA 1mA 10mA 1mA 1mA 1mA 1mA 19 20 21 VDRP FB EAOUT 20V 20V 10V -0.3V -0.3V -0.3V 5mA 1mA 10mA 5mA 1mA 20mA 22 BBFB 20V -0.3V 1mA 1mA 23 VBIAS 20V -0.3V 1mA 1mA 24 VCC 20V -0.3V 1mA 50mA 25 LGND n/a n/a 50mA 1mA 26 RMPOUT 20V -0.3V 1mA 1mA 27 HOTSET 20V -0.3V 1mA 1mA 28 VRHOT 20V -0.3V 1mA 50mA 29 SS/DEL 20V -0.3V 1mA 1mA 30 PWRGD 20V -0.3V 1mA 20mA 31 VIDPGD 20V -0.3V 1mA 20mA 32 VIDDEL 20V -0.3V 1mA 1mA Page 2 of 41 9/30/04 IR3080 ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over: 9.5V ≤ VCC ≤ 14V, 2.0V ≤ VIDPWR ≤ 5.5V and 0 oC ≤ TJ ≤ 100 oC PARAMETER VDAC Reference System Set-Point Accuracy Source Current Sink Current VID Input Threshold VID Pull-up Resistors Regulation Detect Comparator Input Offset Regulation Detect to EAOUT Delay BBFB to FB Bias Current Ratio VID 11111x Blanking Delay VID Step Down Detect Blanking Time VID Down BB Clamp Voltage VID Down BB Clamp Current Error Amplifier Input Offset Voltage FB Bias Current DC Gain Gain-Bandwidth Product Source Current Sink Current Max Voltage Min Voltage VDRP Buffer Amplifier Input Offset Voltage Input Voltage Range Bandwidth (-3dB) Slew Rate IIN Bias Current Page 3 of 41 TEST CONDITION -0.3V ≤ VOSNS- ≤ 0.3V, Connect FB to EAOUT, Measure V(EAOUT) – V(VOSNS-) deviation from Table 1. Applies to all VID codes. RROSC = 41.9kΩ RROSC = 41.9kΩ MIN TYP MAX 0.5 68 47 500 500 -5 0.95 Measure time until PWRGD drives low Measure from VID inputs to EAOUT UNIT % 80 55 600 700 0 92 63 700 1000 5 µA µA mV Ω mV 130 200 ns 1.00 1.05 µA/µA 800 1.7 ns µs Percent of VDAC voltage 70 3.5 75 6.2 80 12 % mA Connect FB to EAOUT, Measure V(EAOUT) – V(DAC). from Table 1. Applies to all VID codes and -0.3V ≤ VOSNS- ≤ 0.3V. Note 2 RROSC = 41.9kΩ Note 1 Note 1 -3 4 8 mV -31 90 4 0.4 0.7 125 30 -29.5 100 7 0.6 1.2 250 100 -28 105 µA dB MHz mA mA mV mV -8 0.8 1 0 VBIAS–VEAOUT (referenced to VBIAS) Normal operation or Fault mode V(VDRP) – V(IIN), 0.8V ≤ V(IIN) ≤ 5.5V Note 1 -2.0 6 10 -0.75 0.8 1.7 375 150 8 5.5 0 9/30/04 mV V MHz V/µs µA IR3080 PARAMETER Oscillator Switching Frequency Peak Voltage (5V typical, measured as % of VBIAS) Valley Voltage (1V typical, measured as % of VBIAS) VBIAS Regulator Output Voltage Current Limit Soft Start and Delay SS/DEL to FB Input Offset Voltage Charge Current Discharge Current Charge/Discharge Current Ratio Charge Voltage Delay Comparator Threshold Discharge Comparator Threshold Over-Current Comparator Input Offset Voltage OCSET Bias Current PWRGD Output Output Voltage Leakage Current VCCVID Regulator Output Voltage Dropout Voltage Current Limit VIDFB Bias Current VCCVID Powergood VCCVID Threshold Voltage Charge Current Discharge Current Charge Voltage Delay Comparator Threshold VIDPGD Output Voltage VIDPGD Leakage Current Page 4 of 41 TEST CONDITION MIN TYP MAX UNIT RROSC = 41.9kΩ RROSC = 41.9kΩ 255 70 300 71 345 74 kHz % RROSC = 41.9kΩ 11 14 16 % -5mA ≤ I(VBIAS) ≤ 0 6.5 -30 6.8 -15 7.1 -6 V mA With FB = 0V, adjust V(SS/DEL) until EAOUT drives high 0.85 1.3 1.5 V 40 4 10 70 6 11.5 100 9 13 µA µA µA/µA 3.7 70 150 4.0 90 200 4.2 110 250 V mV mV -10 -31 0 -29.5 10 -28 mV 150 0 400 10 mV µA 1.215 200 1.0 V mV V mA -225 1.200 130 0.8 -300 -150 1.05 1.10 1.13 V 30 4 3.7 65 66 7 4.0 90 150 0 90 11 4.2 115 400 10 µA mA V mV mV µA Relative to Charge Voltage RROSC = 41.9kΩ I(PWRGD) = 4mA V(PWRGD) = 5.5V -150mA ≤ I(VCCVID) ≤ 0 I(VCCVID) = -20mA I(VCCVID) = -150mA Relative to regulated VCCVID Output Voltage Relative to Charge Voltage I(VIDPGD) = 4mA V(VIDPGD) = 5.5V 1.145 µA µA 9/30/04 IR3080 PARAMETER VCC Under-Voltage Lockout Start Threshold Stop Threshold Hysteresis General VCC Supply Current VIDPWR Supply Current VOSNS- Current VRHOT Comparator HOTSET Bias Current Output Voltage VRHOT Leakage Current Threshold Hysteresis Threshold Voltage (increasing temperature) TEST CONDITION MIN TYP MAX UNIT Start – Stop 8.6 8.4 150 9.1 8.9 200 9.6 9.4 300 V V mV VID0-5 Open, I(VCCVID) = 0 -0.3V ≤ VOSNS- ≤ 0.3V, All VID Codes 8 400 -5.5 11 550 -4.5 14 1000 -3.5 mA µA mA -2 -0.5 300 0 6 1 400 10 10 µA mV µA o C I(VRHOT) = 29mA V(VRHOT) = 5.5V TJ ≥ TJ ≥ 3 85oC o 85 C MIN 4.73mV/ oC x TJ + 1.176V TYP 4.73mV/ oC x TJ + 1.241V MAX 4.73mV/ oC x TJ + 1.356V Note 1: Guaranteed by design, but not tested in production Note 2: VDAC Output is trimmed to compensate for Error Amplifier input offsets errors Page 5 of 41 9/30/04 V IR3080 PIN DESCRIPTION PIN# 1 2 PIN SYMBOL VIDFB VCCVID 3 4-9 VIDPWR VID0-5 10, 11, 13,14 12 15 TRM1-4 VOSNSROSC 16 VDAC 17 OCSET 18 IIN 19 VDRP 20 FB 21 22 EAOUT BBFB 23 VBIAS 24 25 26 27 VCC LGND RMPOUT HOTSET 28 29 VRHOT SS/DEL 30 PWRGD 31 VIDPGD 32 VIDDEL Page 6 of 41 PIN DESCRIPTION Feedback to the VCCVID regulator. Connect to the VCCVID output. 1.2V/150mA Regulator Output. Can also drive external pass transistor to minimize on-chip power dissipation Power for VID Pull-up resistors and VCCVID Regulator Inputs to VID D to A Converter. On-chip 700 ohm pull-up resistors to VIDPWR pin are included. Used for precision post-package trimming of the VDAC voltage. Do not make any connection to these pins. Remote Sense Input. Connect to ground at the Load. Connect a resistor to VOSNS- to program oscillator frequency and FB, OCSET, BBFB, and VDAC bias currents Regulated voltage programmed by the VID inputs. Current Sensing and PWM operation are referenced to this pin. Connect an external RC network to VOSNS- to program Dynamic VID slew rate. Programs the hiccup over-current threshold through an external resistor tied to VDAC and an internal current source. Over-current protection can be disabled by connecting this pin to a DC voltage no greater than 6.5V (do not float this pin as improper operation will occur). Current Sense input from the Phase IC(s). To ensure proper operation bias to at least 250mV (don’t float this pin). Buffered IIN signal. Connect an external RC network to FB to program converter output impedance Inverting input to the Error Amplifier. Converter output voltage is offset from the VDAC voltage through an external resistor connected to the converter output voltage at the load and an internal current source. Output of the Error Amplifier Input to the Regulation Detect Comparator. Connect to converter output voltage and VDRP pin through resistor network to program recovery from VID step-down. Connect to ground to diable Body BrakingTM during transition to a lower VID code. 6.8V/5mA Regulated output used as a system reference voltage for internal circuitry and the Phase ICs Power for internal circuitry Local Ground and IC substrate connection Oscillator Output voltage. Used by Phase ICs to program Phase Delay Inverting input to VRHOT comparator. Connect resistor divider from VBIAS to LGND to program VRHOT threshold. Diode or thermistor may be substituted for lower resistor for enhanced/remote temperature sensing. Applying a voltage exceeding approximately 7.5V disables the oscillator for factory testing. Open Collector output of the VRHOT comparator. Connect external pull-up. Controls Converter Softstart, Power Good, and Over-Current Delay Timing. Connect an external capacitor to LGND to program the timing. An optional resistor can be added in series with the capacitor to reduce the over-current delay time. Open Collector output that drives low during Softstart and any external fault condition. Connect external pull-up. Open Collector output of the VCCVID Power Good circuitry. Connect external pullup. Connect an external capacitor to LGND to program the VCCVID Power Good delay 9/30/04 IR3080 SYSTEM THEORY OF OPERATION XPhaseTM Architecture The XPhaseTM architecture is designed for multiphase interleaved buck converters which are used in applications requiring small size, design flexibility, low voltage, high current and fast transient response. The architecture can control converters of any phase number where flexibility facilitates the design trade-off of multiphase converters. The scalable architecture can be applied to other applications which require high current or multiple output voltages. As shown in Figure 1, the XPhaseTM architecture consists of a Control IC and a scalable array of phase converters each using a single Phase IC. The Control IC communicates with the Phase ICs through a 5-wire analog bus, i.e. bias voltage, phase timing, average current, error amplifier output, and VID voltage. The Control IC incorporates all the system functions, i.e. VID, PWM ramp oscillator, error amplifier, bias voltage, and fault protections etc. The Phase IC implements the functions required by the converter of each phase, i.e. the gate drivers, PWM comparator and latch, over-voltage protection, and current sensing and sharing. There is no unused or redundant silicon with the XPhaseTM architecture compared to others such as a 4 phase controller that can be configured for 2, 3, or 4 phase operation. PCB Layout is easier since the 5 wire bus eliminates the need for point-to-point wiring between the Control IC and each Phase. The critical gate drive and current sense connections are short and local to the Phase ICs. This improves the PCB layout by lowering the parasitic inductance of the gate drive circuits and reducing the noise of the current sense signal. VCCVID (1.2V@150mA) VID PWRGD VR HOT VCC PWRGD 12V 3.3V VID5 VID0 VID1 IR3080 CONTROL IC CIN >> BIAS VOLTAGE VOUT SENSE+ >> PHASE TIMING VID2 << CURRENT SENSE VID3 >> PWM CONTROL VID4 >> VID VOLTAGE CURRENT SHARE IR3086 PHASE IC VOUT+ 0.1uF COUT VOUT- CCS CURRENT SHARE IR3086 PHASE IC VOUT SENSE- 0.1uF CCS CONTROL BUS RCS RCS ADDITIONAL PHASES INPUT/OUTPUT Figure 1. System Block Diagram Page 7 of 41 9/30/04 IR3080 PWM Control Method The PWM block diagram of the XPhaseTM architecture is shown in Figure 2. Feed-forward voltage mode control with trailing edge modulation is used. A high-gain wide-bandwidth voltage type error amplifier in the Control IC is used for the voltage control loop. An external RC circuit connected to the input voltage and ground is used to program the slope of the PWM ramp and to provide the feed-forward control at each phase. The PWM ramp slope will change with the input voltage and automatically compensate for changes in the input voltage. The input voltage can change due to variations in the silver box output voltage or due to wire and PCB-trace voltage drop related to changes in load current. VIN CONTROL IC VPEAK RAMPIN+ RMPOUT RPHS1 VVALLEY PWM LATCH CLOCK PULSE GENERATOR RAMPIN- - EAIN VBIAS + - + + - RVFB X 0.91 + ISHARE FB 10K CURRENT SENSE AMPLIFIER 20mV IROSC X34 RDRP VDRP AMP CSIN+ + IFB VOSNS- CSCOMP - SHARE ADJUST ERROR AMPLIFIER BODY BRAKING COMPARATOR + RAMP DISCHARGE CLAMP SCOMP + ERROR AMP GND - CPWMRMP EAOUT GATEL ENABLE PWMRMP VOSNSVDAC VOUT COUT R + - RPWMRMP VOSNS+ RESET DOMINANT + RPHS2 VDAC VBIAS REGULATOR GATEH S PWM COMPARATOR - RAMP GENERATOR + 50% DUTY CYCLE PHASE IC SYSTEM REFERENCE VOLTAGE BIASIN CCS RCS CCS RCS CSIN- DACIN VDRP + - IIN RAMPIN+ PWM LATCH CLOCK PULSE GENERATOR + RPHS1 PHASE IC SYSTEM REFERENCE VOLTAGE BIASIN - RAMPIN- GATEH S PWM COMPARATOR - EAIN RESET DOMINANT R GATEL + RPHS2 ENABLE PWMRMP + RPWMRMP - X 0.91 - + SHARE ADJUST ERROR AMPLIFIER + ISHARE 10K 20mV CURRENT SENSE AMPLIFIER - CSIN+ + X34 - CSCOMP - CPWMRMP BODY BRAKING COMPARATOR + RAMP DISCHARGE CLAMP SCOMP CSIN- DACIN Figure 2. PWM Block Diagram Frequency and Phase Timing Control The oscillator is located in the Control IC and its frequency is programmable from 150kHz to 1MHZ by an external resistor. The output of the oscillator is a 50% duty cycle triangle waveform with peak and valley voltages of approximately 5V and 1V respectively. This signal is used to program both the switching frequency and phase timing of the Phase ICs. The Phase IC is programmed by resistor divider RPHS1 and RPHS2 connected between the VBIAS reference voltage and the Phase IC LGND pin. A comparator in the Phase ICs detects the crossing of the oscillator waveform over the voltage generated by the resistor divider and triggers a clock pulse that starts the PWM cycle. The peak and valley voltages track the VBIAS voltage reducing potential Phase IC timing errors. Figure 3 shows the Phase timing for an 8 phase converter. Note that both slopes of the triangle waveform can be used for phase timing by swapping the RMPIN+ and RMPIN– pins, as shown in Figure 2. Page 8 of 41 9/30/04 IR3080 50% RAMP DUTY CYCLE RAMP (FROM CONTROL IC) SLOPE = 80mV / % DC VPEAK (5.0V) VPHASE4&5 (4.5V) SLOPE = 1.6mV / ns @ 200kHz SLOPE = 8.0mV / ns @ 1MHz VPHASE3&6 (3.5V) VPHASE2&7 (2.5V) VPHASE1&8 (1.5V) VVALLEY (1.00V) CLK1 PHASE IC CLOCK PULSES CLK2 CLK3 CLK4 CLK5 CLK6 CLK7 CLK8 Figure 3. 8 Phase Oscillator Waveforms PWM Operation The PWM comparator is located in the Phase IC. Upon receiving a clock pulse, the PWM latch is set; the PWMRMP voltage begins to increase; the low side driver is turned off, and the high side driver is then turned on after the nonoverlap time. When the PWMRMP voltage exceeds the Error Amplifier’s output voltage, the PWM latch is reset. This turns off the high side driver and then turns on the low side driver after the non-overlap time; it activates the Ramp Discharge Clamp, which quickly discharges the PWMRMP capacitor to the VDAC voltage of the Control IC until the next clock pulse. The PWM latch is reset dominant allowing all phases to go to zero duty cycle within a few tens of nanoseconds in response to a load step decrease. Phases can overlap and go to 100% duty cycle in response to a load step increase with turn-on gated by the clock pulses. An Error Amplifier output voltage greater than the common mode input range of the PWM comparator results in 100% duty cycle regardless of the voltage of the PWM ramp. This arrangement guarantees the Error Amplifier is always in control and can demand 0 to 100% duty cycle as required. It also favors response to a load step decrease which is appropriate given the low output to input voltage ratio of most systems. The inductor current will increase much more rapidly than decrease in response to load transients. This control method is designed to provide “single cycle transient response” where the inductor current changes in response to load transients within a single switching cycle maximizing the effectiveness of the power train and minimizing the output capacitor requirements. An additional advantage of the architecture is that differences in ground or input voltage at the phases have no effect on operation since the PWM ramps are referenced to VDAC. Figure 4 depicts PWM operating waveforms under various conditions. Page 9 of 41 9/30/04 IR3080 PHASE IC CLOCK PULSE EAIN PWMRMP VDAC 91% VDAC GATEH GATEL STEADY-STATE OPERATION DUTY CYCLE INCREASE DUE TO LOAD INCREASE DUTY CYCLE DECREASE DUE TO VIN INCREASE (FEED-FORWARD) DUTY CYCLE DECREASE DUE TO LOAD DECREASE (BODY BRAKING) OR FAULT (VCC UV, VCCVID UV, OCP, VID=11111X) STEADY-STATE OPERATION Figure 4. PWM Operating Waveforms Body BrakingTM In a conventional synchronous buck converter, the minimum time required to reduce the current in the inductor in response to a load step decrease is; TSLEW = L * ( I MAX − I MIN ) VO The slew rate of the inductor current can be significantly increased by turning off the synchronous rectifier in response to a load step decrease. The switch node voltage is then forced to decrease until conduction of the synchronous rectifier’s body diode occurs. This increases the voltage across the inductor from Vout to Vout + VBODYDIODE. The minimum time required to reduce the current in the inductor in response to a load transient decrease is now; TSLEW = L * ( I MAX − I MIN ) VO + VBODYDIODE Since the voltage drop in the body diode is often higher than output voltage, the inductor current slew rate can be increased by 2X or more. This patent pending technique is referred to as “body braking” and is accomplished through the “0% Duty Cycle Comparator” located in the Phase IC. If the Error Amplifier’s output voltage drops below 91% of the VDAC voltage this comparator turns off the low side gate driver. Lossless Average Inductor Current Sensing Inductor current can be sensed by connecting a series resistor and a capacitor network in parallel with the inductor and measuring the voltage across the capacitor, as shown in Figure 5. The equation of the sensing network is, vC ( s) = vL ( s) 1 RL + sL = iL ( s) 1 + sRCS CCS 1 + sRCS CCS Usually the resistor Rcs and capacitor Ccs are chosen so that the time constant of Rcs and Ccs equals the time constant of the inductor which is the inductance L over the inductor DCR. If the two time constants match, the voltage across Ccs is proportional to the current through L, and the sense circuit can be treated as if only a sense resistor with the value of RL was used. The mismatch of the time constants does not affect the measurement of inductor DC current, but affects the AC component of the inductor current. Page 10 of 41 9/30/04 IR3080 vL iL Current Sense Amp L RL RCS CCS VO CO vCc S CSOUT Figure 5. Inductor Current Sensing and Current Sense Amplifier The advantage of sensing the inductor current versus high side or low side sensing is that actual output current being delivered to the load is obtained rather than peak or sampled information about the switch currents. The output voltage can be positioned to meet a load line based on real time information. Except for a sense resistor in series with the inductor, this is the only sense method that can support a single cycle transient response. Other methods provide no information during either load increase (low side sensing) or load decrease (high side sensing). An additional problem associated with peak or valley current mode control for voltage positioning is that they suffer from peak-to-average errors. These errors will show in many ways but one example is the effect of frequency variation. If the frequency of a particular unit is 10% low, the peak to peak inductor current will be 10% larger and the output impedance of the converter will drop by about 10%. Variations in inductance, current sense amplifier bandwidth, PWM prop delay, any added slope compensation, input voltage, and output voltage are all additional sources of peak-to-average errors. Current Sense Amplifier A high speed differential current sense amplifier is located in the Phase IC, as shown in Figure 5. Its gain decreases with increasing temperature and is nominally 34 at 25ºC and 29 at 125ºC (-1470 ppm/ºC). This reduction of gain tends to compensate the 3850 ppm/ºC increase in inductor DCR. Since in most designs the Phase IC junction is hotter than the inductors these two effects tend to cancel such that no additional temperature compensation of the load line is required. The current sense amplifier can accept positive differential input up to 100mV and negative up to -20mV before clipping. The output of the current sense amplifier is summed with the DAC voltage and sent to the Control IC and other Phases through an on-chip 10KΩ resistor connected to the ISHARE pin. The ISHARE pins of all the phases are tied together and the voltage on the share bus represents the average current through all the inductors and is used by the Control IC for voltage positioning and current limit protection. Average Current Share Loop Current sharing between phases of the converter is achieved by the average current share loop in each Phase IC. The output of the current sense amplifier is compared with the share bus less a 20mV offset. If current in a phase is smaller than the average current, the share adjust amplifier of the phase will activate a current source that reduces the slope of its PWM ramp thereby increasing its duty cycle and output current. The crossover frequency of the current share loop can be programmed with a capacitor at the SCOMP pin so that the share loop does not interact with the output voltage loop. Page 11 of 41 9/30/04 IR3080 IR3080 THEORY OF OPERATION Block Diagram The Block diagram of the IR3080 is shown in Figure 6, and specific features are discussed in the following sections. + IVTTDEL 0.2V - 4V - DELAY COMPARATOR SS/DEL DISCHARGE ON + SS/DEL VDRP VDRP AMP IIN OCSET DISABLE 1.3V + 70uA + OVER-CURRENT COMPARATOR - + + + OFF - OVER CURRENT - + + 90mV 6uA R + - 66uA IDISCHG DISCHARGE COMPARATOR SET DOMINANT - - VIDDEL ICHG PWRGD S VID DELAY COMPARATOR + VCC UVLO COMPARATOR - FAULT LATCH - 9.1V 8.9V VIDPGD START STOP + + + VCC SOFTSTART CLAMP EAOUT ERROR AMP FB - VIDPWR VID = 11111X VID STEP-DOWN VID3 IROSC IROSC IROSC VID CONTROL IROSC VID4 IROSC BBFB IROSC IROSC IROSC VID1 VID2 + - VID0 + IROSC IROSC VID5 1.2V - VID DAC OUTPUT VCCVID REGULATOR + 1.1V + - VCCVID COMPARATOR + VBIAS RAMP GENERATOR IROSC IROSC IOCSET IFB IROSC CURRENT SOURCE GENERATOR ROSC BUFFER AMP - 1.0V 6.85V - + 5.0V LGND + - VBIAS REGULATOR VBIAS RMPOUT VIDFB VDAC VOSNS- 50% DUTY CYCLE VCCVID ROSC VOLTAGE PROPORTIONAL TO ABSOLUTE TEMPERATURE + - VRHOT + START STOP - VRHOT COMPARATOR HOTSET Figure 6. IR3080 Block Diagram VID Control A 6-bit VID voltage compatible with VR 10, as shown in Table 1, is available at the VDAC pin. A detailed block diagram of the VID control circuitry can be found in Figure 7. The VID pins are internally pulled up to VIDPWR pin through 700Ω resistors. The VID input comparators, with 0.6V reference, monitor the VID pins and control the 6 bit Digital-to-Analog Converter (DAC) whose output is sent to the VDAC buffer amplifier. The output of the buffer amplifier is the VDAC pin. The VDAC voltage is post-package trimmed to compensate for the input offsets of the Error Amplifier to provide a 0.5% system set-point accuracy. The actual VDAC voltage does not determine the system accuracy and has a wider tolerance. Page 12 of 41 9/30/04 IR3080 The IR3080 can accept changes in the VID code while operating and vary the DAC voltage accordingly. The sink/source capability of the VDAC buffer amplifier is programmed by the same external resistor that sets the oscillator frequency. The slew rate of the voltage at the VDAC pin can be adjusted by an external capacitor between VDAC pin and the VOSNS- pin. A resistor connected in series with this capacitor is required to compensate the VDAC buffer amplifier. Digital VID transitions result in a smooth analog transition of the VDAC voltage and converter output voltage minimizing inrush currents in the input and output capacitors and overshoot of the output voltage. It is desirable to prevent negative inductor currents in response to a request for a lower VID code. Negative current transforms the buck converter into a boost converter and transfers energy from the output capacitors back into the input voltage. This energy can cause voltage spikes and damage the silver box or other components unless they are specifically designed to handle it. Furthermore, power is wasted during the transfer of energy from the output back to the input. The IR3080 includes circuitry that turns off both control and synchronous MOSFETs in response to a lower VID code so that the load current instead of the inductor discharges the output capacitors. A lower VID code is detected by the VID step-down detect comparator which monitors the “fast” output of the DAC (plus 7mV for noise immunity) compared to the “slow” output of the VDAC pin. If a dynamic VID step down is detected, the body brake latch is set and the output of the error amplifier is pulled down to 75% of the DAC voltage by the VID body brake clamp. This triggers the Body BrakingTM function, which turns off both high side and low side drivers in the phase ICs. The converter’s output voltage needs to be monitored and compared to the VDAC voltage to determine when to resume normal operation. Unfortunately, the voltage on the FB pin can be pulled down by its compensation network during the sudden decrease in the Error Amplifier’s output voltage so an additional pin BBFB is provided. The BBFB pin is connected to the converter output voltage and VDRP pin with resistors of the same value as on the FB pin and therefore provides an un-corrupted representation of converter output voltage. The regulation detect comparator compares the BBFB to the VDAC voltage and resets the body brake latch releasing the error amplifier’s output and allowing normal operation to resume. Body BrakingTM during a transition to a lower VID code can be disabled by connecting the BBFB pin to ground. VIDPWR 800ns BLANKING 700 VID5 VID0 DIGITAL TO ANALOG CONVERTER VID INPUT COMPARATORS (1 OF 6 SHOWN) VID1 VID2 + VDAC BUFFER AMP "FAST" VDAC + VID3 "SLOW" VDAC ISOURCE VID4 + + VID = 11111X DETECT - VDAC ISINK - - 0.6V VOSNS- - EAOUT VID DOWN BB CLAMP 7mV + + 75% - ENABLE VID STEP-DOWN DETECT COMPARATOR + IBBFB + RESET DOMINANT S BODY BRAKE LATCH 1.7us BLANKING R REGULATION DETECT COMPARATOR BBFB - TO ERROR AMP IROSC (From Current Source Generator) Figure 7. VID Control Block Diagram Page 13 of 41 9/30/04 IR3080 Processor Pins (0 = low, 1 = high) VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID3 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID2 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 VID1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 VID0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 VID5 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 Vout (V) 0.8375 0.8500 0.8625 0.8750 0.8875 0.9000 0.9125 0.9250 0.9375 0.9500 0.9625 0.9750 0.9875 1.0000 1.0125 1.0250 1.0375 1.0500 1.0625 1.0750 1.0875 OFF4 OFF4 1.1000 1.1125 1.1250 1.1375 1.1500 1.1625 1.1750 1.1875 1.2000 Processor Pins (0 = low, 1 = high) VID4 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 VID3 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID2 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 VID1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 VID0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 VID5 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 Vout (V) 1.2125 1.2250 1.2375 1.2500 1.2625 1.2750 1.2875 1.3000 1.3125 1.3250 1.3375 1.3500 1.3625 1.3750 1.3875 1.4000 1.4125 1.4250 1.4375 1.4500 1.4625 1.4750 1.4875 1.5000 1.5125 1.5250 1.5375 1.5500 1.5625 1.5750 1.5875 1.6000 Note: 3. Output disabled (Fault mode) Table 1 - Voltage Identification (VID) Adaptive Voltage Positioning Adaptive voltage positioning is needed to reduce the output voltage deviations during load transients and the power dissipation of the load when it is drawing maximum current. The circuitry related to voltage positioning is shown in Figure 8. Resistor RFB is connected between the Error Amplifier’s inverting input pin FB and the converter’s output voltage. An internal current source whose value is programmed by the same external resistor that programs the oscillator frequency pumps current into the FB pin. The error amplifier forces the converter’s output voltage lower to maintain a balance at its inputs. RFB is selected to program the desired amount of fixed offset voltage below the DAC voltage. The voltage at the VDRP pin is a buffered version of the share bus and represents the sum of the DAC voltage and the average inductor current of all the phases. The VDRP pin is connected to the FB pin through the resistor RDRP. Since the Error Amplifier will force the loop to maintain FB to be equal to the VDAC reference voltage, an additional current will flow into the FB pin equal to (VDRP-VDAC) / RDRP. When the load current increases, the adaptive positioning voltage increases accordingly. More current flows through the feedback resistor RFB, and makes the output voltage lower proportional to the load current. The positioning voltage can be programmed by the resistor RDRP so that the droop impedance produces the desired converter output impedance. The offset and slope of the converter output impedance are referenced to and therefore independent of the VDAC voltage. Page 14 of 41 9/30/04 IR3080 Control IC Current Sense Amplifier + ISHARE + EAOUT Vo 10k VDAC - FB RFB RDRP VDRP Amplifier CSIN- Phase IC VDRP Current Sense Amplifier - + ISHARE VDAC - IIN + CSIN+ ... ... IFB - VDAC Phase IC Error Amplifier 10k CSIN+ CSIN- Figure 8. Adaptive voltage positioning Inductor DCR Temperature Correction If the thermal compensation of the inductor DCR provided by the temperature dependent gain of the current sense amplifier is not adequate, a negative temperature coefficient (NTC) thermistor can be used for additional correction. The thermistor should be placed close to the inductor and connected in parallel with the feedback resistor, as shown in Figure 9. The resistor in series with the thermistor is used to reduce the nonlinearity of the thermistor. A similar network must be placed on the BBFB pin to ensure proper operation during a transition to a lower VID code with Body BrakingTM. Control IC VDAC Error Amplifier + EAOUT - IFB AVP Amplifier Vo RFB FB RFB2 Rt RDRP VDRP + IIN Figure 9. Temperature compensation of inductor DCR Remote Voltage Sensing To reduce the effect of impedance in the ground plane, the VOSNS- pin is used for remote sensing and connected directly to the load. The VDAC voltage is referenced to VOSNS- to avoid additional error terms or delay related to a separate differential amplifier. The capacitor connecting the VDAC and VOSNS- pins ensure that high speed transients are fed directly into the error amplifier without delay. Page 15 of 41 9/30/04 IR3080 VCC VID Linear Regulator and VID Power Good The IR3080 integrates a fully protected 1.2V/150mA VCCVID linear regulator with over-current protection. Power for the VCCVID regulator is drawn from the VIDPWR pin which is typically connected to a 3.3V supply. If the linear regulator output voltage is above 1.1V for a time period programmed by a capacitor between VIDDEL and LGND, the VIDPGD pin will send out a VID power good signal and start-up of the converter is allowed. The PWRGD pin is an open-collector output and should be pulled up to a voltage source through a resistor. 26 25 LGND HOTSET 27 28 VRHOT 30 29 SS/DEL PWRGD RMPOUT VDRP 24 23 22 21 20 19 18 VDAC 17 16 TRM4 ROSC 15 13 TRM3 IIN OCSET 14 1uF FB VID3 9 0.1uF VIDPGD VID1 EAOUT VID2 VID4 VCCVID VID5 VID0 VOSNS- 8 BBFB IR3080 CONTROL IC 12 7 VIDPWR TRM1 6 VBIAS TRM2 5 2.5-3.3V VCC VCCVID 11 3 4 VIDFB VIDDEL 32 1.8V 2 10 1 5 OHM 31 An external NPN transistor can be used to enhance the current capability of the linear regulator, as shown in Figure 10. The 5 ohm resistor provides loop compensation and over-current protection. VOUT SENSE- Figure 10. VCC VID linear regulator using external transistor Soft Start, Over-Current Fault Delay, and Hiccup Mode The IR3080 has a programmable soft-start function to limit the surge current during the converter start-up. A capacitor connected between the SS/DEL and LGND pins controls soft start as well as over-current protection delay and hiccup mode timing. A charge current of 70uA and discharge current of 6uA control the up slope and down slope of the voltage at the SS/DEL pin respectively Figure 11 depicts the various operating modes as controlled by the SS/DEL function. If there is no fault, the SS/DEL capacitor pin will begin to be charged. The error amplifier output is clamped low until SS/DEL reaches 1.3V. The error amplifier will then regulate the converter’s output voltage to match the SS/DEL voltage less the 1.3V offset until it reaches the level determined by the VID inputs. The SS/DEL voltage continues to increase until it rises above 3.91V and allows the PWRGD signal to be asserted. SS/DEL finally settles at 4V, indicating the end of the soft start. Under Voltage Lock Out, a VID=11111x, and VCC VID faults immediately set the fault latch causing SS/DEL to begin to discharge. The SS/DEL capacitor will continue to discharge down to 0.2V. If the fault has cleared the fault latch will be reset by the discharge comparator allowing a normal soft start to occur. A delay is included if an over-current condition occurs after a successful soft start sequence. This is required since over-current conditions can occur as part of normal operation due to load transients or VID transitions. If an overcurrent fault occurs during normal operation it will initiate the discharge of the capacitor at SS/DEL but will not set the fault latch immediately. If the over-current condition persists long enough for the SS/DEL capacitor to discharge below the 90mV offset of the delay comparator, the Fault latch will be set pulling the error amplifier’s output low inhibiting switching in the phase ICs and de-asserting the PWRGD signal. The SS/DEL capacitor will continue to discharge until it reaches 0.2V and the fault latch is reset allowing a normal soft start to occur. If an over-current condition is again encountered during the soft start cycle the fault latch will be set without any delay and hiccup mode will begin. During hiccup mode the charge to discharge current ratio results in a fixed 7.9% hiccup mode duty cycle regardless of at what point the over-current condition occurs. However, the hiccup frequency is determined by the load current and over-current set value. Page 16 of 41 9/30/04 IR3080 The over-current delay can be reduced by adding a resistor in series with the SS/DEL capacitor. The delay comparator’s offset voltage is reduced by the drop in the resistor caused by the discharge current. The value of the series resistor should be 10KΩ or less to avoid interference with the soft start function. If SS/DEL pin is pulled below 0.9V, the converter can be disabled. 8.9V UVLO VCC (12V) VIDPWR (3.3V) VCCVID 1.1V (1.2V) 3.91V VIDDEL VIDPGD 3.91V SS/DEL 1.3V VOUT PWRGD OCP THRESHOLD IOUT START-UP (VCCVID GATES FAULT MODE) NORMAL OPERATION (VOUT CHANGES DUE TO LOAD AND VID CHANGES) OCP DELAY HICCUP OVER-CURRENT PROTECTION RE-START AFTER OCP POWER-DOWN (VCC GATES FAULT MODE) Figure 11. Operating Waveforms Under Voltage Lockout (UVLO) The UVLO function monitors the IR3080’s VCC supply pin and ensures that IR3080 has a high enough voltage to power the internal circuit. The IR3080’s UVLO is set higher than the minimum operating voltage of compatible Phase ICs thus providing UVLO protection for them as well. During power-up the fault latch is reset when VCC exceeds 9.1V and there is no other fault. If the VCC voltage drops below 8.9V the fault latch will be set. For converters using a separate 5V supply for gate driver bias an external UVLO circuit can be added to prevent any operation until adequate voltage is present. A diode connected between the 5V supply and the SS/DEL pin provides a simple 5V UVLO function. UVLO of the VIDPWR input is provided by the VID Power Good function. Adequate voltage must be present at VIDPWR pin to allow VCCVID to reach 1.1V. Over Current Protection (OCP) The current limit threshold is set by a resistor connected between the OCSET and VDAC pins. If the IIN pin voltage, which is proportional to the average current plus DAC voltage, exceeds the OCSET voltage, the over-current protection is triggered. Page 17 of 41 9/30/04 IR3080 VID = 11111X Fault VID codes of 111111 and 111110 will set the fault latch and disable the error amplifier. An 800ns delay is provided to prevent a fault condition from occurring during Dynamic VID changes. Power Good Output The PWRGD pin is an open-collector output and should be pulled up to a voltage source through a resistor. During soft start, the PWRGD remains low until the output voltage is in regulation and SS/DEL is above 3.91V. The PWRGD pin becomes low if the fault latch is set. A high level at the PWRGD pin indicates that the converter is in operation and has no fault, but does not ensure the output voltage is within the specification. Output voltage regulation within the design limits can logically be assured however, assuming no component failure in the system. Load Current Indicator Output The VDRP pin voltage represents the average current of the converter plus the DAC voltage. The load current can be retrieved by a differential amplifier which subtracts the VDAC voltage from the VDRP voltage. System Reference Voltage (VBIAS) The IR3081 supplies a 6.8V/5mA precision reference voltage from the VBIAS pin. The oscillator ramp amplitude tracks the VBIAS voltage, which should be used to program the Phase IC trip points to minimize phase delay errors. Thermal Monitoring (VRHOT) The IR3080 senses its own die temperature and produces a voltage at the input of the VRHOT comparator that is proportional to temperature. An external resistor divider connected from VBIAS to the HOTSET pin and ground can be used to program the thermal trip point of the VRHOT comparator. The VRHOT pin is an open-collector output and should be pulled up to a voltage source through a resistor. If the thermal trip point is reached the VRHOT output drives low. Pulling HOTSET above approximately 7.5V will disable the oscillator (used during factory testing). Page 18 of 41 9/30/04 IR3080 APPLICATION INFORMATIONS VCCVID POWERGOOD VRHOT VIDPGD 0.1uF 12V RCS- VGATE QGATE CCS+ RVCC 10 ohm 16 18 19 20 17 CSIN- DACIN BIASIN CSIN+ GATEH PGND GATEL CIN VOUT SENSE+ 15 14 L 13 VOUT+ 12 DISTRIBUTION IMPEDANCE 11 COUT VCC VOUT10 LGND VCCL 9 PWMRMP SCOMP 6 EAIN ISHARE CVCCL RVCC VOUT SENSE- RPWMRMP CVCC RBBDRP 22 21 RCP 20 RCS- CCP CCS+ HOTSET VRHOT RPHASE23 6 SCOMP RPHASE22 16 CSIN+ 17 CSIN- 18 ISHARE CSCOMP GATEH PGND GATEL VCCL CIN 15 14 L 13 12 11 VCC 4 CBST VCCH IR3086 PHASE IC RMPIN- 10 RSHARE 5 CVDAC RMPIN+ LGND 3 RVDAC 19 20 1 RCS+ DBST 9 ROCSET CCS- 20k DACIN RDRP RBIASIN BIASIN 17 CCP1 PHSFLT RDRP1 CDRP 18 PWMRMP 19 EAIN ROSC VRHOT CSCOMP 2 1uF HOTSET RPHASE12 RPHASE13 23 CBST2 VCCH IR3086 PHASE IC RMPIN- RPHASE21 ROSC RFB 24 VDAC 16 15 TRM4 IIN RFB1 RMPIN+ 7 RHOTSETC2 26 VDRP OCSET TRM3 5 25 LGND 27 29 30 31 28 VRHOT SS/DEL HOTSET VIDDEL PWRGD RMPOUT FB VID3 9 VID4 BBFB EAOUT VID2 VOSNS- 8 14 VID3 VID1 13 VID2 7 VBIAS VID0 4 CFB RBBFB VCC IR3080 CONTROL IC VID5 12 6 0.1uF RHOTSETC1 VCCVID VIDPWR 3 CPWMRMP 8 CSS/DEL RSS/DEL CVIDDEL VID1 VIDPGD 32 5 TRM2 4 VID0 TRM1 3 VID5 VID4 3.3V VIDFB 11 2 2 1nF 10 1 1 RCS+ DBST3 PHSFLT RPHASE11 CVCC 0.1uF CCS- 7 DGATE RGATE 20k CPWMRMP 8 RBIASIN CVCCL RVCC RPWMRMP CVCC RCSCCS+ CCS- 20k 16 CSIN+ 17 CSIN- 18 19 20 PGND GATEL CIN 15 14 L 13 12 11 VCC 10 SCOMP CSCOMP GATEH VCCL LGND ISHARE 9 VRHOT 6 RPHASE33 DACIN BIASIN HOTSET RPHASE32 5 PWMRMP 4 CBST VCCH IR3086 PHASE IC RMPIN- EAIN 3 RMPIN+ 7 2 PHSFLT RPHASE31 1 RCS+ DBST CPWMRMP 8 RBIASIN CVCCL RVCC RPWMRMP CVCC RCSCCS+ CCS- 20k 16 CSIN+ 17 CSIN- 18 19 20 PGND GATEL CIN 15 14 L 13 12 11 VCC 10 SCOMP CSCOMP GATEH VCCL LGND ISHARE 9 VRHOT 6 RPHASE43 DACIN BIASIN HOTSET RPHASE42 5 CBST VCCH IR3086 PHASE IC RMPIN- PWMRMP 4 RMPIN+ EAIN 3 7 2 PHSFLT RPHASE41 1 RCS+ DBST CPWMRMP 8 RBIASIN CVCCL RVCC RPWMRMP CVCC RCSCCS+ CCS- 20k 16 CSIN+ 17 CSIN- 18 19 20 PGND GATEL CIN 15 14 L 13 12 11 VCC 10 SCOMP CSCOMP GATEH VCCL LGND ISHARE 9 VRHOT 6 RPHASE53 DACIN BIASIN HOTSET RPHASE52 5 PWMRMP 4 CBST VCCH IR3086 PHASE IC RMPIN- EAIN 3 RMPIN+ 7 2 PHSFLT RPHASE51 1 RCS+ DBST CPWMRMP 8 RBIASIN CVCCL RVCC RPWMRMP CVCC RCSCCS+ 16 CSIN+ 17 CSIN- 18 20 19 DACIN BIASIN PGND GATEL VCCL CIN 15 14 L 13 12 11 VCC SCOMP 6 CSCOMP GATEH 10 ISHARE LGND VRHOT 9 HOTSET RPHASE62 5 PWMRMP 4 CBST VCCH IR3086 PHASE IC RMPIN- EAIN 3 RMPIN+ 7 2 RCS+ DBST PHSFLT RPHASE61 1 RPHASE63 CCS- 20k CPWMRMP 8 RBIASIN CVCCL RVCC RPWMRMP CVCC Figure 12. IR3080/IR3086 Six Phase VRD 10 Converter Page 19 of 41 9/30/04 IR3080 DESIGN PROCEDURES - IR3080 AND IR3086 CHIPSET IR3080 EXTERNAL COMPONENTS Oscillator Resistor Rosc The oscillator of IR3080 generates a triangle waveform to synchronize the phase ICs, and the switching frequency of the each phase converter equals the oscillator frequency, which is set by the external resistor ROSC according to the curve in Figure 13. VID Delay Capacitor CVIDDEL After the VID voltage of the integrated linear regulator is above 1.1V, there is a time delay before the soft start of the converter is initiated. The VID delay time tVID can be programmed by an external capacitor between VIDDEL pin and LGND, and the capacitance is determined by (1). CVIDDEL = 66 *10−6 ∗ tVID 3.91 (1) Soft Start Capacitor CSS/DEL and Resistor RSS/DEL Because the capacitor CSS/DEL programs four different time parameters, i.e. soft start delay time, soft start time, over-current latch delay time, and power good delay time, they should be considered together while choosing CSS/DEL. The SS/DEL pin voltage controls the slew rate of the converter output voltage, as shown in Figure 11. After the VIDDEL pin voltage rises above 3.91V, there is a soft-start delay time tSSDEL, after which the error amplifier output is released to allow the soft start. The soft start time tSS represents the time during which converter voltage rises from zero to VO. tSS can be programmed by an external capacitor, which is determined by Equation (2). C SS / DEL = I CHG * t SS 70 * 10 −6 * t SS = VO VO (2) Once CSS/DEL is chosen, the soft start delay time tSSDEL, the over-current fault latch delay time tOCDEL, and the delay time tVccPG from output voltage (VO) in regulation to Power Good are fixed and shown in Equations (3), (4) and (5) respectively. *1.3 CSS / DEL *1.3 C (3) = tSSDEL = SS / DEL I CHG 70 *10−6 t OCDEL = tVccPG = C SS / DEL * 0.09 C SS / DEL * 0.09 = I DISCHG 6 *10 −6 CSS / DEL * (3.91 − VO − 1.3) CSS / DEL * (3.91 − VO − 1.3) = I CHG 70 *10−6 (4) (5) If faster over-current protection is required, a resistor in series with the soft start capacitor CSS/DEL can be used to reduce the over-current fault latch delay time tOCDEL, and the resistor RSS/DEL is determined by Equation (6). Equation (2) for soft start capacitor CSS/DEL and Equation (5) for power good delay time tVccPG are unchanged, while the equation for soft start delay time tSS/DEL (Equation 3) is changed to Equation (7). Considering the worst case values of charge and discharge current, RSS/DEL should not be greater than 10 kΩ. *I t t ∗ 6 *10−6 0.09 − OCDEL DISCHG 0.09 − OCDEL CSS / DEL CSS / DEL (6) RSS / DEL = − I DISCHG 6 *10−6 Page 20 of 41 9/30/04 IR3080 t SSDEL = C SS / DEL * (1.3 − R SS / DEL * I CHG ) C SS / DEL * (1.3 − R SS / DEL ∗ 70 *10 −6 ) = I CHG 70 *10 − 6 (7) VDAC Slew Rate Programming Capacitor CVDAC and Resistor RVDAC The slew rate of VDAC down-slope SRDOWN can be programmed by the external capacitor CVDAC as defined in Equation (8), where ISINK is the sink current of VDAC pin as shown in Figure 15. The resistor RVDAC is used to compensate VDAC circuit and is determined by Equation (9). The slew rate of VDAC up-slope SRUP is proportional to that of VDAC down-slope and is given by Equation (10), where ISOURCE is the source current of VDAC pin as shown in Figure15. CVDAC = I SINK SR DOWN RVDAC = 0.5 + SRUP = 3.2 ∗ 10 −15 CVDAC 2 I SOURCE CVDAC (8) (9) (10) Over Current Setting Resistor ROCSET The inductor DC resistance is utilized to sense the inductor current. The copper wire of inductor has a constant temperature coefficient of 3850 ppm/°C, and therefore the maximum inductor DCR can be calculated from Equation (11), where RL_MAX and RL_ROOM are the inductor DCR at maximum temperature TL_MAX and room temperature T_ROOM respectively. R L _ MAX = R L _ ROOM ∗ [1 + 3850 * 10 −6 ∗ (T L _ MAX − TROOM )] (11) The current sense amplifier gain of IR3086 decreases with temperature at the rate of 1470 ppm/°C, which compensates part of the inductor DCR increase. The phase IC die temperature is only a couple of degrees Celsius higher than the PCB temperature due to the low thermal impedance of MLPQ package. The minimum current sense amplifier gain at the maximum phase IC temperature TIC_MAX is calculated from Equation (12). GCS _ MIN = GCS _ ROOM ∗ [1 − 1470 * 10 −6 ∗ (TIC _ MAX − TROOM )] (12) The total input offset voltage (VCS_TOFST) of current sense amplifier in phase ICs is the sum of input offset (VCS_OFST) of the amplifier itself and that created by the amplifier input bias currents flowing through the current sense resistors RCS+ and RCS-. VCS _ TOFST = VCS _ OFST + I CSIN + ∗ RCS + − I CSIN − ∗ RCS − (13) The over current limit is set by the external resistor ROCSET as defined in Equation (14), where ILIMIT is the required over current limit. IOCSET, the bias current of OCSET pin, changes with switching frequency setting resistor ROSC and is determined by the curve in Figure 14. KP is the ratio of inductor peak current over average current in each phase and is calculated from Equation (15). ROCSET = [ KP = Page 21 of 41 I LIMIT ∗ RL _ MAX ∗ (1 + K P ) + VCS _ TOFST ]GCS _ MIN / IOCSET n (VI − VO ) ∗ VO /( L ∗ VI ∗ f SW ∗ 2) IO / n (14) (15) 9/30/04 IR3080 No Load Output Voltage Setting Resistor RFB and Adaptive Voltage Positioning Resistor RDRP A resistor between FB pin and the converter output is used to create output voltage offset VO_NLOFST, which is the difference between VDAC voltage and output voltage at no load condition. Adaptive voltage positioning further lowers the converter voltage by RO*IO, where RO is the required output impedance of the converter. RFB is not only determined by IFB, the current flowing out of FB pin as shown in Figure 14, but also affected by the adaptive voltage positioning resistor RDRP and total input offset voltage of current sense amplifiers. RFB and RDRP are determined by (16) and (17) respectively, R FB = R L _ MAX ∗ VO _ NLOFST − VCS _ TOFST ∗ n ∗ RO R DRP = I FB ∗ R L _ MAX R FB ∗ R L _ MAX ∗ GCS _ MIN n ∗ RO (16) (17) Control IC Over Temperature Setting Resistors RHOTSETC1 and RHOTSETC2 The threshold voltage of VRHOT comparator is proportional to the die temperature TJ (ºC) of control IC IR3080, as shown in Equation (18). Determine the relationship between the die temperature of IR3080 and the temperature of the power converter according to the power loss, PCB layout and airflow, etc. Then calculate the VRHOT threshold voltage corresponding to the temperature. VVRHOT = 4.73 *10−3 * TJ + 1.241 (18) Use VBIAS as the reference voltage. If RHOTSETC1 is pre selected, RHOTSETC2 can be calculated according to Equation (19). VVRHOT (19) RHOTSETC 2 = RHOTSETC1 VVBIAS + VVRHOT Body BrakingTM Related Resistors RBBFB and RBBDRP The body brakingTM during Dynamic VID can be disabled by connecting BBFB pin to ground. If the feature is enabled, Resistors RBBFB and RBBDRP are needed to restore the feedback voltage of the error amplifier after Dynamic VID step down. Usually RBBFB and RBBDRP are chosen to match RFB and RDRP respectively. IR3086 EXTERNAL COMPONENTS PWM Ramp Resistor RPWMRMP and Capacitor CPWMRMP PWM ramp is generated by connecting the resistor RPWMRMP between a voltage source and PWMRMP pin as well as the capacitor CPWMRMP between PWMRMP and LGND. Choose the desired PWM ramp magnitude VPWMRMP and the capacitor CPWMRMP in the range of 100pF and 470pF, and then calculate the resistor RPWMRMP from Equation (20). To achieve feed-forward voltage mode control, the resistor RRAMP should be connected to the input of the converter. VO (20) RPWMRMP = VIN * f SW * CPWMRMP * [ln(VIN − VDAC ) − ln(VIN − VDAC − VPWMRMP )] Inductor Current Sensing Capacitor CCS+ and Resistors RCS+ and RCSThe DC resistance of the inductor is utilized to sense the inductor current. Usually the resistor RCS+ and capacitor CCS+ in parallel with the inductor are chosen to match the time constant of the inductor, and therefore the voltage across the capacitor CCS+ represents the inductor current. If the two time constants are not the same, the AC component of the capacitor voltage is different from that of the real inductor current. The time constant mismatch does not affect the average current sharing among the multiple phases, but affect the current signal ISHARE as well as the output voltage during the load current transient if adaptive voltage positioning is adopted. Page 22 of 41 9/30/04 IR3080 Measure the inductance L and the inductor DC resistance RL. Pre-select the capacitor CCS+ and calculate RCS+ as follows. L RL (21) RCS + = C CS + The bias current flowing out of the non-inverting input of the current sense amplifier creates a voltage drop across RCS+, which is equivalent to an input offset voltage of the current sense amplifier. The offset affects the accuracy of converter current signal ISHARE as well as the accuracy of the converter output voltage if adaptive voltage positioning is adopted. To reduce the offset voltage, a resistor RCS- should be added between the amplifier inverting input and the converter output. The resistor RCS- is determined by the ratio of the bias current from the non-inverting input and the bias current from the inverting input. RCS − = I CSIN + ∗ RCS + I CSIN − (22) If RCS- is not used, RCS+ should be chosen so that the offset voltage is small enough. Usually RCS+ should be less than 2 kΩ and therefore a larger CCS+ value is needed. Over Temperature Setting Resistors RHOTSET1 and RHOTSET2 The threshold voltage of VRHOT comparator is proportional to the die temperature TJ (ºC) of phase IC. Determine the relationship between the die temperature of phase IC and the temperature of the power converter according to the power loss, PCB layout and airflow etc, and then calculate HOTSET threshold voltage corresponding to the allowed maximum temperature from Equation (23). V HOTSET = 4.73 * 10 −3 * T J + 1.241 (23) There are two ways to set the over temperature threshold, central setting and local setting. In the central setting, only one resistor divider is used, and the setting voltage is connected to HOTSET pins of all the phase ICs. To reduce the influence of noise on the accuracy of over temperature setting, a 0.1uF capacitor should be placed next to HOTSET pin of each phase IC. In the local setting, a resistor divider per phase is needed, and the setting voltage is connected to HOTSET pin of each phase. The 0.1uF decoupling capacitor is not necessary. Use VBIAS as the reference voltage. If RHOTSET1 is pre-selected, RHOTSET2 can be calculated as follows. RHOTSET 2 = RHOTSET 1 ∗ VHOTSET VBIAS − VHOTSET (24) Phase Delay Timing Resistors RPHASE1 and RPHASE2 The phase delay of the interleaved multiphase converter is programmed by the resistor divider connected at RMPIN+ or RMPIN- depending on which slope of the oscillator ramp is used for the phase delay programming of phase IC, as shown in Figure 3. If the upslope is used, RMPIN+ pin of the phase IC should be connected to RMPOUT pin of the control IC and RMPIN- pin should be connected to the resistor divider. When RMPOUT voltage is above the trip voltage at RMPIN- pin, the PWM latch is set. GATEL becomes low, and GATEH becomes high after the non-overlap time. If down slope is used, RMPIN- pin of the phase IC should be connected to RMPOUT pin of the control IC and RMPIN+ pin should be connected to the resistor divider. When RMPOUT voltage is below the trip voltage at RMPIN- pin, the PWM latch is set. GATEL becomes low, and GATEH becomes high after the non-overlap time. Use VBIAS voltage as the reference for the resistor divider since the oscillator ramp magnitude from control IC tracks VBIAS voltage. Try to avoid both edges of the oscillator ramp for better noise immunity. Determine the ratio of the programming resistors corresponding to the desired switching frequencies and phase numbers. If the resistor RPHASEx1 is pre-selected, the resistor RPHASEx2 is determined as: R PHASEx 2 = Page 23 of 41 RAPHASEx ∗ R PHASEx1 1 − RAPHASEx (25) 9/30/04 IR3080 Combined Over Temperature and Phase Delay Setting Resistors RPHASE1, RPHASE2 and RPHASE3 The over temperature setting resistor divider can be combined with the phase delay resistor divider to save one resistor per phase. Calculate the HOTSET threshold voltage VHOTSET corresponding to the allowed maximum temperature from Equation (23). If the over temperature setting voltage is lower than the phase delay setting voltage, VBIAS*RAPHASEx, connect RMPIN+ or RMPIN- pin between RPHASEx1 and RPHASEx2, and connect HOTSET pin between RPHASEx2 and RPHASEx3. Pre-select RPHASEx1, RPHASEx 2 = ( RAPHASEx ∗ VBIAS − VHOTSET ) * RPHASEx1 VBIAS ∗ (1 − RAPHASEx ) (26) RPHASEx3 = VHOTSET ∗ RPHASEx1 VBIAS * (1 − RAPHASEx ) (27) If the over temperature setting voltage is higher than the phase delay setting voltage, VBIAS*RAPHASEx, connect HOTSET pin between RPHASEx1 and RPHASEx2 and connect RMPIN+ or RMPIN- between RPHASEx2 and RPHASEx3 respectively. Pre-select RPHASEx1, R PHASEx 2 = (V HOTSET − RAPHASEx ∗ V BIAS ) ∗ R PHASEx1 V BIAS − V HOTSET (28) RPHASEx3 = RAPHASEx ∗ VBIAS * RPHASEx1 VBIAS − VHOTSET (29) Bootstrap Capacitor CBST Depending on the duty cycle and gate drive current of the phase IC, a 0.1uF to 1uF capacitor is needed for the bootstrap circuit. Decoupling Capacitors for Phase IC 0.1uF-1uF decoupling capacitors are required at VCC and VCCL pins of phase ICs. VOLTAGE LOOP COMPENSATION The adaptive voltage positioning (AVP) is usually adopted in the computer applications to improve the transient response and reduce the power loss at heavy load. Like current mode control, the adaptive voltage positioning loop introduces extra zero to the voltage loop and splits the double poles of the power stage, which make the voltage loop compensation much easier. Resistors RFB and RDRP are chosen according to Equations (16) and (17), and the selection of compensation types depends on the output capacitors used in the converter. For the applications using Electrolytic, Polymer or ALPolymer capacitors and running at lower frequency, type II compensation shown in Figure 17(a) is usually enough. While for the applications using only ceramic capacitors and running at higher frequency, type III compensation shown in Figure 17(b) is preferred. For applications where AVP is not required, the compensation is the same as for the regular voltage mode control. For converter using Polymer, AL-Polymer, and ceramic capacitors, which have much higher ESR zero frequency, type III compensation is required as shown in Figure 17(b) with RDRP and CDRP removed. Type II Compensation for AVP Applications Determine the compensation at no load, the worst case condition. Choose the crossover frequency fc between 1/10 and 1/5 of the switching frequency per phase. Assume the time constant of the resistor and capacitor across the output inductors matches that of the inductor, and determine RCP and CCP from Equations (30) and (31), where LE and CE are the equivalent inductance of output inductors and the equivalent capacitance of output capacitors respectively. Page 24 of 41 9/30/04 IR3080 CCP1 CCP1 RCP VO+ RFB CCP - VDAC RDRP RFB1 CFB FB CCP - EAOUT EAOUT FB EAOUT EAOUT VDRP RCP RFB VO+ VDRP RDRP VDAC + CDRP + (a) Type II compensation (b) Type III compensation Figure 17. Voltage loop compensation network (2π ∗ fC ) 2 ∗ LE ∗ CE ∗ RFB ∗ VPWMRMP RCP = VO * 1 + (2π * fC * C * RC ) 2 C CP = 10 ∗ L E ∗ C E (30) (31) RCP CCP1 is optional and may be needed in some applications to reduce the jitter caused by the high frequency noise. A ceramic capacitor between 10pF and 220pF is usually enough. Type III Compensation for AVP Applications Determine the compensation at no load, the worst case condition. Assume the time constant of the resistor and capacitor across the output inductors matches that of the inductor, the crossover frequency and phase margin of the voltage loop can be estimated by Equations (32) and (33), where RLE is the equivalent resistance of inductor DCR. f C1 = RDRP 2π * CE ∗ GCS * RFB ∗ RLE θ C1 = 90 − A tan(0.5) ∗ (32) 180 (33) π Choose the desired crossover frequency fc around fc1 estimated by Equation (32) or choose fc between 1/10 and 1/5 of the switching frequency per phase, and select the components to ensure the slope of close loop gain is -20dB /Dec around the crossover frequency. Choose resistor RFB1 according to Equation (34), and determine CFB and RDRP from Equations (35) and (36). 1 R FB 2 R FB1 = CFB = to R FB1 = 1 4π ∗ fC ∗ RFB1 C DRP = ( R FB + R FB1 ) ∗ C FB R DRP 2 R FB 3 (34) (35) (36) RCP and CCP have limited effect on the crossover frequency, and are used only to fine tune the crossover frequency and transient load response. Determine RCP and CCP from Equations (37) and (38). Page 25 of 41 9/30/04 IR3080 RCP = (2π ∗ fC ) 2 ∗ LE ∗ CE ∗ RFB ∗ VPWMRMP VO 10 ∗ L E ∗ C E C CP = (37) (38) RCP CCP1 is optional and may be needed in some applications to reduce the jitter caused by the high frequency noise. A ceramic capacitor between 10pF and 220pF is usually enough. Type III Compensation for Non-AVP Applications Resistor RFB is chosen according to Equations (16), and resistor RDRP and capacitor CDRP are not needed. Choose the crossover frequency fc between 1/10 and 1/5 of the switching frequency per phase and select the desired phase margin θc. Calculate K factor from Equation (39), and determine the component values based on Equations (40) to (44), π θ K = tan[ ∗ ( C + 1.5)] 4 180 RCP = RFB ∗ (39) ( 2π ∗ LE ∗ CE ∗ fC ) 2 ∗ VPWMRMP VO ∗ K (40) CCP = K 2π ∗ fC ∗ RCP (41) CCP1 = 1 2π ∗ fC ∗ K ∗ RCP (42) CFB = K 2π ∗ fC ∗ RFB (43) R FB1 = 1 2π ∗ f C ∗ K ∗ C FB (44) CURRENT SHARE LOOP COMPENSATION The crossover frequency of the current share loop should be at least one decade lower than that of the voltage loop in order to eliminate the interaction between the two loops. A capacitor from SCOMP to ground is usually enough for the share loop compensation. Choose the crossover frequency of current share loop (fCI) based on the crossover frequency of voltage loop (fC), and determine the CSCOMP, CSCOMP = 0.65 * RPWMRMP *VI * I O * GCS _ ROOM * RLE * [1 + 2π * fCI * CE * (VO I O )] * FMI VO ∗ 2π ∗ fCI *1.05 *106 (45) Where FMI is the PWM gain in the current share loop, FMI = Page 26 of 41 RPWMRMP * CPWMRMP * f SW *V PWMRMP (VI − VPWMRMP − VDAC ) * (VI − VDAC ) (46) 9/30/04 IR3080 DESIGN EXAMPLE 1 - VRD 10 CONVERTER SPECIFICATIONS Input Voltage: VI=12 V DAC Voltage: VDAC=1.35 V No Load Output Voltage Offset: VO_NLOFST=20 mV Output Current: IO=105 ADC Maximum Output Current: IOMAX=120 ADC Output Impedance: RO=0.91 mΩ VCC Ready to VCC Power Good Delay: tVccPG=0-10mS VID Delay Time: tVID=2.5mS Soft Start Time: tSS=2mS Over Current Delay: tOCDEL=0.5mS Dynamic VID Down-Slope Slew Rate: SRDOWN=2.5mV/uS Over Temperature Threshold: TPCB=115 ºC POWER STAGE Phase Number: n=6 Switching Frequency: fSW=400 kHz Output Inductors: L=220 nH, RL=0.47 mΩ Output Capacitors: AL-Polymer, C=560uF, RC= 7mΩ, Number Cn=10 IR3080 EXTERNAL COMPONENTS Oscillator Resistor Rosc Once the switching frequency is chosen, ROSC can be determined from the curve in Figure 13. For switching frequency of 400kHz per phase, choose ROSC=30.1kΩ VID Delay Capacitor CVIDDEL Given VID delay time tVID =2.5mS, the capacitor can is, CVIDDEL = 66 *10−6 * tVID 66 *10−6 ∗ 2.5 *10−3 = = 42nF , choose 47nF 3.91 3.91 Soft Start Capacitor CSS/DEL and Resistor RSS/DEL Because faster over-current protection is required, the soft start capacitor CSS/DEL in series with the resistor RSS/DEL is used. Calculate the soft start capacitor from the required soft start time. C SS / DEL = I CHG ∗ t SS 70 * 10 −6 ∗ 2 * 10 −3 = = 0.1uF VO 1.35 − 20 * 10 −3 Calculate the soft start resistor from the required over current delay time tOCDEL, 0.09 − RSS / DEL = tOCDEL ∗ I DISCHG 0.5 *10−3 ∗ 6 *10−6 0.09 − CSS / DEL 0.1 *10− 6 = 10kΩ = I DISCHG 6 *10− 6 The soft start delay time is tSSDEL = CSS / DEL ∗ (1.3 − RSS / DEL ∗ I CHG ) 0.1 *10−6 ∗ (1.3 − 10 *103 * 70 *10−6 ) = = 0.86mS I CHG 70 *10− 6 Page 27 of 41 9/30/04 IR3080 The power good delay time is tVccPG = CSS / DEL * (3.91 − VO − 1.3) 0.1*10−6 * (3.91 − 1.33 − 1.3) = = 1.8ms I CHG 70 *10− 6 VDAC Slew Rate Programming Capacitor CVDAC and Resistor RVDAC From Figure 15, the sink current of VDAC pin corresponding to 400kHz (ROSC=30.1kΩ) is 76uA. Calculate the VDAC down-slope slew-rate programming capacitor from the required down-slope slew rate. CVDAC = I SINK 76 * 10 −6 = = 30.4nF , Choose CVDAC=33nF SR DOWN 2.5 * 10 −3 / 10 −6 Calculate the programming resistor. RVDAC = 0.5 + 3.2 * 10 −15 CVDAC 2 = 0.5 + 3.2 * 10 −15 (33 * 10 −9 ) 2 = 3.5Ω From Figure 15, the source current of VDAC pin is 110uA. The VDAC up-slope slew rate is SRUP = I SOURCE 110 * 10 −6 = = 3.3mV / uS CVDAC 33 * 10 −9 Over Current Setting Resistor ROCSET The room temperature is 25ºC and the target PCB temperature is 100 ºC. The phase IC die temperature is about 1 ºC higher than that of phase IC, and the inductor temperature is close to PCB temperature. Calculate Inductor DC resistance at 100 ºC, RL _ MAX = RL _ ROOM ∗ [1 + 3850*10−6 ∗ (TL _ MAX − TROOM )] = 0.47 *10−3 ∗ [1 + 3850*10−6 ∗ (100 − 25)] = 0.61mΩ The current sense amplifier gain is 34 at 25ºC, and its gain at 101ºC is calculated as, G CS _ MIN = G CS _ ROOM ∗ [1 − 1470 *10 −6 ∗ (T IC _ MAX − T ROOM )] = 34 ∗ [1 − 1470 *10 −6 ∗ (101 − 25)] = 30.2 Set the over current limit at 135A. From Figure 14, the bias current of OCSET pin (IOCSET) is 41uA with ROSC=30.1kΩ. The total current sense amplifier input offset voltage is 0.55mV, which includes the offset created by the current sense amplifier input resistor mismatch. Calculate constant KP, the ratio of inductor peak current over average current in each phase, KP = (V I − VO ) ∗ VO /( L ∗ V I ∗ f SW ∗ 2) (12 − 1.33) ∗ 1.33 /( 220 *10 −9 ∗ 12 ∗ 400 * 10 3 ∗ 2) = = 0.3 I LIMIT / n 135 / 6 ROCSET = [ =( I LIMIT ∗ RL _ MAX ∗ (1 + K P ) + VCS _ TOFST ] ∗ GCS _ MIN / I OCSET n 135 ∗ 0.61 *10 −3 ∗ 1.3 + 0.55 *10 −3 ) ∗ 30.2 /( 41 *10 − 6 ) = 13.3kΩ 6 No Load Output Voltage Setting Resistor RFB and Adaptive Voltage Positioning Resistor RDRP From Figure 14, the bias current of FB pin is 41uA with ROSC=30.1kΩ. RFB = RL _ MAX ∗ VO _ NLOFST − VCS _ TOFST ∗ n ∗ RO Page 28 of 41 I FB ∗ RL _ MAX = 0 .61 * 10 −3 ∗ 20 * 10 −3 − 0 .55 * 10 −3 ∗ 6 ∗ 0 .91 * 10 −3 = 365 Ω 41 * 10 − 6 ∗ 0 .61 * 10 − 3 9/30/04 IR3080 RDRP = RFB ∗ RL _ MAX ∗ GCS _ MIN n ∗ RO = 365 ∗ 0.61 *10−3 ∗ 30.2 = 1.21kΩ 6 ∗ 0.91 *10−3 Control IC Over Temperature Setting Resistors RHOTSETC1 and RHOTSETC2 Set the temperature threshold at 115 ºC, which corresponds to the IC die temperature of 116 ºC. Calculate the HOTSET threshold voltage corresponding to the temperature thresholds. V HOTSET = 4.73 * 10 −3 * TJ + 1.241 = 4.73 * 10 −3 ∗ 116 + 1.241 = 1.79V , Choose RHOTSETC1=20.0kΩ, R HOTSETC 2 = R HOTSETC1 ∗ V HOTSET 20 *10 3 ∗1.79 = 7.15kΩ = 6.8 − 1.79 V BIAS − V HOTSET Body Braking Related Resistors RBBFB and RBBDRP N/A. The body braking during Dynamic VID is disabled. IR3086 EXTERNAL COMPONENTS PWM Ramp Resistor RPWMRMP and Capacitor CPWMRMP Set PWM ramp magnitude VPWMRMP=0.8V. Choose 220pF for PWM ramp capacitor CPWMRMP, and calculate the resistor RPWMRMP, VO RPWMRMP = VIN ∗ f SW ∗ CPWMRMP ∗ [ln(VIN − VDAC ) − ln(VIN − VDAC − VPWMRMP )] = 1.33 12 ∗ 400 *10 3 ∗ 220 * 10 −12 ∗ [ln(12 − 1.35) − ln(12 − 1.35 − 0.8)] = 16.1kΩ , Choose RPWMRMP=16.2kΩ Inductor Current Sensing Capacitor CCS+ and Resistors RCS+ and RCSChoose CCS+=47nF and calculate RCS+, RCS + = L RL 220 *10−9 /(0.47 *10−3 ) = = 10.0kΩ CCS + 47 *10−9 The bias currents of CSIN+ and CSIN- are 0.25uA and 0.4uA respectively. Calculate resistor RCS-, RCS − = 0.25 0.25 ∗ RCS + = ∗ 10.0 *103 = 6.2kΩ , Choose RCS-=6.19kΩ 0.4 0.4 Over Temperature Setting Resistors RHOTSET1 and RHOTSET2 Use central over temperature setting and set the temperature threshold at 115 ºC, which corresponds the IC die temperature of 116 ºC. Calculate the HOTSET threshold voltage corresponding to the temperature thresholds. V HOTSET = 4.73 * 10 −3 * TJ + 1.241 = 4.73 * 10 −3 ∗ 116 + 1.241 = 1.79V Pre-slect RHOTSET1=10.0kΩ, R HOTSET 2 = R HOTSET 1 ∗ V HOTSET 10 *10 3 ∗1.79 = 3.57 kΩ = 6.8 − 1.79 V BIAS − V HOTSET Phase Delay Timing Resistors RPHASE1 and RPHASE2 Use central over temperature setting and set the temperature threshold at 115 ºC, which corresponds the IC die temperature of 116 ºC. Calculate the HOTSET threshold voltage corresponding to the temperature thresholds. Page 29 of 41 9/30/04 IR3080 The phase delay resistor ratios for phases 1 to 6 at 400kHz of switching frequencies are RAPHASE1=0.628, RAPHASE2=0.415, RAPHASE3=0.202, RAPHASE4=0.246, RAPHASE5=0.441 and RAPHASE6=0.637 starting from downslope. Pre-select RPHASE11=RPHASE21=RPHASE31=RPHASE41=RPHASE51= RPHASE61=10kΩ, RPHASE12 = RAPHASE1 0.628 ∗ RPHASE11 = ∗ 10 *103 = 16.9kΩ 1 − RAPHASE1 1 − 0.628 RPHASE22=7.15kΩ, RPHASE32=2.55kΩ, RPHASE42=3.24kΩ, PPHASE52=7.87kΩ, RPHASE62=17.4kΩ Bootstrap Capacitor CBST Choose CBST=0.1uF Decoupling Capacitors for Phase IC and Power Stage Choose CVCC=0.1uF, CVCCL=0.1uF VOLTAGE LOOP COMPENSATION Type II compensation is used for the converter with AL-Polymer output capacitors. Choose the crossover frequency fc=40kHz, which is 1/10 of the switching frequency per phase, and determine Rcp and CCP. RCP = CCP = (2π ∗ fC )2 ∗ LE ∗ CE ∗ RFB ∗ VRAMP VO * 1 + (2π * fC * C * RC )2 10 ∗ LE ∗ CE RCP = = (2π ∗ 40 ∗103 )2 ∗ (220 ∗10−9 / 6) ∗ (560 ∗10−6 ∗10) ∗ 365 ∗ 0.8 (1.35 − 20 ∗10−3 ) * 1 + (2π * 40 *103 * 560 *10−6 * 7 *10−3 )2 10 ∗ (220 ∗ 10−9 / 6) ∗ (560 ∗ 10−6 *10) 2.0 ∗103 = 2.0kΩ = 71nF , Choose CCP=68nF Choose CCP1=47pF to reduce high frequency noise. CURRENT SHARE LOOP COMPENSATION The crossover frequency of the current share loop fCI should be at least one decade lower than that of the voltage loop fC. Choose the crossover frequency of current share loop fCI=4kHz, and calculate CSCOMP, FMI = RPWMRMP * CPWMRMP * f SW *V PWMRMP 16.2 *103 * 220 *10−12 * 400 *103 * 0.8 = = 0.011 (VI − VPWMRMP − VDAC ) * (VI − VDAC ) (12 − 0.8 − 1.35) * (12 − 1.35) CSCOMP = = 0.65 * RPWMRMP *VI * I O * GCS _ ROOM * RLE * [1 + 2π * fCI * CE * (VO I O )] * FMI VO ∗ 2π ∗ fCI *1.05 *106 0.65 *16.2 *10 3 *12 *105 * 34 * (0.47 *10 −3 6) * [1 + 2π * 4 *10 3 * 560 *10 −6 *10 * (1.33 − 105 * 9.1*10 −4 ) 105] * 0.011 (1.33 − 105 * 9.1*10 − 4 ) ∗ 2π ∗ 4 *10 3 *1.05 *10 6 = 31.4nF Choose CSCOMP=33nF. Page 30 of 41 9/30/04 IR3080 DESIGN EXAMPLE 2 - EVRD 10 HIGH FREQUENCY ALL-CERAMIC CONVERTER SPECIFICATIONS Input Voltage: VI=12 V DAC Voltage: VDAC=1.3 V No Load Output Voltage Offset: VO_NLOFST=20 mV Output Current: IO=105 ADC Maximum Output Current: IOMAX=120 ADC Output Impedance: RO=0.91 mΩ VCC Ready to VCC Power Good Delay: tVccPG=0-10mS VID Delay Time: tVID=2.5mS Soft Start Time: tSS=2.9mS Over Current Delay: tOCDEL=2.1mS Dynamic VID Down-Slope Slew Rate: SRDOWN=2.5mV/uS Over Temperature Threshold: TPCB=115 ºC POWER STAGE Phase Number: n=6 Switching Frequency: fSW=800 kHz Output Inductors: L=100 nH, RL=0.5 mΩ Output Capacitors: Ceramic, C=22uF, RC= 2mΩ, Number Cn=62 IR3080 EXTERNAL COMPONENTS Oscillator Resistor Rosc Once the switching frequency is chosen, ROSC can be determined from the curve in Figure 13. For switching frequency of 800kHz per phase, choose ROSC=13.3kΩ VID Delay Capacitor CVIDDEL Given VID delay time tVID =2.5mS, the capacitor can is, CVIDDEL = 66 *10−6 * tVID 66 *10−6 ∗ 2.5 *10−3 = = 42nF , choose 47nF 3.91 3.91 Soft Start Capacitor CSS/DEL and Resistor RSS/DEL Because faster over-current protection is required, the soft start capacitor CSS/DEL in series with the resistor RSS/DEL is used. Calculate the soft start capacitor from the required soft start time. CSS / DEL = I CHG ∗ tSS 70 *10−6 ∗ 2.9 *10−3 = = 0.16uF , choose CSS/DEL=0.15uF VO 1.3 − 20 *10−3 Calculate the soft start resistor from the required over current delay time tOCDEL, 0.09 − RSS / DEL = tOCDEL ∗ I DISCHG 2.1*10−3 ∗ 6 *10−6 0.09 − CSS / DEL 0.15 *10−6 = = 1kΩ I DISCHG 6 *10−6 The soft start delay time is C ∗ (1.3 − R SS / DEL ∗ I CHG ) 0.15 * 10 −6 ∗ (1.3 − 1 * 10 3 * 70 * 10 −6 ) t SSDEL = SS / DEL = = 2.6mS I CHG 70 * 10 −6 Page 31 of 41 9/30/04 IR3080 The power good delay time is C SS / DEL ∗ (3.91 − VO − 1.3) 0.15 * 10 −6 * (3.91 − 1.28 − 1.3) = = 2.85ms I CHG 70 * 10 −6 tVccPG = VDAC Slew Rate Programming Capacitor CVDAC and Resistor RVDAC From Figure 15, the sink current of VDAC pin corresponding to 800kHz (ROSC=13.3kΩ) is 170uA. Calculate the VDAC down-slope slew-rate programming capacitor from the required down-slope slew rate. CVDAC = I SINK 170 * 10 −6 = = 68nF SR DOWN 2.5 * 10 −3 / 10 −6 Calculate the programming resistor. RVDAC = 0.5 + 3.2 * 10 −15 CVDAC 2 = 0.5 + 3.2 * 10 −15 (68 * 10 −9 ) 2 = 1.2Ω From Figure 15, the source current of VDAC pin is 250uA. The VDAC up-slope slew rate is SRUP = I SOURCE 250 *10 −6 = = 3.7 mV / uS CVDAC 68 *10 −9 Over Current Setting Resistor ROCSET The room temperature is 25ºC and the target PCB temperature is 100 ºC. The phase IC die temperature is about 1 ºC higher than that of phase IC, and the inductor temperature is close to PCB temperature. Calculate Inductor DC resistance at 100 ºC, RL _ MAX = RL _ ROOM ∗ [1 + 3850*10−6 ∗ (TL _ MAX − TROOM )] = 0.5 *10−3 ∗ [1 + 3850*10−6 ∗ (100 − 25)] = 0.64mΩ The current sense amplifier gain is 34 at 25ºC, and its gain at 101ºC is calculated as, G CS _ MIN = G CS _ ROOM ∗ [1 − 1470 *10 −6 ∗ (T IC _ MAX − T ROOM )] = 34 ∗ [1 − 1470 *10 −6 ∗ (101 − 25)] = 30.2 Set the over current limit at 135A. From Figure 14, the bias current of OCSET pin (IOCSET) is 90uA with ROSC=13.3kΩ. The total current sense amplifier input offset voltage is 0.55mV, which includes the offset created by the current sense amplifier input resistor mismatch. Calculate constant KP, the ratio of inductor peak current over average current in each phase, KP = (VI − VO ) ∗ VO /( L ∗ VI ∗ f SW ∗ 2) (12 − 1.28) ∗ 1.28 /(100 *10−9 ∗ 12 ∗ 800 *103 ∗ 2) = = 0.32 I LIMIT / n 135 / 6 ROCSET = [ =( RLIMIT ∗ RL _ MAX ∗ (1 + K P ) + VCS _ TOFST ] ∗ GCS _ MIN / I OCSET n 135 ∗ 0.64 *10 − 3 ∗ 1.32 + 0.55 *10 −3 ) * 30.2 /(90 *10 − 6 ) = 6.34 kΩ 6 No Load Output Voltage Setting Resistor RFB and Adaptive Voltage Positioning Resistor RDRP From Figure 14, the bias current of FB pin is 90uA with ROSC=13.3kΩ. RFB = RL _ MAX ∗ VO _ NLOFST − VCS _ TOFST ∗ n ∗ RO Page 32 of 41 I FB ∗ RL _ MAX = 0.64 *10−3 ∗ 20 *10−3 − 0.55 *10−3 ∗ 6 ∗ 0.91 *10−3 = 162Ω 90 *10−6 * 0.64 *10−3 9/30/04 IR3080 RDRP = RFB ∗ RL _ MAX ∗ GCS _ MIN n ∗ RO = 162 ∗ 0.64 *10−3 * 30.2 = 576Ω 6 ∗ 0.91 *10−3 Control IC Over Temperature Setting Resistors RHOTSETC1 and RHOTSETC2 Set the temperature threshold at 115 ºC, which corresponds the IC die temperature of 116 ºC. Calculate the HOTSET threshold voltage corresponding to the temperature thresholds. V HOTSET = 4.73 * 10 −3 * TJ + 1.241 = 4.73 * 10 −3 ∗ 116 + 1.241 = 1.79V , choose RHOTSETC1=10.0kΩ, R HOTSETC 2 = R HOTSETC1 ∗ V HOTSET 10 *10 3 ∗1.79 = = 3.57 kΩ V BIAS − V HOTSET 6.8 − 1.79 Body Braking Related Resistors RBBFB and RBBDRP N/A. The body braking during Dynamic VID is disabled. IR3086 EXTERNAL COMPONENTS PWM Ramp Resistor RPWMRMP and Capacitor CPWMRMP Set PWM ramp magnitude VPWMRMP=0.75V. Choose 100pF for PWM ramp capacitor CPWMRMP, and calculate the resistor RPWMRMP, VO RPWMRMP = VIN * f SW * CPWMRMP * [ln(VIN − VDAC ) − ln(VIN − VDAC − VPWMRMP )] = 1.28 = 18.2kΩ 3 − 12 12 ∗ 800 *10 ∗100 *10 ∗ [ln(12 − 1.3) − ln(12 − 1.3 − 0.75)] Inductor Current Sensing Capacitor CCS+ and Resistors RCS+ and RCSChoose 47nF for capacitor CCS+ and calculate RCS+, RCS + = L RL 100 *10−9 /(0.5 *10−3 ) = = 4.22kΩ CCS + 47 *10−9 The bias currents of CSIN+ and CSIN- are 0.25uA and 0.4uA respectively. Calculate resistor RCS-, RCS − = 0.25 0.25 ∗ RCS + = ∗ 4.22 *10 3 = 2.61kΩ 0.4 0.4 Combined Over Temperature and Phase Delay Setting Resistors RPHASEx1, RPHASEx2 and RPHASEx3 The over temperature setting resistor divider is combined with the phase delay resistor divider. Set the temperature threshold at 115 ºC, which corresponds the IC die temperature of 116 ºC, and calculate the HOTSET threshold voltage corresponding to the temperature thresholds. V HOTSET = 4.73 * 10 −3 ∗ TJ + 1.241 = 4.73 * 10 −3 ∗ 116 + 1.241 = 1.79V The phase delay resistor ratios for phases 1 to 6 at 800kHz of switching frequencies are RAPHASE1=0.665, RAPHASE2=0.432, RAPHASE3=0.198, RAPHASE4=0.206, RAPHASE5=0.401 and RAPHASE6=0.597 starting from downslope. Page 33 of 41 9/30/04 IR3080 The over temperature setting voltage of phases 1, 2, 5, and 6 is lower than the phase delay setting voltage, VBIAS*RAPHASEx. Pre-select RPHASE11=10kΩ, RPHASEx 2 = ( RAPHASEx ∗ VBIAS − VHOTSET ) * RPHASEx1 (0.665 ∗ 6.8 − 1.79) ∗10 *103 = 12.1kΩ = 6.8 ∗ (1 − 0.665) VBIAS ∗ (1 − RAPHASEx ) RPHASEx3 = VHOTSET ∗ RPHASEx1 1.79 ∗ 12.1 *103 = = 7.87 kΩ VBIAS * (1 − RAPHASEx ) 6.8 * (1 − 0.665) RPHASE21=10kΩ, RPHASE22=2.94kΩ, RPHASE23=4.64kΩ RPHASE51=10kΩ, RPHASE52=2.32kΩ, RPHASE53=4.42kΩ RPHASE61=10kΩ, RPHASE62=8.25kΩ, RPHASE63=6.49kΩ The over temperature setting voltage of Phases 3 and 4 is higher than the phase delay setting voltage, VBIAS*RAPHASEx. Pre-select RPHASEX1=10kΩ, R PHASE 32 = (V HOTSET − RAPHASE 3 ∗ V BIAS ) ∗ R PHASE 31 (1.79 − 0.198 ∗ 6.8) ∗10 *10 3 = 887Ω = V BIAS − V HOTSET 6.8 − 1.79 RPHASE 33 = RAPHASE 3 ∗ VBIAS * RPHASE 31 0.198 ∗ 6.8 ∗ 10 *103 = 2.67 kΩ = 6.8 − 1.79 VBIAS − VHOTSET RPHASE41=10kΩ, RPHASE42=768Ω, RPHASE43=2.80kΩ Bootstrap Capacitor CBST Choose CBST=0.1uF Decoupling Capacitors for Phase IC and Power Stage Choose CVCC=0.1uF, CVCCL=0.1uF VOLTAGE LOOP COMPENSATION Type III compensation is used for the converter with only ceramic output capacitors. The crossover frequency and phase margin of the voltage loop can be estimated as follows. f C1 = R DRP 576 = = 146 kHz −6 2π ∗ C E ∗ G CS ∗ R FB ∗ R LE 2π ∗ (62 ∗ 22 *10 ) ∗ 34 ∗ 162 ∗ (0.5 *10 − 3 / 6) θC1 = 90 − A tan(0.5) ∗ Choose RFB1 = 180 π = 63° 2 2 ∗ RFB = ∗ 162 = 110Ω 3 3 Choose the desired crossover frequency fc (=140kHz) around fc1 estimated above, and calculate CFB = 1 4π ∗ fC ∗ RFB1 CDRP = = 1 = 5.2nF , Choose CFB=5.6nF 4π ∗ 140 *103 ∗ 110 ( RFB + RFB1 ) ∗ CFB (162 + 110) ∗ 5.6 *10−9 = = 2.7 nF RDRP 576 Page 34 of 41 9/30/04 IR3080 RCP = CCP = ( 2π ∗ fC ) 2 ∗ LE ∗ CE ∗ RFB ∗ VRAMP (2π ∗140 *103 ) 2 ∗ (100 *10−9 / 6) ∗ (22 *10−6 ∗ 62) ∗162 * 0.75 = = 1.65kΩ VO 1.3 − 20 *10−3 10 ∗ LE ∗ CE RCP = 10 ∗ (100 *10−9 / 6) ∗ ( 22 *10−6 * 62) 1.65 ∗ 103 = 27 nF Choose CCP1=47pF to reduce high frequency noise. CURRENT SHARE LOOP COMPENSATION The crossover frequency of the current share loop fCI should be at least one decade lower than that of the voltage loop fC. Choose the crossover frequency of current share loop fCI=3.5kHz, and calculate CSCOMP, FMI = RPWMRMP * CPWMRMP * f SW *V PWMRMP 18.2 *103 *100 *10−12 * 800 *103 * 0.75 = = 0.011 (VI − VPWMRMP − VDAC ) * (VI − VDAC ) (12 − 0.75 − 1.3) * (12 − 1.3) CSCOMP = = 0.65 * RPWMRMP *VI * I O * GCS _ ROOM * RLE * [1 + 2π * fCI * CE * (VO I O )] * FMI VO ∗ 2π ∗ fCI *1.05 *106 0.65 *18.2 *10 3 *12 *105 * 34 * (0.5 *10 −3 6) * [1 + 2π * 3500 * 22 *10 −6 * 62 * (1.33 − 105 * 9.1*10 −4 ) 105] * 0.011 (1.33 − 105 * 9.1*10 − 4 ) ∗ 2π ∗ 3500 *1.05 *10 6 = 20.6nF Choose CSCOMP=22nF Page 35 of 41 9/30/04 IR3080 LAYOUT GUIDELINES The following layout guidelines are recommended to reduce the parasitic inductance and resistance of the PCB layout, therefore minimizing the noise coupled to the IC. • • • • • • Dedicate at least one middle layer for a ground plane LGND. Connect the ground tab under the control IC to LGND plane through a via. Place the following critical components on the same layer as control IC and position them as close as possible to the respective pins, ROSC, ROCSET, RVDAC, CVDAC, CVCC, CSS/DEL and RCC/DEL. Avoid using any via for the connection. Place the compensation components on the same layer as control IC and position them as close as possible to EAOUT, FB and VDRP pins. Avoid using any via for the connection. Use Kelvin connections for the remote voltage sense signals, VOSNS+ and VOSNS-, and avoid crossing over the fast transition nodes, i.e. switching nodes, gate drive signals and bootstrap nodes. Control bus signals, VDAC, RMPOUT, IIN, VBIAS, and especially EAOUT, should not cross over the fast transition nodes. Page 36 of 41 9/30/04 IR3080 PCB Metal and Component Placement • Lead land width should be equal to nominal part lead width. The minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting. • Lead land length should be equal to maximum part lead length + 0.2 mm outboard extension + 0.05mm inboard extension. The outboard extension ensures a large and inspectable toe fillet, and the inboard extension will accommodate any part misalignment and ensure a fillet. • Center pad land length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be ≥ 0.17mm for 2 oz. Copper (≥ 0.1mm for 1 oz. Copper and ≥ 0.23mm for 3 oz. Copper) • A single 0.30mm diameter via shall be placed in the center of the pad land and connected to ground to minimize the noise effect on the IC. Page 37 of 41 9/30/04 IR3080 Solder Resist • The solder resist should be pulled away from the metal lead lands by a minimum of 0.06mm. The solder resist mis-alignment is a maximum of 0.05mm and it is recommended that the lead lands are all Non Solder Mask Defined (NSMD). Therefore pulling the S/R 0.06mm will always ensure NSMD pads. • The minimum solder resist width is 0.13mm, therefore it is recommended that the solder resist is completely removed from between the lead lands forming a single opening for each “group” of lead lands. • At the inside corner of the solder resist where the lead land groups meet, it is recommended to provide a fillet so a solder resist width of ≥ 0.17mm remains. • The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto the copper of 0.06mm to accommodate solder resist mis-alignment. In 0.5mm pitch cases it is allowable to have the solder resist opening for the land pad to be smaller than the part pad. • Ensure that the solder resist in-between the lead lands and the pad land is ≥ 0.15mm due to the high aspect ratio of the solder resist strip separating the lead lands from the pad land. • The single via in the land pad should be tented with solder resist 0.4mm diameter, or 0.1mm larger than the diameter of the via. Page 38 of 41 9/30/04 IR3080 Stencil Design • The stencil apertures for the lead lands should be approximately 80% of the area of the lead lands. Reducing the amount of solder deposited will minimize the occurrence of lead shorts. Since for 0.5mm pitch devices the leads are only 0.25mm wide, the stencil apertures should not be made narrower; openings in stencils < 0.25mm wide are difficult to maintain repeatable solder release. • The stencil lead land apertures should therefore be shortened in length by 80% and centered on the lead land. • The land pad aperture should be striped with 0.25mm wide openings and spaces to deposit approximately 50% area of solder on the center pad. If too much solder is deposited on the center pad the part will float and the lead lands will be open. • The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back to decrease the incidence of shorting the center land to the lead lands when the part is pushed into the solder paste. Page 39 of 41 9/30/04 IR3080 TYPICAL PERFORMANCE CHARACTERISTICS Oscillator Frequency (kHz) Figure 13 - Oscillator Frequency versus ROSC 1000 950 900 850 800 750 700 650 600 550 500 450 400 350 300 250 200 150 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100 ROSC (K Ohms) Figure 14 - IFB, BBFB, & OCSET Bias Currents vs ROSC 125 115 105 95 85 uA 75 65 55 45 35 25 15 5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100 ROSC (K Ohm) Figure 15 - VDAC Source & Sink Currents vc ROSC (includes OCSET Bias Current) 325 300 275 250 225 uA 200 175 ISINK 150 ISOURCE 125 100 75 50 25 0 10 20 30 40 50 60 70 80 90 100 110 120 130 140 ROSC (K ohm) Figure 16 - Bias Current Accuracy versus ROsC (includes temperature and input voltage variation) 18% +/-3 Sigma Variation (%) 16% 14% FB, BBFB, OCSET Bias Current 12% 10% VDAC Sink Current 8% VDAC Source Current 6% 4% 2% 0% 10 20 30 40 50 60 70 80 90 100 110 120 130 140 ROSC (K Ohm) Page 40 of 41 9/30/04 IR3080 PACKAGE INFORMATION 32L MLPQ (5 x 5 mm Body) – θJA = 30oC/W, θJC = 3oC/W Data and specifications subject to change without notice. This product has been designed and qualified for the Consumer market. Qualification Standards can be found on IR’s Web site. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information.www.irf.com www.irf.com Page 41 of 41 9/30/04