IR3531 - International Rectifier

IR3531
4+1 Phase Dual Output Control IC
FEATURES
DESCRIPTION
 Integrated 6.8V/0.8A Buck Regulator provides
bias to Control and Driver IC(s)
 Adjustable switching frequency from 250 KHz up
to 1.5MHz per phase based on the synchronization
SCLK input
 Sink and source tracking capability
 Margining via SVID for both rails
 Pre-bias compatible
 Soft Stop capability
 0.5% overall system set point accuracy
 Voltage Mode Modulation for excellent transient
performance
 Single NTC thermistor for current reporting, OC
Threshold, and Load Line thermal compensation
 Complete protection including over-current,
over-voltage, over-temperature, open remote
sense and open control loop
 Thermally enhanced 48L 7mm x 7mm MLPQ
package
The IR3531 control IC provides all the necessary control,
communication and protection to support compact dual
output power solutions up to 210W. The IR3531 can be
combined with either discrete IR3535 driver ICs and Direct
FetsTM or our IR35XX family of footprint compatible and
scalable PowIRstagesTM which integrate the MOSFETs and
driver into the same package.
The IR3531 provides overall system control and current
sharing while the Driver IC or power stages senses perphase current locally and communicates it to the Control
IC. The IR3531 has tri-state PWM outputs to allow diode
emulation during light load events.
The IR3531 provides a high performance transient solution
through classic voltage mode control and our non-linear
transient solutions, TurboTM and Body BrakingTM. TurboTM
automatically turns on all phases to minimize load turn-on
transients while Body BrakingTM automatically turns off the
low-side MOSFET to help dissipate stored inductor energy
at load turn-off.
 RoHS compliant
IIN_R1
VDAC1
VDRP1
EA1
FB1
VO1
VOSEN1+
VOSEN1-
TRACK1
IIN4
IIN3
PIN DIAGRAM
PWM_R1
BASIC APPLICATION CIRCUIT
48
47
46
45
44
43
42
41
40
39
38
37
EN
1
36
VRHOT#
2
35
PWM4
VRRDY1
3
34
PWM3
VRRDY
4
VCC
5
SW
6
V12V
7
ALERT#
8
VCLK
BBR1#
33
TSENS
32
ROSC/OVP
31
ADDR
30
ICCP
29
SCLK
9
28
PWM2
VDIO
10
27
PWM1
PHSSHED
11
26
BBR#
IMON_R1
12
25
TRACK
IR3531
48 Pin 7 x 7 MLPQ
Top View
Figure 1: IR3531 Basic Application Circuit,
showing a 4+1 Configuration
1
March 22, 2012 | FINAL | V2.27
13
14
15
16
17
18
19
20
21
22
23
24
IMON
VDAC
VN
VDRP
EA
PSC
FB
VO
VOSEN+
VOSEN-
IIN1
IIN2
49 GND
Figure 2: IR3531 Package Top View
IR3531
4+1 Phase Dual Output Control IC
ORDERING INFORMATION
IR3531 ― M     
Package
PBF – Lead Free
TR – Tape and Reel
Tape & Reel Qty
Part Number
48 Lead MLPQ
(7x7 mm body)
100
IR3531-MPBF
48 Lead MLPQ
(7x7 mm body)
3000
IR3531-MTRPBF
1
1
PWM_R1
IIN_R1
VDAC1
VDRP1
EA1
FB1
VO1
VOSEN1+
VOSEN1-
TRACK1
IIN4
IIN3
Note : Samples only.
48
47
46
45
44
43
42
41
40
39
38
37
EN
1
36
BBR1#
VRHOT#
2
35
PWM4
VRRDY1
3
34
PWM3
VRRDY
4
33
TSENS
VCC
5
32
ROSC/OVP
SW
6
31
ADDR
V12V
7
30
ICCP
ALERT#
8
29
SCLK
VCLK
9
28
PWM2
VDIO
10
27
PWM1
PHSSHED
11
26
BBR#
25
TRACK
IR3531
48 Pin 7 x 7 MLPQ
Top View
49 GND
15
16
17
18
19
20
21
22
23
24
VDRP
EA
PSC
FB
VO
VOSEN+
VOSEN-
IIN1
IIN2
14
VN
13
VDAC
12
IMON
IMON_R1
Figure 3: Package Top View, Enlarged
2
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
FUNCTIONAL BLOCK DIAGRAM
Figure 4: IR3531 Block Diagram
3
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
TYPICAL APPLICATION DIAGRAM
Figure 5: IR3531 Typical Application Diagram
4
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
IR3531
PIN DESCRIPTIONS
PIN #
PIN NAME
1
EN
2
VRHOT#
Open collector output of the VRHOT# comparator which drives low if Rail0 temperature exceeds
the programmed threshold. Connect external pull-up to bias.
3
VDRRY1
Open collector output that drives low during startup and under any external fault condition for
Rail1 regulator. Connect external pull-up to bias.
4
VDRRY
Open collector output that drives low during startup and under any external fault condition for
Rail0 regulator. Connect external pull-up to bias.
5
VCC
Bias buck regulator output, feedback pin, and bias input for internal circuitry.
6
SW
Switching node for bias buck regulator.
7
V12V
8
ALERT#
9
VCLK
SVID Clock Input. Clock is a high impedance input pin. It is driven by the open collector output of a
microprocessor and requires a pull-up resistor.
10
VDIO
SVID Data Input/Output. High impedance input when address, command or data bits are shifted in,
open drain output when acknowledging or sending data back to the microprocessor. Pin requires a
pull up resistor.
11
PHSSHED
Analog signal that represents the number of phases to be disabled. 0% to 25% VCC, no phases
disabled. 25% to 50% VCC, disable 1 phase. 50% to 75% VCC, disable 2 phases. 75% to 100% VCC,
disable 3 phases (if available).
12
IMON_R1
Voltage at this pin is proportional to Rail1 load current. It is also the input to the ADC for output
current register.
13
IMON
Voltage at this pin is proportional to Rail0 load current. It is also the input to the ADC for output
current register.
14
VDAC
Voltage Regulator Rail 0 reference voltage programmed by SVID. VDAC is also used as the A/D
reference during power up for pins ADDR/PSN, TSENS and ICCP.
15
VN
16
VDRP
17
EA
Output of the error amplifier for Rail0.
18
PSC
Node for Power Savings mode compensation input.
19
FB
Inverting input to the Error Amplifier for Rail0.
20
VO
Remote sense amplifier output for Rail0.
21
VOSEN+
Rail0 remote sense amplifier input. Connect to output at the load.
22
VOSEN-
Rail0 remote sense amplifier input. Connect to ground at the load.
23, 24, 37, 38
IIN1-4
Current signals from the driver IC-s of Rail0.
25
TRACK
External tracking reference for Rail0.
26
BBR#
Body-brakingTM bus for Rail0 driver ICs to disable synchronous switches.
27, 28,
34, 35
PWM1-4
29
SCLK
5
PIN DESCRIPTION
Enable input. Grounding this pin shuts down the voltage regulators. Do not float this pin as the
logic state will be undefined.
Power Supply input supply rail.
Output pin for SVID Alert# interrupt. Open collector output that drives low to notify the master.
Node for DCR thermal compensation network.
Buffered, scaled and thermally compensated current signal for Rail0. Connect an external resistor
to FB to program converter output impedance.
PWM outputs for Rail0. Each output is connected to the input of the driver IC. Connecting the
PWMx output to LGND disables the phase, allowing the IR3531 to operate as a 1, 2, 3, or 4 phase
controller.
Synchronization clock input. Program ROSC using ROSC vs. Frequency to match the SCLK frequency.
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
PIN #
PIN NAME
30
ICCP
Program maximum load current for both Rail0 and Rail1.
31
ADDR
Programs SVID address for Rail0 and Rail1, discrete or coupled inductor operation for Rail0,
enable/disable turbo function for Rail0.
32
ROSC/OVP
33
TSENS
Pin for thermal network that senses the temperature of Rail0 and Rail1.
36
BBR1#
Body-brakingTM bus for Rail1 driver ICs to disable synchronous switches.
39
TRACK1
External tracking reference for Rail1.
40
VOSEN1-
Rail1 remote sense amplifier input. Connect to ground at the load.
41
VOSEN1+
Rail1 remote sense amplifier input. Connect to output at the load.
42
VO1
Remote sense amplifier output for Rail1.
43
FB1
Inverting input to the Error Amplifier for Rail1.
44
EA1
Output of the error amplifier for Rail1.
45
VDRP1
Buffered, scaled and thermally compensated current signal for Rail1. Connect an external resistor
to FB1 to program converter output impedance.
46
VDAC1
Buffered Rail1 reference voltage. Voltage can be margined via SVID.
47
IIN_R1
Current signal from Rail1 driver IC.
48
PWM_R1
49
GND
6
PIN DESCRIPTION
Connect a resistor to LGND to program oscillator frequency. Oscillator frequency equals switching
frequency per phase. ROSC/OVP pin is pulled up to VCC when an over voltage event occurs.
PWM output for Rail1.
Local Ground for internal circuitry and IC substrate connection.
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
ABSOLUTE MAXIMUM RATINGS
Storage Temperature Range
-65°C To 150°C
Operating Junction Temperature
0°C To 150°C
ESD Rating
HBM Class 1C JEDEC Standard
MSL Rating
2
Reflow Temperature
260°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the
specifications are not implied.
PIN Number
PIN NAME
VMAX
VMIN
ISOURCE
ISINK
1
EN
3.5V
-0.3V
25mA
1mA
2
VRHOT#
VCC
-0.3V
1mA
50mA
3
VDRRY1
VCC
-0.3V
1mA
20mA
4
VDRRY
VCC
-0.3V
1mA
20mA
5
VCC
8V
-0.3V
1mA
20mA
6
SW
16V
-1.0V
3A
1mA
7
V12V
16V
-0.5V
1mA
1.5A
8
ALERT#
3.5V
-0.3V
1mA
50mA
9
VCLK
3.5V
-0.3V
1mA
1mA
10
VDIO
3.5V
-0.3V
1mA
50mA
11
PHSSHED
VCC
-0.3V
1mA
1mA
12
IMON_R1
3.5V
-0.3V
25mA
1mA
13
IMON
3.5V
-0.3V
25mA
1mA
14
VDAC
3.5V
-0.3V
5mA
35mA
15
VN
VCC
-0.3V
1mA
1mA
16
VDRP
VCC
-0.3V
35mA
1mA
17
EA
VCC
-0.3V
35mA
5mA
18
PSC
VCC
-0.3V
1mA
1mA
19
FB
VCC
-0.3V
1mA
1mA
20
VO
VCC
-0.3V
35mA
5mA
21
VOSEN+
VCC
-0.5V
5mA
1mA
22
VOSEN-
1.0V
-0.5V
5mA
1mA
23
IIN1
VCC
-0.3V
1mA
1mA
24
IIN2
VCC
-0.3V
1mA
1mA
25
TRACK
VCC
-0.3V
1mA
1mA
26
BBR#
VCC
-0.3V
1mA
5mA
27
PWM1
VCC
-0.3V
1mA
5mA
7
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
PIN Number
PIN NAME
VMAX
VMIN
ISOURCE
ISINK
28
PWM2
VCC
-0.3V
1mA
5mA
29
SCLK
3.5V
-0.3V
1mA
5mA
30
ICCP
3.5V
-0.3V
1mA
1mA
31
ADDR
3.5V
-0.3V
1mA
1mA
32
ROSC
VCC
-0.3V
1mA
1mA
33
TSEN
3.5V
-0.3V
1mA
1mA
34
PWM3
VCC
-0.3V
1mA
5mA
35
PWM4
VCC
-0.3V
1mA
5mA
36
BBR1#
VCC
-0.3V
1mA
5mA
37
IIN3
VCC
-0.3V
1mA
1mA
38
IIN4
VCC
-0.3V
1mA
1mA
39
TRACK1
VCC
-0.3V
1mA
1mA
40
VOSEN1-
1.0V
-0.5V
5mA
1mA
41
VOSEN1+
VCC
-0.5V
5mA
1mA
42
VO1
VCC
-0.5V
35mA
5mA
43
FB1
VCC
-0.3V
1mA
1mA
44
EA1
VCC
-0.3V
35mA
5mA
45
VDRP1
VCC
-0.3V
35mA
1mA
46
VDAC1
3.5V
-0.3V
1mA
35mA
47
IIN_R1
VCC
-0.3V
1mA
1mA
48
PWM_R1
VCC
-0.3V
1mA
1mA
49
GND
N/A
N/A
20mA
1mA
8
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
ELECTRICAL SPECIFICATIONS
RECOMMENDED OPERATING CONDITIONS FOR RELIABLE OPERATION WITH MARGIN
The electrical characteristics table lists the spread of values guaranteed within the recommended operating conditions.
Typical values represent the median values, which are related to 25°C. Unless otherwise specified, these specifications
apply over: -0.3V ≤ VOSEN- ≤ 0.3V, 7.75KΩ ≤ ROSC ≤ 50.0 KΩ
Recommended V12V Range
10.8V
12
13.2V
V
Recommended VCC Range
6.6
6.8
7.0
V
VOSEN- and VOSEN1- to LGND offset
-0.3
0
0.3
V
ROSC Resistor Programming Range
7.75
50
KΩ
TJ
100
ºC
MIN
TYP
MAX
UNIT
-0.5
-
0.5
%
0.8 ≤ VID < 1V
-5
-
+5
mV
0.25V ≤ VID < 0.8V
-8
-
+8
mV
Recommended Operating Junction Temperature
0
ELECTRICAL CHARACTERISTICS
PARAMETER
SYMBOL
CONDITIONS
VDAC Reference
System Set-Point Accuracy
SETACC
VID ≥ 1V
Slew Rate – Fast Mode
VIDFAST
15
20
25
mV/µs
Slew Rate – Slow Mode
VIDSLOW
3.75
5
6.25
mV/µs
Default VBOOT Rail 0
VBOOT0
Note 3
-
1.5
-
V
Default VBOOT Rail 1
VBOOT1
Note 3
-
1.5
-
V
VROSC
ROSC = 24.5 KΩ
0.570
0.595
0.620
V
DACOFF
V(VDAC, VDAC1) ― VID code +
VID offset, 0.25V ≤
V(VDAC, VDAC1) ≤ 1.52V,
< 1mA load
-15
0
15
mV
0.25V ≤ V(VDAC1) ≤ 1.52V
0.3
0.44
0.6
0.25V ≤ V(VDAC) ≤ 1.52V
0.9
1.65
2.4
0.5V ≤ V(VDAC1) ≤ 1.52V
2
13
20
0.5
1.5
2
3
15
30
0.5
1.5
3
Unity Gain Bandwidth
-
3.5
-
MHz
Slew Rate
-
1.5
-
V/µs
-14
0
14
mV
Oscillator (Note 4)
ROSC Voltage
VDAC Buffer Amplifier
Input Outset Voltage
Source Current
DACSRC
Sink Current
DACSNK
V(VDAC1) = 0.25V
0.5V ≤ V(VDAC) ≤ 1.52V
V(VDAC) = 0.25V
mA
mA
Thermal Compensation Amplifier (VDRP)
Output Offset Voltage
9
VDRPOUTOFF
March 22, 2012 | FINAL | V2.27
0V ≤ V(IIN) – V(VDAC) ≤ 1.52V,
0.25V ≤ V(VDAC) ≤ 1.52V,
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
SYMBOL
CONDITIONS
Req/R2 = 2
MIN
TYP
MAX
UNIT
mA
Source Current
VDRPSRC
0.25V ≤ V(VDAC) ≤ 1.52V
3
8
15
Sink Current
VDRPSNK
0.5V ≤ V(VDRP) ≤ 1.52V
0.2
0.4
0.7
0.175
0.25
0.4
2
4.5
7
MHz
-
5.5
-
V/µs
-2
0
2
µA
VID = 250 mV
250
350
385
mV
VID = 1.52 V
2
2.15
2.26
V
VID = 250 mV, SF = 500 kHz
60
151
200
VID = 1.52 V, SF = 500 kHz
220
409
480
PS2COTMIN1
VID = 250 mV, SF = 500 kHz
50
100
200
PS2COTMAX1
VID = 1.52 V, SF = 500 kHz
220
358
480
PS1DELAY
PS0 to PS1 only
-
8
-
PWM
Cycle
V(VDRP) = 0.25V
Req/R2 = 2, Note 1
Unity Gain Bandwidth
Slew Rate
V(VN) = 2 V
VN Bias Current
mA
Power Savings Mode Operation
PS2/PS3 Turn-on Threshold
PS2/PS3 Pulse Width Rail0
PS2/PS3 Pulse Width Rail1
PS Mode Enter Delay
PS2THRSH
PS2COT0
ns
ns
Enable Input
Rising Threshold
ENRISE
625
650
675
mV
Falling Threshold
ENFALL
575
600
625
mV
Hysteresis
ENHYST
Bias Current
ENBIAS
Blanking Time
25
50
75
mV
0V ≤ V(ENABLE) ≤ 3.3V
-5
0
5
µA
Noise Pulse < 100ns will not
register an ENABLE state change.
Note 1
75
250
400
ns
15
50
90
mV
-
1
-
MHz
-
1
-
µs
1.00
-2
1.09
0
1.145
2
V
%
-75
0
75
mV
3
8
15
mA
0.2
0.4
0.6
0.175
0.25
0.375
-
9
-
V/V
IMONx Current Report Amplifier
Output Offset Voltage
IMONOFF
Unity Gain Bandwidth
VDRP–VDAC = 0, 225, 450,
900mV
Note 1
Input Filter Time Constant
Max Output Voltage
IMONMAX
Current Report A/D Accuracy
IMONACC
VDRP–VDAC = 900mV
VDRP1OFF
0V≤ V(IIN_R1) - V(VDAC1) ≤ 0.2V
0.25V ≤ V(IIN_R1) - V(VDAC1) ≤
1.52V
Rail1 VDRP Amplifier
Output Outset Voltage
Source Current
VDRP1SRC
VDRP1SNK
Sink Current
0.25V ≤ V(VDAC1) ≤ 1.52V
0.5V≤ V(VDRP1) ≤ 1.52V
mA
Closed Loop Gain
V(VDRP1) = 0.25V
Note 1
Unity Gain Bandwidth
Note 1
0.8
1.5
3
MHz
Slew Rate
Note 1
-
5.5
-
V/µs
Error Amplifier
10
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
SYMBOL
Input Offset Voltage
CONDITIONS
Note 2 (test mode only)
FB Bias Current
MIN
TYP
MAX
UNIT
-
0
-
mV
-1
0
1
µA
DC Gain
Note 1
100
110
120
dB
Unity Gain Bandwidth
Note 1
20
30
40
MHz
Note 1
Slew Rate
7
12
20
V/µs
Sink Current
EASRC
0.40
0.85
1.35
mA
Source Current
EASNK
5
8
12
mA
Maximum Voltage
EAMAX
500
925
1100
mV
Minimum Voltage
EAMIN
-
120
250
mV
Open Voltage Loop Detection
Threshold
EAOPENTHR
100
300
1100
mV
Open Loop Detection Delay
EAOPENDEL
-
8
-
PWM
PS2 Clamp Voltage
EAPS2CLMP
-240
-70
-10
mV
Measure V(VCC) – V(EA), V(EA1)
Measure V(VCCx) - V(EA), V(EA1),
Relative to Error Amplifier
maximum voltage
V(EA), V(EA1) = V(VCC) to
VRRDY = low
With respect to VDAC
Phase Firing Comparators
Input Offset
KEEPOFF
-30
0
30
mV
Propagation Delay
KEEPDEL
-
-
320
ns
µA
Phase Shedding Comparators
Bias Current
PHSDBIAS
Threshold
PHSDTHRS
-2
0
2
Comparator 1
1.3
1.7
2.0
Comparator 2
3.0
3.4
3.85
Comparator 3
4.8
5.1
5.55
42
52.5
57
mV/
%DC
55
70
ns
V
PWM Comparator
PWM Ramp Slope
PWMSLP
V12V= 12V
Minimum Pulse Width
PWMMIN
Note 1
Input Offset Voltage
PWMOFF
Note 1
-5
0
5
mV
SAAOFF
Note 1
-3
0
3
mV
SAAGAIN
CSIN+ = CSIN- = DACIN, Note 1
4
5.0
6
V/V
Note 1
4
8.5
17
kHz
100
180
22 0
mV
-220
-160
-100
mV
1.615
1.65
1.67
V
100
130
150
mV
-
90
180
ns
Share Adjust Amplifier
Input Offset Voltage
Gain
Unity Gain Bandwidth
Maximum PWM Ramp
Floor Voltage
Minimum PWM Ramp
Floor Voltage
MINFLOOR
MAXFLOOR
IOUT = DACIN – 200mV
Measure relative to floor voltage
IOUT = DACIN + 200mV
Measure relative to floor voltage
Over Voltage Protection (OVP) Comparators
Threshold at Power-up
OVPPUP
Threshold during Normal
Operation
OVPTHR
Propagation Delay to OVP
OVPPROP
11
March 22, 2012 | FINAL | V2.27
Compare to VID Voltage +
VID offset
Measure time from V(FB), V(FB1)
> VID voltage + VID offset (250mV
overdrive) to V(PWM) transition
to > 0.5 * V(VCC)
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
Turbo Circuit
Activation Threshold Voltage
SYMBOL
CONDITIONS
TURBACT
Note 1
Compare to EA, Note 1
Turbo Comparator Hysteresis
TURBHYST
Note 1
Note 1
Filter Time Constant
TURBTIME
Note 1
Note 1
Turbo Pulse Width
TURBPW
500kHz 600mV Peak sine wave
on EAIN, measure GATEH pulse
width
Peak Detect Reset
Time Constant
TURBRESET
Over-Current Comparator
Input Filter Time Constant
MIN
TYP
MAX
UNIT
-
390
-
mV
-
90
-
mV
-
8
-
µs
115
230
280
ns
-
400
-
ns
-
2
-
µs
Over-Current Threshold
OCTHRSH
VDRP-VDAC, VDRP1-VDAC1
0.94
1.08
1.18
V
OC Threshold PSI Reduction
Factor
OCPSI
PSI mode, 4ph to 2ph, 2ph to 1ph
450
540
610
PSI mode, 3ph to 1ph
310
360
410
3ph to 2ph
640
720
800
PSI mode, 4ph to 1ph
220
270
310
4ph to 3ph
690
800
900
Delay to OC shutdown
225
256
285
µs
-
4096
-
µs
VCCSTART
5.5
5.85
6.4
V
VCC UVL Stop
VCCSTOP
4.85
5.2
5.65
V
VCC UVL Hysteresis
VCCHYST
515
650
830
mV
OC Delay Time
OCDELAY
OC Hiccup Time
Relaxation Delay
VCC Undervoltage
VCC UVL Start
VRRDY Output
Output Voltage
mV
VRRDYLO
I(VRRDY, VDRRY1) = 4mA
-
150
300
mV
Leakage Current
VRRDYLEAK
V(VRRDY, VDRRY1) = 5.5V
-
0
10
µA
VCC Activation Voltage
VRRDYVCC
1
2
3.6
V
VO-VDAC Undervoltage
Threshold
VOUVRISE
I(VDRRY, VDRRY1) = 4mA,
<300mV
Reference to VDAC
-340
-290
-230
mV
100
150
200
mV
25
60
80
mV
82
90
92
%
0.36
0.40
0.44
V
200
500
700
µA
6.5
6.8
7.1
V
Open Sense Line Detection
Sense Line Detection Active
Comparator Threshold Voltage
OPENACT
Sense Line Detection Active
Comparator Offset Voltage
OPENOFF
V(VO) < [V(VOSEN+) –
V(LGND)] / 2
VOSEN+ Open Sense Line
Comparator Threshold
OPENCOMP+
Compare to V(VCC)
VOSEN- Open Sense Line
Comparator Threshold
Sense Line Detection
Source Currents
VCC Buck Regulator
OPENCOMP-
VCC Output Voltage
12
OPENSRC
V(VO) = 100mV
VCC100
100–400 mA load current
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
Switch Node Rise Time
SYMBOL
SWRISE
MIN
TYP
MAX
UNIT
Note 1
-
5
-
ns
Switch Node Fall Time
SWFALL
Note 1
-
15
-
ns
ADDRBIAS
-2
0
2
µA
ICCP Pin Bias Current
ICCPBIAS
-2
0
2
µA
TSENS Pin Bias Current
TSENBIAS
-2
0
2
µA
A/D Comparator Offset
ADOFFSET
-5
0
5
mV
VCCSTART
8.8
9.6
10.2
V
VCC V12V Stop
VCCSTOP
7.8
8.6
9.2
V
VCC V12V Hysteresis
VCCHYST
0.8
1
1.3
V
-
-
14.3
Ω
A/D Program Inputs
ADDR Pin Bias Current
V12V Undervoltage
VCC V12V Start
SerialVID
ALERT#, VDIO Buffer On
Resistance
ALERT#, VDIO Leakage Current
CONDITIONS
ALERTRES
ALERTLEAK
-10
0
10
µA
VCLK Bias Current
VCLKBIAS
-1
0
1
µA
VDIO Bias Current
VDIOBIAS
-1
0
1
µA
Transmit Data Prop Delay
XMITDELAY
VCLK rising to VDIO change
4
6
12
ns
Comparator Threshold
SVIDTHRSH
VCLK, VDIO rising
500
590
650
VCLK, VDIO falling
450
515
650
mV
Comparator Hysteresis
SVIDHYST
50
75
-
mV
Link States Reset Timer
SVIDTIME
200
-
600
ns
Source Resistance
PWMSRCR
50
144
500
Ω
Sink Resistance
PWMSNKR
75
117
290
Ω
Tri-state Source Impedance
PWMTRIZ
2.0
5.4
7.5
KΩ
Tri-state Bias Current
PWMTRIBIAS
V(PWMx) = 1.65V
-5
0
5
µA
Tri-state Active Pull-up
PWMTRIPUP
V(PWMx) while sourcing
100 µA to GND
0.5
1
1.2
V
Disable Comparator Threshold
PWMDISTHR
0.4
0.6
0.9
V
PWM High Voltage
PWMHIGH
-
-
1
V
PWM Low Voltage
PWMLOW
I(PWM) = -1mA,
measure VCC-PWM
I(PWM) = -1mA
-
-
1
V
BBRTHRFALL
Measure relative to floor voltage
-300
-200
-110
mV
BBRTHRRISE
Measure relative to floor voltage
-200
-100
-10
mV
70
105
130
mV
30
65
90
ns
20
40
75
Ω
PWMx Outputs
Body Braking Comparator
Threshold Voltage with EAIN
Decreasing
Threshold Voltage with EAIN
Increasing
Hysteresis
Propagation Delay
BBR1# Source Resistance
13
BBRTHRHYS
BBRDELAY
BBRSRCRES
March 22, 2012 | FINAL | V2.27
VCC = 5V
Measure time from EAIN <
V(DACIN) (200mV overdrive)
to GATEL transition to < 4V.
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
BBR1# Sink Resistance
SYMBOL
BBRSNKRES
CONDITIONS
BBR1# High Voltage
BBRHIGH
BBR1# Low Voltage
BBRLOW
I(BBR1#) = -1mA, measure V(VCC)
– V(BBR1#)
I(BBR1#) = 1mA
Unity Gain Bandwidth
RSABW
Note 1
Input Outset Voltage
RSAOFF
MIN
TYP
MAX
UNIT
10
35
60
Ω
0
0.4
0.8
V
0
0.35
0.8
V
1.5
3.2
4.5
mV
-5
0
5
mV
0.4
1
2
Remote Sense Differential Amplifier
High Voltage
VOHIGH
0.25V≤ V(VOSEN+) - V(VOSEN-)
≤ 1.52V,
0.25V≤ V(VOSEN1+) - V(VOSEN1-)
≤ 1.52V
0.5V≤ V(VOSEN+) - V(VOSEN-)
≤ 1.52V,
0.5V≤ V(VOSEN1+) - V(VOSEN1-)
≤ 1.52V
V(VOSEN+) - V(VOSEN-) = 0.25V,
V(VOSEN1+) - V(VOSEN1-) =
0.25V
0.25V≤ V(VOSEN+) - V(VOSEN-)
≤ 1.52V,
0.25V≤ V(VOSEN1+) - V(VOSEN1-)
≤ 1.52V
0.25V≤ V(VOSEN+) - V(VOSEN-)
≤ 1.52V,
0.25V≤ V(VOSEN1+) - V(VOSEN1-)
≤ 1.52V
0.25 V < V(VOSEN+) < 1.52V,
0.25 V < V(VOSEN1+) < 1.52V
-0.3V ≤ VOSEN- ≤ 0.3V, All VID
Codes,
-0.3V ≤ VOSEN1- ≤ 0.3V, All VID
Codes
V(VCC) – V(VO), V(VCC) – V(VO1)
1.5
2
2.5
V
Low Voltage
VOLOW
V(VCC) = 7V
-
60
100
mV
VRHTOUT
I(VRHOT#) = 30mA
-
150
400
mV
VRHTLEAK
V(VRHOT#) = 5.5V
-
0
10
µA
PTMTHR
Raise ADDR voltage after VIN
power-up
2.2
2.6
3.1
V
20
-
24
µs
Sink Current
RSASINK
Source Current
RSASRC
Slew Rate
RSASLEW
VOSEN+ Bias Current
VOSNS-BIAS
VOSEN- Bias Current
VOSNS+BIAS
VRHOT# Comparator
Output Voltage
VRHOT# Leakage Current
Platform Test Mode
Comparator Threshold
Link States Reset Timer
VR Settled
Comparator Offset
Delay to ALERT#
PTMTIME
mA
0.225
0.5
0.8
3
9
20
mA
2
4
8
V/µs
-
27
50
µA
-
27
70
µA
VRSTLOFF
Compare FB to VDAC reference
-
20
-
mV
VRSTLDELAY
Delay after DAC settled to within
2 VID steps of final value
-
5
-
µs
Current Inputs
IINx to IINx Impedance
IINRES
-
3000
-
Ω
IINx to IINx Leakage Current
IINLEAK
-1
0
1
µA
-1
0
1
µA
TRACK Inputs
Input Leakage
14
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
PARAMETER
TRACK to FB Offset
SYMBOL
Release Error Voltage
CONDITIONS
Error amp in unity gain
TRACK = VDAC+100mV, VDAC-FB
VO Discharge Comparators
Tri-state Enable Threshold
VO when PWM outputs enter
tri-state
MIN
TYP
MAX
UNIT
15
36
65
mV
-1
0
1
mV
200
250
300
mV
0.8
1.2
1.3
V
SCLK Synchronization Input
Rising Threshold
Falling Threshold
Note 1
0.625
0.85
1.025
V
Input Leakage
-5
0
5
µA
Propagation Delay Rising
-
-
60
ns
-
-
10
pF
3
7
12
mA
Input Capacitance
Note 1
General
VCC Supply Current
VCCBIAS
Notes:
1. Guaranteed by design but not tested in production
2. Error Amplifier input offset is trimmed to within ±1% for optimal system set point accuracy.
3. Final test VBOOT options of 0, 0.9, 1.35 and 1.5V are feasible.
Contact International Rectifier Enterprise Power Business Unit for details.
4. Use of internal oscillator is not recommended, use SCLK input to set PWM frequency.
15
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
THEORY OF OPERATION
SYSTEM DESCRIPTION
The IR3531 Multiphase Buck power system provides
voltage regulation solutions for two individual supply
outputs. The main output, Rail0, controls up to four phases
and produce up to 200A when paired with appropriate
power stages. The secondary output, Rail1, is a single
phase output capable of up to 50A, again with appropriate
power stage. The IR3531 control IC is specialized to allow
external clock synchronization and tracking capability for
each rail. Features include a serial control and telemetry
bus that can control output voltage settings and slew
rates while allowing monitoring of the system thermals
and load currents. The IR3531 control IC contains all
necessary housekeeping, protection and control functions
and communicates a three-level PWM signal to each
power stage.
FREQUENCY AND PHASE TIMING CONTROL
The IR3531 operates with external frequency
synchronization which can be used to control input ripple
from multiple paralleled power supply systems. Systems
can be forced to operate out of phase thereby reducing
instantaneous peak input currents and also controlling
system noise signatures. The internal oscillator is used to
calibrate the PWM ramp slopes and other functions at
power up therefore it is desirable for the externally applied
synchronization frequency to be very near the ROSC
programmed internal frequency times the number of
active phases. Calibration can take up to 1ms. This results
in the PWM gain to be near the desired 50mV/% duty
cycle. Furthermore, it is desired the SCLK input be stable
prior to enabling the IR3531 voltage regulator.
The SCLK input frequency provided needs to equal the
desired base switching frequency multiplied by the active
number of phases. Phase shedding is available however
SCLK needs to be adjusted accordingly to match the
number of active phases.
Phase timing and interleave spacing is automatically
optimized inside the controller and can accommodate
changing phases on the fly (phase shedding). The PHSSHD
pin can be used to dynamically drop from 1-3 phases while
minimizing output voltage transients. Also, phases can be
disabled by grounding the PWM outputs of the IR3531.
Notice the driver ICs should be removed since a PWM low
signal indicates a 0% duty cycle state which turns on the
low-side MOSFETs and can potentially develop large
negative inductor currents. The control IC detects which
PWM pins are grounded during power up to determine the
populated number of phases and automatically optimizes
phase timing for minimal system ripple.
TRACK FUNCTIONALITY
Both Rail outputs of the IR3531 can be independently
controlled through their respective TRACK inputs. TRACK
pins override the internal VDAC reference inputs to the
Error Amplifiers allowing users to control power up and
power down VR output voltage characteristics. The IR3531
is fully soft-stop and pre-bias compatible. The control loop
is full synchronous during soft stop events thereby
preventing COUT capacitor discharge-induced inductive
kicks. The control system allows non-synchronous buck
operation once VO <=250mV ― this allows outputs to
return to their pre-biased operating points if available.
Figure 6: TRACK Operation with Pre-Bias
16
March 22, 2012 | FINAL | V2.27
IR3531
IR3531
4+1 Phase Dual Output Control IC
Figure 7: TRACK Operation without Pre-Bias
The TRACK inputs have a typical 36mV offset from the
closed loop feedback operating point to ensure the error
amplifier is in an off state when TRACK=0V. Furthermore,
TRACK must exceed the respective VDAC by at least 100mV
to ensure VDAC has complete control of the Error Amplifier
as shown in Figures 6 and 7.
across the inductor. Body Braking
overshoot of the converter.
TM
reduces the peak
As a cautionary note the track input provides direct control
of the output PWM duty cycle. The presence of excessive
noise or glitches on TRACK when this input is active can
cause sudden increases in the PWM duty cycle (up to
100%), potentially causing damage to the power converter.
An error amplifier output voltage greater than the
common mode input range of the PWM comparator
results in 100% duty cycle regardless of the voltage of the
PWM ramp. The resulting PWM control loop is capable of
transitioning from 0% duty cycle to 100% duty cycle with
overlapping phases within a few tens of nanoseconds in
response to a load step decrease. Figure 8 on the next
page depicts PWM operating waveforms under various
conditions.
PWM CONTROL METHOD
BODY BRAKINGTM
The steady state control architecture utilized in the IR3531
is feed-forward voltage mode control with trailing edge
modulation. A high-gain wide-bandwidth voltage type
error amplifier is used to achieve accurate voltage
regulation and ultra-fast transient response. Feed-forward
control is established by varying the PWM ramp slope
proportionally to the input voltage resulting in the error
amplifier operating point being independent of the input
voltage. The input voltage can change due to variations in
the silver box output voltage or due to the wire and
PCB-trace voltage drop related to changes in load current.
All PWM ramp slopes are calibrated at initial power-up.
The PWM pulse is terminated once the PWM ramp
exceeds the Error Amplifier output voltage.
In a conventional synchronous buck converter, the
minimum time required to reduce the current in the
inductor in response to a load-step decrease is:
Under dynamic load transitions, the IR3531 utilizes our
TM
patented Body Braking algorithm allows all low-side
MOSFETs to be turned off during a load relaxation event
allowing the MOSFET body diodes to conduct and dissipate
some of the stored inductor energy and also speed up the
inductor current slew rate by introducing a larger voltage
17
March 22, 2012 | FINAL | V2.27
TSLEW 
L * ( I MAX  I MIN )
VO
The slew rate of the inductor current can be significantly
increased by turning off the synchronous rectifier in
response to a load-step decrease. The switch node voltage
is then forced to decrease until conduction of the
synchronous rectifier’s body diode occurs. This increases
the voltage across the inductor from Vout to Vout +
VBODYDIODE. The minimum time required to reduce the
current in the inductor in response to a load transient
decrease is now:
TSLEW 
L * ( I MAX  I MIN )
VO  V BODYDIODE
4+1 Phase Dual Output Control IC
Since the voltage drop in the body diode is often
comparable to the output voltage, the inductor current
IR3531
slew rate can be increased significantly. This patented
PHASE CLOCK
PULSE
EAIN
PWMRMP
FLOOR
GATEH
GATEL
STEADY-STATE
OPERATION
DUTY CYCLE INCREASE
DUE TO LOAD INCREASE
DUTY CYCLE DECREASE
DUE TO V12V INCREASE
(FEED-FORWARD)
DUTY CYCLE DECREASE DUE TO LOAD
DECREASE (BODY BRAKING) OT FAULT
(VCC UV, OCP, VID FAULT)
STEADY-STATE
OPERATION
Figure 8: PWM Operating Waveforms
technique is referred to as “body braking” and is
accomplished through the “body braking comparator.”
If the error amplifier’s output voltage drops below VDAC,
this comparator turns off the low-side gate driver, enabling
the bottom FET body diode to take over. There is 100mV
upslope and 200mV down slope hysteresis for the body
braking comparator.
BODY BRAKINGTM
In a conventional synchronous buck converter, the
minimum time required to reduce the current in the
inductor in response to a load-step decrease is:
TSLEW 
L * ( I MAX  I MIN )
VO
The slew rate of the inductor current can be significantly
increased by turning off the synchronous rectifier in
response to a load-step decrease. The switch node voltage
is then forced to decrease until conduction of the
synchronous rectifier’s body diode occurs. This increases
the voltage across the inductor from Vout to Vout +
VBODYDIODE. The minimum time required to reduce the
current in the inductor in response to a load transient
decrease is now:
TSLEW 
L * ( I MAX  I MIN )
VO  V BODYDIODE
Since the voltage drop in the body diode is often
comparable to the output voltage, the inductor current
18
March 22, 2012 | FINAL | V2.27
slew rate can be increased significantly. This patented
technique is referred to as “body braking” and is
accomplished through the “body braking comparator.”
If the error amplifier’s output voltage drops below VDAC,
this comparator turns off the low-side gate driver, enabling
the bottom FET body diode to take over. There is 100mV
upslope and 200mV down slope hysteresis for the body
braking comparator.
LOSSLESS AVERAGE INDUCTOR
CURRENT SENSING
Inductor current can be sensed by connecting a series
resistor and a capacitor network in parallel with the
inductor and measuring the voltage across the capacitor,
as shown in Figure 8. The equation of the sensing network
is:
1
RL  sL
vC (s)  vL ( s)
 iL ( s)
1  sRCSCCS
1  sRCSCCS
Usually the resistor Rcs and capacitor Ccs are chosen, such
that, the time constant of Rcs and Ccs equals the time
constant of the inductor, which is the inductance L over
the inductor DCR RL. If the two time constants match, the
voltage across Ccs is proportional to the current through L,
and the sense circuit can be treated as if only a sense
resistor with the value of RL was used. The mismatch of the
time constants does not affect the measurement of
inductor DC current, but affects the AC component of the
inductor current.
4+1 Phase Dual Output Control IC
IR3531
The input offset of this amplifier is calibrated to within
+/- 450µV (6 sigma limits) with a 200uV typical LSB
calibration bit. This calibration routine is continuous and
occurs at every 56 PWM cycles.
Figure 9: Inductor Current Sensing and Current Sense Amplifier
The advantage of sensing the inductor current versus
high-side or low-side sensing is that actual output current
being delivered to the load is obtained rather than peak
or sampled information about the switch currents.
The output voltage can be positioned to meet a load line
based on real-time information. Except for a sense resistor
in series with the inductor, this is the only sense method
that can support a single cycle transient response.
Other methods provide no information during either load
increase (low-side sensing) or load decrease (high-side
sensing).
An additional problem associated with peak or valley
current mode control for voltage positioning is that they
suffer from peak-to-average errors. These errors will
appear in many ways but one example is the effect of
frequency variation. If the frequency of a particular unit is
10% low, the peak-to-peak inductor current will be 10%
larger and the output impedance of the converter will drop
by about 10%. Variations in inductance, current sense
amplifier bandwidth, PWM prop delay, any added slope
compensation, input voltage, and output voltage are all
additional sources of peak-to-average errors.
CURRENT SENSE AMPLIFIER
A high speed differential current sense amplifier is located
in our driver ICs, as shown in Figure 9. Its gain is nominally
32.5 over the entire temperature operating range
therefore the 3850 ppm/ºC inductor DCR temperature
coefficient should be compensated in the voltage loop
feedback path. This can be accurately compensated by
using a linearized Negative TC resistor network where
the NTC can be located near the output inductors. The
resulting temperature compensated current information
is used by the control IC for voltage positioning and current
reporting, and over current limit protection.
19
March 22, 2012 | FINAL | V2.27
The current sense amplifier can accept positive differential
input up to 50mV and negative up to -10mV before
clipping. The output of the current sense amplifier is
summed with the VDAC voltage and is returned to the
control IC through the IIN pin. The IIN pins in the control IC
are internally tied together through 3 KOhm resistors to
produce a voltage representative of the average phase
inductor current.
AVERAGE CURRENT SHARE LOOP
A current sharing loop is also incorporated in the IR3531
to ensure balance between the multiphase buck power
stages. Poor current sharing can hamper transient
response and degrade overall system efficiency.
The current information of each phase is compared
against the average phase current through a Share Adjust
Amplifier which then manipulates the respective PWM
ramp start voltage to add or subtract PWM output
duty cycle. The current share amplifier is internally
compensated such that the crossover frequency of the
current share loop is much slower than that of the voltage
loop and the two loops do not interact.
INSTANTANEOUS CURRENT BALANCE
A form of coarse current sharing is also incorporated into
the IR3531 to protect against Synchronized High Load
Repetition Rate transients which can saturate inductors
and cause OVP conditions. The phase firing order of the
multiphase system is continually being re-assessed and
adjusted if required on a cycle-by-cycle basis to prevent
instantaneous phase currents from deviating from each
other. This also improves transient response by ensuring
all phase currents track each other within a few switching
cycles. Individual switch nodes will appear to be variable
frequency however input and output ripple are unaffected
by the varying phase firing order.
4+1 Phase Dual Output Control IC
SVID CONTROL
The SVID bus allows the processor to communicate with
the IR3531. The processor can program the voltage
regulator output voltage and monitor telemetry data the
IR3531 offers such as temperature and both rail currents.
VCLK, VDIO and ALERT# communication lines are designed
for external 50-75 ohm pull up resistors to 1.0-1.2V bias
voltage and should not be floated. Note that ALERT# may
assert twice for VID transitions of 2 VID steps or less.
Addressing is programmed as a percentage of VDAC as
shown by selecting the appropriate ADDR pin resistor
divider combination and supports up to 14 addresses and
2 all call addresses (refer to Table 1). Table 2 provides a list
of supported SVID commands. Table 3 provides a list of
supported required SVID registers. The SVID communicates
VID codes listed in Table 4a and 4b to program the VDAC
set point.
The IR3531 can accept changes in the VID code and will
vary the VDAC voltage accordingly. The slew rate of the
voltage at the VDAC pin can be set by the appropriate
command. The slew rate is internally programmed and no
external pins or components are necessary. Digital VID
transitions result in a smooth analog transition of the
VDAC voltage and converter output voltage minimizing
inrush currents, false over current conditions and
overshoot of the output voltage.
The VID data from the SVID bus is stored in registers and
is sent to the Digital-to-Analog Converter (DAC), whose
output is sent to the VDAC buffer amplifier. The output of
the buffer amplifier is the VDAC pin. To achieve optimal
system setpoint voltage accuracy, first all contributing
offsets of the IR3531 are independently trimmed and lastly
the internal VDAC reference is trimmed to take into
account all sum of all the offset components. Note that the
resulting final VDAC voltage will have a slightly wider
tolerance as it is compensating for the sum of all other
offset components. This results in an overall 0.5% system
set-point accuracy for VID range between 1V to 1.52V.
IR3531
TABLE 1: ADDR/PSN A/D VOLTAGE PROGRAMMING
(AS % OF VDAC)
% of
VDAC
Binary
Code
Address
Name
Sync
Turbo
1.5%
00000
A0/A1
Ext. Sync
Enable
4.7%
00001
A0/A1
Ext. Sync
Disable
7.8%
00010
A0/A1
Int. Clock
Enable
11%
00011
A0/A1
Int. Clock
Disable
14%
00100
A2/A3
Ext. Sync
Enable
17.2%
00101
A2/A3
Ext. Sync
Disable
20.3%
00110
A2/A3
Int. Clock
Enable
23.4%
00111
A2/A3
Int. Clock
Disable
26.5%
01000
A4/A5
Ext. Sync
Enable
29.7%
01001
A4/A5
Ext. Sync
Disable
32.8%
01010
A4/A5
Int. Clock
Enable
36%
01011
A4/A5
Int. Clock
Disable
39%
01100
A6/A7
Ext. Sync
Enable
42.2%
01101
A6/A7
Ext. Sync
Disable
45.3%
01110
A6/A7
Int. Clock
Enable
48.4%
01111
A6/A7
Int. Clock
Disable
51.5%
10000
A8/A9
Ext. Sync
Enable
54.7%
10001
A8/A9
Ext. Sync
Disable
57.8%
10010
A8/A9
Int. Clock
Enable
61%
10011
A8/A9
Int. Clock
Disable
64%
10100
A10/A11
Ext. Sync
Enable
67.2%
10101
A10/A11
Ext. Sync
Disable
70.3%
10110
A10/A11
Int. Clock
Enable
73.4%
10111
A10/A11
Int. Clock
Disable
76.6%
11000
A12/A13
Ext. Sync
Enable
79.7%
11001
A12/A13
Ext. Sync
Disable
82.8%
11010
A12/A13
Int. Clock
Enable
86%
11011
A12/A13
Int. Clock
Disable
Note: A14/A15 are reserved all-call address.
SVID COMMAND STRUCTURE
SVID protocol has two main command groups: the Get and
Set commands. The Get commands retrieve data from the
voltage regulator controller, while the Set commands
make changes to voltage regulator operating points and
power states.
20
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
When the processor (master) issues a Get command, it
transmits the intended controller address and the address
of the register it wants to read. The addressed controller
acknowledges the command and returns the requested
data. Similarly, when the processor issues a Set command,
it transmits the intended controller address and the data it
wants to insert. The only exception is the SetRegADR
command which is used to declare the register address
that SetRegDAT will alter. The controller acknowledges
these commands. Parity checking is not enforced on
SetRegADR/SetRegDAT.
TABLE 2: SUPPORTED COMMAND
Command
Description
SetVIDfast
Slews VOUT to a new Programmed setpoint at
20mV/usec
TABLE 3: SUPPORTED REGISTER
Register
Description
VendorID
Identifies the VR vendor
ProductID
Identifies the product model
ProductRev
Identifies the product revision
SVID Protocol ID
Identifies the version of SVID protocol
VR Capability
Communicates functions the IR3531
supports
Status1 Reg
Stores VR status data
Status2 Reg
Stores SVID bus errors
Temp Zone
Temperature zone from Rail0 sensor
Output Current
Stores output current for Rail0/Rail1
Status2_last_read
Stores previous data of status 2
SetVIDslow
Slews VOUT to a new Programmed setpoint at
5mV/usec
ICC Max
Programs the maximum supported
output current
SetPS
Sets power state
Temp Max
SetRegADR
Declares the address of the register to be
written to
Programs maximum operating
temperature
SR-fast
Stores the fast slew rate value
SetRegDAT
Writes data to the SetRegADR declared register
SR-slow
Stores the slow slew rate value
GetReg
Read data of a specified register
Vboot
Overrides the default Vboot value
TestMode
Test mode is used for final test trimming of the
IR3531 and is not available to users.
Vout Max
Programs the maximum supported
operational Vout
VID Setting
Register contains the current VID setting
Power State
Register contains the current power state
Note: SetVID decay is not supported
VID Offset
1
Allows margining around the VID setpoint
Multi VR Config
Configures other VR-s on the same SVID
bus
SetRegADR
Scratch pad register for temporary
storage of the SetRegADR pointer register
Note 1: VID Offset commands that attempt to push the VID
above 1.52V or below 0V are not acknowledged.
21
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
TABLE 4: VID VALUES
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
00
00000000
0
26
00100110
0.435
4C
01001100
0.625
01
00000001
0.250
27
00100111
0.440
4D
01001101
0.630
02
00000010
0.255
28
00101000
0.445
4E
01001110
0.635
03
00000011
0.260
29
00101001
0.450
4F
01001111
0.640
04
00000100
0.265
2A
00101010
0.455
50
01010000
0.645
05
00000101
0.270
2B
00101011
0.460
51
01010001
0.650
06
00000110
0.275
2C
00101100
0.465
52
01010010
0.655
07
00000111
0.280
2D
00101101
0.470
53
01010011
0.660
08
00001000
0.285
2E
00101110
0.475
54
01010100
0.665
09
00001001
0.290
2F
00101111
0.480
55
01010101
0.670
0A
00001010
0.295
30
00110000
0.485
56
01010110
0.675
0B
00001011
0.300
31
00110001
0.490
57
01010111
0.680
0C
00001100
0.305
32
00110010
0.495
58
01011000
0.685
0D
00001101
0.310
33
00110011
0.500
59
01011001
0.690
0E
00001110
0.315
34
00110100
0.505
5A
01011010
0.695
0F
00001111
0.320
35
00110101
0.510
5B
01011011
0.700
10
00010000
0.325
36
00110110
0.515
5C
01011100
0.705
11
00010001
0.330
37
00110111
0.520
5D
01011101
0.710
12
00010010
0.335
38
00111000
0.525
5E
01011110
0.715
13
00010011
0.340
39
00111001
0.530
5F
01011111
0.720
14
00010100
0.345
3A
00111010
0.535
60
01100000
0.725
15
00010101
0.350
3B
00111011
0.540
61
01100001
0.730
16
00010110
0.355
3C
00111100
0.545
62
01100010
0.735
17
00010111
0.360
3D
00111101
0.550
63
01100011
0.740
18
00011000
0.365
3E
00111110
0.555
64
01100100
0.745
19
00011001
0.370
3F
00111111
0.560
65
01100101
0.750
1A
00011010
0.375
40
01000000
0.565
66
01100110
0.755
1B
00011011
0.380
41
01000001
0.570
67
01100111
0.760
1C
00011100
0.385
42
01000010
0.575
68
01101000
0.765
1D
00011101
0.390
43
01000011
0.580
69
01101001
0.770
1E
00011110
0.395
44
01000100
0.585
6A
01101010
0.775
1F
00011111
0.400
45
01000101
0.590
6B
01101011
0.780
20
00100000
0.405
46
01000110
0.595
6C
01101100
0.785
22
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
21
00100001
0.410
47
01000111
0.600
6D
01101101
0.790
22
00100010
0.415
48
01001000
0.605
6E
01101110
0.795
23
00100011
0.420
49
01001001
0.610
6F
01101111
0.800
24
00100100
0.425
4A
01001010
0.615
70
01110000
0.805
25
00100101
0.430
4B
01001011
0.620
71
01110001
0.810
72
01110010
0.815
99
10011001
1.010
C0
11000000
1.205
73
01110011
0.820
9A
10011010
1.015
C1
11000001
1.210
74
01110100
0.825
9B
10011011
1.020
C2
11000010
1.215
75
01110101
0.830
9C
10011100
1.025
C3
11000011
1.220
76
01110110
0.835
9D
10011101
1.030
C4
11000100
1.225
77
01110111
0.840
9E
10011110
1.035
C5
11000101
1.230
78
01111000
0.845
9F
10011111
1.040
C6
11000110
1.235
79
01111001
0.850
A0
10100000
1.045
C7
11000111
1.240
7A
01111010
0.855
A1
10100001
1.050
C8
11001000
1.245
7B
01111011
0.860
A2
10100010
1.055
C9
11001001
1.250
7C
01111100
0.865
A3
10100011
1.060
CA
11001010
1.255
7D
01111101
0.870
A4
10100100
1.065
CB
11001011
1.260
7E
01111110
0.875
A5
10100101
1.070
CC
11001100
1.265
7F
01111111
0.880
A6
10100110
1.075
CD
11001101
1.270
80
10000000
0.885
A7
10100111
1.080
CE
11001110
1.275
81
10000001
0.890
A8
10101000
1.085
CF
11001111
1.280
82
10000010
0.895
A9
10101001
1.090
D0
11010000
1.285
83
10000011
0.900
AA
10101010
1.095
D1
11010001
1.290
84
10000100
0.905
AB
10101011
1.100
D2
11010010
1.295
85
10000101
0.910
AC
10101100
1.105
D3
11010011
1.300
86
10000110
0.915
AD
10101101
1.110
D4
11010100
1.305
87
10000111
0.920
AE
10101110
1.115
D5
11010101
1.310
88
10001000
0.925
AF
10101111
1.120
D6
11010110
1.315
89
10001001
0.930
B0
10110000
1.125
D7
11010111
1.320
8A
10001010
0.935
B1
10110001
1.130
D8
11011000
1.325
8B
10001011
0.940
B2
10110010
1.135
D9
11011001
1.330
8C
10001100
0.945
B3
10110011
1.140
DA
11011010
1.335
8D
10001101
0.950
B4
10110100
1.145
DB
11011011
1.340
8E
10001110
0.955
B5
10110101
1.150
DC
11011100
1.345
23
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
VID7:VID0
(Hex)
VID7:VID0
(Bin)
Voltage
8F
10001111
0.960
B6
10110110
1.155
DD
11011101
1.350
90
10010000
0.965
B7
10110111
1.160
DE
11011110
1.355
91
10010001
0.970
B8
10111000
1.165
DF
11011111
1.360
92
10010010
0.975
B9
10111001
1.170
E0
11100000
1.365
93
10010011
0.980
BA
10111010
1.175
E1
11100001
1.370
94
10010100
0.985
BB
10111011
1.180
E2
11100010
1.375
95
10010101
0.990
BC
10111100
1.185
E3
11100011
1.380
96
10010110
0.995
BD
10111101
1.190
E4
11100100
1.385
97
10010111
1.000
BE
10111110
1.195
E5
11100101
1.390
98
10011000
1.005
BF
10111111
1.200
E6
11100110
1.395
E7
11100111
1.400
F0
11110000
1.445
F9
11111001
1.490
E8
11101000
1.405
F1
11110001
1.450
FA
11111010
1.495
E9
11101001
1.410
F2
11110010
1.455
FB
11111011
1.500
EA
11101010
1.415
F3
11110011
1.460
FC
11111100
1.505
EB
11101011
1.420
F4
11110100
1.465
FD
11111101
1.510
EC
11101100
1.425
F5
11110101
1.470
FE
11111110
1.515
ED
11101101
1.430
F6
11110110
1.475
FF
11111111
1.520
EE
11101110
1.435
F7
11110111
1.480
EF
11101111
1.440
F8
11111000
1.485
24
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
Adaptive Voltage Positioning (AVP) is a control algorithm
where the output voltage is reduced as the load current
increases. This may also be referred to as VR output
impedance, Voltage Droop or Load Line. AVP is
implemented to reduce the amount of bulk capacitance
for a given load transient and regulation window and
reduces power dissipation at heavy load. The IR3531
implementation of voltage positioning for Rail0 and Rail1
is shown in Figure 10. The output voltage is set by the
VDAC or TRACK reference voltage at the positive input of
the error amplifier.
The VDRP pin is connected to the FB pin through the
resistor RDRP. As load current increases, the VDRP voltage
increases proportionally. Since the error amplifier will
force the loop to maintain FB to be equal to the VDAC
reference voltage, the additional RDRP current has to flow
through the RFB resistor which introduces an offset
voltage that is proportional to the load current. The RFB
current is equal to (VDRP-VDAC)/RDRP. The positioning
voltage can be programmed by the resistors RDRP and RFB
so that the droop impedance produces the desired
converter output impedance. The offset and slope of the
converter output impedance are referenced to and
therefore independent of the VDAC voltage.
INDUCTOR DCR TEMPERATURE COMPENSATION
CURRENT MONITOR (IMON)
The load current information for all the phases is fed back
to the control IC through the Driver IC IOUT pins where
this information is averaged and buffered to the Thermal
Compensation Amplifier. The gain of the Thermal
Compensation Amplifier is modified by temperature by
introducing a negative temperature coefficient (NTC)
thermistor (RTHERM1) and linearizing resistor network
(RTCMP1 and 2) connected between the VN and VDRP
pins. The thermistor should be placed close to the power
stage to accurately sense the thermal performance of the
inductor DCR.
The control IC generates a current monitor signal IMON
using the VDRP voltage and the VDAC reference, also
shown in Figure 10. The voltage at this pin reports the
average load current information referenced to LGND.
The slope of the IMON signal with respect to the load
current can be adjusted with the resistors RTCMP2 and
RTCMP3. The IMON signal is clamped at 1.09V in order to
facilitate direct interfacing with the master.
ADAPTIVE VOLTAGE POSITIONING
Figure 10: Adaptive voltage positioning with thermal compensation
25
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
REMOTE VOLTAGE SENSING
PROTECTION
The remote sense differential amplifier in the IR3531 is a
high speed, low input offset unity gain buffer that provides
accurate voltage sensing and fast transient response.
VOSEN+ and VOSEN- are the remote-sensing Kelvin
connections that are tied directly to the load. Internal
resistors to the differential amplifier produce VOSEN+ and
VOSEN- bias currents of up to 50µA maximum and limits
the size series resistors for acceptable regulation of the
output voltage. Open sense lead detection is also included
in this amplifier and is discussed further in the fault
section.
The Fault Table below describes the different faults that
can occur and how the IR3531 reacts to protect the supply
and the load from possible damage. The fault types that
can occur are listed in row 1. Row 2 has the method that
a fault is cleared. The first 3 faults are latched in the UV
fault latch and the VCC power has to be recycled to clear.
An over voltage fault can be cleared by recycling either
VCC or the Enable signal. The rest of the faults (except for
UVLO VOUT and SVID faults) are temporarily latched in the
SS fault latch until the fault condition clears. Most faults
disable the error amplifier (except for SVID and VOUT
UVLO). Most faults (except SVID) flag VRRDY. VRRDY
returns to active high when all faults are cleared. The delay
row shows reaction time after detecting a fault condition.
Delays are provided to minimize the possibility of nuisance
faults. The table applies for both rails of the IC.
PHASE SHEDDING
IR3531 allows phases to be disabled through the PHSSHED
pin. Shedding can be performed either statically at power
up or can be exercised dynamically during normal
operation. One, two or three phases can be disabled to
help enhance light load efficiency. The internal clock
frequency is automatically adjusted to achieve graceful
transition. Phase shedding is not recommended if an
external synchronization clock is being applied.
TABLE 5: PHASE SHEDDING PROGRAMMING THRESHOLDS
Threshold
Action
PHSSHED < 0.25VCC
No Phases Shed
0.25VCC < PHSSHED < 0.5VCC
Shed 1 Phase
0.5VCC < PHSSHED < 0.75VCC
Shed 2 Phases
PHSSHED < 0.75VCC
Shed 3 Phases
POWER STATES AND HIGH EFFICIENCY MODE
AT LOW LOADS
System processors can request the VR to enter higher
efficiency Power Savings modes. The IR3531 enters single
phase operation when a PS1 command is issued from
the processor. This mode is intended for loads less than
20A. There is an 8 switching cycle delay before the VR
transitions from PS0 to PS1. PS2 mode is not supported.
PLATFORM TEST MODE
Platform test mode allows users to test the VR solution
when the default VBOOT voltage programmed on IR3531
is 0V and there is no communication capability to send
commands. The address pin needs to be pulled up to 3.3V
for IR3531 to go into platform test mode. IR3531 will boot
to 1V in this mode.
26
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
TABLE 6: FAULT OPERATION
FAULT TYPE
Open
Control Loop
Fault Clearing
Method
Open
Sense Line
Yes
ROSC/OVP
drives high
until OV clears
No
VRRDY Low?
If fault occurs
on Rail0 will
Rail1 continue
to operate?
If fault occurs
on Rail1 will
Rail0 continue
to operate?
VCC
UVLO
Over
Current
VO
UVLO
Resume Normal Operation when Condition Clears
Yes
Yes
No
No
No
Yes
Transition to 250mV and holds until fault is cleared
Cycles from
VBOOT to
250mV
No
Change
No
No
No
Yes
No
No
No
Hiccup
Yes
No
No
No
Yes
No
No
No
Hiccup
Yes
No
4 SVID
Clock
Cycles to
send NAK
250 ns
Blank
Time
No
256μs OC
duration,
4ms off
duration
No
8 PWM
Cycles
Delay
V12V
UVLO
No
Yes
VDAC
Response?
Enable
Low
SVID
Recycle VCC
or Enable
Recycle VCC
Error Amp
Disabled
Over
Voltage
No
No
ENABLE INPUT
OPEN REMOTE SENSE LINE PROTECTION
The Enable pin has a 0.6V falling threshold that sets the
Fault Latch, a 650mV rising threshold that clears the fault
latch and has a 250ns filter to prevent chatter due to
system noise.
The VOSEN+ and VOSEN- remote sense line impedances
are checked prior to power up to verify they are connected
to low impedances. If high impedance is detected, an Open
Sense Line fault is latched and requires VCC to be recycled
to clear. During normal operation, the remote sense amp
operating environment is monitored to ensure the remote
sense lines are connected. Again, if an abnormal mode is
detected, the sense line impedances are again checked.
If high impedance is detected, an Open Sense Line fault is
latched and requires VCC to be recycled to clear.
OPEN VOLTAGE LOOP DETECTION
If for some reason the control loop fails during operation,
the system protects itself by latching an open loop
fault that requires VCC recycling to clear. Detection is
performed by monitoring the output of the error amplifier.
The fault is latched if EAOUT operates above VCC-1.08V for
8 switching cycles indicating the control loop is broken.
V12V AND VCC UNDER VOLTAGE LOCKOUT
(UVLO)
The IR3531 monitors the converter input voltage rails
(V12V and VCC) and issues a UVLO fault if either voltage is
below the desired operating range. The maximum power
up clear thresholds are 10.2V for V12V and 6.2V for VCC.
27
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
Figure 12 shows two different power-up responses where
Enable going high is gating the first VDAC slew and the
calibration routine is gating the second VDAC slew. The
default slew rate is 5mV/µsec. The control loop ensures
the regulator output voltage will track VDAC. The soft start
sequence finishes when VOUT is settled to the VBOOT set
point and VRRDY is asserted.
START-UP AND SHUT-DOWN SEQUENCE
The IR3531 has a programmable, digitally controlled softstart function to limit the surge current during the voltage
regulator start-up. The default boot voltage for Rail0 rail is
0.9V, for Rail1 it is 1.5V. Figure 11 depicts an Enable gated
power-up and V12V UVLO shutdown followed by a V12V
UVLO gated power up and an Enable low shutdown.
The IR3531 has soft stop capability which allows the
voltage regulator to power down in a controlled fashion
without producing negative undershoots resulting from
fast discharge of output capacitance. Pre-biased outputs
are also supported as shown in Figure 13.
The IR3531 requires less than 1ms to perform calibration
routines once V12V (VIN) UVLO is cleared. Note VDAC is
forced to 1.52V during calibration and A/D sampling and
settles to 250mV once calibration is complete.
V12V
UVLO Threshold
1.52V During Pin Program Sensing
1.52V During Pin Program Sensing
VDAC=0.9V
VDAC=0.9V
250mV
0V
ENABLE
tmax=(1.52-0.25)/5mV=254usec
Allow 1msec after VIN UVLO to
allow A/D pin sensing and
internal calibration routines to occur
before attempting power-up.
Figure 12: V12V Power and Enable Cycling
TRACK
VOUT=PREBIAS
VOUT= VDAC
VOUT=>PREBIAS
VDAC
VDAC
250mV
VRRDY
DIODE EMULATION
NOT ALLOWED
DIODE EMULATION
NOT ALLOWED
ENABLE
Figure 13: Enable Power Cycling Under Pre-bias
28
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
OVER-CURRENT CONTROL DURING SOFT-START
Over current protection is performed internally by
comparing the VDRP pin voltage against an OC offset
voltage that is added to the respective VDAC pin voltage.
This OC offset voltage is adjusted to match the active
number of phases since VDRP represents average perphase current. This ensures that the current limit is
correctly adjusted during phase shedding operation.
The OC offset voltage is set as percentages of 1.025V
above VDAC.
An over current condition is registered if the VDRP pin
voltage, which is proportional to the average current plus
VDAC voltage, exceeds the VDAC+ OC offset voltage.
IR3531
Figure 14 shows the over-current control with delay during
various soft start events. The delay time is fixed at 256μs.
The delay is required since over-current conditions can
occur as part of normal operation due to inrush current.
If an over-current occurs during soft start (before VRRDY
is asserted), the control IC will not react until the over
current delay time has elapsed. If the over-current
condition persists after delay time is reached, the fault
latch will be set pulling the error amplifier’s output low
and inhibiting switching in the driver ICs. The VDAC voltage
will slowly ramp down until it reaches 0.25V and the fault
latch is reset allowing a normal soft start to occur. If an
over-current condition is again encountered during the
soft start cycle, the constant over-current control actions
will repeat and the converter will be in hiccup mode.
ENABLE
INTERNAL
OC DELAY
TRACK
4ms DELAY
4ms DELAY
4ms DELAY
VDAC
EA
VOUT
VRRDY
OCP
THRESHOLD
IOUT
START-UP WITH
OUTPUT SHORTED
HICCUP OVER-CURRENT
PROTECTION
(OUTPUT SHORTED)
NORMAL
START-UP
OCP
DELAY
NORMAL
OPERATION
OUTPUT
SHORTED
OVER-CURRENT
PROTECTION
(OUTPUT
SHORTED)
Figure 14: Over-Current Waveforms during and after start-up
29
March 22, 2012 | FINAL | V2.27
NORMAL
NORMAL
START-UP OPERATION
POWER
DOWN
IR3531
4+1 Phase Dual Output Control IC
ICCP (ICC MAX) PROGRAMMING
TEMPERATURE TELEMETRY
SVID register ICC MAX contains information on the
maximum allowable current supported by the voltage
regulator solution and can be equivalent to the CPU’s
ICC_MAX. The CPU reads this register for platform
compatibility during boot and uses this data in conjunction
with the IOUT register for performance management.
This data is in an 8-bit binary formant equivalent to amps,
i.e. 75A=4Bh.
The maximum temperature TMAX (22h) value is factory
programmed to 110C. This register contains the maximum
temperature the VR supports prior to issuing a thermal
alert or VR_Hot. The master reads this register and uses
this data in conjunction with the Temperature Zones for
performance management. Factory trim options are listed
in Table 8.
TABLE 8: TEMP MAX (PROGRAMMED AT FINAL TEST)
30
Binary Code
00000
00001
00010
00011
00100
00101
00110
00111
01000
01001
01010
01011
01100
01101
01110
01111
10000
10001
10010
10011
10100
10101
10110
10111
11000
11001
11010
11011
11100
11101
11110
11111
Current Level
60A/25A
60A/35A
70A/25A
70A/35A
80A/25A
80A/35A
90A/25A
90A/35A
100A/25A
100A/35A
110A/25A
110A/35A
120A/25A
120A/35A
130A/25A
130A/35A
140A/25A
140A/35A
150A/25A
150A/35A
160A/25A
160A/35A
170A/25A
170A/35A
180A/25A
180A/35A
190A/25A
190A/35A
200A/25A
200A/35A
225A/25A
225A/35A
March 22, 2012 | FINAL | V2.27
Binary Code
Temperature
0000
90 Deg C
1000
106 Deg C
0001
92 Deg C
1001
108 Deg C
0010
94 Deg C
1010
110 Deg C
0011
96 Deg C
1011
112 Deg C
0100
98 Deg C
1100
114 Deg C
0101
100 Deg C
1101
116 Deg C
0110
102 Deg C
1110
118 Deg C
0111
104 Deg C
1111
120 Deg C
THERMAL MONITORING (VRHOT#)
The IR3531 provides two methods of thermal monitoring:
a VRHOT# pin which flags an over temperature event and
temperature telemetry is available through the SVID bus
and the Temperature Zone register.
A thermal sense network which includes an NTC thermistor
provides board temperature information at TSENS pin
as shown in Figure 15. The thermistor is usually placed in
a temperature sensitive region of the converter and is
linearized by a resistor network. VRHOT# will be active
low once the voltage on TSENS crosses Zone 7, or 56.3%
of VDAC. VRHOT# will de-assert once TSENS falls below
Zone 5. The VRHOT# pin is an open-collector output and
should be pulled up to a voltage source through a resistor.
VDAC
RTHERM2
RHOTSET1
Control IC
RHOTSET2
%VDAC
1.5
4.7
7.8
11
14
17.2
20.3
23.4
26.5
29.7
32.8
36
39
42.2
45.3
48.4
51.5
54.7
57.8
61
64
67.2
70.3
73.4
76.6
79.7
82.8
86
89
92.2
95.3
98.4
Temperature
VRHOT#
TSENS
RHOTSET3
VDAC
Zone 5: 53.1%
Zone 7: 56.3%
-
TABLE 7: ICCP (ICC MAX) A/D VOLTAGE PROGRAMMING
(AS % OF VDAC)
Binary Code
+
The voltage is programmed by an external resistor divider
string referenced to VDAC. Table 7 lists the available
current thresholds
Figure 15: Over Temperature Detection Circuit
IR3531
4+1 Phase Dual Output Control IC
The IR3531 compares the TSENS pin voltage against fixed
percentages of VDAC thresholds as indicated in Table 9.
The user can program the external TSENS network to
achieve a desired offset and slope to associate a zone
(stored in register 12h) with a desired temperature.
Zones correspond to the bit number of this 8-bit register,
i.e. Zone 0=bit 0 and Zone 3=bit3 and therefore register
12h behaves like a thermometer. Notice that the zones 1
through 7 thresholds are equally spaced (~1.6% between
thresholds) and the separation between Zone0 and Zone1
is approximately double. Since these zone thresholds are
fixed and equally separated, the respective zone
temperature values will also be equally separated for a
TSENS voltage which has a linear slope vs. temperature.
The SVID status register bit#1 and the ALERT# serve as
thermal warning flags when zones 5 and 6 are crossed as
indicated in Table 9. These warning flags may be used by
the system to reduce the load, increase airflow, and
prevent the system from entering thermal shutdown.
The VRHOT# pin is asserted as zones 6 and 7 are crossed
and can be used as a thermal shutdown flag.
TMAX is merely a reference point to communicate with
downstream system monitors what temperature a zone
equates to. For example, the TMAX register is defaulted in
the IR3531 as 110°C. The micro processor can perform a
GetReg on TMAX and is now able to associate a Zone 4
declaration by the IR3531 to equate to 100.1°C
TABLE 9: TEMPERATURE ZONES
Temperature Zone
TSENS Threshold % VDAC
% of TMAX
Degrees C based on 110°C TMAX
Zone 0
43.8%
75%
82.5C
Zone 1
46.9%
82%
90.2
Zone 2
48.4%
85%
93.5
Zone 3
50%
88%
96.8
Zone 4
51.6%
91%
100.1 Falling,
Status bit 1 de-asserted, ALERT#.
Zone 5
53.1%
94%
103.4 Falling, VRHOT#
de-asserted
Zone 6
54.7%
97%
106.7 Rising,
Status bit 1 asserted, ALERT#.
Zone 7
56.3%
100%
110 Rising, VRHOT# asserted
OVER VOLTAGE PROTECTION (OVP)
The IR3531 offers multilevel output over-voltage
protection to ensure no conflicts occur during pre-biased
conditioned power-up or no/light load soft stop. OVP is
sensed through the FB which allows users to externally use
FB resistor dividers if output voltages greater than 1.52V
are desired. The OVP threshold is set to 1.65V during
power up until VR Settled is reached, then the threshold
is reduced to VDAC+130mV. This OVP threshold is
maintained during normal operation and remains until
VO, the output of the remote sense amplifier, reaches
250mV with respect to ground. This ensures OVP
protection during soft stop events or down tracking
events. The OVP threshold then returns to 1.65V on the
FB pin to allow pre-bias startup.
The over voltage condition also sets the over voltage fault
latch which ensures the voltage regulator is off. OVP
overrides the normal PWM operation and will regulate the
output voltage by modulating the low-side MOSFET within
approximately 150ns to prevent the FB pin from exceeding
the OVP threshold. The OVP fault condition can only be
cleared by cycling VCC UVLO or ENABLE.
OV
THRESHOLD
VDAC + 130mV
VDAC + 130mV
VDAC
NORMAL
OPERATION
IR3531 drives the ROSC/OVP pin above V(VCC)–1V to
indicate an over voltage event has occurred. This ROSC/
OVP flag can be used by the system designer to shut the
input if desired.
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March 22, 2012 | FINAL | V2.27
VID DOWN
VID LOW
VID UP
NORMAL
OPERATION
Figure 16: Over Voltage Protection during SETVID Fast/Slow
IR3531
4+1 Phase Dual Output Control IC
DESIGN PROCEDURES
RICCP1 
IR3531 EXTERNAL COMPONENTS
Switching Frequency Setting
Use of internal oscillator mode is not recommended. Use
the SCLK input to set PWM frequency. When SCLK is used,
ROSC should be present, and selected for the per phase
switching frequency in use. The chart below shows the
relationship between the per-phase switching frequency
and the ROSC value.
%VDAC
100 *100 * RICCP 2
%VDAC
1
where, %VDAC is the desired percentage of VDAC found in
Table 7.
PHASE SHEDDING IMPLEMENTATION CIRCUITS
The following is a proposed circuit to implement phase
shedding. Two signals (S1 and S2) drive logic level
MOSFETs to produce a four level PHSSHED signal.
The operation is described in Table 6.
Figure 17: RROSC vs. Per-phase Switching Frequency
Figure 18: Phase Shedding Implementation
ADDRESS AND PHASE NUMBER PROGRAMMING
RESISTORS RADDR1 AND RADDR2
The ADDR pin is multi-function: SVID addressing for Rail0
and Rail1, internal/external clock synchronization and
Turbo enable/disable is selected through this pin. Choose
RADDR2 and apply the following equation to determine
RADDR1:
%VDAC
100 * 100 * RADDR2
RADDR1 
%VDAC
TABLE 10: PHASE SHEDDING CONTROL
S1
S2
V(PHSSHED)
Phases
0
0
VCC
Drop 3 Phases
0
1
0.625* VCC
Drop 2 Phases
1
0
0.31* VCC
Drop 1 Phases
1
1
0V
Drop 0 Phases
1
IMON AND IMON1 CAPACITORS
where, %VDAC is the desired percentage of VDAC found
in Table 1.
Use 100nF for CIMON and CIMON1 to provide an
approximate 1ms filtered time constant for current
reporting data.
ICCP PROGRAMMING RESISTORS
RICCP1 AND RICCP2
VCC BIAS REGULATOR POWER STAGE
COMPONENTS
The ICCP programming resistors are used to program the
maximum currents Rail0 and Rail1 can support. Choose
RICCP2 and follow the equation below to calculate RICCP1.
Use a 10 µH inductor with a current rating no less than 2 A.
Use a Schottky diode with operating current of 1 A or
higher and capable of withstanding 2 A for short periods of
time. A 10 µF capacitor ceramic capacitor rated for 16V is
recommended for charge storage and filtering.
32
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
TEMPERATURE SENSING
The TSENS pin is used to provide temperature information
of the voltage regulator by providing temperature zone
information to the microprocessor through the SVID.
This information is also used to flag VRHOT#. Temperature
is sensed via a linearized NTC resistor network. Temperature sensing and temperature zones are represented as
a percentage of the reference voltage VDAC as required by
the processor specification. A properly designed network
will get the TSENS voltage very close to the required target.
1% thermistors are highly recommended to achieve the
specified accuracy. Thermistor Beta is the biggest factor
in attaining accuracy. The target and TSENS voltages are
calculated from the equations below. The analysis is done
at VDAC of 1.5, because that is where the biggest error
occurs.
VTARGET 
VTSENSE 
RTSeq 
0.11 *1.5
0.11 *1.5
* T  0.453 *1.5 
* T min
T max  T min
T max  T min
RHOTSET 3
*1.5
RHOTSET 3  RTSeq
( RHOTSET1  RTHERM 2) * RHOTSET 2
RHOTSET1  RHOTSET 2  RTHERM 2
RTHERM 2  RTHERM 2 ROOM * exp(beta (
1
1

))
T TROOM
where RTHERM2ROOM is the thermistor value at room
temperature, beta is the thermistor coefficient, Tmax and
Tmin are the temperatures of the highest and lowest
temperature zone respectively. The temperature sensing
components are chosen by finding an approximate
solution that brings the target and TSENS as close to each
other as possible. This can be done using an optimization
routine of your choice such as the IR3531 excel design tool.
RAIL0 THERMAL COMPENSATION
Thermal compensation is required to counter the effect of
the inductor DCR positive temperature coefficient. Failure
to compensate results in large current reporting errors and
poor load line regulation. Thermal compensation is done
using a NTC thermistor and a linearizing resistor network.
A properly design network is necessary to achieve the
required accuracy targets. 1% thermistors are highly
recommended to achieve the specified accuracy.
Thermistor Beta is the biggest factor in attaining accuracy.
33
March 22, 2012 | FINAL | V2.27
IR3531
The goal is to keep VDRP-VDAC at 900 mV for all
temperatures at the maximum current. Thus, the equation
below has to be satisfied.
1 DCR * Gcs
RTCeq
VDRP  VDAC  * (
) * (1 
) * Im ax  900mV
3
n
RTCMP3
RTCeq 
RTCMP 2 * ( RTCMP1  RTHERM 1)
RTCMP1  RTCMP 2  RTHERM 1
RTHERM 1  RTHERM 1ROOM * exp(beta (
1
1

))
T TROOM
DCR  DCRROOM * (1  3850e  6 * (T  TROOM ))
where RTHERM1ROOM is the thermistor value at room
temperature, beta is the thermistor coefficient, Tmax and
Tmin are the temperatures of the highest and lowest
temperature zone respectively, Gcs is the typical current
sense amplifier gain of 32.5, and DCRROOM is the inductor
series resistance at room temperature. The temperature
sensing components are chosen by finding an approximate
solution that results in VDRP-VDAC=900mV over the entire
temperature operating range. This can be done using an
optimization routine of your choice such as the IR3531
excel design tool.
RAIL0 DROOP RESISTOR CALCULATION
RDRP in combination with the feedback resistor RFB sets
the load line of Rail0. RFB is first chosen with a typical
suggested value of 2kOhm. The following equation
calculates RDRP.
RDRP 
RFB * DCRROOM * Gcs
RTCeq ROOM
* (1 
)
3 * Ro * n
RTCMP3
where Ro is the load line, DCRROOM is the inductor series
resistance at room temperature, Gcs is the typical current
sense amplifier gain of 32.5, n is the number of phases and
RTCeqROOM is the same as RTCeq in section Rail0 Thermal
Compensation with RTHERM1 value at room temperature.
4+1 Phase Dual Output Control IC
RAIL1 THERMAL COMPENSATION
RSCALE1, RSCALE2, RSCALE3 and RTHERM3 are used to
provide current reporting thermal compensation for Rail1.
The purpose is to keep VDRP1-VDAC1 equal to 900mV for
all temperatures at the maximum load current. This is
expressed mathematically in the following equation.
VDRP1  VDAC1  9 * DCR * Gcs * Im ax *
( RSCALE1  RTHERM 3) * RSCALE 2




RSCALE
1

RTHERM
3

RSCALE
3


( RSCALE1  RTHERM 3) * RSCALE 3 

 RSCALE 2 

RSCALE1  RTHERM 3  RSCALE 3 

 900 mV
IR3531
COMPENSATION NETWORKS
IR3531 utilizes voltage mode control for small signal loop
regulation. The compensation scheme is a classic type 3
system consisting of components RFB(1), CFB(1), RCFB(1),
CEA(1), CCP(1) and RCP(1).
The system dynamics can change significantly when
transitioning from 4 phases to 1 phase. Loop 0 has an
additional component, RPSC, that is inserted in the loop
when in PS1 mode (single phase) to optimize phase
margin. RPSC adds to RCP thereby reducing the system
bandwidth if desired. To disable this feature, place RPSC
as a zero ohm resistor. The IR3531 excel design tool can
be used to calculate an initial starting point.
Note RDRP needs to be recalculated if RFB is changed.
where DCR and RTHERM3 are expressed in section Rail0
Thermal Compensation. Imax is the maximum current for
Rail1 and Gcs is the typical current sense amplifier gain of
32.5. The temperature sensing components are chosen by
finding an approximate solution that results in VDRP1VDAC1=900mV over the entire temperature operating
range. This can be done using an optimization routine of
your choice such as the IR3531 excel design tool.
LAYOUT GUIDELINES

VCC bias inductor LVCC must be close to SW pin.
VCC bias bulk cap COUTVCC must be located near
LVCC and connections for COUTVCC must be as
short as possible.

For both rails, all components connected to EA,
FB, VDRP, and VO pins must be located on the
same layer as the IR3531 as close to these pins as
possible.

Insert 9 equally spaced connection vias to GND
tab of IR3531.

V12V decoupling cap must be near pin of IR3531
with GND connection as short as possible.

ROSC must be located close to pin of IR3531.

RTHERM1 and RTHERM3 must be located close to
inductor of associated voltage regulator. Locate
RTHERM2 to provide overall temperature reading
of the power converter.
RAIL 1 DROOP RESISTOR CALCULATION
RDRP1 in combination with the feedback resistor RFB1
sets the load line of Rail1. RFB1 is first chosen with a
typical suggested value of 2kOhm. The equation below
calculates RDRP1.
RFB1 * DCRROOM * Gcs
*
Ro
( RSCALE1  RTHERM 3 ROOM ) * RSCALE 3




RSCALE
1

RTHERM
3

RSCALE
3


ROOM

RSCALE1  RTHERM 3 ROOM ) * RSCALE 3 
 RSCALE 2 

RSCALE1  RTHERM 3 ROOM  RSCALE 3 

RDRP1  9 *
where Ro is the load line, DCRROOM is the inductor series
resistance at room temperature, Gcs is the typical current
sense amplifier gain of 32.5, RTHERM3ROOM value at room
temperature.
34
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
METAL AND COMPONENT PLACEMENT
 Lead land width should be equal to nominal part
lead width. The minimum lead to lead spacing
should be ≥ 0.2mm to minimize prevent shorting.
 Lead land length should be equal to maximum
part lead length + 0.3 mm outboard extension +
0.05mm inboard extension. The outboard
extension ensures a large and inspectable toe fillet,
and the inboard extension will accommodate any
part misalignment and ensure a fillet.
 Center pad land length and width should be
equal to maximum part pad length and width.
However, the minimum metal to metal spacing
should be ≥ 0.17mm for 2 oz. Copper (≥ 0.1mm for
1 oz. Copper and ≥ 0.23mm for 3 oz. Copper)
 A single 0.30mm diameter via shall be placed in the
center of the pad land and connected to ground to
minimize the noise effect on the IC.
 No PCB traces should routed nor Vias placed under
any of the 4 corners of the IC package. Doing so
can cause the IC to rise up from the PCB resulting
in poor solder joints to the IC leads.
Figure 19: Metal and Component Placement
* Contact International Rectifier to receive an electronic PCB Library file in your preferred format.
35
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
SOLDER RESIST
 The solder resist should be pulled away from
the metal lead lands by a minimum of 0.06mm.
The solder resist misalignment is a maximum
of 0.05mm and it is recommended that the lead
lands are all Non Solder Mask Defined (NSMD).
Therefore pulling the S/R 0.06mm will always
ensure NSMD pads.
 The minimum solder resist width is 0.13mm.
 At the inside corner of the solder resist where
the lead land groups meet, it is recommended
to provide a fillet so a solder resist width of
≥ 0.17mm remains.
 The land pad should be Solder Mask Defined (SMD),
with a minimum overlap of the solder resist onto the
copper of 0.06mm to accommodate solder resist
miss-alignment. In 0.5mm pitch cases it is allowable
to have the solder resist opening for the land pad to
be smaller than the part pad.
 Ensure that the solder resist in-between the lead
lands and the pad land is ≥ 0.15mm due to the high
aspect ratio of the solder resist strip separating the
lead lands from the pad land.
 The vias in the large center pad should be tented or
plugged from bottom board side with solder resist.
Figure 20: Solder Resist
* Contact International Rectifier to receive an electronic PCB Library file in your preferred format.
36
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
IR3531
STENCIL DESIGN
 The land pad aperture should be approximately 70%
area of solder on the center pad. If too much solder
is deposited on the center pad the part will float and
the lead lands will be open.
 The stencil apertures for the lead lands should be
approximately 80% of the area of the lead lands.
Reducing the amount of solder deposited will
minimize the occurrence of lead shorts. Since for
0.5mm pitch devices the leads are only 0.25mm
wide, the stencil apertures should not be made
narrower; openings in stencils < 0.25mm wide
are difficult to maintain repeatable solder
release.
 The stencil lead land apertures should therefore
be shortened in length by 80% and centered on
the lead land.
 The maximum length and width of the land pad
stencil aperture should be equal to the solder resist
opening minus an annular 0.2mm pull back to
decrease the incidence of shorting the center land
to the lead lands when the part is pushed into the
solder paste.
.
Figure 21: Stencil Design
* Contact International Rectifier to receive an electronic PCB Library file in your preferred format.
37
March 22, 2012 | FINAL | V2.27
4+1 Phase Dual Output Control IC
MARKING INFORMATION
3531
?YWW?
XXXXX
SITE/DATE/MARKING CODE
LOT CODE
Figure 22: Package Marking
PACKAGE INFORMATION
48L MLPQ (7 x 7 mm Body) θJA = 23.5 ºC/W, θJC = 1 ºC/W
Figure 23: Package Dimensions
38
March 22, 2012 | FINAL | V2.27
IR3531
4+1 Phase Dual Output Control IC
IR3531
Data and specifications subject to change without notice.
This product will be designed and qualified for the Consumer market.
Qualification Standards can be found on IR’s Web site.
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
Visit us at www.irf.com for sales contact information.
www.irf.com
39
March 22, 2012 | FINAL | V2.27