Data Sheet No.PD94714 revA IR3624MPBF HIGH FREQUENCY SYNCHRONOUS PWM BUCK CONTROLLER Features Description • • • • The IR3624 is a PWM controller designed for • • • • • Internal 600kHz Oscillator Operates with Single 5V or 12V Supply Programmable Over Current Protection Hiccup Current Limit Using MOSFET RDS(on) sensing Precision Reference Voltage (0.6V) Programmable Soft-Start Pre-Bias Start-up Thermal Protection 10-Lead MLPD Package high performance synchronous Buck DC/DC applications. The IR3624 drives a pair of external N-MOSFETs using a fixed 600kHz switching frequency allowing the use of small external components. The output voltage can be precisely regulated using the internal 0.6V reference voltage for low voltage applications. Protection such as Pre-Bias startup, hiccup current limit and thermal shutdown are provided to give required Applications • • • • • system level security in the event of fault Embedded Telecom Systems Distributed Point of Load Power Architectures Computing Peripheral Voltage Regulator Graphics Card General DC/DC Converters conditions. Vin Vc Vcc HDrv OCSet Vout U1 Comp LDrv PGnd SS / SD Gnd Fb Fig. 1: Typical application Circuit ORDERING INFORMATION PKG DESIG M M PACKAGE DESCRIPTION IR3624MPBF IR3624MTRPBF PIN PARTS PARTS COUNT PER TUBE PER REEL 10 121 ------10 -------3000 T&R ORIANTAION Figure A IR3624MPBF ABSOLUTE MAXIMUM RATINGS (Voltages referenced to GND) • Vcc Supply Voltage ................................................… -0.5V to 16V • Vc Supply Voltage …………………………………….. -0.5V to 30V • Storage Temperature Range ..................................... -65°C To 150°C • Operating Junction Temperature Range ................... -40°C To 150°C • ESD Classification …………………………………..… JEDEC, JESD22-A114 • Moisture Sensitivity Level ……………………………. JEDEC Level 1 @ 260oC Caution: Stresses above those listed in “Absolute Maximum Rating” may cause permanent damage to the device. These are stress ratings only and function of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to “Absolute Maximum Rating” conditions for extended periods may affect device reliability. Package Information VCC 1 10 OCSet LDrv 2 9 SS/SD PGnd 3 8 Gnd HDrv 4 7 Comp VC 5 6 Fb Exposed Pad 10-Lead MLPD, 3x3mm ΘJA = 33o C/W * ΘJC = 2.1o C/W *Exposed pad on underside is connected to a copper pad through vias for 4-layer PCB board design 2 IR3624MPBF Block Diagram Vcc 3V Bias Generator 3V 3V 0.6V Vc 20uA Thermal Shutdown 40uA SS / SD 3uA OCP UVLO POR HDrv POR Vc 0.6V PWM Comp Error Amp Vcc R Q S Fb LDrv Comp R 0.3V Ramp Oscillator Set Q PGnd PBias S SS POR OCP Gnd OCSet 20uA Fig. 2: Simplified block diagram of the IR3624 3 IR3624MPBF Pin Description Pin Name Description 1 Vcc This pin provides power for the internal blocks of the IC as well as powers the low side driver. A minimum of 0.1uF, high frequency capacitor must be connected from this pin to power ground. Output driver for low side MOSFET 2 LDrv 3 PGnd Power Ground. This pin serves as a separate ground for the MOSFET drivers and should be connected to the system’s power ground plane. 4 HDrv Output driver for high side MOSFET 5 Vc 6 Fb 7 Comp 8 Gnd 9 SS/SD 10 OCSet This pin powers the high side driver and must be connected to a voltage higher than bus voltage. A minimum of 0.1uF, high frequency capacitor must be connected from this pin to power ground. Inverting input to the error amplifier. This pin is connected directly to the output of the regulator via resistor divider to set the output voltage and provide feedback to the error amplifier. Output of error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. Signal ground for internal reference and control circuitry. Soft start / shutdown. This pin provides user programmable soft-start function. Connect an external capacitor from this pin to ground to set the start up time of the output voltage. The converter can be shutdown by pulling this pin below 0.3V. Current limit set point. A resistor from this pin to drain of low side MOSFET will set the current limit threshold. 4 IR3624MPBF Recommended Operating Conditions Symbol Definition Min Vcc Vc Fs Tj Supply Voltage Supply Voltage Operating Frequency Junction Temperature Max 4.5 10 14 28 660 125 -40 Units V V kHz o C Electrical Specifications Unless otherwise specified, these specification apply over Vcc=Vc=12V, 0oC<Tj< 105oC Parameter SYM Test Condition Min TYP MAX Units Reference Voltage Feedback Voltage VFB 0.6 o Accuracy o 0 C<Tj<105 C o o -40 C<Tj<105 C, Note1 -1.5 -2.5 V +1.5 +1.5 % % Supply Current VCC Supply Current (Static) VCC Supply Current (Dynamic) ICC(Static) ICC(Dynamic) VC Supply Current (Static) IC(Static) VC Supply Current (Dynamic) IC(Dynamic) SS=0V, No Switching 6 8 mA Fs=600kHz, CLOAD=1.5nF 10 15 mA SS=0V, No Switching 3 6 mA Fs=600kHz, CLOAD=1.5nF 17 25 mA 0.2 4.4 4.1 0.3 3.5 3.25 0.25 V V V V V V 600 660 kHz Under Voltage Lockout VCC-Start-Threshold VCC-Stop-Threshold VCC-Hysteresis VC-Start-Threshold VC-Stop-Threshold VC-Hysteresis VCC_UVLO(R) VCC_UVLO(F) VC_UVLO(R) VC_UVLO(F) Supply ramping up Supply ramping down Supply ramping up and down Supply ramping up Supply ramping down Supply ramping up and down 4.0 3.7 0.15 3.1 2.85 0.15 0.25 Oscillator Frequency FS 540 Ramp Amplitude Vramp Note2 Min Duty Cycle Dmin Fb=1V 0 % Min Pulse Width Dmin(ctrl) Fs=600kHz, Note2 80 ns Max Duty Cycle Dmax Fs=600kHz, Fb=0.5V 1.25 71 V % Note1: Cold temperature performance is guaranteed via correlation using statistical quality control. Not tested for production. Note2: Guaranteed by Design but not tested for production. 5 IR3624MPBF Parameter SYM Test Condition Min TYP MAX Units -0.1 -0.5 µA Error Amplifier Input Bias Current IFB SS=3V Source/Sink Current I(source/Sink) 50 70 90 µA Transconductance gm 1000 1300 1600 µmho 15 20 25 µA 0.25 V 25 µA µA % Soft Start/SD Soft Start Current ISS Shutdown Threshold SD Output SS=0V Over Current Protection OCSET Current Hiccup Current Hiccup Duty Cycle IOCSET IHiccup Hiccup(duty) 15 20 3 15 Note2 IHiccup / ISS , Note2 Thermal Shutdown Thermal Shutdown Threshold Note2 Thermal Shutdown Hysteresis o C 135 o Note2 20 CL=1.5nF, Fs=600kHz See Fig 3 CL=1.5nF, Fs=600kHz See Fig 3 30 60 ns 30 60 ns C Output Drivers LO, Drive Rise Time Tr(Lo) HI Drive Rise Time Tr(Hi) LO Drive Fall Time Tf(Lo) CL=1.5nF, Fs=600kHz See Fig 3 30 60 ns HI Drive Fall Time Tf(Hi) 30 60 ns Dead Band Time Tdead CL=1.5nF, Fs=600kHz See Fig 3 See Fig 3 50 100 ns Note2: Guaranteed by Design but not tested for production. Tr Tf 9V High Side Driver (HDrv) 2V Tr Tf 9V Low Side Driver (LDrv) 2V Deadband H_to_L Deadband L_to_H Fig. 3: Definition of Rise/Fall time and Deadband Time 6 IR3624MPBF TYPICAL OPERATING CHARACTERISTICS ISS (mA) Vfb (mV) 603 602 601 [uA] [mV] 600 599 598 597 596 595 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 100 110 120 -15 -16 -17 -18 -19 -20 -21 -22 -23 -24 -25 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 100 110 120 60 70 80 90 100 110 120 70 80 90 100 110 120 70 80 90 100 110 120 Tem p ['C] Tem p ['C] Iccq(mA) Icq(mA) 8 6 7.5 5.5 7 5 [mA] [mA] 6.5 6 5.5 4.5 4 5 3.5 4.5 4 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 3 -40 -30 -20 -10 90 100 110 120 0 10 20 Tem p ['C] 40 50 Tem p ['C] Transconductance (gm) (mMHO) Frequency (kHz) 1.5 620 1.4 610 600 1.3 [KHz] [mMHO] 30 1.2 590 580 1.1 1 -40 -30 -20 -10 570 0 10 20 30 40 50 60 70 80 560 -40 -30 -20 -10 90 100 110 120 0 10 20 Tem p ['C] 30 40 50 60 Tem p ['C] Dead Time (nS) IOCSET (uA) -16 100 -17 -18 90 80 -19 [nS] [uA] -15 -20 -21 -22 60 50 40 -23 -24 -40 -30 -20 -10 70 0 10 20 30 40 50 Tem p ['C] 60 70 80 90 100 110 120 30 -40 -30 -20 -10 0 10 20 30 40 50 60 Tem p ['C] 7 IR3624MPBF Circuit Description THEORY OF OPEARTION Shutdown Introduction The output can be shutdown by pulling the softstart pin below 0.3V. This can be easily done by using an external small signal transistor. During shutdown both MOSFET drivers will be turned off. Normal operation will resume by cycling soft start pin. The IR3624 is a voltage mode PWM synchronous controller and operates with a fixed 600kHz switching frequency, allowing the use of small external components. The output voltage is set by feedback pin (Fb) and the internal reference voltage (0.6V). These are two inputs to error amplifier. The error signal between these two inputs is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulses of variable duty-cycle (D) which drivers N-channel external MOSFETs. The timing of the IC is controlled by an internal oscillator circuit that uses on-chip capacitor to set the switching frequency. The IR3624 operates with single input voltage from 4.5V to 12V allowing an extended operating input voltage range. The current limit is programmable and uses onresistance of the low-side MOSFET, eliminating the need for external current sense resistor. Under-Voltage Lockout The under-voltage lockout circuit monitors the two input supplies (Vcc and Vc) and assures that the MOSFET driver outputs remain in the off state whenever the supply voltage drops below set thresholds. Lockout occurs if Vc or Vcc fall below 3.3V and 4.2V respectively. Normal operation resumes once Vc and Vcc rise above the set values. Thermal Shutdown Temperature sensing is provided inside IR3624. The trip threshold is typically set to 145oC. When trip threshold is exceeded, thermal shutdown turns off both MOSFETs. Thermal shutdown is not latched and automatic restart is initiated when the sensed temperature drops within the operating range. There is a 20oC hysteresis in the thermal shutdown threshold. Error Amplifier The IR3624 is a voltage mode controller. The error amplifier is of transconductance type. The amplifier is capable of operating with Type III compensation control scheme using low ESR output capacitance. Pre-Bias Startup IR3624 is able to start up into pre-charged output, which prevents oscillation and disturbances of the output voltage. The output starts in asynchronous fashion and keeps the synchronous MOSFET off until the first gate signal for control MOSFET is generated. Figure 4 shows a typical Pre-Bias condition at start up. Depends on system configuration, specific amount of output capacitors may be required to prevent discharging the output voltage. Volt Vo Pre-Bias Voltage (Output Voltage before startup) Time Fig. 4: Pre-Bias start up 8 IR3624MPBF Soft-Start The IR3624 has programmable soft-start to control the output voltage rise and limit the inrush current during start-up. To ensure correct start-up, the soft-start sequence initiates when Vcc and Vc rise above their threshold and generate the Power On Ready (POR) signal. The soft-start function operates by sourcing current to charge an external capacitor to about 3V. Initially, the soft-start function clamps the output of error amplifier by injecting a current (40uA) into the Fb pin and generates a voltage about 0.96V (40ux24K) across the negative input of error amplifier (see figure 5). The magnitude of the injected current is inversely proportional to the voltage at the soft-start pin. As the soft-start voltage ramps up, the injected current decreases linearly and so does the voltage at negative input of error amplifier. When the soft-start capacitor is around 1V, the voltage at the positive input of the error amplifier is approximately 0.6V. The output of error amplifier will start increasing and generating the first PWM signal. As the softstart capacitor voltage continues to go up, the current flowing into the Fb pin will keep decreasing. The feedback voltage increases linearly as the soft start voltage ramps up. When soft-start voltage is around 2V the output voltage is reached the steady state and the injected current is zero. 3V 20uA SS/SD 40uA POR Comp 24K 0.6V Error Amp 24K Fb Fig. 5: Soft-Start circuit for IR3624 Output of UVLO POR 3V ≅2V ≅1V Soft-Start Voltage Current flowing into Fb pin 0V 40uA 0uA Voltage at negative input ≅0.96V of Error Amp Figure 6 shows the theoretical operational waveforms during soft-start. 0.6V The output voltage start-up time is the time period when soft-start capacitor voltage increases from 1V to 2V. 0.6V The start-up time will be dependent on the size of the external soft-start capacitor and can be estimate by: 20µA ∗ Tstart = 2V −1V Css Voltage at Fb pin 0V Fig. 6: Theoretical operation waveforms during soft-start For a given start-up time, the soft-start capacitor (nF) can be estimated as: CSS ≅ 20µA * Tstart (ms) --(1) 9 IR3624MPBF Over-Current Protection 28uA The over current protection is performed by sensing current through the RDS(on) of low side MOSFET. This method enhances the converter’s efficiency and reduce cost by eliminating a current sense resistor. As shown in figure 7, an external resistor (RSET) is connected between OCSet pin and the drain of low side MOSFET (Q2) which sets the current limit set point. The internal current source develops a voltage across RSET. When the low side MOSFET is turned on, the inductor current flows through the Q2 and results a voltage which is given by: VOCSet = (IOCSet ∗ ROCSet ) − (RDS(on) ∗ IL ) --(2 ) IOCSET IR3624 Q1 L1 OCSet RSET VOUT OCP 20uA SS1 / SD 20 3uA Fig. 8: 3uA current source for discharging soft-start capacitor during hiccup The OCP circuit starts sampling current when the low gate drive is about 3V. The OCSet pin is internally clamped during deadtime to prevent false trigging, figure 9 shows the OCSet pin during one switching cycle. As it is shown there is about 150ns delay to mask the deadtime, since this node contains switching noises, this delay also functions as a filter. Q2 Hiccup Control Fig. 7: Connection of over current sensing resistor Deadtime The critical inductor current can be calculated by setting: VOCSet = (IOCSet ∗ ROCSet ) − (RDS(on) ∗ IL ) = 0 ISET = IL(critical) = ROCSet ∗ IOCSet RDS(on) Blanking time IOCSet*ROCSet Clamp voltage --(3 ) An over current is detected if the OCSet pin goes below ground. This trips the OCP comparator and cycles the soft start function in hiccup mode. The hiccup is performed by charging and discharging the soft-start capacitor in certain slope rate. As shown in figure 8 a 3uA current source is used to discharge the soft-start capacitor. The OCP comparator resets after every soft start cycles, the converter stays in this mode until the overload or short circuit is removed. The converter will automatically recover. Fig. 9: OCset pin during normal condition Ch1: Inductor point, Ch2:Ldrv, Ch3:OCSet The value of RSET should be checked in an actual circuit to ensure that the over current protection circuit activates as expected. The IR3624 current limit is designed primarily as disaster preventing, "no blow up" circuit, and doesn't operate as a precision current regulator. 10 IR3624MPBF Application Information Soft-Start Programming Design Example: The soft-start timing can be programmed by selecting the soft-start capacitance value. The start-up time of the converter can be calculated by using: The following example is a typical application for IR3624. The application circuit is shown in page17. Vin = 12V , (13.2V , max) CSS ≅ 20µA * Tstart --(1) ∆Vo ≤ 30 mV Where Tstart is the desired start-up time (ms) For a start-up time of 5ms, the soft-start capacitor will be 0.1uF. Choose a ceramic capacitor at 0.1uF. Fs = 600 kHz Vc supply for single input voltage Vo = 1.8V Io = 6 A Output Voltage Programming Output voltage is programmed by reference voltage and external voltage divider. The Fb pin is the inverting input of the error amplifier, which is internally referenced to 0.6V. The divider is ratioed to provide 0.6V at the Fb pin when the output is at its desired value. The output voltage is defined by using the following equation: ⎛ R ⎞ Vo = Vref ∗ ⎜⎜1 + 8 ⎟⎟ R9 ⎠ ⎝ --( 4 ) When an external resistor divider is connected to the output as shown in figure 10. VOUT IR3624 R8 Fb R9 Fig. 10: Typical application of the IR3624 for programming the output voltage To drive the high side switch, it is necessary to supply a gate voltage at least 4V grater than the bus voltage. This is achieved by using a charge pump configuration as shown in figure 11. This method is simple and inexpensive. The operation of the circuit is as follows: when the lower MOSFET is turned on, the capacitor (C1) is pulled down to ground and charges, up to VBUS value, through the diode (D1). The bus voltage will be added to this voltage when upper MOSFET turns on in next cycle, and providing supply voltage (Vc) through diode (D2). Vc is approximately: VC ≅ 2 ∗Vbus − (VD1 + VD2 ) --(6 ) Capacitors in the range of 0.1uF is generally adequate for most applications. The diodes must be a fast recovery device to minimize the amount of charge fed back from the charge pump capacitor into VBUS. The diodes need to be able to block the full power rail voltage, which is seen when the high side MOSFET is switched on. For low voltage application, schottky diodes can be used to minimize forward drop across the diodes at start up. VBUS Equation (4) can be rewritten as: D1 C3 D2 ⎛ Vref R9 = R8 ∗ ⎜⎜ ⎝ V O−Vref ⎞ ⎟⎟ ⎠ VBUS Vc --( 5 ) C2 C1 Q1 L For the calculated values of R8 and R9 see feedback compensation section. IR3624 HDrv Q2 Fig. 11: Charge pump circuit to generate Vc voltage 11 IR3624MPBF Input Capacitor Selection The input filter capacitor should be selected based on how much ripple the supply can tolerate on the DC input line. The ripple current generated during the on time of upper MOSFET should be provided by input capacitor. The RMS value of this ripple is expressed by: IRMS = Io ∗ D ∗ (1 − D ) --(7 ) V D= o Vin Where: D is the Duty Cycle IRMS is the RMS value of the input capacitor current. Io is the output current. If ∆i ≈ 50%(Io ), then the output inductor will be: L = 0.82uH The ACT STS703 series provides a range of inductors in different values, low profile suitable for large currents. Output Capacitor Selection The voltage ripple and transient requirements determines the output capacitors types and values. The criteria is normally based on the value of the Effective Series Resistance (ESR). However the actual capacitance value and the Equivalent Series Inductance (ESL) are other contributing components, these components can be described as: For Io=6A and D=0.13, the IRMS=2.0A. ∆Vo = ∆Vo(ESR) + ∆Vo(ESL) + ∆Vo(C ) Ceramic capacitors are recommended due to their peak current capabilities, they also feature low ESR and ESL at higher frequency which enhance better efficiency, ∆Vo(ESR) = ∆IL * ESR Use one 10uF, 25V ceramic capacitor from Panasonic. ∆Vo(ESL) = ⎜ Inductor Selection The inductor is selected based on output power, operating frequency and efficiency requirements. Low inductor value causes large ripple current, resulting in the smaller size, faster response to a load transient but poor efficiency and high output noise. Generally, the selection of inductor value can be reduced to desired maximum ripple current in the inductor ( ∆i ) . The optimum point is usually found between 20% and 50% ripple of the output current. For the buck converter, the inductor value for desired operating ripple current can be determined using the following relation: Vin − Vo = L ∗ ∆i 1 ; ∆t = D ∗ Fs ∆t L = (Vin − Vo ) ∗ Vo Vin ∗ ∆i * Fs Where: Vin = Maximum input voltage Vo = Output Voltage ∆i = Inductor ripple current F s= Switching frequency ∆t = Turn on time D = Duty cycle --(8 ) - -(9) ⎛Vin ⎞ ⎟ * ESL ⎝L⎠ ∆Vo(C ) = ∆IL 8 * Co * Fs ∆Vo = Output voltage ripple ∆IL = Inductor ripple current Since the output capacitor has major role in overall performance of converter and determine the result of transient response, selection of capacitor is critical. The IR3624 can perform well with all types of capacitors. As a rule the capacitor must have low enough ESR to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The goal for this design is to meet the voltage ripple requirement in smallest possible capacitor size. Therefore ceramic capacitor is selected due to low ESR and small size. Two of the Panasonic ECJ2FB0J226M (22uF, 6.3V, X5R and EIA 0805 case size) is a good choice. In the case of tantalum or low ESR electrolytic capacitors, the ESR dominates the output voltage ripple, equation (9) can be used to calculate the required ESR for the specific voltage ripple. 12 IR3624MPBF Power MOSFET Selection The IR3624 uses two N-Channel MOSFETs. The selections criteria to meet power transfer requirements is based on maximum drain-source voltage (VDSS), gate-source drive voltage (Vgs), maximum output current, On-resistance RDS(on) and thermal management. The MOSFET must have a maximum operating voltage (VDSS) exceeding the maximum input voltage (Vin). The gate drive requirement is almost the same for both MOSFETs. Logic-level transistor can be used and caution should be taken with devices at very low Vgs to prevent undesired turn-on of the complementary MOSFET, which results a shootthrough current. The total power dissipation for MOSFETs includes conduction and switching losses. For the Buck converter the average inductor current is equal to the DC load current. The conduction loss is defined as: Pcond = (upper switch)= I ∗ Rds(on) ∗ D ∗ ϑ Pcond = (lower switch)= I ∗ Rds(on) ∗ (1 − D) ∗ϑ 2 load 2 load switching losses in synchronous Buck converter. The synchronous MOSFET turns on under zero voltage conditions, therefore, the turn on losses for synchronous MOSFET can be neglected. With a linear approximation, the total switching loss can be expressed as: Psw = Vds(off ) tr + tf * * Iload - - - (10) 2 T Where: V ds(off) = Drain to source voltage at the off time tr = Rise time tf = Fall time T = Switching period Iload = Load current The switching time waveforms is shown in figure12. VDS 90% ϑ = Rds(on) temperature dependency The RDS(on) temperature dependency should be considered for the worst case operation. This is typically given in the MOSFET data sheet. Ensure that the conduction losses and switching losses do not exceed the package ratings or violate the overall thermal budget. For this design, IRF8910 is a good choice. The device provides two N-MOSFETs in a compact SO-8 package. The IRF8910 has the following data: Vds = 20V, Id = 10A Rds(on) = 13.4mΩ @Vgs = 10V The conduction losses will be: Pcon=0.724W The switching loss is more difficult to calculate, even though the switching transition is well understood. The reason is the effect of the parasitic components and switching times during the switching procedures such as turn-on / turnoff delays and rise and fall times. The control MOSFET contributes to the majority of the 10% VGS td(ON) tr td(OFF) tf Fig. 12: switching time waveforms From IRF8910 data sheet: tr = 10ns tf = 4.1ns These values are taken under a certain condition test. For more details please refer to the IRF8910 data sheet. By using equation (10), we can calculate the switching losses. Psw=0.37W 13 IR3624MPBF Feedback Compensation The IR3624 is a voltage mode controller; the control loop is a single voltage feedback path including error amplifier and error comparator. To achieve fast transient response and accurate output regulation, a compensation circuit is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin (greater than 45o). The output LC filter introduces a double pole, – 40dB/decade gain slope above its corner resonant frequency, and a total phase lag of 180o (see figure 13). The resonant frequency of the LC filter expressed as follows: FLC = The ESR zero of the output capacitor expressed as follows: 1 FESR = - - - (12) 2 ∗ π * ESR * Co VOUT R8 Fb E/A R9 Gain VREF R3 CPOLE Gain(dB) H(s) dB FZ Frequency Fig. 14: TypeII compensation network and its asymptotic gain plot The transfer function (Ve/Vo) is given by: ⎛ R9 ⎞ 1 + sR3C4 ⎟⎟ * H(s) = ⎜⎜ gm * - - - (13) R sC4 9 + R8 ⎠ ⎝ Phase 0 0dB Ve C4 1 - - - (11) 2 ∗ π Lo ∗ Co Figure 13 shows gain and phase of the LC filter. Since we already have 180o phase shift just from the output filter, the system risks being unstable. Comp -40dB/decade The (s) indicates that the transfer function varies as a function of frequency. This configuration introduces a gain and zero, expressed by: FLC Frequency -180 FLC Frequency [H(s)] = ⎛⎜⎜ g ⎝ Fig. 13: Gain and Phase of LC filter The IR3624’s error amplifier is a differential-input transconductance amplifier. The output is available for DC gain control or AC phase compensation. The error amplifier can be compensated either in type II or typeIII compensation. When it is used in typeII compensation the transconductance properties of the error amplifier become evident and can be used to cancel one of the output filter poles. This will be accomplished with a series RC circuit from Comp pin to ground as shown in figure 14. This method requires that the output capacitor should have enough ESR to satisfy stability requirements. In general the output capacitor’s ESR generates a zero typically at 5kHz to 50kHz which is essential for an acceptable phase margin. Fz = m * R9 ⎞ ⎟ * R3 - - - (14) R9 + R8 ⎟⎠ 1 2π * R3 * C4 - - - (15) The gain is determined by the voltage divider and error amplifier’s transconductance gain. First select the desired zero-crossover frequency (Fo): Fo > FESR and Fo ≤ (1/5 ~ 1/10) * Fs Use the following equation to calculate R4: R3 = Vosc * Fo * FESR * (R8 + R9 ) Vin * FLC2 * R9 * gm - - - (15) Where: Vin = Maximum Input Voltage Vosc = Oscillator Ramp Voltage Fo = Crossover Frequency FESR = Zero Frequency of the Output Capacitor FLC = Resonant Frequency of the Output Filter R8 and R9 = Feedback Resistor Dividers gm = Error Amplifier Transconductance 14 IR3624MPBF To cancel one of the LC filter poles, place the zero before the LC filter resonant frequency pole: VOUT ZIN Fz = 75%FLC C7 1 Fz = 0.75 * 2π Lo * Co - - - (16) Using equations (15) and (16) to calculate C9. One more capacitor is sometimes added in parallel with C4 and R3. This introduces one more pole which is mainly used to suppress the switching noise. The additional pole is given by: 1 1 = ≅ 1 π * R 3 * Fs π * R3 * Fs − C4 Fs 2 For a general solution for unconditionally stability for any type of output capacitors, in a wide range of ESR values we should implement local feedback with a compensation network (typeIII). The typically used compensation network for voltage-mode controller is shown in figure 15. In such configuration, the transfer function is given by: Ve 1 − g m Zf = Vo 1 + g m ZIN The error amplifier gain is independent of the transconductance under the following condition: gm * Z f >> 1 and gm * Z in >> 1 - - - (17) By replacing Zin and Zf according to figure 15, the transformer function can be expressed as: H (s ) = (1 + sR3C4 ) * [1 + sC7 (R8 + R10 )] 1 * sR8 (C4 + C3 ) ⎡ ⎛ C4 * C3 ⎞⎤ ⎟⎟⎥ * (1 + sR10C7 ) ⎢1 + sR3 ⎜⎜ ⎝ C4 + C3 ⎠⎦ ⎣ C4 R8 Zf Fb R9 E/A Comp Ve VREF Gain(dB) H(s) dB FZ1 The pole sets to one half of switching frequency which results in the capacitor CPOLE: For FP << R3 R10 1 FP = C *C 2π * R3 * 4 POLE C4 + CPOLE CPOLE C3 FZ2 FP2 FP3 Frequency Fig.15: Compensation network with local feedback and its asymptotic gain plot As known, transconductance amplifier has high impedance (current source) output, therefore, consider should be taken when loading the error amplifier output. It may exceed its source/sink output current capability, so that the amplifier will not be able to swing its output voltage over the necessary range. The compensation network has three poles and two zeros and they are expressed as follows: FP1 = 0 FP 2 = 1 2π * R10 * C7 FP 3 = 1 1 ≅ ⎛ C * C3 ⎞ 2π * R3 * C3 ⎟⎟ 2π * R3 ⎜⎜ 4 ⎝ C4 + C3 ⎠ Fz1 = 1 2π * R3 * C4 Fz 2 = 1 1 ≅ 2π * C7 * (R8 + R10 ) 2π * C7 * R8 Cross over frequency is expressed as: Fo = R3 * C7 * Vin 1 * Vosc 2π * Lo * Co 15 IR3624MPBF Based on the frequency of the zero generated by output capacitor and its ESR versus crossover frequency, the compensation type can be different. The table below shows the compensation types and location of crossover frequency. Compensator type FESR vs. Fo Output capacitor TypII(PI) FLC<FESR<Fo<Fs/2 Electrolytic , Tantalum TypeIII(PID) Method A FLC<Fo<FESR<Fs/2 Tantalum, ceramic TypeIII(PID) Method B FLC<Fo<Fs/2<FESR Ceramic Table1- The compensation type and location of FESR versus Fo The following design rules will give a crossover frequency approximately one-tenth of the switching frequency. The higher the band width, the potentially faster the load transient response. The DC gain will be large enough to provide high DC-regulation accuracy (typically -5dB to -12dB). The phase margin should be greater than 45o for overall stability. Desired Phase Margin: Θmax = π 3 1 − SinΘ 1 + SinΘ FZ 2 = 16kHz FZ 2 = Fo * 1 + SinΘ 1 − SinΘ FP 2 = 224kHz FP 2 = Fo * The details of these compensation types are discussed in application note AN-1043 which can be downloaded from IR Web-Site. Select : FZ1 = 0.5 * FZ 2 and FP3 = 0.5 * Fs For this design we have: R3 ≥ Vin=13.2V Vo=1.8V Vosc=1.25V Vref=0.6V gm=1000umoh Lo=0.82uH Co=2x22uF, ESR=1.5mOhm Fs=600kHz Calculate C4 , C3 and C7 : C4 = 1 ; C4 = 3.96nF, Select : C4 = 3.9nF 2π * FZ1 * R 3 C3 = 1 ; C3 = 106pF, Select : C3 = 100pF 2π * FP 3 * R3 C7 = 2π * Fo * Lo * Co * Vosc ; C10 = 0.26nF, R3 * Vin These result to: FLC=26.5kHz FESR=2.4MHz Fs/2=300kHz Select crossover frequency: Fo < FESR and Fo ≤ (1/5 ~ 1/10) * Fs Fo=60kHz 2 ; R3 ≥ 2KΩ ; Select : R3 = 5KΩ gm Select : C7 = 0.33nF Calculate R10, R8 and R9 : R10 = 1 ; R10 = 2.1KΩ, Select : R10 = 2KΩ 2π * C7 * FP 2 R8 = 1 − R10; R8 = 28KΩ, Select : R8 = 28KΩ 2π * C7 * FZ 2 R9 = Vref * R8 ; R9 = 14KΩ , Select : R9 = 14KΩ Vo − Vref Since: FLC<Fo<Fs/2<FESR, typeIII method B is selected to place the poles and zeros. 16 IR3624MPBF Programming the Current-Limit The Current-Limit threshold can be set by connecting a resistor (RSET) from drain of low side MOSFET to the OCSet pin. The resistor can be calculated by using equation (3). The RDS(on) has a positive temperature coefficient and it should be considered for the worse case operation. This resistor must be placed close to the IC, place a small ceramic capacitor from this pin to ground for noise rejection purposes. ISET = IL(critical) = ROCSet ∗ IOCSet RDS(on) --(3 ) RDS( on ) = 13.4 mΩ ∗1.5 = 20.1mΩ ISET ≅ Io ( LIM ) = 6 A ∗1.5 = 9 A (50% over nominal output current) ROCSet = R7 = 9KΩ Layout Consideration The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results. Start to place the power components, make all the connection in the top layer with wide, copper filled areas. The inductor, output capacitor and the MOSFET should be close to each other as possible. This helps to reduce the EMI radiated by the power traces due to the high switching currents through them. Place input capacitor directly to the drain of the high-side MOSFET, to reduce the ESR replace the single input capacitor with two parallel units. The feedback part of the system should be kept away from the inductor and other noise sources. The critical bypass components such as capacitors for Vcc and Vc should be close to respective pins. It is important to place the feedback components include feedback resistors and compensation components close to Fb and Comp pins. In multilayer PCB use one layer as power ground plane and have a control circuit ground (analog ground), to which all signals are referenced. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. These two grounds must be connected together on the PC board layout at a single point. The MLPD is thermal enhanced package, based on thermal performance it is recommended to use 4-layers PCB. To effectively remove heat from the device the exposed pad should be connected to ground plane using vias. 17 IR3624MPBF D1 BAT54 13.2V Vc C9 10uF ceramic C5 0.1uF Vcc C1 C6 0.1uF 0.1uF HDrv 0.82uH R7 OCSet IR3624 SS / SD C2 1.8V L1 9K Q1 IRF8910 LDrv 0.1uF PGnd Gnd Comp R8 28K Fb C4 R3 3.9nF 5K R9 14K R10 2K C11,C12 2x22uF Ceramic C7 330pF C3 100pF Fig.16: Application circuit for 13.2V to 1.8V Using ceramic output capacitor with typeIII compensation 18 IR3624MPBF TYPICAL OPERATING WAVEFORMS Vin=13.2V, Vo=1.8V, Io=0-6A, Fs=600 kHz, Room Temperature, No Air Flow Fig.17: Start up at 6A Load Ch1:Vin, Ch2:Vss, Ch3:Vo Fig.19: Output Voltage Ripple at 6A load Ch1: Vo, Ch2:Inductor Point (Lx) Fig.18: Pre-Bias Start up Ch1:Vin, Ch2:Vss, Ch3:Vo Fig.20: Gate signals at 6A load Ch1:HDrv, Ch2:Lx, Ch3:LDrv Fig.21: Transient Response Fig.22: Shorted Output, Hiccup Condition Ch1:Vo, Ch4:Istep (0-3A) Ch1:Vo, Ch2:Vss, Ch4:Io 19 IR3624MPBF TYPICAL OPERATING WAVEFORMS Vin=13.2V, Vo=1.8V, Io=0-6A, Fs=600 kHz, Room Temperature, No Air Flow Fig.23: Bode Plot at 10%of load shows a bandwidth of 70kHz and phase margin of 62 degree IR3624, 600 kHz 12V to 1.8V 85 Efficiency (%) 80 75 70 Series1 65 60 55 50 0 1 2 3 4 5 6 7 Io(A) Fig.24: Efficiency at room temperature, no air flow 20 IR3624MPBF (IR3624M) MLPD Package 3x3-10Lead D E/2 E S Y M B O L MIN NOM A 0.80 0.90 A1 0.00 0.02 VEED-5 A3 A A3 SEATING PLANE A1 Terminal 1 Identifier NOM 1.00 .032 .035 .039 0.05 .000 .0008 .0019 0.20 REF MAX .008 REF 0.18 0.25 0.30 .0071 .0098 .0118 D2 2.20 _ 2.70 .087 _ .106 3.00 BSC .118 BSC 3.00 BSC E E2 MIN MAX b D D2 INCHES MILLIMETERS .118 BSC E2 1.40 _ 1.75 .055 _ .068 L 0.30 0.40 0.50 .012 .016 .019 e 0.50 PITCH .020 PITCH N 10 10 ND 5 5 Leads on 2 sides e b L (ND-1) x e TAPE & REEL ORIENTATION 1 1 1 Figure A IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 This product has been designed and qualified for the Industrial market. Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 02/01 21