Technology Licensed from International Rectifier APU3073 SYNCHRONOUS PWM CONTROLLER WITH OVER-CURRENT PROTECTION / LDO CONTROLLER FEATURES DESCRIPTION Synchronous Controller plus one LDO controller Current Limit using MOSFET Sensing Single 5V/12V Supply Operation Programmable Switching Frequency up to 400KHz Soft-Start Function Fixed Frequency Voltage Mode Precision Reference Voltage Available Uncommitted Error Amplifier available for DDR voltage tracking application The APU3073 controller IC is designed to provide a low cost synchronous Buck regulator for on-board DC to DC converter for multiple output applications. The outputs can be programmed as low as 0.8V for low voltage applications. Selectable over-current protection is provided by using external MOSFET's on-resistance for optimum cost and performance. This device features a programmable frequency set from 200KHz to 400KHz, under-voltage lockout for all input supplies, an external programmable soft-start function as well as output under-voltage detection that latches off the device when an output short is detected. APPLICATIONS DDR memory source sink VTT application Low cost on-board DC to DC such as 12V/5V to output voltages as low as 0.8V Graphic Card Hard Disk Drive Multi-Output Applications RoHS Compliant TYPICAL APPLICATION Vcc 3.3V Q1 Drv2 Fb2 R1 VOUT2 C2 VcH R2 U1 APU3073 L1 C1 VcL +5V C6 C7 Q4 HDrv VP1 C3 0.1uF C4 R8 12V D1 VREF OCSet C9 Comp R9 L2 R7 VOUT1 LDrv Q5 C10 Rt SS/SD C11 R10 Fb1 Gnd PGnd R11 Figure 1 - Typical application of APU3073. PACKAGE ORDER INFORMATION TA (°C) 0 To 70 DEVICE APU3073O Data and specifications subject to change without notice. PACKAGE 16-Pin TSSOP 200815061-1/17 APU3073 ABSOLUTE MAXIMUM RATINGS Vcc Supply Voltage ................................................... VcL, VcH Supply Voltage .......................................... Storage Temperature Range ...................................... Operating Junction Temperature Range ..................... -0.5 - 25V -0.5 - 25V -65°C To 150°C 0°C To 125°C CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. PACKAGE INFORMATION 16-Pin TSSOP (O) Fb2 1 16 OCSet Drv2 2 15 VcH 14 HDrv Rt 3 SS/SD 4 13 Gnd Comp 5 12 PGnd Fb1 6 11 LDrv VP1 7 10 VcL VREF 8 9 Vcc uJA=908C/W ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over Vcc=5V, VcL=VcH=12V and TA=0°C to 70°C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER Feedback Voltage Fb Voltage Fb Voltage Line Regulation Reference Voltage Ref Voltage Initial Accuracy Drive Current UVLO UVLO Threshold - Vcc UVLO Hysteresis - Vcc UVLO Threshold - VcH UVLO Hysteresis - VcH UVLO Threshold - Fb1 UVLO Hysteresis - Fb1 Supply Current Vcc Dynamic Supply Current Vc Dynamic Supply Current Vcc Static Supply Current Vc Static Supply Current Soft-Start Section Charge Current SYM TEST CONDITION MIN TYP MAX UNITS 0.784 0.8 0.2 0.816 0.625 V % 0.784 0.8 2 0.816 V mA UVLO VCC Supply Ramping Up 3.9 4.8 UVLO VCH Supply Ramping Up 3.3 UVLO Fb1 Fb Ramping Down 0.3 4.4 0.25 3.5 0.2 0.4 0.1 V V V V V V 5 5 3.5 3 10 15 10 5 mA mA mA mA 25 30 mA VFB LREG 5<Vcc<12 VREF IREF Note 1 Dyn ICC Dyn IC ICCQ ICQ SS IB Freq=200KHz, CL=1500pF Freq=200KHz, CL=1500pF SS=0V SS=0V SS=0V 10 3.7 0.5 2/17 APU3073 PARAMETER Error Amp Fb Voltage Input Bias Current Fb Voltage Input Bias Current VP Voltage Range Transconductance Oscillator Frequency SYM Ramp Amplitude Output Drivers Rise Time Fall Time Dead Band Time Max Duty Cycle Min Duty Cycle LDO Controller Section Drive Current Fb Voltage Input Bias Current Thermal Shutdown Current Limit OC Threshold Set Current OC Comp Off-Set Voltage VRAMP IFB1 IFB2 VP TEST CONDITION SS=3V SS=0V Note 1 MIN TYP MAX UNITS -5 35 0.8 -0.1 55 +5 75 1.5 mA mA V mmho 210 400 1.25 240 460 KHz 50 50 100 90 100 100 700 Freq Tr Tf TDB DMAX DMIN Rt=100K Rt=50K Note 1 180 340 CLOAD =1500pF CLOAD =1500pF Fb=0.7V, Freq=200KHz Fb=0.9V Drv1 85 0 ns ns ns % % 40 0.784 -1 65 0.8 -0.1 150 0.816 +1 mA V mA 8C 20 -5 30 0 40 +5 mA mV Note 1 IOCSET VOC(OFFSET) VPP Note 1: Guaranteed by design but not tested in production. PIN DESCRIPTIONS PIN# 1 2 3 PIN SYMBOL PIN DESCRIPTION Fb2 These pins provide feedback for the linear regulator controllers. Drv2 Outputs of the linear regulator controllers. Rt A resistor should be connected from this pin to ground for setting the switching frequency. 4 SS / SD 5 Comp 6 Fb1 7 8 9 VP1 VREF Vcc 10 VcL 11 LDrv This pin provides soft-start for the switching regulator. An internal current source charges an external capacitor that is connected from this pin to ground which ramps up the output of the switching regulator, preventing it from overshooting as well as limiting the input current. The converter can be shutdown by pulling this pin down below 0.4V. Compensation pin of the error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. This pin is connected directly to the output of the switching regulator via resistor divider to provide feedback to the Error amplifier. Non-inverting input of error amplifier. Reference voltage. This pin provides biasing for the internal blocks of the IC as well as powers the LDO controller. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. This pin powers the low side output driver and can be connected either to Vcc or separate supply. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Output driver for the synchronous power MOSFET. 3/17 APU3073 PIN# 12 PIN SYMBOL PIN DESCRIPTION PGnd This pin serves as the separate ground for MOSFET's driver and should be connected to system's ground plane. Gnd This pin serves as analog ground for internal reference and control circuitry. A high frequency capacitor must be connected from Vcc pin to this pin for noise free operation. HDrv Output driver for the high side power MOSFET. This pin should not go negative (below ground), this may cause problem for the gate drive circuit. It can happen when the inductor current goes negative (Source/Sink), soft-start at no load and for the fast load transient from full load to no load. To prevent negative voltage at gate drive, a low forward voltage drop diode might be connected between this pin and ground. VcH This pin is connected to a voltage that must be at least 4V higher than the bus voltage of the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. OCSet This pin is connected to the Drain of the lower MOSFET via an external resister and it provides the positive sensing for the internal current sensing circuitry. The external resistor programs the current limit threshold depending on the RDS(ON) of the power MOSFET. An external capacitor can be placed in parallel with the programming resistor to provide high frequency noise filtering. 13 14 15 16 BLOCK DIAGRAM Vcc 9 Bias Generator 3V 1.25V 3V 0.8V 8 VREF 1.25V 20uA POR Vcc VcH SS/SD 4 3V 3 Rt 15 VcH 3.5V / 3.3V Rt Ct Oscillator En 14 HDrv S Error Comp Comp 5 Fb1 6 UVLO 64uA Max POR VP1 7 4.2V / 4.0V 25K Q Error Amp R 10 VcL Reset Dom 25K 11 LDrv 0.4V 20uA CS Comp POR OCSet 16 12 PGnd FbLo Comp Vcc 0.8V TSD 2 Drv2 Fb2 1 13 Gnd Figure 2 - Simplified block diagram of the APU3073. 4/17 APU3073 THEORY OF OPERATION Introduction The APU3073 is designed for a two output application and it includes one synchronous buck controller and a linear regulator controller. The PWM section is a fixed frequency, voltage mode and consists of a precision reference voltage, an uncommitted error amplifier, an internal oscillator, a PWM comparator, an internal regulator, a comparator for current limit, gate drivers, soft-start and shutdown circuits (see Block Diagram). The output voltage of the synchronous converter is set and controlled by the output of the error amplifier; this is the amplified error signal from the sensed output voltage and the voltage on non-inverting input of error amplifier(V P). This voltage is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulses of variable duty-cycle, which drives the two N-channel external MOSFETs. The timing of the IC is provided through an internal oscillator circuit which uses on-chip capacitor. The oscillation frequency is programmable between 200KHz to 400KHz by using an external resistor. Figure 14 shows switching frequency vs. external resistor (Rt). Soft-Start The APU3073 has a programmable soft-start to control the output voltage rise and limit the current surge at the start-up. To ensure correct start-up, the soft-start sequence initiates when the input supplies rise above their threshold and generates the Power On Reset (POR) signal. Soft-start function operates by sourcing an internal current to charge an external capacitor to about 3V. Initially, the soft-start function clamps the E/A’s output of the PWM converter and disables the short circuit protection. During the power up of the buck converter, the output starts at zero and voltage at Fb1 is below 0.4V. The feedback UVLO is disabled during this time by injecting a current (64mA) into the Fb1. This generates a voltage about 1.6V (64mA325K) across the negative input of E/A and positive input of the feedback UVLO comparator (see Fig3). 3V 20uA HDrv SS/SD 64uA Max POR Comp 0.8V Fb1 25K Error Amp LDrv 25K 0.4V 64uA325K=1.6V When SS=0 POR Feeback UVLO Comp Figure 3 - APU3073 soft-start diagram. The magnitude of this current is inversely proportional to the voltage at soft-start pin. The 20mA current source starts to charge up the external capacitor. In the mean time, the soft-start voltage ramps up, the current flowing into Fb1 pin starts to decrease linearly and so does the voltage at the positive pin of feedback UVLO comparator and the voltage negative input of E/A. When the soft-start capacitor is around 1V, the current flowing into the Fb1 pin is approximately 32mA. The voltage at the positive input of the E/A is approximately: 32mA325K = 0.8V The E/A will start to operate and the output voltage starts to increase. As the soft-start capacitor voltage continues to go up, the current flowing into the Fb1 pin will keep decreasing. Because the voltage at pin of E/A is regulated to reference voltage 0.8V, the voltage at the Fb1 is: VFB1 = 0.8-25K3(Injected Current) 5/17 APU3073 The feedback voltage increases linearly as the injecting current goes down. The injecting current drops to zero when soft-start voltage is around 2V and the output voltage goes into steady state. As shown in Figure 4, the positive pin of feedback UVLO comparator is always higher than 0.4V, therefore, feedback UVLO is not functional during soft-start. Output of UVLO POR 3V ≅2V Soft-Start Voltage Current flowing into Fb1 pin Supply Voltage Under-Voltage Lockout The under-voltage lockout circuit assures that the MOSFET driver outputs, remain in the off state whenever the supply voltage drops below set parameters. Lockout occurs if Vcc or VcH fall below 4.0V and 3.3V respectively. Normal operation resumes once these voltages rise above the set values. ≅1V 0V 64uA 0uA Voltage at negative input ≅1.6V of Error Amp and Feedback UVLO comparator 0.8V 0.8V Voltage at Fb1 pin LDO Controller The LDO section is powered directly from Vcc. The output of LDO can be set as low as 0.8V and can be programmed to higher voltages by using two external resistors. 0V Figure 4 - Theoretical operation waveforms during Soft-Start. From this analysis, the output start-up time is defined as when soft-start capacitor voltage increases from 1V to 2V. The start-up time will be dependent on the size of the external soft-start capacitor and can be estimated by: Shutdown The PWM section can be shutdown by pulling the softstart pin below 0.4V. The control MOSFET turns off and the synchronous MOSFET turns on during shutdown. Over-Current Protection Over-current protection is achieved with a cycle by cycle scheme and it is performed by sensing current through the RDS(ON) of low side MOSFET. As shown in Figure 5, an external resistor (RSET) is connected between OCSet pin and the drain of low side MOSFET (Q2) and sets the current limit set point. The internal current source develops a voltage across RSET. When the low side switch is turned on, the inductor current flows through the Q2 and results a voltage which is given by: VOCSET = IOCSET3RSET-RDS(ON)3iL I OCSET APU3073 CSS = 20mA3TSTART/1V MOSFET Drivers The driver capabilities of both high and low side drivers are optimized to maintain fast switching transitions. They are sized to drive a MOSFET that can deliver up to 20A output current. The low side MOSFET diver is supplied directly by VCC while the high side driver is supplied by VC. An internal dead time control is implemented to prevent cross-conduction and allows the use of several kinds of MOSFETs. Q1 L1 OCSet RSET 20mA3TSTART/CSS = 2V-1V For a given start up time, the soft-start capacitor can be calculated as: ---(1) VOUT Q2 Osc Figure 5 - Diagram of the over current sensing. When voltage VOCSET is below zero, the current sensing comparator flips and disables the oscillator. The high side MOSFET is turned off and the low side MOSFET is turned on until the inductor current reduces to below current set value. The critical inductor current can be calculated by setting: VOCSET = IOCSET3RSET - RDS(ON)3IL = 0 ISET = IL(CRITICAL) = RSET3IOCSET RDS(ON) ---(2) 6/17 APU3073 If the over-current condition is temporary and goes away quickly, the APU3073 will resume its normal operation. If output is shorted or over-current condition persists, the output voltage will keep going down until it is below 0.4V. Then the output under-voltage lock out comparator goes high and turns off both MOSFETs. The operation waveforms are shown in Figure 6. Inductor i L(PEAK) iL(AVG) Current ISET=iL(VALLEY) Current Limit Comparator Output HDrv Feedback VREF voltage 0.4V Switching frequency High Side MOSFET turn on time (t ON) tOFF tON FS(NOM) IOUT IOUT DMAX/FS(NOM) VOUT FS(NOM)3 VIN IOUT <IL>=IOUT Normal operation From Figure 7, the average inductor current during the current limit mode is: DIPK-PK(LIM) IO(LIM) = ISET + ---(4) 2 The inductor's ripple current can be expressed as: DIPK-PK(LIM) = (V IN - VOUT)3VOUT VIN3L3fS Combination of above equation and (4) results in: Average Inductor Current IO(LIM) Figure 7 - Operation waveforms during current limit. IO(MAX) IOUT Over Current Shutdown Limit Mode by UVLO ISET = IO(LIM) - RSET = During over-current mode, the valley inductor current is: iL(VALLEY) = ISET IN OUT OUT S IN ---(5) Combination of equations (5) and (2) results in the relationship between RSET and output current limit: Figure 6 - Diagram of over-current operation. Operation in current limit is shown in Figure 7, the high side MOSFET is turned off and inductor current starts to decrease. Because the output inductor current is higher than the current limit setpoint (ISET), the over-current comparator keeps high until the inductor current decreases to be below ISET. Then another cycle starts. )3V ((V23f-V 3L3V ) [ ( )] RDS(ON) (V IN-V OUT)3VOUT 3 IO(LIM) IOCSET 23fS3L3VIN ---(6) Where: IO(LIM) = The Output Current Limit -typical is 50% higher than nominal output current. VIN = Maximum Input Voltage VOUT = Output Voltage fS = Switching Frequency L = Output Inductor RDS(ON) = RDS(ON) of Low Side MOSFET IOCSET = OC Threshold Set Current The peak inductor current is given as: IL(PEAK) = ISET+(V IN-V OUT)3tON/L ---(3) To avoid undesirable trigger of over-current protection, this relationship must be satisfied: ISET / IO(NOM) - DIPK-PK(NOM) 2 From the above analysis, the current limit is not only dependent on the current setting resistor RSET and RDS(ON) of low side MOSFET but it is also dependent on the input voltage, output voltage, inductance and switching frequency as well. The cycle-by-cycle over-current limit will hold for a certain amount of time, until the output voltage drops below 0.4V, the under-voltage lock out activates and latches off the output driver. The operation waveform is shown in Figure 4. Normal operation will resume after APU3073 is powered up again. 7/17 APU3073 APPLICATION INFORMATION Design Example: The following example is a typical application for APU3073, the schematic is Figure 17 on page 16. Supply Voltage VCC=VCL=VCH=12V Switcher VIN = 5V VOUT = 2.5V IOUT = 8A DVOUT = 50mV fS = 200KHz Linear Regulator VIN = 2.5V VOUT = 1.6V IOUT = 2A Output Voltage Programming Output voltage is programmed by reference voltage and external voltage divider. The Fb pin is the inverting input of the error amplifier, which is referenced to the voltage on non-inverting pin of error amplifier. For this application, this pin (V P) is connected to reference voltage (V REF). The output voltage is defined by using the following equation: R6 VOUT = VP 3 1 + ---(7) R5 ( ) VP = VREF = 0.8V When an external resistor divider is connected to the output as shown in Figure 8. VOUT APU3073 VREF R6 Fb R5 VP Figure 8 - Typical application of the APU3039 for programming the output voltage. Equation (7) can be rewritten as: R6 = R5 3 ( VV OUT P ) -1 Choose R5 = 1K. This will result to R6 = 2.15K If the high value feedback resistors are used, the input bias current of the Fb pin could cause a slight increase in output voltage. The output voltage set point can be more accurate by using precision resistor. Soft-Start Programming The soft-start timing can be programmed by selecting the soft-start capacitance value. The start-up time of the converter can be calculated by using: Css ≅ 203tSTART (mF) Where ---(8) tSTART is the desirable start-up time (s) For a start-up time of 5ms, the soft-start capacitor will be 0.1mF. Choose a ceramic capacitor at 0.1mF. Supply VcL and VcH To drive the high side switch, it is necessary to supply a gate voltage at least 4V greater than the Bus voltage. For this application, VcL and VcH are biased with a separate 12V supply. Input Capacitor Selection The input filter capacitor should be based on how much ripple the supply can tolerate on the DC input line. The ripple current generated during the on time of upper MOSFET should be provided by input capacitor. The RMS value of this ripple is expressed by: IRMS = IOUT D3(1-D) ---(9) Where: D is the Duty Cycle, D=VOUT/VIN. IRMS is the RMS value of the input capacitor current. IOUT is the output current for each channel. For VIN=5V, IOUT=8A and D=0.5, the IRMS=4A For higher efficiency, a low ESR capacitor is recommended. Choose two Poscap from Sanyo 6TPB47M (16V, 47mF) with a max allowable ripple current of 5.2A. Inductor Selection The inductor is selected based on operating frequency, transient performance and allowable output voltage ripple. Low inductor value results to faster response to step load (high di/dt) and smaller size but will cause larger output ripple due to increase of inductor ripple current. As a rule of thumb, select an inductor that produces a ripple current of 10-40% of full load DC. For the buck converter, the inductor value for desired operating ripple current can be determined using the following relation: Di 1 VOUT ; Dt = D3 ;D= Dt fS VIN VOUT L = (V IN - VOUT)3 ---(11) VIN3Di3fS Where: VIN = Maximum Input Voltage VOUT = Output Voltage ∆i = Inductor Ripple Current fS = Switching Frequency ∆t = Turn On Time D = Duty Cycle VIN - VOUT = L3 8/17 APU3073 If Di = 25%(IO), then the output inductor will be: L = 3.125mH The Coilcraft DO5022HC series provides a range of inductors in different values, low profile suitable for large currents. 3.3mH is a good choice for this application. This will result to a ripple approximately 23% of output current. The total power dissipation for MOSFETs includes conduction and switching losses. For the Buck converter, the average inductor current is equal to the DC load current. The conduction loss is defined as: 2 PCOND(Upper Switch) = ILOAD 3RDS(ON)3D3q 2 PCOND(Lower Switch) = ILOAD 3RDS(ON)3(1 - D)3q q = RDS(ON) Temperature Dependency Output Capacitor Selection The criteria to select the output capacitor is normally based on the value of the Effective Series Resistance (ESR). In general, the output capacitor must have low enough ESR to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The ESR of the output capacitor is calculated by the following relationship: DVO ESR [ ---(10) DIO Where: DVO = Output Voltage Ripple Di = Inductor Ripple Current DVO = 50mV and DI ≅ 23% of 8A = 1.89A This results to: ESR=26.5mV The Sanyo TPC series, Poscap capacitor is a good choice. The 6TPC330M, 330mF, 6.3V has an ESR 40mV. Selecting two of these capacitors in parallel, results to an ESR of ≅ 20mV which achieves our low ESR goal. The capacitor value must be high enough to absorb the inductor's ripple current. The larger the value of capacitor, the lower will be the output ripple voltage. Power MOSFET Selection The APU3073 uses two N-Channel MOSFETs. The selections criteria to meet power transfer requirements is based on maximum drain-source voltage (V DSS), gatesource drive voltage (V GS), maximum output current, Onresistance RDS(ON) and thermal management. The MOSFET must have a maximum operating voltage (V DSS) exceeding the maximum input voltage (V IN). The gate drive requirement is almost the same for both MOSFETs. Logic-level transistor can be used and caution should be taken with devices at very low VGS to prevent undesired turn-on of the complementary MOSFET, which results a shoot-through current. The RDS(ON) temperature dependency should be considered for the worst case operation. This is typically given in the MOSFET data sheet. Ensure that the conduction losses and switching losses do not exceed the package ratings or violate the overall thermal budget. Choose IRF7832 for both control MOSFET and synchronous MOSFET. This device provides low on-resistance in a compact SOIC 8-Pin package. The MOSFETs have the following data: IRF7832 VDSS = 30V ID = 16A @ 708C RDS(ON) = 4mV The total conduction losses will be: PCON(TOTAL) = PCON(UPPER) + PCON(LOWER) PCON(TOTAL) = 0.38W The switching loss is more difficult to calculate, even though the switching transition is well understood. The reason is the effect of the parasitic components and switching times during the switching procedures such as turn-on / turnoff delays and rise and fall times. The control MOSFET contributes to the majority of the switching losses in synchronous Buck converter. The synchronous MOSFET turns on under zero voltage conditions, therefore, the turn on losses for synchronous MOSFET can be neglected. With a linear approximation, the total switching loss can be expressed as: VDS(OFF) tr + tf 3 3 ILOAD ---(12) 2 T Where: VDS(OFF) = Drain to Source Voltage at off time tr = Rise Time tf = Fall Time T = Switching Period ILOAD = Load Current PSW = The switching time waveform is shown in Figure 9. 9/17 APU3073 VDS 90% The output LC filter introduces a double pole, –40dB/ decade gain slope above its corner resonant frequency, and a total phase lag of 1808 (see Figure 10). The Resonant frequency of the LC filter is expressed as follows: 1 FLC = ---(13) 2p3 LO3CO 10% VGS td(ON) tr td(OFF) tf Figure 9 - Switching time waveforms. From IRF7832 data sheet we obtain: Figure 10 shows gain and phase of the LC filter. Since we already have 1808 phase shift just from the output filter, the system risks being unstable. Gain Phase 08 0dB -40dB/decade IRF7832 tr = 12.3ns tf = 21ns These values are taken under a certain condition test. For more details please refer to the IRF7832 datasheet. FLC Frequency -1808 F LC Frequency Figure 10 - Gain and phase of LC filter. By using equation (12), we can calculate the total switching losses. PSW(TOTAL) = 133mW Programming the Over-Current Limit The over-current threshold can be set by connecting a resistor (RSET) from drain of low side MOSFET to the OCSet pin. The resistor can be calculated by using equation (2). The RDS(ON) has a positive temperature coefficient and it should be considered for the worse case operation. RDS(ON) = 4mV31.5 = 6mV ISET ≅ IO(LIM) = 8A31.5 = 12A (50% over nominal output current) This results to: RSET ≅ 4.8KV Select: RSET = 5KV Feedback Compensation The APU3073 is a voltage mode controller; the control loop is a single voltage feedback path including error amplifier and error comparator. To achieve fast transient response and accurate output regulation, a compensation circuit is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin (greater than 458). The APU3073’s error amplifier is a differential-input transconductance amplifier. The output is available for DC gain control or AC phase compensation. The E/A can be compensated with or without the use of local feedback. When operated without local feedback, the transconductance properties of the E/A become evident and can be used to cancel one of the output filter poles. This will be accomplished with a series RC circuit from Comp pin to ground as shown in Figure 11. Note that this method requires that the output capacitor should have enough ESR to satisfy stability requirements. In general, the output capacitor’s ESR generates a zero typically at 5KHz to 50KHz which is essential for an acceptable phase margin. The ESR zero of the output capacitor expressed as follows: FESR = 1 2p3ESR3Co ---(14) 10/17 APU3073 VOUT R6 For: VIN = 5V VOSC = 1.25V Fo = 20KHz FESR = 12KHz Fb R5 Vp=VREF E/A Comp Ve C9 R4 FLC = 3.41KHz R5 = 1K R6 = 2.15K gm = 700mmho This results to R4=23.14K Choose R4=24K CPOLE Gain(dB) To cancel one of the LC filter poles, place the zero before the LC filter resonant frequency pole: H(s) dB FZ ≅ 75%FLC Frequency FZ Figure 11 - Compensation network without local feedback and its asymptotic gain plot. FZ ≅ 0.753 1 LO 3 CO 2p For: Lo = 3.3mH Co = 660mF ---(19) FZ = 2.5KHz R4 = 24K The transfer function (Ve / VOUT) is given by: ( H(s) = gm3 R5 1 + sR4C9 3 R6 + R5 sC9 ) Using equations (17) and (19) to calculate C9, we get: ---(15) The (s) indicates that the transfer function varies as a function of frequency. This configuration introduces a gain and zero, expressed by: R5 |H(s=j32p3FO)| = gm3 3R4 ---(16) R63R5 FZ = 1 2p3R43C9 First select the desired zero-crossover frequency (Fo): Fo > FESR and FO [ (1/5 ~ 1/10)3fS 1 FP = ---(18) Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Fo = Crossover Frequency FESR = Zero Frequency of the Output Capacitor FLC = Resonant Frequency of the Output Filter R5 and R6 = Resistor Dividers for Output Voltage Programming gm = Error Amplifier Transconductance C93CPOLE C9 + CPOLE The pole sets to one half of switching frequency which results in the capacitor CPOLE: CPOLE = Use the following equation to calculate R4: 1 VOSC Fo3FESR R5 + R6 3 3 3 2 g m VIN FLC R5 One more capacitor is sometimes added in parallel with C9 and R4. This introduces one more pole which is mainly used to suppress the switching noise. The additional pole is given by: 2p3R43 ---(17) |H(s)| is the gain at zero cross frequency. R4 = C9 ≅ 2590pF; Choose C9 =2200pF 1 ≅ 1 p3R43fS p3R43fS - 1 C9 fS for FP << 2 For a general solution for unconditionally stability for ceramic capacitor with very low ESR and any type of output capacitors, in a wide range of ESR values we should implement local feedback with a compensation network. The typically used compensation network for voltage-mode controller is shown in Figure 12. 11/17 APU3073 VOUT ZIN C10 R7 R8 FZ1 = C12 C11 R6 Zf 1 2p3R73C11 1 1 FZ2 = 2p3C103(R6 + R8) ≅ 2p3C103R6 Cross Over Frequency: Fb E/A R5 Comp Ve Gain(dB) H(s) dB FZ2 FP2 FP3 Frequency Figure 12 - Compensation network with local feedback and its asymptotic gain plot. In such configuration, the transfer function is given by: Ve 1 - gmZf = VOUT 1 + gmZIN The error amplifier gain is independent of the transconductance under the following condition: gmZf >> 1 and gmZIN >>1 ---(20) By replacing ZIN and Zf according to Figure 7, the transformer function can be expressed as: H(s) = (1+sR7C11)3[1+sC10(R6+R8)] 1 3 sR6(C12+C11) C12C11 1+sR7 C12+C11 3(1+sR8C10) [ ( )] As known, transconductance amplifier has high impedance (current source) output, therefore, consider should be taken when loading the E/A output. It may exceed its source/sink output current capability, so that the amplifier will not be able to swing its output voltage over the necessary range. The compensation network has three poles and two zeros and they are expressed as follows: FP1 = 0 FP2 = FP3 = 1 2p3R83C10 ---(21) The stability requirement will be satisfied by placing the poles and zeros of the compensation network according to following design rules. The consideration has been taken to satisfy condition (20) regarding transconductance error amplifier. These design rules will give a crossover frequency approximately one-tenth of the switching frequency. The higher the band width, the potentially faster the load transient speed. The gain margin will be large enough to provide high DC-regulation accuracy (typically -5dB to 12dB). The phase margin should be greater than 458 for overall stability. Based on the frequency of the zero generated by ESR versus crossover frequency, the compensation type can be different. The table below shows the compensation type and location of crossover frequency. Compensator Location of Zero Typical Type Crossover Frequency Output (FO) Capacitor Type II (PI) FPO < FZO < FO < fS/2 Electrolytic, Tantalum Type III (PID) FPO < FO < FZO < fS/2 Tantalum, Method A Ceramic Type III (PID) FPO < FO < fS/2 < FZO Ceramic Method B Table - The compensation type and location of zero crossover frequency. Detail information is dicussed in application Note AN1043 which can be downloaded from the IR Web-Site. 1 ( CC 3C +C ) 2p3R73 VIN 1 3 VOSC 2p3Lo3Co Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Lo = Output Inductor Co = Total Output Capacitors Vp=VREF FZ1 FO = R73C103 12 12 11 ≅ 1 2p3R73C12 11 12/17 APU3073 LDO Section Output Voltage Programming Output voltage for LDO is programmed by reference voltage and external voltage divider. The Fb2 pin is the inverting input of the error amplifier, which is internally referenced to 0.8V. The divider is ratioed to provide 0.8V at the Fb2 pin when the output is at its desired value. The output voltage is defined by using the following equation ( VOUT2 = VREF3 1+ R7 R10 ) For: VOUT2 = 1.6V VREF = 0.8V R10 = 1KV Results to R7=1KV VOUT2 APU3073 R7 Fb2 R10 Layout Consideration The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results. Start to place the power components. Make all the connections in the top layer with wide, copper filled areas. The inductor, output capacitor and the MOSFET should be close to each other as possible. This helps to reduce the EMI radiated by the power traces due to the high switching currents through them. Place input capacitor directly to the drain of the high-side MOSFET. To reduce the ESR, replace the single input capacitor with two parallel units. The feedback part of the system should be kept away from the inductor and other noise sources and be placed close to the IC. In multilayer PCB, use one layer as power ground plane and have a separate control circuit ground (analog ground), to which all signals are referenced. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. These two grounds must be connected together on the PC board layout at a single point. 500 Figure 13 - Programming the output voltage for LDO. 450 RDS(ON) = VIN(LDO) - VOUT2 IOUT2 For: VIN(LDO) = 2.5V VOUT2 = 1.6V IOUT2 = 2A Frequency (KHz) 400 LDO Power MOSFET Selection The first step in selecting the power MOSFET for the linear regulator is to select the maximum RDS(ON) based on the input to the dropout voltage and the maximum load current. 350 300 250 200 150 100 50 0 0 50 100 150 200 250 300 350 400 450 500 550 Rt (K V) Figure 14 - Switching Frequency vs. Rt. Results to: RDS(ON)(MAX) = 0.45V Note that since the MOSFET RDS(ON) increases with temperature, this number must be divided by ~1.5 in order to find the RDS(ON)(MAX) at room temperature. The IRLR2703 has a maximum of 0.065V RDS(ON) at room temperature, which meets our requirements. 13/17 APU3073 TYPICAL APPLICATION 2.5V Vcc Q3 IRLR2703 C13 150uF C16 1uF Drv2 VcL 1.6V @ 2A R2 U1 VcH APU3073 VP1 C10 0.1uF R7 24K D2 BAT54 VREF C2 33pF C7 2200pF C11 1uF HDrv OCSet Comp LDrv Gnd C1 47uF L2 R4 3.3uH 5.1K Q2 IRF7832 SS/SD Fb1 PGnd +5V C3 0.1uF Q1 IRF7832 Rt C6 0.1uF L1 1uH D3 BAT54 C19 1uF Fb2 1K C14 R14 150uF 1K C2A,B,C=47uF C9B C9C C12 330uF 330uF 1uF 2.5V @ 8A R9 R10 1K 2.15K Figure 15 - Typical application of APU3073 for single 5V. 14/17 APU3073 TYPICAL APPLICATION 3.3V VcH Q3 IRLR2703 C13 150uF 12V C11 1uF Drv2 Vcc 1.6V @ 1A R2 1K C14 R14 150uF 1K 24K Fb2 VcL U1 APU3073 VP1 C10 0.1uF R7 C16 1uF HDrv D2 BAT54 VREF C2 33pF C7 2200pF C19 1uF OCSet Comp LDrv C2A,B,C=47uF 1uH L2 R4 3.3uH 5.1K Q2 IRF7832 SS/SD Gnd Fb1 PGnd +5V C1 47uF Q1 IRF7832 Rt C6 0.1uF L1 C9B C9C C12 330uF 330uF 1uF 2.5V @ 8A R9 R10 1K 2.15K Figure 16 - Typical application of APU3073. 15/17 APU3073 APPLICATION EXPERIMENTAL WAVEFORMS Figure 18 - Normal condition at no load. Ch1: HDrv Ch2: LDrv Ch4: Inductor Current Figure 19 - Gate signals when SS pin pulls low. Ch1: HDrv Ch2: LDrv Figure 20 - Soft-Start. Ch1: VIN (5V) Ch2: Bias Voltage (12V) Ch3: VOUT1 (PWM) Ch4: VOUT2 (LDO) 16/17 APU3073 APPLICATION EXPERIMENTAL WAVEFORMS Figure 21 - Output Shorted at start-up. Ch1: VOUT Ch4: IOUT Figure 22 - Load Transient Response (PWM Section). Ch1: VOUT1 Ch4: IOUT1 (0-8A) Figure 23 - Load Transient Response (LDO Section). Ch2: VOUT2 Ch4: IOUT2 (0-2A) 17/17