IRF IR3622MPBF

Data Sheet No.PD94722 revA
IR3622MPbF
HIGH FREQUENCY 2-PHASE, SINGLE OR DUAL OUTPUT SYNCHRONOUS STEP
DOWN CONTROLLER WITH OUTPUT TRACKING AND SEQUENCING
Description
Features
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Dual Synchronous Controller with 180o
Out of Phase Operation
Configurable to 2-Independent Outputs or Current
Shared Single Output
Output Voltage Tracking
Power up / down Sequencing
Current Sharing Using Inductor’s DCR
+/-1% Accurate Reference Voltage
Programmable Switching Frequency up 600kHz
Programmable Over Current Protection
Hiccup Current Limit Using MOSFET Rds(on) sensing
Latched Overvoltage Protection
Dual Programmable Soft-Starts
Programmable Enable Input
Pre-Bias Start-up
Dual Power Good Outputs
On Board Regulator
External Frequency Synchronization
Thermal Protection
32-Lead MLPQ Package
Applications
•
•
•
•
•
Embedded Telecom Systems
Distributed Point of Load Power Architectures
Computing Peripheral Voltage Regulators
Graphics Cards
General DC/DC Converters
The IR3622 IC integrates a dual synchronous
Buck controller, providing a high performance and
flexible solution. The IR3622 can be configured
as 2-independent outputs or as current shared
single output. The current share configuration is
ideal for high current applications.
The IR3622 enables output tracking and
sequencing of multiple rails in either ratiometric or
simultaneous fashion. The IR3622 features 180o
out of phase operation which reduces the
required input/output capacitance and results in
lower number of capacitors. The switching
frequency is programmable from 200kHz to
600kHz per phase using one external resistor. In
addition, IR3622 also allows the switching
frequency to be synchronized to an external clock
signal.
Other key features offered by this device include
two independent programmable soft starts, two
independent power good outputs, precision
enable input, and under voltage lockout function.
The current limit is provided by sensing the lower
MOSFET's on-resistance for optimum cost and
performance. The output voltages are monitored
through dedicated pins to protect against open
circuit, and enhance faster response to an
overvoltage event.
Vin
Rt
IR3622
Comp2
Vo1
Vo2
Vo2
HDrv1
OCSet1
Comp1
Vo1
Vout1
LDrv1
Ratiometric Powerup
PGnd1
Vin
SS1 / SD
Ratiometric Powerdown
Vo1
Vo1
Vo2
Vo2
SS2 / SD
Vout1
HDrv2
Seq
Track
OCSet2
Gnd
PGnd2
Vout2
LDrv2
Simultaneous Powerup
Simultaneous Powerdown
ORDERING INFORMATION
PKG
DESIG
M
M
www.irf.com
PACKAGE
DESCRIPTION
IR3622MPbF
IR3622MTRPbF
PIN
PARTS
PARTS
COUNT PER TUBE PER REEL
32
73
------32
-------3000
T&R
ORIANTAION
Fig A
03/15/07
IR3622MPbF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND)
Vcc, VcL Supply Voltage .....................................… -0.5V to 16V
VcH1,VcH2 ……………………….………….……… -0.5V to 30V
PGood1, PGood2 ………. .…………………………. -0.5V to 16V
HDrv1, HDrv2 ………………………………………… -0.5V to 30V (-2V for 100ns)
LDrv1, LDrv2 ………………………………………… -0.5V to 16V (-2V for 100ns)
Gnd to PGnd ……………………………………….. +/- 0.3V
Storage Temperature Range .................................. -65°C To 150°C
Operating Junction Temperature Range ................ -40°C To 125°C
ESD Classification ………………………………..… JEDEC, JESD22-A114 (1KV)
Moisture Sensitivity Level ………………………….. JEDEC, Level 3 @ 260oC
Caution: Stresses above those listed in “Absolute Maximum Rating” may cause permanent damage to the
device. These are stress ratings only and function of the device at these or any other conditions beyond those
indicated in the operational sections of the specifications is not implied. Exposure to “Absolute Maximum Rating”
conditions for extended periods may affect device reliability.
VP
28 27
26
25
1
VP
29
2
PG
oo
d1
Gn
d
VR
30
31
EF
Vc
c
32
UT
3
Tra
ck
VO
Package Information
Rt
1
24 PGood2
VSEN2
2
23 Sync
3
Comp2 4
22 VSEN1
Fb2
SS2/SD2/Mode
21 Fb1
Pad
5
20 Comp1
OCSet2 6
19 SS1/SD1
VcH2
7
18 OCSet1
HDrv2
8
17
12
VcL
10
13
14
15
16
LD
rv1
PG
nd
1
Se
q
HD
rv1
11
En
ab
le
PG
nd
2
LD
rv2
9
VcH1
ΘJA = 36o C/W*
ΘJC = 1o C/W
*Exposed pad on underside is connected to a copper
pad through vias for 4-layer PCB board design
www.irf.com
2
IR3622MPbF
Recommended Operating Conditions
Symbol
Definition
Vcc, VcL
VcH1, VcH2
Fs
Tj
Supply Voltage
Supply Voltage
Operating frequency
Junction temperature
Min
Max
Units
7.5
Converter Voltage + 5V
200
-40
14.5
28
600
125
V
V
kHz
o
C
Electrical Specifications
Unless otherwise specified, these specification apply over Vcc=VcL=VcH1=VcH2=12V, 0oC<Tj<105oC
Parameter
SYM
Test Condition
Min
TYP
MAX
Units
-1
+1
%
-1.5
+1.5
%
Output Voltage Accuracy
FB1,2 Voltage
VFB
Vp1=Vp2=Vref=0.8V
o
Accuracy
o
0 C <Tj< 125 C
o
o
-40 C <Tj< 125 C; Note2
0.8
V
Supply Current
VCC Supply Current
(Static)
VCC Supply Current
(Dynamic)
ICC (Static)
ICC (Dynamic)
VCL Supply Current
(Static)
ICL (Static)
VCL Supply Current
(Dynamic)
ICL (Dynamic)
VCH1,2 Supply Current
(Static)
ICH (Static)
VCH1,2 Supply Current
(Dynamic)
ICH (Dynamic
SS=0V, No Switching
13
18
mA
Fs=300kHz, CLOAD= 3.3nF
20
30
mA
SS=0V, No Switching
8
10
mA
Fs=300kHz, CLOAD= 3.3nF
30
42
mA
SS=0V, No Switching
9
11
mA
Fs=300kHz, CLOAD= 3.3nF
30
42
mA
V
Under Voltage Lockout, Enable
VCC-Start Threshold
VCC_UVLO(R)
Supply ramping up
6.85
7.1
7.5
VCC-Stop Threshold
VCC_UVLO(F)
Supply ramping down
6.1
6.4
6.7
V
Supply ramping up and down
0.5
0.8
1.0
V
4.6
4.95
5.3
V
VCC-Hysteresis
VCH1,2-Start Threshold
VCH_UVLO(R)
Supply ramping up
VCH1,2-Stop Threshold
VCH_UVLO(F)
Supply ramping down
4.1
4.4
4.7
V
Supply ramping up and down
0.32
0.6
0.8
V
Supply ramping up
1.14
1.24
1.34
V
Supply ramping up and down
0.15
0.2
0.33
V
200
600
kHz
-12
+12
%
VCH1,2-Hysteresis
Enable-Threshold
En_UVLO
Enable-Hysteresis
Oscillator
Frequency Range
FS
Accuracy
Ramp Amplitude
Fs=300kHz
Vramp
Min Duty Cycle
Dmin
Min Pulse Width
Ton(min)
Max duty Cycle
Dmax
Sync Frequency Range
Sync Pulse Duration
Sync High Level Threshold
Sync Low Level Threshold
www.irf.com
Sync(F)
Note1
1.25
Fb=1V
FS=300kHz, Note1
FS=300kHz, Fb=0.6V
V
0
%
150
ns
84
%
20% above free running Freq
1200
Sync(Pulse)
200
Sync(H)
Sync(L)
2
300
kHz
ns
0.6
V
V
3
IR3622MPbF
Electrical Specifications
Parameter
SYM
Test Condition
Min
TYP
MAX
Units
-0.1
-0.5
µA
200
280
µA
4500
µmho
+4
mV
Vcc-2
V
7.7
V
Error Amplifier 1, 2
Fb Voltage Input
Bias Current
E/A Source/Sink
Current
Transconductance
IFB
SS=3V
I(source/Sink)
120
gm1,2
Input offset
Voltage
VP Voltage Range
Voffset
VP
3000
Fb to Vref
-4
Note1
0.4
0
Internal Regulator
Output Accuracy
Vout3
Dropout
Vdrop
Current Limit
6.7
7.2
Vcc(min)=9V, Isource=100mA
Ishort
2
110
V
mA
Soft Start/SD
Soft Start Current
ISS
Shutdown
Threshold
SD
Source/Sink
18
23
28
µA
0.25
V
Over Current Protection
OCSET Current
Hiccup Duty Cycle
IOCSET
Hiccup(duty)
16
20
5
24
µA
%
1.1Vref
1.15Vref
1.2Vref
V
5
µs
Ihiccup / Iocset, Note1
Over Voltage Protection
OVP Trip Threshold
OVP Fault Prop
Delay
OVP(trip)
OVP(delay)
Output Forced to 1.25Vref
Thermal Shutdown
Note1
Thermal shutdown
Thermal shutdown
Hysteresis
o
140
20
Note1
o
C
C
Power Good
Vsen Lower Trip
point
PGood Output Low
Voltage
Vsen(trip)
PG(voltage)
Vsen Ramping Down
0.8Vref
IPGood=2mA
0.9Vref
0.95Vref
V
0.1
0.5
V
Output Drivers
LO, Drive Rise Time
Tr(Lo)
CLOAD=3.3nF, Fs=300KHz, 2V to 9V
25
50
ns
LO Drive Fall Time
Tf(Lo)
CLOAD =3.3nF, Fs=300KHz, 9V to 2V
25
50
ns
HI Drive Rise Time
Tr(Hi)
25
50
ns
HI Drive Fall Time
Tf(Hi)
CLOAD =3.3nF, Fs=300KHz, 9V to 2V
25
50
ns
Dead Band Time
Tdead
See Figure1
20
60
100
ns
On
Off
2.0
CLOAD =3.3nF, Fs=300KHz, 2V to 9V
Seq Input
Seq Threshold
Seq
V
0.3
Tracking
Track voltage range
www.irf.com
TK
Note1
0
Vcc
V
4
IR3622MPbF
Note1: Guaranteed by design but not tested in production.
Note2: Cold temperature performance is guaranteed via correlation using statistical quality
control. Not tested in production.
Tr
Tf
9V
High Side Driver
(HDrv)
2V
Tr
Tf
9V
Low Side Driver
(LDrv)
2V
Deadband
H_to_L
Deadband
L_to_H
Fig. 1: Rise / Fall and deadband time for driver section
www.irf.com
5
IR3622MPbF
Pin#
Pin Name
Description
1
Rt
2
VSEN2
Connecting a resistor from this pin to ground sets the switching frequency
(see figure 16 on page 17 for selecting resistor value)
Sense pin for OVP2 and Power Good2, Channel 2
3
Fb2
Inverting input to the error amplifier2
4
5
Comp2
SS2/SD2/Mode
6
OCSet2
Compensation pin for the error amplifier2
Soft start for channel 2, can be used as SD pin. Float this pin for current
share single output application
Current limit set point for channel2
7, 17
VcH2, VcH1
8,16
HDrv2, HDrv1
Supply voltage for the high side output drivers. These are connected to
voltage that must be typically 6V higher than their bus voltages. A 0.1uF
high frequency capacitor must be connected from these pins to PGND to
provide peak drive current capability
Output drivers for the high side power MOSFETs
Enable
Enable pin, recycling this pin will reset OV, SS and Prebias latch
10, 14
PGnd2, PGnd1
11, 13
LDrv2 , LDrv1
These pins serve as the separate grounds for MOSFET drivers and should
be connected to the system’s ground plane
Output drivers for the synchronous power MOSFETs
12
VcL
Supply voltage for the low side output drivers
15
Seq
18
OCSet1
Enable pin for tracking and sequencing. If this pin is not used
connect it to Vout3
Current limit set point for Channel 1
19
SS1/SD1
Soft start for Channel 1, can be used as SD pin
20
Comp1
Compensation pin for the error amplifier1
21
Fb1
Inverting input to the error amplifier1
22
VSEN1
Sense pin for OVP1 and Power Good1, Channel 1
23
Sync
External synchronization pin
24
PGood2
25
VP1
Power Good pin output for channel 2, open collector. This pin needs to be
externally pulled high
Non inverting input of error amplifier1
26
VP2
Non inverting input of error amplifier2
27
VREF
Reference Voltage
28
Gnd
IC’s Ground
29
PGood1
30
Vcc
31
Vout3
32
Track
Power Good pin output for Channel 1, open collector. This pin needs to be
externally pulled high
Supply voltage for the internal blocks of the IC. A 0.1uF high frequency
capacitor must be connected from this pin to Gnd.
Output of the internal regulator. A 0.1uF high frequency capacitor must be
connected from this pin to PGnd.
Sets the type of power up / down sequencing (ratiometric or
simultaneous). If this pin is not used connect it to Vout3
9
www.irf.com
6
IR3622MPbF
Block Diagram
0.3V
Enable
S
POR
SS1
Vcc
Q
23uA 23uA
Seq
Mode
SS2 / SD
3V
Bias
Generator
64uA
UVLO
SS1 / SD
VcH1
0.8V
64uA
PBias1
R
POR
HDrv1
VcH1
VcH2
POR
PWM Comp1
Thermal
Shutdown
Error Amp1
OVP1
LDrv1
3uA
R
VP1
VCL
SS1
PBias1
PGnd1
Q
Fb1
Set1
Ramp1
Comp1
20uA
Reset Dom
Two Phase
Oscillator
Rt
OCSet1
S
Set2
Ramp2
VcH2
Reset Dom
Sync
S
PWM Comp2
0.8V
VREF
Q
R
Error Amp2
Track
HDrv2
SS1
Hiccup
Control
SS2
Mode
OVP2
VP2
LDrv2
0.3V
Fb2
S
PBias2
Q
Comp2
SS2
POR
R
S
VSEN1
Q
1.15Vref
POR
PGnd2
SS2
OVP1
HDrv1 OFF / LDrv1 ON
OCSet2
3uA
20uA
R
PGood1
0.90Vref
SS1 / SD Vcc
Q
1.15Vref
POR
VOUT3
Tracking
Seq
23uA
S
VSEN2
Regulator
OVP2
HDrv2 OFF / LDrv2 ON
R
PGood2
0.90Vref
Gnd
Fig. 2: Simplified block diagram of the IR3622
www.irf.com
7
IR3622MPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC-125oC)
VFb2 vs Temperature
VFb1 vs Temperature
0.802
0.802
0.8015
0.8015
0.801
VFb2 (V)
VFb1 (V)
0.801
0.8005
0.8
0.7995
0.799
0.8005
0.8
0.7995
0.799
0.7985
0.7985
0.798
0.798
0.7975
0.7975
-40
-20
0
20
40
60
80
100
-40
120
-15
SS1 Current vs Temperature
60
85
110
SS2 Current vs Temperature
-19
-19
SS2 Current (uA)
SS1 Cu rren t (u A)
35
Tem perature (C)
Te m pe rature (C)
-20
-21
-22
-23
-24
-25
-40
-15
10
35
60
85
-20
-21
-22
-23
-24
-25
-40
110
-15
10
35
60
85
110
Te m pe rature (C)
Te m pe rature (C)
Vcc_UVLO vs Temperature
Vout3 vs Temperature
7.3
7.25
7.28
7.23
7.26
7.21
7.19
VOut3 (V)
Vcc_U VL O (V)
10
7.24
7.22
7.2
7.18
7.16
7.14
7.17
7.15
7.13
7.11
7.09
7.07
7.12
7.05
7.1
-40
-15
10
35
60
Te m pe rature (C)
www.irf.com
85
110
-40
-15
10
35
60
85
110
Tem perature (C)
8
IR3622MPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC-125oC)
IOCSET2 vs Temperature
IOCSET1 vs Temperature
22
IOCSET2 (uA)
IO CSET1 (u A)
22
21.5
21
20.5
20
19.5
21.5
21
20.5
20
19.5
19
18.5
18
19
-40
-15
10
35
60
85
-40
110
-15
GM1 vs Temperature
60
85
110
85
110
GM2 vs Temperature
4000
4100
3900
4000
GM2 (umho)
GM1 (umho)
35
Tem perature (C)
Te m pe rature (C)
3800
3700
3600
3500
3900
3800
3700
3600
3500
3400
3400
-40
-15
10
35
60
85
-40
110
-15
Freq 300KHz vs Temperature
300
290
280
10
35
60
Tem perature (C)
www.irf.com
85
110
M ax D u ty C ycle (% )
310
-15
35
60
Max Duty Cycle vs Temperature
320
-40
10
Tem perature (C)
Tem perature (C)
F re q (K H z )
10
96
94
92
90
88
86
84
-40
-15
10
35
60
85
110
Te m pe rature
9
IR3622MPbF
Circuit Description
THEORY OF OPERATION
Introduction
The IR3622 is a versatile device for high
performance buck converters. It consists of two
synchronous buck controllers which can be
operated either in two independent outputs mode
or in current share single output mode for high
current applications.
The timing of the IC is provided by an internal
oscillator circuit which generates two 180o-out-ofphase clock signals that can be externally
programmed up to 600kHz per phase.
Programmable Enable Input
The enable features another level of flexibility for
start up. The Enable has precise threshold which
is internally monitored by under-voltage lockout
circuit.
It’s threshold can be externally programmed to
desired level by using two external resistors, so
the converter doesn’t start up until the input
voltage is sufficiently high (see figure 3).
Under-Voltage Lockout
The under-voltage lockout circuit monitors four
signals (Vcc, VcH1, VcH2 and Enable). This
ensures the correct operation of the converter
during power up and power down sequence. The
driver outputs remain in the off state whenever
one of these signals drop below set thresholds.
Normal operation resumes once these signals
rise above the set values. Figure 3 shows a
typical start up sequence.
12V
11V
12V
7.2V
7.2V
Vbus
Vcc
Vout3
Seq
Enable OK (IC's POR)
3V
Enable
SS
Fig. 3: Normal Start up, Enable threshold is externally set to 11V
Seq pin is pulled to Vout3 prior to start up
www.irf.com
10
IR3622MPbF
In addition, the 180o out of phase contributes to
input current cancellation. This results in much
smaller input capacitor’s RMS current and
reduces the number of required input capacitors.
Figure 5 shows the equivalent RMS current.
Internal Regulator
RMS Current Normalized (IRMS/Iout)
The IR3622 features an on-board 7.2V regulator
with short circuit protection. The regulator is
capable of sourcing current up to 100mA. This
integrated regulator can be used to generate the
necessary bias voltage for drivers, an example of
how this can be used is shown in figure 23,
page26.
Out-of-Phase Operation
The IR3622 drives its two output stages 180o outof-phase. In current share mode single output,
the two inductor ripple currents cancel each other
and result in a reduction of the output current
ripple and yield a smaller output capacitor for the
same voltage ripple requirement. Figure 4 shows
two channels inductor current and the resulting
voltage ripple at the output.
Single Phase
2 Phase
Duty Cycle (Vo/Vin)
Fig. 5: Input RMS value vs. Duty Cycle
HDRV1
Mode Selection
0
DT
The IR3622 can operate as a dual output
independently regulated buck converter, or as a
2 phase single output buck converter (current
share mode). The SS2 pin is used for mode
selection. In current share mode this pin should
be floating. In the dual output mode, a soft start
capacitor must be connected from this pin to the
ground to program the start time for the second
output.
T
HDRV2
IL1
IL2
Independent Mode
Ic
Io
Fig. 4: Current ripple cancellation for output
www.irf.com
In this mode the IR3622 provides control to two
independent output power supplies with either
common or different input voltages. The output
voltage of each individual channel is set and
controlled by the output of the error amplifier,
which is the amplified error signal from the
sensed output voltage and the reference voltage.
The error amplifier output voltage is compared to
the ramp signal thus generating fixed frequency
pulses of variable duty-cycle, (PWM) which are
applied to the internal MOSEFT drivers. Figure
24 shows a typical schematic for such
application.
11
IR3622MPbF
Master Phase
Vin
Current Share Mode
IL1
This feature allows to connect both outputs
together to increase current handling capability of
the converter to support a common load. In the
current sharing mode, error amplifier 1 becomes
the master which regulates the common output
voltage and the error amplifier 2 performs the
current sharing function, figure 6 shows the
configuration of error amplifier 2.
In this mode, IR3622 makes sure the master
channel starts first followed by slave channel to
prevent any glitch during start up. This is done by
clamping the output of slave’s error amplifier until
the master channel generates the first PWM
signal.
L1
+
Q2
The IR3622 uses a lossless current sensing for
current share purposes. The inductor current is
sensed by connecting a series resistor and a
capacitor network in parallel with the inductor
and by measuring the voltage across the
capacitor. The measured voltage is proportional
to the inductor current. This is shown figure 6.
The voltage across the inductor’s DCR can be
expressed by:
R L1
V RL 1 ( s ) = (V in − V out ) *
- - - -(1 )
R L1 + sL 1
V RL 1 ( s ) = I L1 * R L1
- - - -( 2 )
VL1 (s) C1
+ VC1(s) VP2
VOUT
FB2
Vin
Q3
At no load condition the slave channel may be
kept off depending on the offset of the error
amplifier.
Lossless Inductor Current Sensing
R1
RL1
R2
L2
C2
RL2
Q4
Slave Phase
Fig. 6: Loss Less inductor current sensing
and current sharing
the sense circuit can be treated as if only a
sense resistor with the value RL1 was used.
If : R 1 * C 1 =
L1
R L1
VC ( s ) ≈ I L1 * R L1
The mismatch of the time constant does not
affect the measurements of inductor DC current,
but affects the AC component of the inductor
current.
The voltage across the C1 can expressed by:
Soft-Start
1
VC 1 ( s ) = (V in − V out
sC 1
)*
R1 + 1
sC 1
- - - -( 3 )
Combining equations (1),(2) and (3) result in the
following expression for VC1:
VC 1 ( s ) = I L1 *
R L1 + sL 1
1 + sR 1 * C 1
- - - -( 4 )
Usually the resistor R1 and C1 are chosen so that
the time constant of R1 and C1 equals the time
constant of the inductor which is the inductance
L1 over the inductor’s DCR (RL1). If the two time
constants match, the voltage across C1 is
proportional to the current through L1, and
www.irf.com
The IR3622 has programmable soft-start to
control the output voltage rise and limit the inrush
current during start-up. It provides a separate
soft-start function for each output. This will
enable to sequence the outputs by controlling the
rise time of each output through selection of
different value soft-start capacitors.
To ensure correct start-up, the soft-start
sequence initiates when the Vcc, VcH1, VcH2
and Enable rise above their threshold and
generate the Power On Reset (POR) signal.
Soft-start function operates by sourcing an
internal current to charge an external capacitor to
about 3V. Initially, the soft-start function clamps
the error amplifier’s output of the PWM
converter.
12
IR3622MPbF
3V
Soft-Start (cont.)
During power up, the converter output starts at
zero and thus the voltage at Fb is about 0V. An
internal voltage-controlled current source (64uA)
injects current into the Fb pin and generates a
voltage about 1.6V (64ux25K) across the
negative input of error amplifier, see figure 7.
This keeps the output of the error amplifier low.
The magnitude of this current is inversely
proportional to the voltage at the soft-start pin.
The 23uA current source starts to charge up the
external capacitor. In the mean time, the softstart voltage ramps up, the current flowing into
Fb pin starts to decrease linearly and so does the
voltage at the negative input of error amplifier.
ISS1 = 23uA
64uA
OCP1
SS1/SD1
Ihiccup1 = 3uA
POR
Seq
E/A1
Fb1
VP1
When the soft-start voltage reaches about 1V,
the voltage at the negative input of the error
amplifier is approximately 0.8V.
As the soft-start capacitor voltage charges up,
the current flowing into the Fb pin keeps
decreasing.
The feedback voltage increases linearly as the
injecting current goes down. The injecting current
drops to zero when soft-start voltage is around
1.8V and the output voltage goes into steady
state. Figure 8 shows the theoretical operational
waveforms during soft-start.
The output start-up time is the time period when
soft-start capacitor voltage increases from 1V to
1.8V. The start-up time will be dependent on the
size of the external soft-start capacitor. The startup time can be estimated by:
23µA ∗
Tstart
= 1.8V − 1V
Css
3V
ISS2 = 23uA
64uA
SS2/SD2
OCP2
Ihiccup2 = 3uA
POR
E/A2
Fb2
VP2
Track
Fig. 7: Soft-Start circuit for IR3622
Output of POR
3V
≅1.8V
For a given start up time, the soft-start capacitor
(nF) can be estimated as:
C SS ≅
23 ( µ A ) * T start ( ms )
0 . 8 (V )
- - - -( 5 )
For normal start up the Seq pin should be pulled
high (usually can be connected to Vout3).
Soft-Start
Voltage
Current flowing
into Fb pin
≅1V
0V
64uA
0uA
Voltage at negative input ≅1.6V
of Error Amp
0.8V
0.8V
Voltage at Fb pin
0V
Fig. 8: Theoretical operation waveforms
during soft-start
www.irf.com
13
IR3622MPbF
Output Voltage Tracking and
Sequencing
The IR3622 can accommodate a full spectrum of
user programmable tracking and sequencing
options using Track, Seq, Enable and Power
Good pins.
Through these pins both simple voltage tracking
such as that required by the DDR memory
application or more sophisticated sequencing
such as ratiometric or simultaneous can be
implemented.
The Seq pin controls the internal current sources
to set the power up or down sequencing. Toggle
this pin high for power up, and toggle this pin low
for power down.
The Track pin is used to determine the second
channel output for either ratiometric or
simultaneous by using two external resistors.
Figure 9 shows how these pins are configured for
different sequencing mode.
In general the RA and RB set the output voltage
for the first output and RC and RD set the output
voltage for the second output.
For simultaneous vs. ratiometric, RE and RF can
be selected according to the table below:
Track Pin
Simultaneously
Ratiometric
RE=RC , RF=RD
RE =RA , RF=RB
3V
ISS1 = 23uA
64uA
SS1/SD1
OCP1
CSS1
Ihiccup1 = 3uA
POR
Seq
Vo1
RA
Fb1
RB
VP1
Fig. 10: Ratiometric Power Up / down
E/A1
VREF
3V
ISS2 = 23uA
64uA
SS2/SD2
Floating
OCP2
POR
Vo2
RC
RD
Vo1
RE
RF
Fb2
Ihiccup2 = 3uA
E/A2
Track
VP2
VREF
Fig. 9: Using Seq and Track pin for different sequencing
www.irf.com
Fig. 11: Simultaneously Power up /down
14
IR3622MPbF
Fault Protection
The IR3622 monitors the output voltage for over voltage protection and power good indication. It senses
the Rds(on) of low side MOSFET for over current protection. It also protects the output for prebias
conditions. Figure 12 shows the IC’s operating waveforms under different fault conditions.
POR
3V
1.8V
1.0V
SS
Set Voltage
90%Vfb
Pre_Bias Voltage
Vo
PGood
OCP Threshold
Iout
OV
t0
t1 t2 t3
t4
t5
t6
t7
t8
t9
t10
t11
Fig. 12: Fault Conditions
t0 – t1: Vcc, VcH1,VcH2 and Enable signals passed their respective UVLO threshold. Soft start sequence starts.
t1 – t2: Power Good signal flags high.
t1 – t3: Output voltage ramps up and reaches the set voltage.
t4 – t5: OC event, SS ramps down. IC in Hiccup mode.
t5– t6: OC is removed, recovery sequence, fresh SS.
t6 –t7: Output voltage reaches the set voltage.
t8: OVP event. HDrv turns off and LDrv turns on. The IC latches off.
t9 –t10: Manually recycled the Vcc after latched OVP. PreBias start up.
t10 –t11: New Soft Start sequence
www.irf.com
15
IR3622MPbF
Over-Current Protection
The over current protection is performed by
sensing current through the Rds(on) of the low side
MOSFET (Q2). This method enhances the
converter’s efficiency and reduce cost by
eliminating a current sense resistor. As shown in
figure 13, an external resistor (RSET) is connected
between the OCSet pin and the drain of Q2
which sets the current limit set point.
The internal current source develops a voltage
across RSET. When the Q2 is turned on, the
inductor current flows through the Q2 and results
in a voltage drop which is given by:
VOCSet = (IOCSet ∗ ROCSet ) − (Rds(on) ∗ IL )
- - - -( 6 )
IOCSET
Q1
IR3622
L1
OCSet RSET
VOUT
28uA
OCP
23uA
SS1 / SD
20
3uA
Fig. 14: 3uA current source for discharging
soft-start capacitor during hiccup
The OCP circuit starts sampling current when the
low gate drive is about 3V. The OCSet pin is
internally clamped to approximately 1.4V during
deadtime to prevent false trigging. Figure 15
shows the OCSet pin during one switching cycle.
There is about 150ns delay to mask out the
deadtime, since this node contains switching
noise, this delay also functions as a filter.
Q2
Hiccup
Control
Fig. 13: Connection of over current sensing resistor
The critical inductor current can be calculated by
setting:
Deadtime
VOCSet = (IOCSet ∗ ROCSet ) − (Rds(on) ∗ IL ) = 0
ISET = IL(critical)
R
∗I
= OCSet OCSet
Rds(on)
Blanking time
Iocset * Rocset
Clamp Voltage
- - - -(7 )
An over current is detected if the OCSet pin goes
below ground. This trips the OCP comparator
and cycles the soft start function in hiccup mode.
The hiccup is performed by charging and
discharging the soft-start capacitor at a certain
slope rate. As shown in figure 14 the 3uA current
source is used to discharge the soft-start
capacitor.
The OCP comparator resets after every soft start
cycles, and the converter stays in this mode until
the overload or short circuit is removed. The
converter will automatically recover.
www.irf.com
Fig. 15: OCset pin during normal condition
Ch1: Inductor point, Ch2:LDrv, Ch3:OCSet
The value of RSET should be checked in an actual
circuit to ensure that the over current protection
circuit activates as expected. The IR3622 current
limit is designed primarily as short circuit
protection, "no blow up" circuit, and doesn't
operate as a precision current regulator.
When the SS2 is floating, an over current
condition on either phase would result in hiccup
current protection.
16
IR3622MPbF
Pre-Bias
Operating Frequency Selection
The IR3622 is able to start up into pre-charged
output,
which
prevents
oscillation
and
disturbances of the output voltage.
The output starts in asynchronous fashion and
keeps the synchronous MOSFET off until the first
gate signal for control MOSFET is generated.
Figure below shows a typical Pre-Bias condition
at start up.
Depending on system configuration, specific
amount of output capacitance may be required to
prevent discharging the output voltage.
The switching frequency is determined by
connecting an external resistor (Rt) to ground.
Figure 16 provides a graph of oscillator
frequency
versus
Rt.
The
maximum
recommended channel frequency is 600kHz.
Vo
Pre-Bias Voltage
(Output Voltage before startup)
600
500
Fsw (kHz)
V
700
400
300
200
100
Time
0
5
Over Voltage Protection
Over-voltage is sensed through two dedicated
sense pins VSEN1, VSEN2. A separate OVP circuit
is provided for each channel.
The OVP threshold is user programmable and
can be set by two external resistors. Upon overvoltage condition of either one of the outputs, the
OVP forces a latched shutdown on the fault
output. In this mode, the upper FET driver turns
off and the lower FET drivers turn on, thus
crowbaring the output. Reset is performed by
recycling the Vcc or Enable.
Power Good
The IR3622 provides two separate open collector
power good signals which report the status of the
outputs. The outputs are sensed through the two
dedicated VSEN1 and VSEN2 pins.
Once the IR3622 is enabled and the outputs
reach the set value (90% of the Vout set point)
the power good signals go open and stay open
as long as the outputs stay within the set values.
These pins need to be externally pulled high.
Shutdown using Soft Start pins
The outputs can be shutdown by pulling the softstart pins below 0.3V. This can be easily done by
using an external small signal transistor. During
shutdown both MOSFET drivers will be turned
off. Normal operation will resume by cycling soft
start pin.
www.irf.com
10
15
20
25
30
35
40
45
50
55
60
65
Rt (Kohm)
Fig. 16: Switching Frequency vs. External Resistor (Rt)
Frequency Synchronization
The IR3622 is capable of accepting an external
digital synchronization signal. Synchronization
will be enabled by the rising edge at an external
clock. Per –channel switching frequency is set
by external resistor (Rt). The free running
frequency oscillator frequency is twice the perchannel frequency. During synchronization, Rt is
selected such that the free running frequency is
20% below the synchronization frequency.
Synchronization capability is provided for both
single output current share mode and dual
output configuration. The sync pin is noise
immune, when unused it should be left floating.
Thermal Shutdown
Temperature sensing is provided inside IR3622.
The trip threshold is typically set to 140oC.
When trip threshold is exceeded, thermal
shutdown turns off both MOSFETs. Thermal
shutdown is not latched and automatic restart is
initiated when the sensed temperature drops to
the normal range. There is a 20oC (typical)
hysteresis in the shutdown threshold.
17
IR3622MPbF
Application Information
Design Example:
Soft-Start Programming
The following is a design of typical single output
current share application for IR3622. The
application circuit is shown on page 26.
The soft-start timing can be programmed by
selecting the soft-start capacitance value. The
start-up time of the converter can be calculated
using the following expression:
Vin = 12V , ( ±10%)
CSS (nF ) ≅ 28.75(µA) * Tstart (ms)
Vo = 1.8V
Io = 40 A
- - - -(10 )
Where Tstart is the desired start-up time (ms)
For a start-up time of 5ms, the soft-start
capacitor will be 0.15uF. Choose a ceramic
capacitor at 0.15uF.
∆Vo ≤ 30 mV
Fs = 375 kHz
Output Voltage Programming
Output voltage is programmed by reference
voltage and external voltage divider. As shown in
figure 17 the Fb1 pin is the inverting input of the
error amplifier, which is internally referenced to
0.8V. The divider is set to provide 0.8V at the Fb
pin when the output is at its desired value. The
output voltage is defined by the following
equation:
VOUT
IR3622
The 180o out of phase will reduce the RMS value
of the ripple current seen by input capacitors.
This reduces numbers of input capacitors. The
input capacitors must be able to handle both the
maximum ripple RMS current at the highest
ambient temperature, as well as the maximum
input voltage. The RMS value of current ripple for
a duty cycle under 50% is expressed by:
IRMS =
(I D (1 − D ) + I D (1 − D ) − 2I I D D )
2
1
1
1
2
2
2
2
R5
Fig. 17: Typical application of the IR3622 for
programming the output voltage
⎛
R ⎞
Vo = VREF ∗ ⎜⎜1 + 6 ⎟⎟
R5 ⎠
⎝
- - - -( 8 )
Equation (8) can be rewritten as:
⎞
⎟⎟
⎠
- - - -( 9 )
For the calculated values of R5 and R6 see
feedback compensation section.
1
2
- - - -(11)
Where:
-IRMS is the RMS value of the input capacitor
current
-D1 and D2 are duty cycle for each channel
-I1 and I2 are the output current for each channel
For Io=40A and D=0.16 (1.8V/10.8V),
the IRMS= 9.43A.
Ceramic capacitors are recommended due to
their peak current capabilities. They also feature
low ESR and ESL at higher frequency, which
enhance circuit efficiency.
Use 10x22uF, 16V ceramic capacitor from TDK
(C3225X5R1C226M).
For the single output application when the duty
cycle is larger than 50% the following equation
can be used to calculate the total RMS current
for the input capacitor current:
IRMS = IO (2D(1 − D) + (2 − 2D))
www.irf.com
1 2
R6
Fb1
⎛ V
R5 = R6 ∗ ⎜⎜ ref
⎝ V o−Vref
Input Capacitor Selection
D > 0.5
18
IR3622MPbF
Inductor Selection
Output Capacitor Selection
The inductor is selected based on output power,
operating frequency and efficiency requirements.
Low inductor value results in large ripple current,
smaller size, faster response to a load transient
but poor efficiency and high output noise.
Generally, the selection of inductor value can be
reduced to desired maximum ripple current in the
inductor ( ∆i ) . The optimum point is usually found
between 20% and 50% ripple of the output
current.
The voltage ripple and transient requirements
determine the output capacitors types and
values. The criteria is normally based on the
value of the Equivalent Series Resistance (ESR).
However the actual capacitance value and the
Equivalent Series Inductance (ESL) are other
contributing factors. The overall output voltage
ripple can be expressed as:
For the buck converter, the inductor value for
desired operating ripple current can be
determined using the following relation:
Vin − Vo = L ∗
L = (Vin − Vo ) ∗
∆i
1
; ∆t = D ∗
Fs
∆t
Vo
Vin ∗ ∆i * Fs
Vin = Maximum input voltage
Vo = Output Voltage
∆i = Inductor ripple current
F s= Switching frequency
∆t = Turn on time
D = Duty cycle
For 2-phase single output application the inductor
ripple current is chosen between 20-50% of
maximum phase current
If ∆i ≈ 50%(Io ) , then the output inductor will be:
L = 0.41uH
The Coilcraft MLC1260-401ML (L1=0.4uH, 20A,
RL1=0.93mOhm) is a low profile inductor suitable
for this application.
Use the following equation to calculate C1 and R1
for current sensing:
(refer to figure 6 on page 12)
L1
R L1
This results to C1=1uF and R1=0.432K
www.irf.com
where:
∆Vo(ESR) = ∆IL * ESR
- - - -(13)
⎛Vin ⎞
⎟ * ESL
⎝L⎠
∆Vo(ESL) = ⎜
- - - -(12 )
Where:
R1 * C1 =
∆Vo = ∆Vo(ESR) + ∆Vo(ESL) + ∆Vo(C )
∆Vo(C ) =
∆IL
8 * Co * Fs
∆Vo = Output voltage ripple
∆IL = Inductor ripple current
Therefore it is recommended to select output
capacitor with low enough ESR to meet output
ripple and step load transient requirements.
The output ripple is highest at maximum input
voltage since ∆i increases with input voltage.
Special Polymer capacitors offers low ESR with
large storage capacity per unit volume. These
capacitors offer a cost effective output capacitor
solution and are ideal choice when combined
with a controller having high loop bandwidth.
The IR3622 can perform well with all types of
capacitors.
Panasonic EEFSXOD221R (SP, 220F, 2V,
9mOhm) is selected for this design.
Equation (13) can be used to calculate the
required ESR for the specific voltage ripple.
Four SP capacitors would meet the voltage ripple
requirement.
19
IR3622MPbF
Power MOSFET Selection
The IR3622 uses two N-Channel MOSFETs per
channel. The selection criteria to meet power
transfer requirements are based on maximum
drain-source voltage (VDSS), gate-source drive
voltage (Vgs), maximum output current, Onresistance RDS(on), and thermal management.
The MOSFET must have a maximum operating
voltage (VDSS) exceeding the maximum input
voltage (Vin).
The gate drive requirement is almost the same
for both MOSFETs. Logic-level transistor can be
used and caution should be taken with devices at
very low gate threshold voltage (Vgs) to prevent
undesired turn-on of the complementary
MOSFET, which results a shoot-through current.
The total power dissipation for MOSFETs
includes conduction and switching losses. For
the Buck converter the average inductor current
is equal to the DC load current. The conduction
loss is defined as:
switching losses in synchronous Buck converter.
The synchronous MOSFET turns on under zero
voltage conditions, therefore, the turn on losses
for synchronous MOSFET can be neglected.
With a linear approximation, the total switching
loss can be expressed as:
Psw =
Vds(off ) tr + tf
*
* Iload - - - (13A)
2
T
Where:
V ds(off) = Drain to source voltage at the off time
tr = Rise time
tf = Fall time
T = Switching period
Iload = Load current
The switching time waveforms is shown in
figure18.
VDS
90%
2
Pcond = (upper switch)= Iload
∗ Rds(on) ∗ D ∗ ϑ
2
Pcond = (lower switch)= Iload
∗ Rds(on) ∗ (1 − D) ∗ϑ
ϑ = Rds(on) temperature dependency
The RDS(on) temperature dependency should be
considered for the worst case operation. This is
typically given in the MOSFET data sheet.
Ensure that the conduction losses and switching
losses do not exceed the package ratings or
violate the overall thermal budget.
For this design, IRF6622 is selected for control
FET and IRF6629 is selected for synchronous
FET. These devices provide low on resistance in
a compact Direct FET package.
The MOSFETs have the following data:
ControlFET(IRF6622):
Vds = 25V,Qg = 18.7 nC @10Vgs
Rds(on) = 6.3mΩ @Vgs = 10V
SyncFET(IRF6629):
Vds = 25V,Qg = 51nC @10Vgs
Rds(on) = 2.1mΩ @Vgs = 10V
The conduction losses will be: Pcon=1.1W/Phase
The switching loss is more difficult to calculate,
even though the switching transition is well
understood. The reason is the effect of the
parasitic components and switching times during
the switching procedures such as turn-on / turnoff delays and rise and fall times. The control
MOSFET contributes to the majority of the
www.irf.com
10%
VGS
td(ON)
tr
td(OFF)
tf
Fig. 18: switching time waveforms
From IRF6622 data sheet:
tr = 13ns
tf = 14ns
These values are taken under a certain condition
test. For more details please refer to the IRF6622
data sheet.
By using equation (13A), we can calculate the
switching losses. Psw=2.8W
The reverse recovery loss is also another
contributing factor in control FET switching
losses. This is equivalent to extra current
requires to remove the minority charges from
synchronous FET. The reverse recovery loss can
be expressed as:
PQrr = Qrr * trr * Fs
Qrr : ReverseRecoveryCharge
trr : ReverseRecoveryTime
Fs : SwitchingFrequency
20
IR3622MPbF
Feedback Compensation
The IR3622 is a voltage mode controller; the
control loop is a single voltage feedback path
including error amplifier and error comparator. To
achieve fast transient response and accurate
output regulation, a compensation circuit is
necessary. The goal of the compensation
network is to provide a closed loop transfer
function with the highest 0dB crossing frequency
and adequate phase margin (greater than 45o).
FESR =
VOUT
1
2 ∗ π Lo ∗ Co
Gain
R5
VREF
-40dB/decade
-180
CPOLE
Gain(dB)
H(s) dB
Frequency
Fig. 20: TypeII compensation network
and its asymptotic gain plot
The transfer function (Ve/Vo) is given by:
- - - -(16)
The (s) indicates that the transfer function varies
as a function of frequency. This configuration
introduces a gain and zero, expressed by:
FLC
Frequency
[H(s)] = ⎛⎜⎜ g
Fig. 19: Gain and Phase of LC filter
The IR3622’s error amplifier is a differential-input
transconductance amplifier. The output is
available for DC gain control and AC phase
compensation.
The E/A can be compensated either in type II or
type III compensation. When it is used in type II
compensation the transconductance properties of
the E/A become evident and can be used to
cancel one of the output filter poles. This will be
accomplished with a series RC circuit from Comp
pin to ground as shown in figure 20.
This method requires that the output capacitor
has enough ESR to satisfy stability requirements.
In general the output capacitor’s ESR generates
a zero typically at 5kHz to 50kHz which is
essential for an acceptable phase margin. The
ESR zero of the output capacitor expressed as
follows:
www.irf.com
Ve
R4
⎛
R5 ⎞ 1 + sR4C9
⎟*
H(s) = ⎜⎜ gm *
R5 + R6 ⎟⎠
sC9
⎝
0
0dB
Comp
C9
FZ
Phase
FLC Frequency
E/A
- - - -(14)
Figure 19 shows gain and phase of the LC filter.
Since we already have 180o phase shift just from
the output filter, the system risks being unstable.
- - - -(15)
R6 Fb
The output LC filter introduces a double pole, –
40dB/decade gain slope above its corner
resonant frequency, and a total phase lag of 180o
(see figure 19). The resonant frequency of the LC
filter expressed as follows:
FLC =
1
2 ∗ π * ESR * Co
⎝
Fz =
m
*
R5 ⎞
⎟ * R4
R5 + R6 ⎟⎠
1
2π * R4 * C9
- - - -(17)
- - - -(18)
The gain is determined by the voltage divider and
E/A’s transconductance gain.
First select the desired zero-crossover frequency
(Fo):
Fo > FESR and Fo ≤ (1/5 ~ 1/10) * Fs
Use the following equation to calculate R4:
R4 =
Vosc * Fo * FESR * (R5 + R6 )
Vin * FLC2 * R5 * gm
- - - -(19)
Where:
Vin = Maximum Input Voltage
Vosc = Oscillator Ramp Voltage
Fo = Crossover Frequency
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
gm = Error Amplifier Transconductance
21
IR3622MPbF
To cancel one of the LC filter poles, place the
zero before the LC filter resonant frequency pole:
Fz = 75%FLC
Fz = 0.75 *
1
2π Lo * Co
VOUT
ZIN
C12
C10
- - - -(20)
R8
R7
C11
R6
Zf
Using equations (18) and (20) to calculate C9.
C9 =
Fb
1
2π * R4 * Fz
R5
One more capacitor is sometimes added in
parallel with C9 and R4. This introduces one more
pole which is mainly used to suppress the
switching noise.
The additional pole is given by:
FP =
1
C *C
2π * R4 * 9 POLE
C9 + CPOLE
CPOLE =
1
π * R4 * Fs −
For FP <<
1
C9
≅
1
π * R4 * Fs
Fs
2
For a general solution for unconditional stability
for any type of output capacitors in a wide range
of ESR values, we should implement local
feedback with a compensation network (typeIII).
The typically used compensation network for
voltage-mode controller is shown in figure 21.
In such configuration, the transfer function is
given by:
Ve 1 − g m Zf
=
Vo 1 + g m ZIN
- - - -(21)
By replacing Zin and Zf according to figure 15, the
transformer function can be expressed as:
H (s ) =
(1 + sR7C11 ) * [1 + sC10 (R6 + R8 )]
1
*
sR6 (C11 + C12 ) ⎡
⎛ C11 * C12 ⎞⎤
⎟⎟⎥ * (1 + sR8C10 )
⎢1 + sR7 ⎜⎜
⎝ C11 + C12 ⎠⎦
⎣
www.irf.com
Ve
H(s) dB
FZ2
FP2
FP3
Frequency
Fig. 21: Compensation network with local
feedback and its asymptotic gain plot
As known, transconductance amplifier has high
impedance (current source) output, which needs
to be considered when loading the E/A output. If
the source/sink output current capability is
exceeded the amplifier will not be able to swing
its output voltage over the necessary range.
The compensation network has three poles and
two zeros and they are expressed as follows:
FP1 = 0
FP 2 =
1
2π * R8 * C10
1
1
≅
⎛ C11 * C12 ⎞ 2π * R7 * C12
⎟⎟
2π * R7 ⎜⎜
⎝ C11 + C12 ⎠
1
Fz1 =
2π * R7 * C11
FP 3 =
The error amplifier gain is independent of the
transconductance under the following condition:
g m * Zf >> 1 and g m * Zin >> 1
Comp
VREF
Gain(dB)
FZ1
The pole sets to one half of switching frequency
which results in the capacitor CPOLE:
E/A
Fz 2 =
1
1
≅
2π * C10 * (R6 + R8 ) 2π * C10 * R6
Cross over frequency is expressed as:
Fo = R7 * C10 *
Vin
1
*
Vosc 2π * Lo * Co
22
IR3622MPbF
Based on the frequency of the zero generated by
output capacitor and its ESR versus crossover
frequency, the compensation type can be
different. The table below shows the
compensation types and location of crossover
frequency.
Compensator
type
FESR vs. Fo
Output
capacitor
TypII(PI)
FLC<FESR<Fo<Fs/2
Electrolytic
, Tantalum
TypeIII(PID)
Method A
FLC<Fo<FESR<Fs/2
Tantalum,
ceramic
TypeIII(PID)
Method B
FLC<Fo<Fs/2<FESR
Ceramic
Table1- The compensation type and location
of FESR versus Fo
The following design rules will give a crossover
frequency approximately one-sixth of the
switching frequency. The higher the band width,
the potentially faster the load transient response.
The DC gain will be large enough to provide high
DC-regulation accuracy (typically -5dB to -12dB).
The phase margin should be greater than 45o for
overall stability.
2
; R7 ≥ 0.67KΩ ; Select : R7 = 6.04KΩ
gm
R7 ≥
Calculate C11 , C12 and C10 :
FZ1 = 0.75 * FLC
1
; C11 = 2.90nF, Select : C11 = 2.8nF
2π * FZ1 * R 7
C11 =
FP3 = Fs
The details of these compensation types are
discussed in application note AN-1043 which can
be downloaded from IR Web-Site.
C12 =
1
; C12 = 70pF, Select : C12 = 56pF
2π * FP 3 * R7
For this design we have:
C10 =
2π * Fo * Lo * Co * Vosc
; C10 = 1.03nF,
R7 * Vin
Vin=13.2V
Vo=1.8V
Vosc=1.25V
Vref=0.8V
gm=3000umoh
Lo=0.4uH, DCR=0.930mOhm
Co=4x220uF, ESR= 2.25mOhm
Fs=375kHz
Select : C10 = 1.5nF
Calculate R8 , R6 and R5 :
R8 =
1
; R8 = 1.32KΩ, Select : R8 = 1KΩ
2π * C10 * FP 2
R6 =
1
− R8 ; R6 = 7.84KΩ, Select : R6 = 7.87KΩ
2π * C10 * FZ 2
R5 =
Vref
* R6 ; R5 = 6.30KΩ , Select : R5 = 6.34KΩ
Vo − Vref
These result in:
FLC=12kHz
(Replace L to L/2 in formula#14 for current share configuration)
FESR=80.38kHz
Fs/2=185kHz
Select crossover frequency:
Fo < FESR and Fo ≤ (1/5 ~ 1/10) * Fs
Fo=60kHz
Since: FLC<Fo<FESR<Fs/2, typeIII method A is
selected to place the pole and zeros.
www.irf.com
Check :
R8 R6 R5
0.78kΩ >
>
1
gm
1
= 0.33 KΩ
gm
1
OK!
gm
If this condition is not met, then iteration may be
required by selecting larger R7.
23
IR3622MPbF
Compensation for
(slave channel)
Current
Loop
The slave error amplifier is differential
transconductance
amplifier,
in
2-phase
configuration the main goal for the slave channel
feedback loop is to control the inductor current to
match the master channel inductor current as
well provides highest bandwidth and adequate
phase margin for overall stability. The following
analysis is valid for both using external current
sense resistors and using DCR of the inductor.
The transfer function of power stage is
expressed by:
G(s) =
IL2 (s)
Vin
=
Ve
sL2 * Vosc
Select a zero frequency for current loop (Fo2) 1.2
times larger than zero cross frequency for
voltage loop (Fo1).
FO2 ≅ 1.25% * FO1
H(FO2 ) = gm * Rs1 * R2 *
Where:
Vin=Input voltage
L2=Output inductor
Vosc=Oscillator Peak Voltage
As shown the G(s) is a function of inductor
current. The transfer function for compensation
network is given by equation (23), when using a
series RC circuit as shown in figure22.
IL2
1
2π * FO2 * L2 * Vosc
*
gm * Rs1
Vin
Vin=13.2V
Vosc=1.25V
gm=3000umoh
L2=0.4uH
Rs1=DCR=0.930mOhm
Fo2=72kHz
This results to : R2=6.14K
Select R2=6.09K
Fb2
Vp2
- - - -( 25 )
The power stage of current loop has a dominant
pole (Fp) at frequency expressed by:
L2
RS2
- - - -( 24 )
From (24), R2 can be expressed as:
R2 =
- - - -( 22 )
Vin
=1
2π * FO2 * L2 * Vosc
E/A2
FP =
Comp2
Ve
Req = Rds(on1) * D + Rds(on2 ) * (1 − D) + RL
R2
RS1
L1
Req
2π * L2
C2
IL1
Fig. 22: The Compensation network for current loop
Where Rds(on1) is the on-resistance of control
FET, Rds(on2) is the on-resistance of synchronous
FET, RL is the DCR of output inductance and D
is the duty cycle
Req=3.7mOhm
T (s) =
Ve (s) ⎛
R ⎞ ⎛1 + sC2R2 ⎞
⎟
= ⎜ gm * s1 ⎟⎟ * ⎜⎜
Rs2 ⎜⎝
Rs2 ⎠ ⎝ sC2 ⎟⎠
- - - -( 23 )
Set the zero of compensator at 10 times the
dominant pole frequency FP, the compensator
capacitor, C2 can be expressed as:
The loop gain function is:
Fz = 10 * FP
H(s) = [G(s) * T (s) * Rs2 ]
⎛
R ⎞ ⎛1 + sR2C2 ⎞ ⎛ Vin ⎞
⎟
⎟*⎜
H(s) = Rs2 * ⎜⎜ gm * s1 ⎟⎟ * ⎜⎜
R
sC2 ⎟⎠ ⎜⎝ sL2 * Vosc ⎟⎠
s2 ⎠ ⎝
⎝
www.irf.com
C2 =
C2=1.8nF
1
2π * R2 * Fz
All design should be tested for stability
to verify the calculated values.
24
IR3622MPbF
Programming the Current-Limit
The Current-Limit threshold can be set by
connecting a resistor (RSET) from drain of low
side MOSFET to the OCSet pin. The resistor
can be calculated by using equation (7).
The Rds(on) has a positive temperature coefficient
and it should be considered for the worse case
operation. This resistor must be placed close to
the IC, place a small ceramic capacitor from this
pin to ground for noise rejection purposes.
ISET = IL(critical) =
ROCSet ∗ IOCSet
Rds(on)
- - - -(7 )
Rds ( on ) = 2.1mΩ ∗1.5 = 3.15 mΩ
ISET ≅ Io ( LIM ) = 20 A ∗1.5 = 30 A
(50% over nominal output current)
ROCSet = R3 = R4 = 4KΩ
www.irf.com
Layout Consideration
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Start to place the power components, make all
the connection in the top layer with wide, copper
filled areas. The inductor, output capacitor should
be close to each other as possible. This helps to
reduce the EMI radiated by the power traces due
to the high switching currents through them.
Place input capacitor close to control FETs, to
reduce the ESR replace the single input capacitor
with two parallel units. The feedback part of the
system should be kept away from the inductor
and other noise sources, and be placed close to
the IC. In multilayer PCB use one layer as power
ground plane and have a control circuit ground
(analog ground), to which all signals are
referenced. The goal is to localize the high
current path to a separate loop that does not
interfere with the more sensitive analog control
function. These two grounds must be connected
together on the PC board layout at a single point.
The exposed pad of IC should be connected to
analog ground.
25
IR3622MPbF
Typical Application
D1
C12
12V
C11
C3
C4
C5
R11
VCL
VcH1 VOUT3 VcH2
Vcc
HDrv1
OCSet1
Enable
VREF
VP1
Rt
R2
Vout3
R3
C8
R4
C9
U1
IR3622
Comp1
LDrv1
R10
SS1 / SD
SS2 / SD
Vout3
L3
Q3
R5
Vout
C15
LDrv2
C16
R7
VSEN2
Fb1
Fb2
HDrv2
R13
R14
VSEN1
OCSet2
PGood1
C10
Q2
C17
Comp2
PGood1
R1
C14
PGnd1
VP2
Sync
R12
C13
R6
Q4
R8
R9
C18
L4
Q5
PGnd2
Gnd
Track
Seq
PGood2
Fig. 23: Application circuit for 12V to 1.8V @ 40A
www.irf.com
26
IR3622MPbF
Typical Application
D1
BAT54S
C12
12V
C11
C3
C4
VCL
R26
R2
R3
R24
Rt
C8
R4
Comp2
PGood1
PGood2
PGood2
Vout1
R21
R7
VSEN1
VSEN1
VSEN2 VSEN2
Fb1
Fb2
HDrv2
OCSet2
R6
LDrv2
SS1 / SD
C15
D2 VSEN1
BAT54S
C30
R8
R5
C17
C9
PGood1
C10
C16
R20
VOUT3
Comp1
R25
Q3
PGnd1
Sync
U1
IR3622
L3
Vout1
LDrv1
Enable
VP1
VREF
VP2
Q2
R1
OCSet1
R27
VOUT3
HDrv1
Vcc
C5
C14
C13
VcH1 VOUT3 VcH2
Q4
Vout2
C18
Q5
R22
PGnd2
SS2 / SD
R9
L4
VSEN2
R23
Gnd
Seq
RA
Track
RB
Fig. 24: Application circuit for Dual output application
Tracking and sequencing using Track pin
Track Pin
Simultaneously
Ratiometric
www.irf.com
RA=R9 , RB=R5
RA =R7 , RB=R8
27
IR3622MPbF
PCB Metal and Components Placement
ƒ Lead land width should be equal to nominal part lead width. The minimum lead to lead spacing
should be ≥ 0.2mm to minimize shorting.
ƒ Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension +
0.05mm inboard extension. The outboard extension ensures a large and inspectable toe fillet, and the
inboard extension will accommodate any part misalignment and ensure a fillet.
ƒ Center pad land length and width should be equal to maximum part pad length and width. However,
the minimum metal to metal spacing should be ≥ 0.17mm for 2 oz. Copper (≥ 0.1mm for 1 oz. Copper
and ≥ 0.23mm for 3 oz. Copper).
ƒ A single 0.30mm diameter via shall be placed in the center of the pad land and connected to ground to
minimize the noise effect on the IC.
www.irf.com
28
IR3622MPbF
Solder Resist
ƒ The solder resist should be pulled away from the metal lead lands by a minimum of 0.06mm. The
solder resist mis-alignment is a maximum of 0.05mm and it is recommended that the lead lands are all
Non Solder Mask Defined (NSMD). Therefore pulling the S/R 0.06mm will always ensure NSMD pads.
ƒ The minimum solder resist width is 0.13mm.
At the inside corner of the solder resist where the lead land groups meet, it is recommended to provide
a fillet so a solder resist width of ≥ 0.17mm remains.
ƒ The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto
the copper of 0.06mm to accommodate solder resist mis-alignment. In 0.5mm pitch cases it is
allowable to have the solder resist opening for the land pad to be smaller than the part pad.
ƒEnsure that the solder resist in-between the lead lands and the pad land is ≥ 0.15mm due to the high
aspect ratio of the solder resist strip separating the lead lands from the pad land.
ƒ The single via in the land pad should be tented or plugged from bottom boardside with solder resist.
www.irf.com
29
IR3622MPbF
Stencil Design
ƒ The stencil apertures for the lead lands should be approximately 80% of the area of the lead lands.
Reducing the amount of solder deposited will minimize the occurrence of lead shorts. Since for 0.5mm
pitch devices the leads are only 0.25mm wide, the stencil apertures should not be made narrower;
openings in stencils < 0.25mm wide are difficult to maintain repeatable solder release.
ƒ The stencil lead land apertures should therefore be shortened in length by 80% and centered on the
lead land.
ƒ The land pad aperture should be striped with 0.25mm wide openings and spaces to deposit
approximately 50% area of solder on the center pad. If too much solder is deposited on the center pad
the part will float and the lead lands will be open.
ƒ The maximum length and width of the land pad stencil aperture should be equal to the solder resist
opening minus an annular 0.2mm pull back to decrease the incidence of shorting the center land to the
lead lands when the part is pushed into the solder paste.
www.irf.com
30
IR3622MPbF
(IR3622M) MLPQ Package; 5x5-32 Lead
Feed Direction
Figure A
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
This product has been designed and qualified for the Industrial market.
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 6/15/2007
www.irf.com
31