LM3150 SIMPLE SWITCHER® CONTROLLER, 42V Synchronous Step-Down General Description Features SWITCHER® The LM3150 SIMPLE Controller is an easy to use and simplified step down power controller capable of providing up to 12A of output current in a typical application. Operating with an input voltage range of 6V-42V, the LM3150 features an adjustable output voltage down to 0.6V. The switching frequency is adjustable up to 1 MHz and the synchronous architecture provides for highly efficient designs. The LM3150 controller employs a Constant On-Time (COT) architecture with a proprietary Emulated Ripple Mode (ERM) control that allows for the use of low ESR output capacitors, which reduces overall solution size and output voltage ripple. The Constant On-Time (COT) regulation architecture allows for fast transient response and requires no loop compensation, which reduces external component count and reduces design complexity. Fault protection features such as thermal shutdown, undervoltage lockout, over-voltage protection, short-circuit protection, current limit, and output voltage pre-bias startup allow for a reliable and robust solution. The LM3150 SIMPLE SWITCHER® concept provides for an easy to use complete design using a minimum number of external components and National’s WEBENCH® online design tool. WEBENCH® provides design support for every step of the design process and includes features such as external component calculation with a new MOSFET selector, electrical simulation, thermal simulation, and Build-It boards for prototyping. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ PowerWise® step-down controller 6V to 42V Wide input voltage range Adjustable output voltage down to 0.6V Programmable switching frequency up to 1 MHz No loop compensation required Fully WEBENCH® enabled Low external component count Constant On-Time control Ultra-fast transient response Stable with low ESR capacitors Output voltage pre-bias startup Valley current limit Programmable soft-start Typical Applications ■ ■ ■ ■ ■ Telecom Networking Equipment Routers Security Surveillance Power Modules Typical Application 30053101 SIMPLE SWITCHER® is a registered trademark of National Semiconductor Corporation © 2008 National Semiconductor Corporation 300531 www.national.com LM3150 SIMPLE SWITCHER® CONTROLLER, 42V Synchronous Step-Down October 17, 2008 LM3150 Connection Diagram 30053102 eTSSOP-14 Ordering Information Order Number Package Type NSC Package Drawing LM3150MH Supplied As 94 Units per Anti-Static Tube LM3150MHE eTSSOP-14 MXA14A LM3150MHX 250 Units in Tape and Reel 2500 Units in Tape and Reel Pin Descriptions Pin Name Description Function 1 VCC Supply Voltage for FET Drivers Nominally regulated to 5.95V. Connect a 1.0 µF to 2.2 µF decoupling capacitor from this pin to ground. 2 VIN Input Supply Voltage Supply pin to the device. Nominal input range is 6V to 42V. 3 EN Enable To enable the IC apply a logic high signal to this pin greater than 1.26V typical or leave floating. To disable the part, ground the EN pin. 4 FB Feedback Internally connected to the regulation, over-voltage, and short-circuit comparators. The regulation setting is 0.6V at this pin. Connect to feedback resistor divider between the output and ground to set the output voltage. 5,9 SGND Signal Ground Ground for all internal bias and reference circuitry. Should be connected to PGND at a single point. 6 SS Soft-Start An internal 7.7 µA current source charges an external capacitor to provide the softstart function. 7 RON On-time Control An external resistor from VIN to this pin sets the high-side switch on-time. 8 ILIM Current Limit Monitors current through the low-side switch and triggers current limit operation if the inductor valley current exceeds a user defined value that is set by RLIM and the Sense current, ILIM-TH, sourced out of this pin during operation. 10 SW Switch Node Switch pin of controller and high-gate driver lower supply rail. A boost capacitor is also connected between this pin and BST pin 11 HG High-Side Gate Drive Gate drive signal to the high-side NMOS switch. The high-side gate driver voltage is supplied by the differential voltage between the BST pin and SW pin. 12 BST Connection for Bootstrap Capacitor High-gate driver upper supply rail. Connect a 0.33 µF-0.47 µF capacitor from SW pin to this pin. An internal diode charges the capacitor during the high-side switch offtime. Do not connect to an external supply rail. 13 LG Low-Side Gate Drive Gate drive signal to the low-side NMOS switch. The low-side gate driver voltage is supplied by VCC. 14 PGND Power Ground Synchronous rectifier MOSFET source connection. Tie to power ground plane. Should be tied to SGND at a single point. EP EP Exposed Pad Exposed die attach pad should be connected directly to SGND. Also used to help dissipate heat out of the IC. www.national.com 2 If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN, RON to GND SW to GND BST to SW BST to GND Operating Ratings -0.3V to 47V -3V to 47V -0.3V to 7V -0.3V to 52V -0.3V to 7V 2 kV -65°C to +150°C (Note 1) VIN Junction Temperature Range (TJ) EN 6V to 42V −40°C to + 125°C 0V to 5V Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 18V. Symbol Parameter Conditions Min Typ Max Units 5.65 5.95 6.25 V Start-Up Regulator, VCC CVCC = 1 µF, 0 mA to 40 mA VCC IVCC = 2 mA, VIN = 5.5V 40 IVCC = 30 mA, VIN = 5.5V 330 VIN - VCC VIN - VCC Dropout Voltage IVCCL VCC Current Limit (Note 3) VCC = 0V VCC Under-Voltage Lockout Threshold (UVLO) VCC Increasing VCCUVLO-HYS VCC UVLO Hysteresis VCC Decreasing tCC-UVLO-D VCC UVLO Filter Delay IIN Input Operating Current No Switching, VFB = 1V 3.5 5 mA Input Operating Current, Device Shutdown VEN = 0V 32 55 µA Boost Pin Leakage VBST – VSW = 6V 2 nA HG Drive Pull–Up On-Resistance IHG Source = 200 mA 5 Ω HG Drive Pull–Down On-Resistance IHG Sink = 200 mA 3.4 Ω LG Drive Pull–Up On-Resistance ILG Source = 200 mA 3.4 Ω LG Drive Pull–Down On-Resistance ILG Sink = 200 mA 2 Ω SS Pin Source Current VSS = 0V VCCUVLO IIN-SD 65 100 4.75 5.1 mV mA 5.40 V 475 mV 3 µs GATE Drive IQ-BST RDS-HG-Pull-Up RDS-HG-Pull-Down RDS-LG-Pull-Up RDS-LG-Pull-Down Soft-Start ISS ISS-DIS 5.9 SS Pin Discharge Current 7.7 9.5 200 µA µA Current Limit ILIM-TH Current Limit Sense Pin Source Current 75 85 95 µA ON/OFF Timer tON ON Timer Pulse Width tON-MIN ON Timer Minimum Pulse Width tOFF OFF Timer Minimum Pulse Width VIN = 10V, RON = 100 kΩ, VFB = 0.6V 1.02 VIN = 18V, RON = 100 kΩ, VFB = 0.6V 0.62 VIN = 42V, RON = 100 kΩ, VFB = 0.6V 0.36 (Note 4) 200 µs ns 370 525 ns 1.20 1.26 V Enable Input VEN VEN-HYS EN Pin Input Threshold Trip Point VEN Rising EN Pin Threshold Hysteresis VEN Falling 3 1.14 120 mV www.national.com LM3150 All Other Inputs to GND ESD Rating (Note 2) Storage Temperature Range Absolute Maximum Ratings (Note 1) LM3150 Symbol Parameter Conditions Min Typ Max Units 0.588 0.600 0.612 V 0.690 0.720 0.748 Regulation and Over-Voltage Comparator VFB VFB-OV IFB In-Regulation Feedback Voltage VSS > 0.6V Feedback Over-Voltage Threshold Feedback Bias Current 20 V nA Boost Diode Vf Forward Voltage IBST = 2 mA 0.7 IBST = 30 mA 1 V Thermal Characteristics TSD θJA θJC Thermal Shutdown Rising 165 °C Thermal Shutdown Hysteresis Falling 15 °C 4 Layer JEDEC Printed Circuit Board, 9 Vias, No Air Flow 40 2 Layer JEDEC Printed Circuit Board. No Air Flow 140 Junction to Ambient Junction to Case No Air Flow °C/W 4 °C/W Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but does not guarantee specific performance limits. For guaranteed specifications and conditions, see the Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test Method is per JESD-22-A114. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: See Applications section for minimum on-time when using MOSFETs connected to gate drivers. www.national.com 4 LM3150 Simplified Block Diagram 30053103 5 www.national.com LM3150 Typical Performance Characteristics Boost Diode Forward Voltage vs. Temperature ILIM-TH vs. Temperature 30053140 30053141 Quiescent Current vs. Temperature Soft-Start Current vs. Temperature 30053142 30053143 tON vs. Temperature tON vs. Temperature 30053144 www.national.com 30053145 6 LM3150 tON vs. Temperature VCC Current Limit vs. Temperature 30053147 30053146 VCC Dropout vs. Temperature VCC vs. Temperature 30053148 30053149 7 www.national.com LM3150 Programming the Output Voltage Theory of Operation The output voltage is set by two external resistors (RFB1,RFB2). The regulated output voltage is calculated as follows: SWITCHER® The LM3150 synchronous step-down SIMPLE Controller utilizes a Constant On-Time (COT) architecture which is a derivative of the hysteretic control scheme. COT relies on a fixed switch on-time to regulate the output. The ontime of the high-side switch can be set manually by adjusting the size of an external resistor (RON). To maintain a relatively constant switching frequency as VIN varies, the LM3150 automatically adjusts the on-time inversely with the input voltage. Assuming an ideal system and VIN is much greater than 1V, the following approximations can be made: The on-time, tON: (5) Where RFB2 is the top resistor connected between VOUT and FB, and RFB1 is the bottom resistor connected between FB and GND. Regulation Comparator The feedback voltage at FB is compared to the internal reference voltage of 0.6V. In normal operation (the output voltage is regulated), an on-time period is initiated when the voltage at FB falls below 0.6V. The high-side switch stays on for the on-time, causing the FB voltage to rise above 0.6V. After the on-time period, the high-side switch stays off until the FB voltage falls below 0.6V. (1) Where constant K = 100 pC The RON resistance value can be calculated as follows: Over-Voltage Comparator (2) Where fs is the desired switching frequency. Control is based on a comparator and the on-timer, with the output voltage feedback (FB) compared with an internal reference of 0.6V. If the FB level is below the reference, the highside switch is turned on for a fixed time, tON, which is determined by the input voltage and the resistor RON. Following this on-time, the switch remains off for a minimum off-time, tOFF, as specified in the Electrical Characteristics table or until the FB pin voltage is below the reference, then the switch turns on again for another on-time period. The switching will continue in this fashion to maintain regulation. During continuous conduction mode (CCM), the switching frequency ideally depends on duty-cycle and on-time only. In a practical application however, there is a small delay in the time that the HG goes low and the SW node goes low that also affects the switching frequency that is accounted for in the typical application curves. The duty-cycle and frequency can be approximated as: The over-voltage comparator is provided to protect the output from over-voltage conditions due to sudden input line voltage changes or output loading changes. The over-voltage comparator continuously monitors the voltage at the FB pin and compares it to a 0.72V internal reference. If the voltage at FB rises above 0.72V, the on-time pulse is immediately terminated. This condition can occur if the input or the output load changes suddenly. Once the over-voltage protection is activated, the HG and LG signals remain off until the voltage at FB pin falls below 0.72V. Current Limit Current limit detection occurs during the off-time by monitoring the current through the low-side switch using an external resistor, RLIM. If during the off-time the current in the low-side switch exceeds the user defined current limit value, the next on-time cycle is immediately terminated. Current sensing is achieved by comparing the voltage across the low side FET with the voltage across the current limit set resistor RLIM. If the voltage across RLIM and the voltage across the low-side FET are equal then the current limit comparator will terminate the next on-time cycle. The RLIM value can be approximated as follows: (3) (4) Typical COT hysteretic controllers need a significant amount of output capacitor ESR to maintain a minimum amount of ripple at the FB pin in order to switch properly and maintain efficient regulation. The LM3150 however, utilizes a proprietary Emulated Ripple Mode control scheme (ERM) that allows the use of low ESR output capacitors. Not only does this reduce the need for high output capacitor ESR, but also significantly reduces the amount of output voltage ripple seen in a typical hysteretic control scheme. The output ripple voltage can become so low that it is comparable to voltage-mode and current-mode control schemes. www.national.com (6) (7) Where IOCL is the user-defined average output current limit value, RDS(ON)max is the resistance value of the low-side FETat the expected maximum FET junction temperature, and ILIMTH is an internal current supply of 85 µA typical. Figure 1 illustrates the inductor current waveform. During normal operation, the output current ripple is dictated by the switching of the FETs. The current through the low-side switch, Ivalley, is sampled at the end of each switching cycle and compared to the current limit, ICL, current. The valley current can be calculated as follows: 8 The LM3150 will sense a short-circuit on the output by monitoring the output voltage. When the feedback voltage has fallen below 60% of the reference voltage, Vref x 0.6 (≈ 0.36V), short-circuit mode of operation will start. During short-circuit operation, the SS pin is discharged and the output voltage will fall to 0V. The SS pin voltage, VSS, is then ramped back up at the rate determined by the SS capacitor and ISS until VSS reaches 0.7V. During this re-ramp phase, if the short-circuit fault is still present the output current will be equal to the set current limit. Once the soft-start voltage reaches 0.7V the output voltage is sensed again and if the VFB is still below Vref x 0.6 then the SS pin is discharged again and the cycle repeats until the short-circuit fault is removed. (8) Where IOUT is the average output current and ΔIL is the peakto-peak inductor ripple current. If an overload condition occurs, the current through the lowside switch will increase which will cause the current limit comparator to trigger the logic to skip the next on-time cycle. The IC will then try to recover by checking the valley current during each off-time. If the valley current is greater than or equal to ICL, then the IC will keep the low-side FET on and allow the inductor current to further decay. Throughout the whole process, regardless of the load current, the on-time of the controller will stay constant and thereby the positive ripple current slope will remain constant. During each on-time the current ramps-up an amount equal to: Soft-Start The soft-start (SS) feature allows the regulator to gradually reach a steady-state operating point, which reduces start-up stresses and current surges. At turn-on, while VCC is below the under-voltage threshold, the SS pin is internally grounded and VOUT is held at 0V. The SS capacitor is used to slowly ramp VFB from 0V to 0.6V. By changing the capacitor value, the duration of start-up can be changed accordingly. The start-up time can be calculated using the following equation: (9) The valley current limit feature prevents current runaway conditions due to propagation delays or inductor saturation since the inductor current is forced to decay following any overload conditions. Current sensing is achieved by either a low value sense resistor in series with the low-side FET or by utilizing the RDS(ON) of the low-side FET. The RDS(ON) sensing method is the preferred choice for a more simplified design and lower costs. The RDS(ON) value of a FET has a positive temperature coefficient and will increase in value as the FET’s temperature increases. The LM3150 controller will maintain a more stable current limit that is closer to the original value that was set by the user, by positively adjusting the ILIM-TH value as the IC temperature increases. This does not provide an exact temperature compensation but allows for a more tightly controlled current limit when compared to traditional RDS(ON) sensing methods when the RDS(ON) value can change typically 140% from room to maximum temperature and cause other components to be over-designed. The temperature compensated ILIM-TH is shown below where TJ is the die temperature of the LM3150 in Celsius: ILIM-TH(TJ) = ILIM-TH x [1 + 3.3 x 10-3 x (TJ - 27)] (11) Where tSS is measured in seconds, Vref = 0.6V and ISS is the soft-start pin source current, which is typically 7.7 µA (refer to electrical table). An internal switch grounds the SS pin if VCC is below the under-voltage lockout threshold, if a thermal shutdown occurs, or if the EN pin is grounded. By using an externally controlled switch, the output voltage can be shut off by grounding the SS pin. During startup the LM3150 will operate in diode emulation mode, where the low-side gate LG will turn off and remain off when the inductor current falls to zero. Diode emulation mode will allow start-up into a pre-biased output voltage. When softstart is greater than 0.7V, the LM3150 will remain in continuous conduction mode. During diode emulation mode at current limit the low-gate will remain off when the inductor current is off. (10) To calculate the RLIM value with temperature compensation, substitute equation (10) into ILIM-TH in equation (7). 30053112 FIGURE 1. Inductor Current - Current Limit Operation 9 www.national.com LM3150 Short-Circuit Protection LM3150 The soft-start time should be greater than the input voltage rise time and also satisfy the following equality to maintain a smooth transition of the output voltage to the programmed regulation voltage during startup. tSS ≥ (VOUT x COUT) / (IOCL - IOUT) Design Guide The design guide provides the equations required to design with the LM3150 SIMPLE SWITCHER® Controller. WEBENCH® design tool can be used with or in place of this section for a more complete and simplified design process. 1. Define Power Supply Operating Conditions a. Required Output Voltage b. Maximum and Minimum DC Input Voltage c. Maximum Expected Load Current during Normal Operation d. Soft-Start Time 2. Set Output Voltage With Feedback Resistors (12) Enable/Shutdown The EN pin can be activated by either leaving the pin floating due to an internal pull up resistor to VIN or by applying a logic high signal to the EN pin of 1.26V or greater. The LM3150 can be remotely shut down by taking the EN pin below 1.02V. Low quiescent shutdown is achieved when VEN is less than 0.4V. During low quiescent shutdown the internal bias circuitry is turned off. The LM3150 has certain fault conditions that can trigger shutdown, such as over-voltage protection, current limit, undervoltage lockout, or thermal shutdown. During shutdown, the soft-start capacitor is discharged. Once the fault condition is removed, the soft-start capacitor begins charging, allowing the part to start-up in a controlled fashion. In conditions where there may be an open drain connection to the EN pin, it may be necessary to add a 1 nF bypass capacitor to this pin. This will help decouple noise from the EN pin and prevent false disabling. (13) where RFB1 is the bottom resistor and RFB2 is the top resistor. 3. Determine RON and fs The available frequency range for a given input voltage range, is determined by the duty-cycle, D = VOUT/VIN, and the minimum tON and tOFF times as specified in the electrical characteristics table. The maximum frequency is thus, fsmax = Dmin/ tON-MIN. Where Dmin=VOUT/VIN-MAX, is the minimum duty-cycle. The off-time will need to be less than the minimum off-time tOFF as specified in the electrical characteristics table plus any turn off and turn on delays of the MOSFETs which can easily add another 200 ns. The minimum off-time will occur at maximum duty cycle Dmax and will determine if the frequency chosen will allow for the minimum desired input voltage. The requirement for minimum off-time is tOFF= (1–Dmax)/fs ≥ (tOFFMIN + 200 ns). If tOFF does not meet this requirement it will be necessary to choose a smaller switching frequency fS. Choose RON so that the switching frequency at your typical input voltage matches your fS chosen above using the following formula: Thermal Protection The LM3150 should be operated such that the junction temperature does not exceed the maximum operating junction temperature. An internal thermal shutdown circuit, which activates at 165°C (typical), takes the controller to a low-power reset state by disabling the buck switch and the on-timer, and grounding the SS pin. This feature helps prevent catastrophic failures from accidental device overheating. When the junction temperature falls back below 150°C the SS pin is released and device operation resumes. RON = [(VOUT x VIN) - VOUT] / (VIN x K x fS) + ROND (14) ROND = - [(VIN - 1) x (VIN x 16.5 + 100)] - 1000 (15) 4. Determine Inductor Required Using Figure 2 To use the nomograph in Figure 2, calculate the inductor voltmicrosecond constant ET from the following formula: (16) Where fs is in kHz units. The intersection of the Load Current and the Volt-microseconds lines on the chart below will determine which inductors are capable for use in the design. The chart shows a sample of parts that can be used. The offline calculator tools and WEBENCH® will fully calculate the requirements for the components needed for the design. www.national.com 10 LM3150 30053152 FIGURE 2. Inductor Nomograph TABLE 1. Inductor Selection Table Inductor Designator Inductance (µH) Current (A) Part Name Vendor L01 47 7-9 L02 33 7-9 SER2817H-333KL COILCRAFT L03 L04 22 7-9 SER2814H-223KL COILCRAFT 15 7-9 7447709150 WURTH L05 10 7-9 RLF12560T-100M7R5 TDK L06 6.8 7-9 B82477-G4682-M EPCOS L07 4.7 7-9 B82477-G4472-M EPCOS L08 3.3 7-9 DR1050-3R3-R COOPER L09 2.2 7-9 MSS1048-222 COILCRAFT L10 1.5 7-9 SRU1048-1R5Y BOURNS L11 1 7-9 DO3316P-102 COILCRAFT L12 0.68 7-9 DO3316H-681 COILCRAFT L13 33 9-12 L14 22 9-12 SER2918H-223 COILCRAFT L15 15 9-12 SER2814H-153KL COILCRAFT L16 10 9-12 7447709100 WURTH L17 6.8 9-12 SPT50H-652 COILCRAFT L18 4.7 9-12 SER1360-472 COILCRAFT L19 3.3 9-12 MSS1260-332 COILCRAFT L20 2.2 9-12 DR1050-2R2-R COOPER 11 www.national.com LM3150 Inductor Designator Inductance (µH) Current (A) Part Name L21 1.5 9-12 DR1050-1R5-R COOPER L22 1 9-12 DO3316H-102 COILCRAFT L23 0.68 9-12 L24 0.47 9-12 L25 22 12-15 SER2817H-223KL COILCRAFT L26 15 12-15 L27 10 12-15 SER2814L-103KL COILCRAFT L28 6.8 12-15 7447709006 WURTH L29 4.7 12-15 7447709004 WURTH L30 3.3 12-15 L31 2.2 12-15 L32 1.5 12-15 MLC1245-152 COILCRAFT L33 1 12-15 L34 0.68 12-15 DO3316H-681 COILCRAFT L35 0.47 12-15 L36 0.33 12-15 DR73-R33-R COOPER L37 22 15- L38 15 15- SER2817H-153KL COILCRAFT L39 10 15- SER2814H-103KL COILCRAFT L40 6.8 15- L41 4.7 15- SER2013-472ML COILCRAFT L42 3.3 15- SER2013-362L COILCRAFT L43 2.2 15- L44 1.5 15- HA3778–AL COILCRAFT L45 1 15- B82477-G4102-M EPCOS L46 0.68 15- L47 0.47 15- L48 0.33 15- www.national.com 12 Vendor LM3150 5. Determine Output Capacitance Typical hysteretic COT converters similar to the LM3150 require a certain amount of ripple that is generated across the ESR of the output capacitor and fed back to the error comparator. Emulated Ripple Mode control built into the LM3150 will recreate a similar ripple signal and thus the requirement for output capacitor ESR will decrease compared to a typical Hysteretic COT converter. The emulated ripple is generated by sensing the voltage signal across the low-side FET and is then compared to the FB voltage at the error comparator input to determine when to initiate the next on-time period. COmin = 70 / (fs2 x L) (17) The maximum ESR allowed to prevent over-voltage protection during normal operation is: ESRmax = (80 mV x L x Af) / ETmin (18) ETmin is calculated using VIN-MIN 30053181 Af = VOUT / 0.6 if there is no feed-forward capacitor used FIGURE 3. Typical MOSFET Gate Charge Curve Af = 1 if there is a feed-forward capacitor used See following design example for estimated power dissipation calculation. The minimum ESR must meet both of the following criteria: ESRmin ≥ (15 mV x L x Af) / ETmax (19) ESRmin ≥ [ ETmax / (VIN - VOUT) ] x (Af / CO) (20) 8. Calculate Input Capacitance The main parameters for the input capacitor are the voltage rating, which must be greater than or equal to the maximum DC input voltage of the power supply, and its rms current rating. The maximum rms current is approximately 50% of the maximum load current. ETmax is calculated using VIN-MAX. Any additional parallel capacitors should be chosen so that their effective impedance will not negatively attenuate the output ripple voltage. 6. Determine The Use of Feed-Forward Capacitor (26) Certain applications may require a feed-forward capacitor for improved stability and easier selection of available output capacitance. Use the following equation to calculate the value of Cff. ZFB = (RFB1 x RFB2)/(RFB1 + RFB2) (21) Cff = VOUT/(VIN-MIN x fS x ZFB) (22) Where, ΔVIN-MAX is the maximum allowable input ripple voltage. A good starting point for the input ripple voltage is 5% of VIN. When using low ESR ceramic capacitors on the input of the LM3150 a resonant circuit can be formed with the impedance of the input power supply and parasitic impedance of long leads/PCB traces to the LM3150 input capacitors. It is recommended to use a damping capacitor under these circumstances, such as aluminum electrolytic that will prevent ringing on the input. The damping capacitor should be chosen to be approximately 5 times greater than the parallel ceramic capacitors combination. The total input capacitance should be greater than 10 times the input inductance of the power supply leads/pcb trace. The damping capacitor should also be chosen to handle its share of the rms input current which is shared proportionately with the parallel impedance of the ceramic capacitors and aluminum electrolytic at the LM3150 switching frequency. The CBYP capacitor should be placed directly at the VIN pin. The recommended value is 0.1 µF. 9. Calculate Soft-Start Capacitor 7. MOSFET and RLIM Selection The high-side and low-side FETs must have a drain to source (VDS) rating of at least 1.2 x VIN. Use the following equations to calculate the desired target value of the low-side FET RDS(ON) for current limit. (23) ILIM-TH(Tj) = ILIM-TH x [1 + 3.3 x 10-3 x (Tj - 27)] (24) The gate drive current from VCC must not exceed the minimum current limit of VCC. The drive current from VCC can be calculated with: IVCCdrive = Qgtotal x fS (25) Where, Q gtotal is the combined total gate charge of the highside and low-side FETs. The plateau voltage of the FET VGS vs Qg curve, as shown in Figure 3, must be less than VCC - 750 mV. (27) Where tss is the soft-start time in seconds and Vref = 0.6V. 10. CVCC, CBST and CEN CVCC should be placed directly at the VCC pin with a recommended value of 1 µF to 2.2 µF. For input voltage ranges that include voltages below 8V a 1 µF capacitor must be used for CVCC. CBST creates a voltage used to drive the gate of the 13 www.national.com LM3150 high-side FET. It is charged during the SW off-time. The recommended value for CBST is 0.47 µF. The EN bypass capac- itor, CEN, recommended value is 1000 pF when driving the EN pin from open drain type of signal. Design Example 30053161 FIGURE 4. Design Example Schematic fs < (1 - D)/725 ns 1. Define Power Supply Operating Conditions a. VOUT = 3.3V fS < (1 - 0.55)/725 ns = 620 kHz b. VIN-MIN = 6V, VIN-TYP = 12V, VIN-MAX = 24V A switching frequency is arbitrarily chosen at 500 kHz which should allow for reasonable size components and satisfies the requirements above. fS = 500 kHz Using fS = 500 kHz RON can be calculated as follows: c. Typical Load Current = 12A, Max Load Current = 15A d. Soft-Start time tSS = 5 ms 2. Set Output Voltage with Feedback Resistors RON = [(VOUT x VIN) - VOUT] / (VIN x K x fS) + ROND ROND = - [(VIN - 1) x (VIN x 16.5 + 100)] - 1000 ROND = - [(12 - 1) x (12 x 16.5 + 100)] -1000 ROND = -4.3 kΩ RON = [(3.3 x 12) - 3.3] / (12 x 100 pC x 500 kHz) - 4.3 kΩ RFB2 = 22.455 kΩ RON = 56.2 kΩ RFB2 = 22.6 kΩ, nearest 1% standard value. 4. Determine Inductor Required 3. Determine RON and fS a. ET = (24-3.3) x (3.3/24) x (1000/500) = 5.7 V µs Dmin = VOUT/VIN-MAX b. From the inductor nomograph a 12A load and 5.7 V µs calculation corresponds to a L44 type of inductor. Dmin = 3.3V/24V = 0.137 Dmax = 3.3V / 6V = 0.55 c. Using the inductor designator L44 in Table 1 the Coilcraft HA3778–AL 1.65 µH inductor is chosen. fsmax = 0.137/ 200 ns = 687 kHz Dmax = VOUT/VIN-MIN 5. Determine Output Capacitance The voltage rating on the output capacitor should be greater than or equal to the output voltage. As a rule of thumb most capacitor manufacturers suggests not to exceed 90% of the capacitor rated voltage. In the case of multilayer ceramics the capacitance will tend to decrease dramatically as the applied voltage is increased towards the capacitor rated voltage. The capacitance can decrease by as much as 50% when the applied voltage is only 30% of the rated voltage. The chosen tOFF = (1-0.55)/687 kHz = 654 ns tOFF should meet the following criteria: tOFF > tOFF-MIN + 200 ns tOFF > 725 ns At the maximum switching frequency of 687 kHz, which is limited by the minimum on-time, the off-time of 654 ns is less than 725 ns. Therefore the switching frequency should be reduced and meet the following criteria: www.national.com 14 Let Cff = 270 pF, which is the closest next standard value. 7. MOSFET and RLIM Selection The LM3150 is designed to drive N-channel MOSFETs. For a maximum input voltage of 24V we should choose N-channel MOSFETs with a maximum drain-source voltage, VDS, greater than 1.2 x 24V = 28.8V. FETs with maximum VDS of 30V will be the first option. The combined total gate charge Qgtotal of the high-side and low-side FET should satisfy the following: For this design the chosen ripple current ratio, r = 0.3, represents the ratio of inductor peak-to-peak current to load current IOUT. A good starting point for ripple ratio is 0.3 but it is acceptable to choose r between 0.25 to 0.5. The nomographs in this datasheet all use 0.3 as the ripple current ratio. Qgtotal ≤ IVCCL / fs Qgtotal ≤ 65 mA / 500 kHz Qgtotal ≤ 130 nC Where IVCCL is the minimum current limit of VCC, over the temperature range, specified in the electrical characteristics table. The MOSFET gate charge Qg is gathered from reading the VGS vs Qg curve of the MOSFET datasheet at the VGS = 5V for the high-side, M1, MOSFET and VGS = 6V for the lowside, M2, MOSFET. The Renesas MOSFET RJK0305DPB has a gate charge of 10 nC at VGS = 5V, and 12 nC at VGS = 6V. This combined gate charge for a high-side, M1, and low-side, M2, MOSFET 12 nC + 10 nC = 22 nC is less than 130 nC calculated Qgtotal. Irmsco = 1A tON = (3.3V/12V)/500 kHz = 550 ns Minimum output capacitance is: COmin = 70 / (fs2 x L) COmin = 70 / (500 kHz2 x 1.65 µH) = 169 µF The maximum ESR allowed to prevent over-voltage protection during normal operation is: The calculated MOSFET power dissipation must be less than the max allowed power dissipation, Pdmax, as specified in the MOSFET datasheet. An approximate calculation of the FET power dissipated Pd, of the high-side and low-side FET is given by: High-Side MOSFET ESRmax = (80 mV x L x Af) / ET Af = VOUT / 0.6 without a feed-forward capacitor Af = 1 with a feed-forward capacitor For this design a feed-forward capacitor will be used to help minimize output ripple. ESRmax = (80 mV x 1.65 µH x 1) / 5.7 V µs ESRmax = 23 mΩ The minimum ESR must meet both of the following criteria: ESRmin ≥ (15 mV x L x Af) / ET ESRmin ≥ [ ET / (VIN - VOUT) ] x (Af / CO) ESRmin ≥ (15 mV x 1.65 µH x 1) / 5.7 V µs = 4.3 mΩ ESRmin ≥ [5.7 V µs / (12 - 3.3) ] x (1 / 169 µF) = 3.9 mΩ Based on the above criteria two 150 µF polymer aluminum capacitors with a ESR = 12 mΩ each for a effective ESR in parallel of 6 mΩ was chosen from Panasonic. The part number is EEF-UE0J101P. The max power dissipation of the RJK0305DPB is rated as 45W for a junction temperature that is 125°C higher than the case temperature and a thermal resistance from the FET junction to case, θJC, of 2.78°C/W. When the FET is mounted onto the PCB, the PCB will have some additional thermal resistance such that the total system thermal resistance of the FET package and the PCB, θJA, is typically in the range of 30° C/W for this type of FET package. The max power dissipation, Pdmax, with the FET mounted onto a PCB with a 125°C junction temperature rise above ambient temperature and θJA = 30°C/W, can be estimated by: 6. Determine Use of Feed-Forward Capacitor From step 5 the capacitor chosen in ESR is small enough that we should use a feed-forward capacitor. This is calculated from: Pdmax = 125°C / 30°C/W = 4.1W The system calculated Pdh of 0.674W is much less than the FET Pdmax of 4.1W and therefore the RJK0305DPB max allowable power dissipation criteria is met. Low-Side MOSFET Primary loss is conduction loss given by: 15 www.national.com LM3150 capacitor should also be able to handle the rms current which is equal to: LM3150 Pdl = Iout2 x RDS(ON) x (1-D) = 122 x 0.01 x (1-0.275) = 1W 10. CVCC, CEN, and CBST CVCC = 1 µF ceramic with a voltage rating greater than 10V Pdl is also less than the Pdmax specified on the RJK0305DPB MOSFET datasheet. However, it is not always necessary to use the same MOSFET for both the high-side and low-side. For most applications it is necessary to choose the high-side MOSFET with the lowest gate charge and the low-side MOSFET is chosen for the lowest allowed RDS(ON). The plateau voltage of the FET VGS vs Qg curve must be less than VCC - 750 mV. The current limit resistor, RLIM, is calculated by estimating the RDS(ON) of the low-side FET at the maximum junction temperature of 100°C. By choosing to go into current limit when the average output load current is 20% higher than the output load current of 12A while the inductor ripple current ratio is 1/3 of the load current will make ICL= 10.4A. Then the following calculation of RLIM is: CEN = 1000 pF ceramic with a voltage rating greater than 10V CBST = 0.47 µF ceramic with a voltage rating greater than 10V RLIM = (10.4 x 0.014) / (75 x 10-6) = 1.9 kΩ Let RLIM = 1.91 kΩ which is the next standard value. 8. Calculate Input Capacitance The input capacitor should be chosen so that the voltage rating is greater than the maximum input voltage which for this example is 24V. Similar to the output capacitor, the voltage rating needed will depend on the type of capacitor chosen. The input capacitor should also be able to handle the input rms current, which is a maximum of approximately 0.5 x IOUT. For this example the rms input current is approximately 0.5 x 12A = 6A. The minimum capacitance with a maximum 5% input ripple ΔVIN-MAX = (0.05 x 12) = 0.6V: CIN = [12 x 0.275 x (1-0.275)] / [500 kHz x 0.6] = 8 µF To handle the large input rms current 2 ceramic capacitors are chosen at 10 µF each with a voltage rating of 50V and case size of 1210. Each ceramic capacitor is capable of handling 3A of rms current. A aluminum electrolytic of 5 times the combined input capacitance, 5 x 20 µF = 100 µF, is chosen to provide input voltage filter damping because of the low ESR ceramic input capacitors. CBYP = 0.1µF ceramic with a voltage rating greater than maximum VIN 9. Calculate Soft-Start Capacitor The soft start-time should be greater than the input voltage rise time and also satisfy the following equality to maintain a smooth transition of the output voltage to the programmed regulation voltage during startup. The desired soft-start time, tss, of 5 ms also needs to satisfy the equality in equation 12, by using the chosen component values through the previous steps as shown below: 5 ms > (3.3V x 300 µF) / (1.2 x 12A - 12A) 5 ms > 0.412 ms Since the desired soft-start time satisfies the equality in equation 12, the soft start capacitor is calculated as: CSS = (7.7 µA x 5 ms) / 0.6V = 0.064 µF Let CSS = 0.068 µF, which is the next closest standard value. This should be a ceramic cap with a voltage rating greater than 10V. www.national.com 16 Designator Value Parameters Manufacturer Part Number CBST 0.47 µF Ceramic, X7R, 16V, 10% TDK C2012X7R1C474K CBYP 0.1 µF Ceramic, X7R, 50V, 10% TDK C2012X7R1H104K CEN 1000 pF Ceramic, X7R, 50V, 10% TDK C1608X7R1H102K CFF 270 pF Ceramic, C0G, 50V, 5% Vishay-Bccomponents VJ0805A271JXACW1BC CIN1, CIN2 10 µF Ceramic, X5R, 35V, 20% Taiyo Yuden GMK325BJ106KN-T COUT1, COUT2 150 µF Polymer Aluminum, , 6.3V, 20% Panasonic EEF-UE0J151R CSS 0.068 µF Ceramic, 0805, 25V, 10% Vishay VJ0805Y683KXXA CVCC 1 µF Ceramic, X7R, 16V, 10% Kemet C0805C105K4RACTU L1 1.65 µH Shielded Drum Core, 2.53 mΩ Coilcraft HA3778–AL M1, M2 30V 8 nC, RDS(ON) @4.5V=10 mΩ Renesas RJK0305DPB RFB1 4.99 kΩ 1%, 0.125W Vishay-Dale CRCW08054k99FKEA RFB2 22.6 kΩ 1%, 0.125W Vishay-Dale CRCW080522k6FKEA RLIM 1.91 kΩ 1%, 0.125W Vishay-Dale CRCW08051K91FKEA RON 56.2 kΩ 1%, 0.125W Vishay-Dale CRCW080556K2FKEA U1 LM3150 National Semiconductor LM3150MH 17 www.national.com LM3150 Bill of Materials LM3150 PCB Layout Considerations It is good practice to layout the power components first, such as the input and output capacitors, FETs, and inductor. The first priority is to make the loop between the input capacitors and the source of the low-side FET to be very small and tie the grounds of the low-side FET and input capacitor directly to each other and then to the ground plane through vias. As shown in Figure 5 when the input capacitor ground is tied directly to the source of the low-side FET, parasitic inductance in the power path, along with noise coupled into the ground plane, are reduced. The switch node is the next item of importance. The switch node should be made only as large as required to handle the load current. There are fast voltage transitions occurring in the switch node at a high frequency, and if the switch node is made too large it may act as an antennae and couple switching noise into other parts of the circuit. For high power designs, it is recommended to use a multi-layer board. The FETs are going to be the largest heat generating devices in the design, and as such, care should be taken to remove the heat. On multi-layer boards using exposed-pad packages for the FETs such as the power-pak SO-8, vias should be used under the FETs to the same plane on the interior layers to help dissipate the heat and cool the FETs. For the typical single FET Power-Pak type FETs, the high-side FET DAP is VIN. The VIN plane should be copied to the other interior layers to the bottom layer for maximum heat dissipation. Likewise, the DAP of the low-side FET is connected to the SW node and the SW node shape should be duplicated to the other PCB layers for maximum heat dissipation. See the Evaluation Board application note AN-1900 for an example of a typical multi-layer board layout, and the Demonstration Board Reference Design Application Note for a typical 2 layer board layout. Each design allows for single sided component mounting. 30053158 FIGURE 5. Schematic of Parasitics 30053180 FIGURE 6. PCB Placement of Power Stage www.national.com 18 LM3150 Physical Dimensions inches (millimeters) unless otherwise noted 14-Lead eTSSOP Package NS Package Number MXA14A 19 www.national.com LM3150 SIMPLE SWITCHER® CONTROLLER, 42V Synchronous Step-Down Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Design Support Amplifiers www.national.com/amplifiers WEBENCH www.national.com/webench Audio www.national.com/audio Analog University www.national.com/AU Clock Conditioners www.national.com/timing App Notes www.national.com/appnotes Data Converters www.national.com/adc Distributors www.national.com/contacts Displays www.national.com/displays Green Compliance www.national.com/quality/green Ethernet www.national.com/ethernet Packaging www.national.com/packaging Interface www.national.com/interface Quality and Reliability www.national.com/quality LVDS www.national.com/lvds Reference Designs www.national.com/refdesigns Power Management www.national.com/power Feedback www.national.com/feedback Switching Regulators www.national.com/switchers LDOs www.national.com/ldo LED Lighting www.national.com/led PowerWise www.national.com/powerwise Serial Digital Interface (SDI) www.national.com/sdi Temperature Sensors www.national.com/tempsensors Wireless (PLL/VCO) www.national.com/wireless THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness. National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other brand or product names may be trademarks or registered trademarks of their respective holders. Copyright© 2008 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Technical Support Center Email: [email protected] Tel: 1-800-272-9959 www.national.com National Semiconductor Europe Technical Support Center Email: [email protected] German Tel: +49 (0) 180 5010 771 English Tel: +44 (0) 870 850 4288 National Semiconductor Asia Pacific Technical Support Center Email: [email protected] National Semiconductor Japan Technical Support Center Email: [email protected]