ONSEMI NCP1410DMR2

NCP1410
250 mA Sync-Rect PFM
Step-Up DC-DC Converter
with Low-Battery Detector
NCP1410 is a monolithic micropower high frequency Boost
(step–up) voltage switching converter IC specially designed for
battery operated hand–held electronic products up to 250 mA loading.
It integrates Synchronous Rectifier for improving efficiency as well as
eliminating the external Schottky Diode. High switching frequency
(up to 600 kHz) allows low profile inductor and output capacitor being
used. Low–Battery Detector, Logic–Controlled Shutdown and
Cycle–by–Cycle Current Limit provide value–added features for
various battery–operated applications. With all these functions ON,
the device quiescent supply current is only 9.0 µA typical. This device
is available in space saving compact Micro8 package.
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MARKING
DIAGRAM
8
Micro8
DM SUFFIX
CASE 846A
8
A1
AYW
1
1
Features
•
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•
•
•
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High Efficiency up to 92%
Very Low Device Quiescent Supply Current of 9.0 A Typical
Allows use of Small Size Inductor and Capacitor
Built–in Synchronous Rectifier (PFET) Eliminates One External
Schottky Diode
High Switching Frequency (up to 600 kHz) Allows Use of Small
Size Inductor and Capacitor
High Accuracy Reference Output, 1.19 V ± 0.6% @ 25°C, can
supply more than 2.5 mA when VOUT ≥ 3.3 V
1.0 V Startup at No Load Guaranteed
Output Voltage from 1.5 V to 5.5 V Adjustable
Output Current up to 250 mA @ Vin = 2.5 V, Vout = 3.3 V
Logic–Controlled Shutdown
Open Drain Low–Battery Detector Output
1.0 A Cycle–by–Cycle Current Limit
Low Profile and Minimum External Parts
Compact Micro8 Package
Typical Applications
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A1 = Device Marking
A = Assembly Location
Y = Year
W = Work Week
PIN CONNECTIONS
FB 1
8 OUT
LBI 2
7 LX
LBO 3
6 GND
REF 4
5 SHDN
(Top View)
ORDERING INFORMATION
Device
NCP1410DMR2
Package
Shipping
Micro8
4000 Tape & Reel
Personal Digital Assistant (PDA)
Handheld Digital Audio Product
Camcorders and Digital Still Camera
Hand–held Instrument
Conversion from One or Two NiMH or NiCd, or One Li–ion Cell to
3.3 V/5.0 V
 Semiconductor Components Industries, LLC, 2001
December, 2001 – Rev. 2
1
Publication Order Number:
NCP1410/D
NCP1410
Input
1.0 V to VOUT
10 µF
150 pF
22 µF
500 k
360 k
+
200 k
LBI
Low Battery
Open Drain
Output
Output 1.5 V to 5.5 V
IOUT typical up to
33 µF 250 mA at 3.3 V Output
and 2.5 V Input
VOUT
FB
Low Battery
Sense Input
NCP1410
LX
LBO
GND
REF
SHDN
150 nF
56 nF
Shutdown
Open Drain
Input
Figure 1. Typical Operating Circuit
MAXIMUM RATINGS (Note 1)
Rating
Symbol
Value
Unit
VOUT
–0.3 to 6.0
V
VIO
–0.3 to 6.0
V
PD
RθJA
520
240
mW
°C/W
Operating Junction Temperature Range
TJ
–40 to +150
°C
Operating Ambient Temperature Range
TA
–40 to +85
°C
Storage Temperature Range
Tstg
–55 to +150
°C
Device Power Supply (Pin 8)
Input/Output Pins
Pin 1–5, Pin 7
Thermal Characteristics
Micro8 Plastic Package
Maximum Power Dissipation @ TA = 25°C
Thermal Resistance Junction to Air
1. This device series contains ESD protection and exceeds the following tests:
Human Body Model (HBM) 2.0 kV per JEDEC standard: JESD22–A114.
Machine Model Method (MM) 200 V per JEDEC standard: JESD22–A115.
2. The maximum package power dissipation limit must not be exceeded.
TJ(max) TA
PD RJA
3. Latch–up Current Maximum Rating: 150 mA per JEDEC standard: JESD78.
4. Moisture Sensitivity Level: MSL 1 per IPC/JEDEC standard: J–STD–020A.
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NCP1410
ELECTRICAL CHARACTERISTICS (VOUT = 3.3 V, TA = 25°C for typical value, –40°C ≤ TA ≤ 85°C for min/max values
unless otherwise noted.)
Characteristics
Operating Voltage
Output Voltage Range (Adjusted by external feedback)
Reference Voltage (CREF = 150 nF, under no loading, TA = 25°C)
Symbol
Min
Typ
Max
Unit
VIN
1.0
–
5.5
V
VOUT
VIN
–
5.5
V
VREF_NL
1.183
1.190
1.197
V
VREF_NL_A
1.178
–
1.202
V
TCVREF
–
0.03
–
mV/°C
Reference Voltage Load Current (VOUT = 3.3 V,
VREF = VREF_NL ±1.5%, CREF = 1.0 F) (Note 5)
IREF
2.5
–
–
mA
Reference Voltage Load Regulation (VOUT = 3.3 V,
IREF = 0 to 100 A, CREF = 1.0 F)
VREF_LOAD
–
0.015
1.0
mV
Reference Voltage Line Regulation (VOUT from 1.5 V to 5.5 V,
CREF = 1.0 F)
VREF_LINE
–
0.03
1.0
mV/V
FB, LBI Input Threshold (ILOAD = 0 mA)
VFB, VLBI
1.174
1.190
1.200
V
N–FET ON Resistance
RDS(ON)–N
0.6
P–FET ON Resistance
RDS(ON)–P
0.9
Reference Voltage (CREF = 150 nF, under no loading,
–40°C ≤ TA ≤ 85°C)
Reference Voltage Temperature Coefficient
LX Switch Current Limit (NFET)
ILIM
–
1.0
–
A
Operating Current into OUT (VFB = 1.4 V, i.e. No switching,
VOUT = 3.3 V)
IQ
–
9.0
14
A
Shutdown Current into OUT (SHDN = GND)
ISD
–
0.05
1.0
A
LX Switch MAX. ON–Time (VFB = 1.0 V, VOUT = 3.3 V)
tON
1.2
1.4
1.8
S
LX Switch MIN. OFF–Time (VFB = 1.0 V, VOUT = 3.3 V)
tOFF
0.25
0.31
0.37
S
FB Input Current
IFB
–
1.5
9.0
nA
LBI Input Current
ILBI
–
1.5
8.0
nA
VLBO_L
–
–
0.05
V
ISHDN
–
1.5
8.0
nA
SHDN Input Threshold, Low
VSHDN_L
–
–
0.3
V
SHDN Input Threshold, High
VSHDN_H
0.6
–
–
V
LBO Low Output Voltage (VLBI = 0, ISINK = 1.0 mA)
SHDN Input Current
5. Loading capability increases with VOUT.
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NCP1410
PIN FUNCTION DESCRIPTIONS
Pin #
Symbol
Pin Description
1
FB
Output Voltage Feedback Input.
2
LBI
Low–Battery Detector Input.
3
LBO
Open–Drain Low–Battery Detector Output. Output is LOW when VLBI is 1.178 V. LBO is high
impedance during shutdown.
4
REF
1.190 V Reference Voltage Output, bypass with 150 nF capacitor if this pin is not loaded, bypass with
1.0 F if this pin is loaded up to 2.5 mA @ VOUT = 3.3 V.
5
SHDN
6
GND
7
LX
8
OUT
Shutdown Input. HIGH ( 0.6 V) = operating; LOW ( 0.3 V) = shutdown.
Ground.
N–Channel and P–Channel Power MOSFET Drain Connection.
Power Output. OUT provides bootstrap power to the IC.
Vbat
L
ZLC
+
–
Vbat
5
RSHDN
SHDN
_ZCUR
PFM
+
–
OUT
CONTROL
LOGIC
VDD
CFB1 RFB1
SENSEFET
_MAINSW2ON
_CEN
GND
_PFM
REF
CREF
Voltage
Reference
COUT
6
M1
RFB2
VDD
_MAINSWOFD
GND
_SYNSW2ON
4
GND
_VREFOK
_SYNSWOFD
_ILIM
+
–
+
RSENSE
ILIM
3
GND
2
VOUT
8
M2
Chip
Enable
_PWGONCE
FB
VDD LX
20 mV
CSHDN
1
7
+
LBO
+
–
LBI
GND
Figure 2. Simplified Functional Diagram
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NCP1410
TYPICAL OPERATING CHARACTERISTICS
1.195
VOUT = 3.3 V
L = 22 µH
CIN = 10 µF
COUT = 33 µF
CREF = 1.0 µF
TA = 25°C
1.208
1.204
REFERENCE VOLTAGE, VREF/V
REFERENCE VOLTAGE, VREF/V
1.212
VIN = 2.2 V
VIN = 1.8 V
1.2
1.196
VIN = 3.0 V
1.192
1.188
1.0
1000
10
100
OUTPUT CURRENT, ILOAD/mA
1.193
IREF = 0 mA
1.190
1.188
IREF = 2.5 mA
1.185
CREF = 1.0 F
TA = 25°C
1.183
1.180
1.5
Figure 3. Reference Voltage vs. Output Current
SWITCH ON RESISTANCE, RDS(ON)/
REFERENCE VOLTAGE, VREF/V
1.192
1.190
1.188
VOUT = 3.3 V
CREF = 150 nF
IREF = 0 mA
–20
0
20
40
60
80
100
5.5
1.8
1.6
1.4
1.2
P–FET (M2)
1.0
0.8
0.6
N–FET (M1)
0.4
0.2
0
–40
–20
0
20
40
60
80
100
AMBIENT TEMPERATURE, TA/°C
AMBIENT TEMPERATURE, TA/°C
Figure 5. Reference Voltage vs. Temperature
Figure 6. Switch ON Resistance vs. Temperature
1.8
2.0
1.7
1.8
MINIMUM STARTUP BATTERY
VOLTAGE, VBATT/V
LX SWITCH MAXIMUM ON TIME, tON/S
1.184
–40
2.5
3.0
3.5
4.0
5.0
4.5
INPUT VOLTAGE AT OUT PIN, VOUT,/V
Figure 4. Reference Voltage vs. Input Voltage
at OUT pin
1.194
1.186
2.0
1.6
1.5
1.4
1.3
1.2
–40
–20
0
20
40
60
80
100
Without Schottky Diode
1.6
1.4
1.2
With Schottky Diode
(MBR0502)
1.0
0.8
0.6
0
20
40
60
80
100
120
AMBIENT TEMPERATURE, TA/°C
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 7. LX Switch Maximum ON Time vs.
Temperature
Figure 8. Minimum Startup Battery Voltage vs.
Loading Current
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NCP1410
TYPICAL OPERATING CHARACTERISTICS
100
80
L = 22 µH
90
EFFICIENCY (%)
EFFICIENCY (%)
90
100
L = 15 µH
L = 10 µH
70
60
VIN = 1.8 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
50
1.0
L = 22 µH
80
70
60
10
100
Figure 9. Efficiency vs. Load Current
Figure 10. Efficiency vs. Load Current
L = 27 µH
EFFICIENCY (%)
EFFICIENCY (%)
90
L = 15 µH
L = 10 µH
VIN = 2.2 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
50
1.0
L = 22 µH
80
70
60
10
100
10
100
OUTPUT LOADING CURRENT, ILOAD/mA
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 11. Efficiency vs. Load Current
Figure 12. Efficiency vs. Load Current
100
L = 22 µH
1000
L = 27 µH
90
EFFICIENCY (%)
90
EFFICIENCY (%)
VIN = 3.0 V
VOUT = 5.0 V
CIN = 10 µF
COUT = 33 µF
50
1.0
1000
100
L = 15 µH
L = 10 µH
70
50
1.0
1000
100
70
60
100
OUTPUT LOADING CURRENT, ILOAD/mA
L = 22 µH
80
10
OUTPUT LOADING CURRENT, ILOAD/mA
90
60
VIN = 2.2 V
VOUT = 5.0 V
CIN = 10 µF
COUT = 33 µF
50
1.0
1000
100
80
L = 27 µH
VIN = 3.0 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
L = 22 µH
80
70
60
10
100
1000
50
1.0
OUTPUT LOADING CURRENT, ILOAD/mA
VIN = 4.5 V
VOUT = 5.0 V
CIN = 10 µF
COUT = 33 µF
10
100
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 13. Efficiency vs. Load Current
Figure 14. Efficiency vs. Load Current
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1000
NCP1410
TYPICAL OPERATING CHARACTERISTICS
3
OUTPUT VOLTAGE CHANGE (%)
OUTPUT VOLTAGE CHANGE (%)
3
2
1
3.0 V
0
2.2 V
–1
VIN = 1.8 V
L = 22 H
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
–2
–3
1.0
10
100
3.0 V
0
2.2 V
–1
VIN = 1.8 V
L = 15 H
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
–2
10
100
1000
OUTPUT LOADING CURRENT, ILOAD/mA
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 15. Output Voltage Change vs.
Load Current
Figure 16. Output Voltage Change vs.
Load Current
200
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
L = 22 H
180
160
140
RIPPLE VOLTAGE, VRIPPLE/mVp–p
RIPPLE VOLTAGE, VRIPPLE/mVp–p
1
–3
1.0
1000
200
120
100
200 mA
80
100 mA
60
40
20
0
1.0
NO LOAD OPERATING CURRENT, IBATT/µA
2
1.5
2.0
160
140
120
100
80
200 mA
60
40
20
100 mA
0
1.0
3.0
2.5
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
L = 15 H
180
1.5
2.0
2.5
BATTERY INPUT VOLTAGE, VBATT/V
BATTERY INPUT VOLTAGE, VBATT/V
Figure 17. Output Ripple Voltage vs.
Battery Input Voltage
Figure 18. Output Ripple Voltage vs.
Battery Input Voltage
14
12
10
8.0
6.0
4.0
2.0
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 100 mA; L = 22 µH,
COUT = 33 µF)
0
0
1.0
2.0
3.0
4.0
5.0
INPUT VOLTAGE AT OUT PIN, VOUT/V
6.0
Upper Trace: Output Voltage Waveform, 2.0 V/Division
Lower Trace: Shutdown Pin Waveform, 1.0 V/Division
Figure 19. No Load Operating Current vs.
Input Voltage at OUT Pin
Figure 20. Startup Transient Response
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3.0
NCP1410
TYPICAL OPERATING CHARACTERISTICS
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 10 mA; L = 22 µH,
COUT = 33 µF)
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 10 mA; L = 22 µH,
COUT = 33 µF)
Upper Trace: Voltage at LX pin, 2.0 V/Division
MiddleTrace Otuput Voltage Ripple, 50 mV/Division
Lower Trace: Inductor Current, IL, 100 mA/Division
Upper Trace: Voltage at LX pin, 2.0 V/Division
MiddleTrace Otuput Voltage Ripple, 50 mV/Division
Lower Trace: Inductor Current, IL, 100 mA/Division
Figure 21. Continuous Conduction Mode
Switching Waveform
Figure 22. Discontinuous Conduction Mode
Switching Waveform
(VIN = 1.8 V, to 3.0 V, L = 22 µH, COUT = 33 µF)
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
Upper Trace: Battery Voltage, VIN, 1.0 V/Division
Lower Trace: Output Voltage Ripple, 100 mV/Division
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Figure 23. Line Transient Response for VOUT = 3.3 V
Figure 24. Load Transient Response for VIN = 1.8 V
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Figure 25. Load Transient Response for VIN = 2.4 V
Figure 26. Load Transient Response for VIN = 3.3 V
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NCP1410
DETAILED OPERATION DESCRIPTIONS
When the main regulator is operating in CCM, as M2 is
being turned OFF, and M1 is just turned ON with M2 not
being completed OFF, the above mentioned situation will
occur. So dead time is introduced to make sure M2 is
completed OFF before M1 is being turned ON.
When the regulator is operating in DCM, as coil current
is dropped to zero, M2 is supposed to be OFF. Fail to do so,
reverse current will flow from the output bulk capacitor
through M2 and then the inductor to the battery input. It
causes damage to the battery. So the ZLC comparator comes
with fixed offset voltage to switch M2 OFF before any
reverse current builds up. However, if M2 is switch OFF too
early, large residue coil current flows through the body diode
of M2 and increases conduction loss. Therefore,
determination on the offset voltage is essential for optimum
performance.
With the implementation of synchronous rectification,
efficiency can be as high as 92%. For single cell input
voltage, use an external Schottky diode such as MBR0520
connected from pin 7 to pin 8 to ensure start–up.
NCP1410 is a monolithic micropower high frequency
step–up voltage switching converter IC specially designed
for battery operated hand–held electronic products up to
250 mA loading. It integrates Synchronous Rectifier for
improving efficiency as well as eliminating the external
Schottky Diode. High switching frequency (up to 600 kHz)
allows low profile inductor and output capacitor being used.
Low–Battery Detector, Logic–Controlled Shutdown and
Cycle–by–Cycle Current Limit provide value–added
features for various battery–operated application. With all
these functions ON, the quiescent supply current is only
9.0 µA typical. This device is available in a compact Micro8
package.
PFM Regulation Scheme
From the simplified Functional Diagram (Figure 2), the
output voltage is divided down and fed back to pin 1 (FB).
This voltage goes to the non–inverting input of the PFM
comparator whereas the comparator’s inverting input is
connected to REF. A switching cycle is initiated by the
falling edge of the comparator, at the moment, the main
switch (M1) is turned ON. After the maximum ON–time
(typical 1.4 µS) elapses or the current limit is reached, M1
is turned OFF, and the synchronous switch (M2) is turned
ON. The M1 OFF time is not less than the minimum
OFF–time (typical 0.31 µS), this is to ensure energy transfer
from the inductor to the output capacitor. If the regulator is
operating at continuous conduction mode (CCM), M2 is
turned OFF just before M1 is supposed to be ON again. If the
regulator is operating at discontinuous conduction mode
(DCM), which means the coil current will decrease to zero
before the next cycle, M1 is turned OFF as the coil current
is almost reaching zero. The comparator (ZLC) with fixed
offset is dedicated to sense the voltage drop across M2 as it
is conducting, when the voltage drop is below the offset, the
ZLC comparator output goes HIGH, and M2 is turned OFF.
Negative feedback of closed loop operation regulates
voltage at pin 1 (FB) equal to the internal voltage reference
(1.190 V).
Cycle–by–Cycle Current Limit
From Figure 2, SENSEFET is applied to sample the coil
current as M1 is ON. With that sample current flowing
through a sense resistor, sense–voltage is developed.
Threshold detector (ILIM) detects whether the
sense–voltage is higher than preset level. If it happens,
detector output signifies the CONTROL LOGIC to switch
OFF M1, and M1 can only be switched ON as next cycle
starts after the minimum OFF–time (typical 0.31 µS). With
properly sizing of SENSEFET and sense resistor, the peak
coil current limit is set at 1.0 A typically.
Voltage Reference
The voltage at REF is set typically at +1.190 V. It can
output up to 2.5 mA with load regulation ±1.5%, at VOUT
equal to 3.3 V. If VOUT is increased, the REF load capability
can also be increased. A bypass capacitor of 0.15 µF is
required for proper operation when REF is not loaded. If
REF is loaded, 1.0 F capacitor at REF is needed.
Shutdown
Synchronous Rectification
The IC is shutdown when the voltage at pin 5 (SHDN) is
pulled lower than 0.3 V. During shutdown, M1 and M2 are
both switched OFF, however, the body diode of M2 allows
current flow from battery to the output, the IC internal circuit
will consume less than 0.05 µA current typically. If the pin
5 voltage is pull higher than 0.6 V, for example, by a resistor
connected to VIN, the IC is enabled, and the internal circuit
will only consume 9.0 µA current typically from the OUT
pin. Refer to Figure 2, the product of RSHDN and CSHDN
must be larger than (500 k • 56 nF, i.e. 28 msec). This is to
provide reset pulse for startup as battery is plugged in.
Synchronous Rectifier is used to replace Schottky Diode
for eliminating the conduction loss contributed by forward
voltage of the latter. Synchronous Rectifier is normally
realized by powerFET with gate control circuitry which,
however, involved relative complicated timing concerns.
As main switch M1 is being turned OFF, if the
synchronous switch M2 is just turned ON with M1 not being
completed turned OFF, current will be shunt from the output
bulk capacitor through M2 and M1 to ground. This power
loss lowers overall efficiency. So a certain amount of dead
time is introduced to make sure M1 is completely OFF
before M2 is being turned ON.
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NCP1410
Low–Battery Detection
Capacitors Selection
A comparator with 30 mV hysteresis is applied to perform
the low–battery detection function. When pin 2 (LBI) is at
a voltage, which can be defined by a resistor divider from the
battery voltage, lower than the internal reference voltage,
1.190 V, the comparator output will cause a 50 Ohm low side
switch to be turned ON. It will pull down the voltage at pin
3 (LBO) which has a hundreds kilo–Ohm of pull–high
resistance. If the pin 2 voltage is higher than 1.190 V + 30
mV, the comparator output will cause the 50 Ohm low side
switch to be turned OFF, pin 3 will become high impedance,
and its voltage will be pulled high by the external resistor.
In all switching mode boost converter applications, both
the input and output terminals sees pulsating voltage/current
waveforms. The currents flowing into and out of the
capacitors multiplying with the Equivalent Series
Resistance (ESR) of the capacitor producing ripple voltage
at the terminals. During the syn–rect switch off cycle, the
charges stored in the output capacitor is used to sustain the
output load current. Load current at this period and the ESR
combined and reflected as ripple at the output terminals. For
all cases, the lower the capacitor ESR, the lower the ripple
voltage at output. As a general guide line, low ESR
capacitors should be used. Ceramic capacitors have the
lowest ESR, but low ESR tantalum capacitors can also be
used as a cost effective substitute.
APPLICATIONS INFORMATION
Output Voltage Setting
The output voltage of the converter is determined by the
external feedback network comprised of RFB1 and RFB2 and
the relationship is given by:
Optional Startup Schottky Diode for Low Battery
Voltage
In general operation, no external Schottky diode is
required, however, in case you are intended to operate the
device close to 1 V level, a Schottky diode connected
between the LX and OUT pins as shown in Figure 27 can
help during startup of the converter. The effect of the
additional Schottky was shown in Figure 8.
R
VOUT 1.190 V 1 FB1
RFB2
where RF2 and RF1 are the upper and lower feedback
resistors respectively.
Low Battery Detect Level Setting
The Low Battery Detect Voltage of the converter is
determined by the external divider network comprised of
RLB1 and RLB2 and the relationship is given by:
L
MBR0502
VOUT
R
VLB 1.190 V 1 LB1
RLB2
OUT
where RLB1 and RLB2 are the upper and lower divider
resistors respectively.
NCP1410
LX
COUT
Inductor Selection
The NCP1410 is tested to produce optimum performance
with a 22 µH inductor at VIN = 3 V, VOUT = 3.3 V supplying
output current up to 250 mA. For other input/output
requirements, inductance in the range 10 µH to 47 µH can
be used according to end application specifications.
Selecting an inductor is a compromise between output
current capability and tolerable output voltage ripple. Of
course, the first thing we need to obey is to keep the peak
inductor current below its saturation limit at maximum
current and the ILIM of the device. In NCP1410, ILIM is set
at 1 A. As a rule of thumb, low inductance values supply
higher output current, but also increase the ripple at output
and reducing efficiency, on the other hand, high inductance
values can improve output ripple and efficiency, however it
also limit the output current capability at the same time. One
other parameter of the inductor is its DC resistance, this
resistance can introduce unwanted power loss and hence
reduce overall efficiency, the basic rule is selecting an
inductor with lowest DC resistance within the board space
limitation of the end application.
Figure 27. Schottky Device Between LX and
OUT Pins
PCB Layout Recommendations
Good PCB layout plays an important role in switching
mode power conversion. Careful PCB layout can help to
minimize ground bounce, EMI noise and unwanted
feedback that can affect the performance of the converter.
Hints in the following paragraphs, can be used as guidelines
in most situations.
Grounding
Star–ground connection should be used to connect the
output power return ground, the input power return ground
and the device power ground together at one point. All high
current running paths must be thick enough for current
flowing through and producing insignificant voltage drop
along the path. Feedback signal path must be separated with
the main current path and sensing directly at the anode of the
output capacitor.
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NCP1410
Components Placement
Feedback Network
Power components, i.e. input capacitor, inductor and
output capacitor, must be placed as close together as
possible. All connecting traces must be short, direct and
thick. High current flowing and switching paths must be
kept away from the feedback (FB, pin 1) terminal to avoid
unwanted injection of noise into the feedback path.
Feedback of the output voltage must be a separate trace
detached from the power path. External feedback network
must be placed very close to the feedback (FB, pin 1) pin and
sensing the output voltage directly at the anode of the output
capacitor.
TYPICAL APPLICATION CIRCUIT
RFB1
355 K
VIN = 1.8 V to 3.0 V
CFB
150 pF
L
22 µH
VBATT
VOUT = 3.3 V/250 mA max.
VOUT
+ CIN
10 µF/
10 V
GND
RLB1
225 K
1 FB
2 LBI
RLB2
330 K
RFB2
200 K
VOUT 8
LX 7
NCP1410
3 LBO
GND 6
4 REF
SHDN 5
RSHDN
560 K
+ COUT
33 µF/
10 V
CSHDN
56 nF
CREF
150 nF
Figure 28. Typical Application Schematic for 2 Alkaline Cells Supply
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GND
NCP1410
GENERAL DESIGN PROCEDURES
Switching mode converter design is considered as black
magic to most engineers, some complicate empirical
formulae are available for reference usage. Those formulae
are derived form the assumption that the key components,
i.e. power inductor and capacitors are available with no
tolerance. Practically, its not true, the result is not a matter
of how accurate the equations you are using to calculate the
component values, the outcome is still somehow away from
the optimum point. In below a simple method base on the
most basic first order equations to estimate the inductor and
capacitor values for NCP1410 operate in Continuous
Conduction Mode is introduced. The component value set
can be used as a starting point to fine tune the circuit
operation. By all means, detail bench testing is needed to get
the best performance out of the circuit.
Determine the Steady State Duty Ratio, D for typical VIN,
operation will be optimized around this point:
VOUT
1
VIN
1D
D1
VIN
1 2.4 V 0.273
3.3 V
VOUT
Determine the average inductor current, ILAVG at
maximum IOUT:
I
250 mA
ILAVG OUT 344 mA
1 0.273
1D
Determine the peak inductor ripple current, IRIPPLE–P and
calculate the inductor value:
Assume IRIPPLE–P is 20% of ILAVG, the inductance of the
power inductor can be calculated as in below:
Design Parameters:
IRIPPLE–P = 0.20 x 344 mA = 68.8 mA
VIN = 1.8 V to 3.0 V, Typical 2.4 V
VOUT = 3.3 V
IOUT = 200 mA (250 mA max)
VLB = 2.0 V
VOUT–RIPPLE = 40 mVP–P at IOUT = 250 mA
L
Standard value of 22 µH is selected for initial trial.
Determine the output voltage ripple, VOUT–RIPPLE and
calculate the output capacitor value:
VOUT–RIPPLE = 40 mVP–P at IOUT = 250 mA
Calculate the feedback network:
Select RFB2 = 200 K
RFB1 RFB2
VVOUT
1
REF
RFB1 200 K
3.3 V 1 355 K
1.19
V
COUT VVLB
RLB1 330 K
2.0 V 1 225 K
1.19
V
REF
IOUT tON
VOUT–RIPPLE IOUT ESRCOUT
where tON = 1.4 µS and ESRCOUT = 0.1 Ω,
COUT Calculate the Low Battery Detect divider:
VLB = 2.0 V
Select RLB2 = 330 K
RLB1 RLB2
VIN tON
2.4 V 0.4 S
24.4 H
2(68.8 mA)
2 IRIPPLE P
250 mA 0.4 S
23.33 F
40 mV 250 mA 0.1 From above calculation, you need at least 23.33 F in
order to achieve the specified ripple level at conditions
stated. Practically, a one level larger capacitor will be used
to accommodate factors not take into account in the
calculation, therefore a capacitor value of 33 F is selected.
1
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NCP1410
PACKAGE DIMENSIONS
Micro8
DM SUFFIX
CASE 846A–02
ISSUE E
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT
EXCEED 0.15 (0.006) PER SIDE.
4. DIMENSION B DOES NOT INCLUDE INTERLEAD
FLASH OR PROTRUSION. INTERLEAD FLASH OR
PROTRUSION SHALL NOT EXCEED 0.25 (0.010)
PER SIDE.
–A–
–B–
K
PIN 1 ID
G
D 8 PL
0.08 (0.003)
–T–
M
T B
S
A
S
SEATING
PLANE
0.038 (0.0015)
C
H
L
J
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DIM
A
B
C
D
G
H
J
K
L
MILLIMETERS
MIN
MAX
2.90
3.10
2.90
3.10
--1.10
0.25
0.40
0.65 BSC
0.05
0.15
0.13
0.23
4.75
5.05
0.40
0.70
INCHES
MIN
MAX
0.114
0.122
0.114
0.122
--0.043
0.010
0.016
0.026 BSC
0.002
0.006
0.005
0.009
0.187
0.199
0.016
0.028
NCP1410
Notes
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NCP1410
Notes
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NCP1410
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NCP1410//D