ONSEMI NCP1417

NCP1417
200 mA DC-DC Step-up
Converter with Dual Low
Battery Protection
NCP1417 is a monolithic micropower high frequency Boost
(step–up) voltage switching converter IC specially designed for
battery operated hand–held electronic products up to 200 mA loading.
It integrates Synchronous Rectifier for improving efficiency as well as
eliminating the external Schottky Diode. High switching frequency
(up to 600 kHz) allows use of a low profile inductor and output
capacitor. Dual Low–Battery Detectors and Cycle–by–Cycle Current
Limit provide value–added features for various battery–operated
applications. With all these functions ON, the quiescent supply current
is only 9.0 A typical. This device is available in a space saving
compact Micro8 package.
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MARKING
DIAGRAM
8
Micro8
DM SUFFIX
CASE 846A
8
1
1417
AYW
1
Features
1417
A
Y
W
• High Efficiency, Up to 92%, Typical
• Very Low Device Quiescent Supply Current of 9.0 A Typical
• Built–in Synchronous Rectifier (P–FET) Eliminates One External
= Device Marking
= Assembly Location
= Year
= Work Week
Schottky Diode
• High Switching Frequency (Up to 600 kHz) Allows Small Size
•
•
•
•
•
•
•
•
•
Inductor and Capacitor
High Accuracy Reference Output, 1.19 V ± 0.6% @ 25C, Can
Supply More Than 2.5 mA when VOUT ≥ 3.3 V
1.0 V Startup at No Load Guaranteed
Output Voltage from 1.5 V to 5.5 V Adjustable
Output Current Up to 200 mA @ Vin = 2.5 V, Vout = 3.3 V
Multi–Function LBI/Shutdown Control Pin
Dual Open Drain Low–Battery Detector Outputs
1.0 A Cycle by Cycle Current Limit
Low Profile and Minimum External Parts
Compact Micro8 Package
PIN CONNECTIONS
FB
1
8
OUT
LBI/SHDN
2
7
LX
LBO1
3
6
GND
REF
4
5
LBO2
(Top View)
ORDERING INFORMATION
Device
NCP1417DMR2
Package
Shipping
Micro8
4000 Tape & Reel
Applications
•
•
•
•
•
Personal Digital Assistants (PDA)
Handheld Digital Audio Product
Camcorders and Digital Still Camera
Handheld Instrument
Conversion from One or Two NiMH or NiCd or One Lithium–ion
Cells to 3.3 V/5.0 V
 Semiconductor Components Industries, LLC, 2002
April, 2002 – Rev. 1
1
Publication Order Number:
NCP1417/D
NCP1417
Input
1.0 V to
VOUT
10 F
22 H
150 pF
355 K
+
VOUT
FB
VOUT
200 K
Low Battery
Sense Input
33 F
LX
LBI/SHDN
Output 1.5 V to 5.5 V
IOUT typical up to
200 mA at 3.3 V Output
and 2.5 V Input
NCP1417
Shutdown
Input
LBO1
GND
REF
LBO2
56 nF
Low Battery
Open Drain
Output 1
Low Battery
Open Drain
Output 2
150 nF
Figure 1. Typical Operating Circuit
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
VOUT
–0.3 to 6.0
V
VIO
–0.3 to 6.0
V
PD
RJA
520
240
mW
C/W
Operating Junction Temperature Range
TJ
–40 to +150
C
Operating Ambient Temperature Range
TA
–40 to +85
C
Storage Temperature Range
Tstg
–55 to +150
C
Power Supply (Pin 8)
Input/Output Pins
Pin 1–5, Pin 7
Thermal Characteristics
Micro8 Plastic Package
Maximum Power Dissipation @ TA = 25°C
Thermal Resistance Junction to Air
1. This device contains ESD protection and exceeds the following tests:
Human Body Model (HBM) 2.0 kV per JEDEC standard: JESD22–A114.
Machine Model (MM) 200 V per JEDEC standard: JESD22–A115.
2. The maximum package power dissipation limit must not be exceeded.
TJ(max) TA
PD RJA
3. Latch–up Current Maximum Rating: 150 mA per JEDEC standard: JESD78.
4. Moisture Sensitivity Level: MSL 1 per IPC/JEDEC standard: J–STD–020A.
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NCP1417
ELECTRICAL CHARACTERISTICS (VOUT = 3.3 V, TA = 25°C for typical value, –40°C ≤ TA ≤ 85C for min/max values unless
otherwise noted.)
Characteristics
Operating Voltage
Output Voltage Range (Adjusted by External Feedback)
Reference Voltage (CREF = 150 nF, Under No Loading, TA = 25C)
Symbol
Min
Typ
Max
Unit
VIN
1.0
–
5.5
V
VOUT
VIN
–
5.5
V
VREF_NL
1.183
1.190
1.197
V
VREF_NL_A
1.178
–
1.202
V
TCVREF
–
0.03
–
mV/C
IREF
2.5
–
–
mA
Reference Voltage Load Regulation
(VOUT = 3.3 V, ILOAD = 0 to 100 A, CREF = 1.0 F)
VREF_LOAD
–
0.015
1.0
mV
Reference Voltage Line Regulation
VREF_LINE
–
0.03
1.0
mV/V
FB, LBI Input Threshold
VFB, VLBI
1.172
1.190
1.200
V
Internal NFET ON–Resistance (ILX = 100 mA)
RDS(ON)_N
–
0.65
–
Internal PFET ON–Resistance (ILX = 100 mA)
RDS(ON)_P
–
1.3
–
ILIM
–
1.0
–
A
IQ
–
9.0
14
A
Reference Voltage (CREF = 150 nF, Under No Loading,
–40C ≤ TA ≤ 85C)
Reference Voltage Temperature Coefficient
Reference Voltage Load Current
(VOUT = 3.3 V, VREF = VREF_NL 1.5%, CREF = 1.0 F) (Note 5)
LX Switch Current Limit (NFET)
Operating Current into OUT
(VFB = 1.4 V, i.e. No Switching, VOUT = 3.3 V)
Shutdown Current into OUT (SHDN = GND)
ISD
–
0.05
1.0
A
LX Switch MAX. ON–Time (VFB = 1.0 V, VOUT = 3.3 V)
tON
0.8
1.4
2.0
S
LX Switch MIN. OFF–Time (VFB = 1.0 V, VOUT = 3.3 V)
tOFF
0.22
0.25
0.46
S
FB Input Current
LBI/SHDN Input Current
LBO1/LBO2 Low Output Voltage (VLBI = 0, ISINK = 1.0 mA)
LBI/SHDN Input Threshold for LBO1
LBI/SHDN Input Threshold for LBO2
IFB
–
1.5
20
nA
ILBI, ISHDN
–
1.5
8.0
nA
VLBO_L1
VLBO_L2
–
–
–
–
0.08
0.08
V
VLBI1
1.172
1.190
1.200
V
VLBI2
0.904
0.944
0.965
V
LBI/SHDN Input Threshold, Low
VSHDN_L
–
–
0.3
V
LBI/SHDN Input Threshold, High
VSHDN_H
0.6
–
–
V
5. Loading capability decreases with VOUT.
PIN FUNCTION DESCRIPTION
Pin No.
Pin Name
Pin Description
1
FB
2
LBI/SHDN
3
LBO1
Open–Drain Low–Battery Detector Output. Output is LOW when VLBI is < 1.172 V. LBO1 is high
impedance during shutdown.
4
REF
1.190 V Reference Voltage Output, bypassing with 150 nF capacitor if this pin is not loaded,
bypassing with 1 µF if this pin is loaded up to 2.5 mA @ VOUT = 3.3 V.
5
LBO2
Open–Drain Low–Battery Detector Output. Output is LOW when VLBI is < 0.904 V. LBO2 is high
impedance during shutdown.
6
GND
Ground
7
LX
8
OUT
Output Voltage Feedback Input.
Low–Battery Detector Input and Shutdown Control input multi–function pin.
N–Channel and P–Channel Power MOSFET Drain Connection.
Power Output. OUT provides bootstrap power to the IC.
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NCP1417
VBAT
+
ZLC
–
CHIP
ENABLE
7
LX
20 mV
+
M2
VDD 8
OUT
_ZCUR
SenseFET
_PWGONCE
1
_CEN
FB
+
PFM
–
_PFM
VDD
_MAINSW2ON
CONTROL
LOGIC
M1
6
GND
GND
_MAINSWOFD
VDD
_SYNSW2ON
4
REF
VOLTAGE
REFERENCE
_ILIM
2
LBI/SHDN
GND
_SYNSWOFD
_VREFOK
VREF
+
3
LBO1
GND
CP1
+
–
30 mV
5
LBO2
GND
CP2
–
0.8 x VREF
+
ILIM
–
+
30 mV
Figure 2. Simplified Functional Diagram
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4
GND
VOUT
NCP1417
1.195
VOUT = 3.3 V
L = 22 µH
Cin = 10 µF
Cout = 33 µF
CREF = 1 µF
TA = 25C
1.215
1.210
REFERENCE VOLTAGE, VREF/V
REFERENCE VOLTAGE, VREF/V
1.220
VIN = 1.8 V
1.192
1.189
1.205
VIN = 2.2 V
IREF = 2.5 mA
1.186
1.200
1
10
100
1000
1
1.5
2
3.5
4
4.5
5
Figure 4. Reference Voltage versus
Input Voltage at OUT Pin
SWITCH ON RESISTANCE, RDS(ON)/Ω
1.188
VOUT = 3.3 V
CREF = 150 nF
IREF = 0 mA
–20
0
20
40
60
80
1.5
P–FET (M2)
1
N–FET (M1)
0.5
VOUT = 3.3 V
0
–40
100
–20
1.6
1.5
1.4
1.3
40
60
80
100
MIN. STARTUP BATTERY VOLTAGE, VBATT/V
1.7
20
20
40
60
80
100
100
120
Figure 6. Switch ON Resistance
versus Temperature
1.8
0
0
AMBIENT TEMPERATURE, TA/C
Figure 5. Reference Voltage versus
Temperature
–20
5.5
2
AMBIENT TEMPERATURE, TA/C
Lx SWITCH MAX. ON TIME, tON/µS
3
Figure 3. Reference Voltage versus
Output Current
1.19
1.2
–40
2.5
INPUT VOLTAGE at OUT PIN, VOUT/V
1.192
1.184
–40
1.180
OUTPUT CURRENT, ILOAD/mA
1.194
1.186
CREF = 1 µF
TA = 25C
1.183
VIN = 3 V
1.195
1.190
REFERENCE VOLTAGE, VREF/V
IREF = 0 mA
2.1
1.8
Without Schottky Diode
1.5
1.2
With Schottky Diode
(MBR0502)
0.9
0.6
0
20
40
60
80
AMBIENT TEMPERATURE, TA/C
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 7. Lx Switch Max. ON Time
versus Temperature
Figure 8. Min. Startup Battery Voltage
versus Loading Current
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NCP1417
100
100
L = 22 µH
80
L = 15 µH
L = 10 µH
70
VIN = 1.8 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
60
50
1
10
100
50
1000
1
10
100
1000
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 9. Efficiency versus Load Current
Figure 10. Efficiency versus Load Current
80
L = 15 µH
L = 10 µH
70
VIN = 2.2 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
1
L = 27 µH
90
EFFICIENCY/%
EFFICIENCY/%
VIN = 2.2 V
VOUT = 5 V
CIN = 10 µF
COUT = 33 µF
OUTPUT LOADING CURRENT, ILOAD/mA
L = 22 µH
60
L = 22 µH
80
70
VIN = 3 V
VOUT = 5 V
CIN = 10 µF
COUT = 33 µF
60
10
100
50
1000
1
10
100
1000
OUTPUT LOADING CURRENT, ILOAD/mA
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 11. Efficiency versus Load Current
Figure 12. Efficiency versus Load Current
100
100
L = 27 µH
L = 22 µH
90
L = 10 µH
80
EFFICIENCY/%
90
EFFICIENCY/%
70
100
90
L = 15 µH
70
VIN = 3 V
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
60
50
L = 22 µH
80
60
100
50
L = 27 µH
90
EFFICIENCY/%
EFFICIENCY/%
90
1
L = 22 µH
80
70
VIN = 4.5 V
VOUT = 5 V
CIN = 10 µF
COUT = 33 µF
60
10
100
1000
50
1
10
100
1000
OUTPUT LOADING CURRENT, ILOAD/mA
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 13. Efficiency versus Load Current
Figure 14. Efficiency versus Load Current
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NCP1417
3
OUTPUT VOLTAGE CHANGE/%
OUTPUT VOLTAGE CHANGE/%
3
2
1
3V
0
2.2 V
–1
L = 22 H
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
–2
–3
1.8 V
1
10
100
2
1
3V
0
L = 15 H
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
–2
–3
1000
2.2 V
–1
1
OUTPUT LOADING CURRENT, ILOAD/mA
1000
100
RIPPLE VOLTAGE, VRIPPLE/mVp–p
RIPPLE VOLTAGE, VRIPPLE/mVp–p
100
Figure 16. Output Voltage Change versus
Load Current
100
NO LOAD OPERATING CURRENT, I BATT/µA
10
OUTPUT LOADING CURRENT, ILOAD/mA
Figure 15. Output Voltage Change
versus Load Current
100 mA
80
150 mA
60
40
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
L = 22 H
20
0
1.8 V
1
2
1.5
2.5
150 mA
60
40
VOUT = 3.3 V
CIN = 10 µF
COUT = 33 µF
L = 15 H
20
0
3
100 mA
80
1
1.5
2
2.5
BATTERY INPUT VOLTAGE, VBATT/V
BATTERY INPUT VOLTAGE, VBATT/V
Figure 17. Battery Input Voltage versus
Output Ripple Voltage
Figure 18. Battery Input Voltage versus
Output Ripple Voltage
3
20
16
12
8
4
0
0
1
2
3
4
6
5
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 100 mA; L = 22 µH,
COUT = 33 µF)
INPUT VOLTAGE AT OUT PIN, VOUT/V
Upper Trace: Output Voltage Waveform, 1.0 V/Division
Lower Trace: Shutdown Pin Waveform, 1.0 V/Division
Figure 20. Startup Transient Response
Figure 19. No Load Operating Current
versus Input Voltage at OUT Pin
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NCP1417
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 100 mA; L = 22 µH,
COUT = 33 µF)
(VIN = 2.2 V, VOUT = 3.3 V, ILOAD = 30 mA; L = 22 µH,
COUT = 33 µF)
Upper Trace: Voltage at LX pin, 2.0 V/Division
Middle Trace: Output Voltage Ripple, 50 mV/Division
Lower Trace: Inductor Current, IL, 100 mA/Division
Upper Trace: Voltage at LX pin, 2.0 V/Division
Middle Trace: Output Voltage Ripple, 50 mV/Division
Lower Trace: Inductor Current, IL, 100 mA/Division
Figure 21. Continuous Conduction
Mode Switching Waveform
Figure 22. Discontinuous Conduction
Mode Switching Waveform
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
(VIN = 1.8 V, VOUT = 3.0 V, L = 22 µH, COUT = 33 µF)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Battery Voltage, VIN, 1.0 V/Division
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Figure 24. Load Transient Response for VIN = 1.8 V
Figure 23. Line Transient Response for VOUT = 3.3 V
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
(VOUT = 3.3 V, ILOAD = 10 mA to 100 mA; L = 22 µH,
COUT = 33 µF)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, ILOAD, 50 mA/Division
Figure 25. Load Transient Response for VIN = 2.4 V
Figure 26. Load Transient Response for VIN = 3.3 V
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NCP1417
DETAILED OPERATION DESCRIPTIONS
situation will occur. So dead time is introduced to make sure
M2 is completely turned OFF before M1 is being turned ON.
When the regulator is operating in DCM, as coil current
is dropped to zero, M2 is supposed to be OFF. Fail to do so,
reverse current will flow from the output bulk capacitor
through M2 and then the inductor to the battery input. It
causes damage to the battery. So the ZLC comparator comes
with fixed offset voltage to switch M2 OFF before any
reverse current builds up. However, if M2 is switch OFF too
early, large residue coil current flows through the body diode
of M2 and increases conduction loss. Therefore,
determination on the offset voltage is essential for optimum
performance.
With the implementation of synchronous rectification,
efficiency can be as high as 92%. For single cell input
voltage, use an external schottky diode such as MBR0520
connected from pin 7 to pin 8 to ensure quick start–up.
NCP1417 is a monolithic micropower high frequency
step–up voltage switching converter IC specially designed
for battery operated hand–held electronic products up to
200 mA loading. It integrates Synchronous Rectifier for
improving efficiency as well as eliminating the external
Schottky Diode. High switching frequency (up to 600 kHz)
allows low profile inductor and output capacitor being used.
Dual Low–Battery Detectors, Logic–Controlled Shutdown
and Cycle–by–Cycle Current Limit provide value–added
features for various battery–operated application. With all
these functions ON, the quiescent supply current is only
9 µA typical. This device is available in compact Micro8
package.
PFM Regulation Scheme
From the simplified Functional Diagram (Figure 2), the
output voltage is divided down and fed back to pin 1 (FB).
This voltage goes to the non–inverting input of the PFM
comparator whereas the comparator’s inverting input is
connected to REF. A switching cycle is initiated by the
falling edge of the comparator, at the moment, the main
switch (M1) is turned ON. After the maximum ON–time
(typical 1.4 S) elapses or the current limit is reached, M1
is turned OFF, and the synchronous switch (M2) is turned
ON. The M1 OFF time is not less than the minimum
OFF–time (typical 0.25 S), this is to ensure energy transfer
from the inductor to the output capacitor. If the regulator is
operating at continuous conduction mode (CCM), M2 is
turned OFF just before M1 is supposed to be ON again. If the
regulator is operating at discontinuous conduction mode
(DCM), which means the coil current will decrease to zero
before the next cycle, M1 is turned OFF as the coil current
is almost reaching zero. The comparator (ZLC) with fixed
offset is dedicated to sense the voltage drop across M2 as it
is conducting, when the voltage drop is below the offset, the
ZLC comparator output goes HIGH, and M2 is turned OFF.
Negative feedback of closed loop operation regulates
voltage at pin 1 (FB) equal to the internal voltage reference
(1.190 V).
Cycle–by–Cycle Current Limit
From Figure 2, SenseFET is applied to sample the coil
current as M1 is ON. With that sample current flowing
through a sense resistor, sense–voltage is developed.
Threshold detector (ILIM) detects whether the
sense–voltage is higher than preset level. If it happens,
detector output signifies the CONTROL LOGIC to switch
OFF M1, and M1 can only be switched ON as next cycle
starts after the minimum OFF–time (typical 0.25 S). With
properly sizing of SenseFET and sense resistor, the peak coil
current limit is set at 1.0 A typically.
Voltage Reference
The voltage at REF is set typically at +1.190 V. It can
deliver up to 2.5 mA with load regulation ±1.5%, at VOUT
equal to 3.3 V. If VOUT is increased, the REF load
capability can also be increased. A bypass capacitor of
0.15 F is required for proper operation when REF is not
loaded. If REF is loaded, 1.0 F capacitor at REF is needed.
Shutdown
The IC is shutdown when the voltage at pin 2
(LBI/SHDN) is pulled lower than 0.3 V via an open drain
transistor. During shutdown, M1 and M2 are both switched
OFF, however, the body diode of M2 allows current flow
from battery to the output, the IC internal circuit will
consume less than 0.05 A current typically. If the pin 2 pull
low is released, the IC will be enabled. The internal circuit
will only consume 9.0 A current typically from the OUT
pin.
Synchronous Rectification
Synchronous Rectifier is used to replace Schottky Diode
to eliminate the conduction loss contributed by forward
voltage drop of the latter. Synchronous Rectifier is normally
realized by powerFET with gate control circuitry which,
however, involved relative complicated timing concerns.
As main switch M1 is being turned OFF, if the
synchronous switch M2 is just turned ON with M1 not being
completed turned OFF, current will be shunt from the output
bulk capacitor through M2 and M1 to ground. This power
loss lowers overall efficiency. So a certain amount of dead
time is introduced to make sure M1 is completely OFF
before M2 is being turned ON.
When the main regulator is operating in CCM, as M2 is
being turned OFF, and M1 is just turned ON with M2 not
being completely turned OFF, the above mentioned
Dual Low–Battery Detection
Two comparators with 30 mV hysteresis are applied to
perform the dual low–battery detection function. When pin
2 (LBI) is at a voltage, which can be defined by a resistor
divider from the battery voltage, lower than the internal
reference voltage, 1.190 V, the first comparator, CP1 output
will cause a 50 low side switch to be turned ON. It will
pull down the voltage at pin 3 (LBO1) which has a hundreds
kilo–Ohm of pull–high resistance. If the pin 2 voltage is
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NCP1417
out of the capacitors multiplying with the Equivalent Series
Resistance (ESR) of the capacitor producing ripple voltage
at the terminals. During the syn–rect switch off cycle, the
charges stored in the output capacitor is used to sustain the
output load current. Load current at this period and the ESR
combined and reflected as ripple at the output terminal. For
all cases, the lower the capacitor ESR, the lower the ripple
voltage at output. As a general guide line, low ESR
capacitors should be used. Ceramic capacitors have the
lowest ESR, but low ESR tantalum capacitors can also be
used as a cost effective substitute.
higher than 1.190 V + 30 mV, the comparator output will
cause the 50 low side switch to be turned OFF, pin 3 will
become high impedance, and its voltage will be pulled high.
The second low–battery detector functions in the same
manner, the second comparator, CP2 with a lower triggering
reference point derived from the internal reference is used
instead, typical 0.944 V. This configuration provides two
levels of low battery warning to the target system.
APPLICATIONS INFORMATION
Output Voltage Setting
The output voltage of the converter is determined by the
external feedback network comprised of RFB1 and RFB2 and
the relationship is given by:
Optional Startup Schottky Diode for Low Battery
Voltage
In general operation, no external schottky diode is
required, however, in case you are intended to operate the
device close to 1.0 V level, a schottky diode connected
between the LX and OUT pins as shown in Figure 27 can
help during startup of the converter. The effect of the
additional schottky is shown in Figure 8.
R
VOUT 1.190 V 1 FB1
RFB2
where RFB1 and RFB2 are the upper and lower feedback
resistors respectively.
Low Battery Detect Level Setting
The Low Battery Detect Voltages of the converter are
determined by the external divider network comprised of
RLB1 and RLB2 and the relationship is given by:
L
MBR0520
VOUT
R
VLB1 1.190 V 1 LB1
RLB2
OUT
NCP1417
where RLB1 and RLB2 are the upper and lower divider
resistors respectively. By setting the VLB1, the second low
battery detection point, VLB2 will be fixed automatically.
COUT
LX
Inductor Selection
The NCP1417 is tested to produce optimum performance
with a 22 H inductor at VIN = 3.0 V, VOUT = 3.3 V
supplying output current up to 200 mA. For other
input/output requirements, inductance in the range 10 H to
47 H can be used according to end application
specifications. Selecting an inductor is a compromise
between output current capability and tolerable output
voltage ripple. Of course, the first thing we need to obey is
to keep the peak inductor current below its saturation limit
at maximum current and the ILIM of the device. In NCP1417,
ILIM is set at 1.0 A. As a rule of thumb, low inductance
values supply higher output current, but also increase the
ripple at output and reducing efficiency, on the other hand,
high inductance values can improve output ripple and
efficiency, however it also limit the output current capability
at the same time. One other parameter of the inductor is its
DC resistance, this resistance can introduce unwanted
power loss and hence reduce overall efficiency, the basic
rule is selecting an inductor with lowest DC resistance
within the board space limitation of the end application.
Figure 27.
PCB Layout Recommendations
Good PCB layout plays an important role in switching
mode power conversion. Careful PCB layout can help to
minimize ground bounce, EMI noise and unwanted
feedback that can affect the performance of the converter.
Hints suggested in below can be used as a guide line in most
situations.
Grounding
Star–ground connection should be used to connect the
output power return ground, the input power return ground
and the device power ground together at one point. All high
current running paths must be thick enough for current
flowing through and producing insignificant voltage drop
along the path. Feedback signal path must be separated with
the main current path and sensing directly at the anode of the
output capacitor.
Components Placement
Power components, i.e. input capacitor, inductor and
output capacitor, must be placed as close together as
possible. All connecting traces must be short, direct and
thick. High current flowing and switching paths must be
Capacitors Selection
In all switching mode boost converter applications,
both the input and output terminals sees impulsive
voltage/current waveforms. The currents flowing into and
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NCP1417
must be placed very close to the feedback (FB, pin 1) pin and
sensing the output voltage directly at the anode of the output
capacitor.
kept away from the feedback (FB, pin 1) terminal to avoid
unwanted injection of noise into the feedback path.
Feedback Network
Feedback of the output voltage must be a separate trace
detached from the power path. External feedback network
TYPICAL APPLICATION CIRCUIT
Input
1 V to
VOUT
CIN
10 µF
L
22 µH
CFR1
RFB2
200 K
308 K
RLB1
Shutdown
Open Drain
Input
56 nF
CSHDN
150 pF
RFB1
355 K
1
2
3
4
RLB2
330 K
NCP1417
FB
LBI/SHDN
OUT
LX
LB01
GND
REF
LB02
8
7
6
5
150 nF
CREF
33 µF
COUT
+
VOUT = 3.3 V/200 mA max.
Low Battery
Open Drain
Output 2
Low Battery
Open Drain
Output 1
Figure 28. Typical Application Schematic for 2 Alkaline Cells Supply
GENERAL DESIGN PROCEDURES
Design Parameters:
VIN = 1.8 V to 3.0 V, Typical 2.4 V
VOUT = 3.3 V
IOUT = 150 mA (200 mA max)
VLB1 = 2.3 V; VLB2 0.8 VLB1 = 1.84 V
VOUT–RIPPLE = 40 mVP–P at IOUT = 200 mA
Switch mode converter design is considered as black
magic to most engineers, some complicate empirical
formulae are available for reference usage. Those formulae
are derived from the assumption that the key components,
i.e. power inductor and capacitors are available with no
tolerance. Practically, its not true, the result is not a matter
of how accurate the equations you are using to calculate the
component values, the outcome is still somehow away from
the optimum point. Following, is a simple method based on
the most basic first order equations to estimate the inductor
and capacitor values for NCP1417 operating in Continuous
Conduction Mode. The component value set can be used as
a starting point to fine tune the circuit operation. By all
means, detail bench testing is needed to get the best
performance out of the circuit.
Calculate the feedback network:
Select RFB2 = 200 K
RFB1 RFB2
VVOUT
1
REF
RFB1 200 K 3.3 V 1 355 K
1.19 V
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NCP1417
With the feedback resistor divider, additional small
capacitor, CFB1 in parallel with RFB1 is required to ensure
stability. The value can be in between 68 nF to 220 nF, the
rule is to select the lowest capacitance to ensure stability.
Also a small capacitor, CFB2 in parallel with RFB2 may also
be needed to lower the feedback ripple hence improve
output ripple and regulation. In this example, only CFB1 is
used and the value is 150 nF.
I
ILAVG OUT 200 mA 275 mA
1 0.273
1D
Determine the peak inductor ripple current, IRIPPLE–P and
calculate the inductor value:
Assume IRIPPLE–P is 25% of ILAVG, the inductance of the
power inductor can be calculated as follows:
IRIPPLE–P 0.25 275 mA 68.8 mA
L
Calculate the Low Battery Detect divider:
VLB1 = 2.3 V
Select RLB2 = 330 K
RLB1 RLB2
Standard value of 22 H is selected for initial trial.
Determine the output voltage ripple, VOUT–RIPPLE and
calculate the output capacitor value:
VVLB1 1
REF
VIN tON
2.4 V 1.4 S
24.4 H
2(68.8 mA)
2IRIPPLE–P
VOUT RIPPLE 40 mVP–P at IOUT 200 mA
RLB1 330 K 2.3 V 1 308 K
1.19 V
COUT IOUT tON
VOUT–RIPPLE IOUT ESRCOUT
Once the VLB1 is set, the next low battery detection point,
VLB2 will be fixed automatically.
Determine the Steady State Duty Ratio, D for typical VIN,
operation will be optimized around this point:
where tON 1.4 S and ESRCOUT 0.15 ,
VOUT
1
1D
VIN
From above calculation, you need at least 28 F in order
to achieve the specified ripple level at conditions stated.
Practically, a one level larger capacitor will be used to
accommodate factors not take into account in the
calculation. So a capacitor value of 33 F is selected as
initial trial.
D1
COUT VIN
1 2.4 V 0.273
3.3 V
VOUT
Determine the average inductor current, ILAVG at maximum
IOUT:
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200 mA 1.4 S
28 F
40 mV 200 mA 0.15 NCP1417
PACKAGE DIMENSIONS
Micro8
DM SUFFIX
CASE 846A–02
ISSUE E
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT
EXCEED 0.15 (0.006) PER SIDE.
4. DIMENSION B DOES NOT INCLUDE INTERLEAD
FLASH OR PROTRUSION. INTERLEAD FLASH OR
PROTRUSION SHALL NOT EXCEED 0.25 (0.010)
PER SIDE.
–A–
–B–
K
PIN 1 ID
G
D 8 PL
0.08 (0.003)
–T–
M
T B
S
A
S
SEATING
PLANE
0.038 (0.0015)
C
H
L
J
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DIM
A
B
C
D
G
H
J
K
L
MILLIMETERS
MIN
MAX
2.90
3.10
2.90
3.10
--1.10
0.25
0.40
0.65 BSC
0.05
0.15
0.13
0.23
4.75
5.05
0.40
0.70
INCHES
MIN
MAX
0.114
0.122
0.114
0.122
--0.043
0.010
0.016
0.026 BSC
0.002
0.006
0.005
0.009
0.187
0.199
0.016
0.028
NCP1417
Notes
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NCP1417
Notes
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NCP1417
Micro8 is a trademark of International Rectifier
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NCP1417/D