INTERSIL ISL62381

ISL62381, ISL62382, ISL62383, ISL62381C,
ISL62382C, ISL62383C
Data Sheet
May 13, 2011
FN6665.5
High-Efficiency, Quad or Triple-Output
System Power Supply Controller for
Notebook Computers
Features
The ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C
and ISL62383C family of controllers generate supply voltages
for battery-powered systems. These controllers include two
pulse-width modulation (PWM) controllers, adjustable from
0.6V to 5.5V, and two linear regulators, LDO5 and LDO3, that
generate a fixed 5V and an adjustable output respectively, and
each can deliver up to 100mA. The ISL62383 and ISL62383C
have the same outputs as the ISL62381, ISL62382, ISL62381C
and ISL62382C but without LDO3 linear regulator. The channel
2 switching regulator will automatically take over the LDO5 load
when programmed to 5V output. This provides a large power
savings and boosts efficiency. These controllers include onboard power-up sequencing, two power-good (PGOOD)
outputs, digital soft-start, and internal soft-stop output discharge
that prevent negative voltages on shutdown.
• ±1% Output Voltage Accuracy: -10°C to +100°C
The patented R3 PWM control scheme provides a low jitter
• Thermal Monitor and Protection
• High Performance R3 Technology
• Fast Transient Response
system with fast response to load transients. Light-load
efficiency is improved with period-stretching discontinuous
conduction mode (DCM) operation. To eliminate noise in
audio frequency applications, an ultrasonic DCM mode is
included, which limits the minimum switching frequency to
approximately 28kHz.
The ISL62381, ISL62382, ISL62381C and ISL62382C are
available in a 32 Ld 5x5 TQFN package, and the ISL62383
and ISL62383C are available in a 28 Ld 4x4 TQFN package.
This family of controllers can operate over the extended
temperature range (-10°C to +100°C).
• Two Fully Programmable Switch-Mode Power Supplies with
Independent Operation
• Programmable Switching Frequency
• Integrated MOSFET Drivers and Bootstrap Diode
• Adjustable (+1.2V to +5V) LDO Output
• Fixed +5V LDO Output with Automatic Switchover to SMPS2
• Internal Soft-Start and Soft-Stop Output Discharge
• Wide Input Voltage Range: +5.5V to +25V
• Full and Ultrasonic Pulse-Skipping Mode
• Power-Good Indicator
• Overvoltage, Undervoltage and Overcurrent Protection
• Fault Identification by PGOOD Pull-Down Resistance
• Pb-Free (RoHS Compliant)
Applications
• Notebook and Sub-Notebook Computers
• PDAs and Mobile Communication Devices
• 3-Cell and 4-Cell Li+ Battery-Powered Devices
• General Purpose Switching Buck Regulators
Ordering Information
TEMP
RANGE
(°C)
PART
NUMBER
(Notes 1, 2, 3)
PART
MARKING
ISL62381HRTZ
62381 HRTZ
-10 to +100 32 Ld 5x5 TQFN L32.5x5A
ISL62382HRTZ
62382 HRTZ
-10 to +100 32 Ld 5x5 TQFN L32.5x5A
ISL62383HRTZ
623 83HRTZ
-10 to +100 28 Ld 4x4 TQFN L28.4x4
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL62381CHRTZ 62381 CHRTZ -10 to +100 32 Ld 5x5 TQFN L32.5x5A
ISL62382CHRTZ 62382 CHRTZ -10 to +100 32 Ld 5x5 TQFN L32.5x5A
ISL62383CHRTZ 62383 CHRTZ -10 to +100 28 Ld 4x4 TQFN L28.4x4
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on
reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pbfree material sets, molding compounds/die attach materials, and 100%
matte tin plate plus anneal (e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations). Intersil Pb-free products are MSL classified at Pb-free
peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information
page for ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C,
ISL62383C. For more information on MSL please see techbrief TB363.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2008, 2011. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Pinouts
ISL62383, ISL62383C
(28 LD TQFN)
TOP VIEW
FB2
VOUT2
ISEN2
OCSET2
EN2
PHASE2
UGATE2
BOOT2
FB2
VOUT2
ISEN2
OSCET2
EN2
PHASE2
UGATE2
ISL62381, ISL62382, ISL62381C, ISL62382C
(32 LD TQFN)
TOP VIEW
32
31
30
29
28
27
26
25
28
27
26
25
24
23
22
PGOOD2
1
21
BOOT2
PGND
FSET2
2
20
LGATE2
22
LDO5
FCCM
3
19
PGND
21
VIN
VCC2
4
18
LDO5
VCC1
5
17
VIN
FSET1
6
16
LGATE1
PGOOD1
7
15
BOOT1
PGOOD2
1
24
LGATE2
FSET2
2
23
FCCM
3
VCC2
4
GND
17
LGATE1
9
10
11
12
2
13
14
15
16
8
9
10
11
12
13
14
UGATE1
8
PHASE1
PGOOD1
EN1
LDO3FB
OCSET1
18
ISEN1
7
VOUT1
FSET1
BOOT1
LDO3
UGATE1
19
PHASE1
6
EN1
LDO3EN
OSCET1
LDO3IN
ISEN1
20
VOUT1
5
FB1
VCC1
FB1
GND
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Absolute Maximum Ratings
Thermal Information
VIN to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
VCC1,2, PGOOD1,2, LDO5 to GND. . . . . . . . . . . . . . -0.3V to +7.0V
EN1,2, LDO3EN . . . . . . . . . . . . . . . . . . -0.3V to GND, VCC1 + 0.3V
VOUT1,2, FB1,2, LDO3FB, FSET1,2 . . -0.3V to GND, VCC1 + 0.3V
PHASE1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . (DC) -0.3V to +28V
(<100ns Pulse Width, 10µJ) . . . . . . . . . . . . . . . . . . . . . . . . . -5.0V
BOOT1,2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT1,2 to PHASE1,2 . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
UGATE1,2 . . . . . . . . . . . (DC) -0.3V to PHASE1,2, BOOT1,2 + 0.3V
(<200ns Pulse Width, 20µJ) . . . . . . . . . . . . . . . . . . . . . . . . -4.0V
LGATE1,2 . . . . . . . . . . . . . . . . . . . (DC) -0.3V to GND, VCC1 + 0.3V
(<100ns Pulse Width, 4µJ) . . . . . . . . . . . . . . . . . . . . . . . . . . -2.0V
LDO3, LDO5 Output Continuous Current . . . . . . . . . . . . . . +100mA
Thermal Resistance (Typical, Notes 4, 5) JA (°C/W) JC (°C/W)
32 Ld TQFN Package . . . . . . . . . . . . .
30
1.75
28 Ld TQFN Package . . . . . . . . . . . . .
37
3
Junction Temperature Range. . . . . . . . . . . . . . . . . .-55°C to +150C
Operating Temperature Range . . . . . . . . . . . . . . . .-10C to +100C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65C to +150C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Ambient Temperature Range. . . . . . . . . . . . . . . . . .-10°C to +100°C
Supply Voltage (VIN to GND) . . . . . . . . . . . . . . . . . . . . 5.5V to 25V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
4. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379
5. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
These specifications apply for TA = -10°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C,
VIN = 12V. Boldface limits apply over the operating temperature range, -10°C to +100°C.
Electrical Specifications
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
Rising Threshold
5.3
5.4
5.5
V
Hysteresis
20
80
150
mV
PARAMETER
CONDITIONS
VIN
VIN Power-on Reset (POR)
VIN Shutdown Supply Current
EN1 = EN2 = GND or Floating, LDO3EN = GND
-
6
15
µA
VIN Standby Supply Current
EN1 = EN2 = GND or Floating, LDO3EN = VCC1
-
150
250
µA
I_LDO5 = 0
4.9
5.0
5.1
V
I_LDO5 = 100mA (Note 6)
4.9
5.0
5.1
V
LINEAR REGULATOR
LDO5 Output Voltage
LDO5 Short-Circuit Current (Note 6)
LDO5 = GND
-
190
-
mA
LDO5 UVLO Threshold Voltage (Note 6)
Rising edge of LDO5
-
4.35
-
V
Falling edge of LDO5
-
4.15
-
V
4.63
4.80
4.93
V
SMPS2 to LDO5 Switchover Resistance (Note 6) VOUT2 to LDO5, VOUT2 = 5V
-
2.5
3.2

LDO3 Reference Voltage (Note 6)
-
1.2
-
V
1.2
-
5
V
-
180
-
mA
SMPS2 to LDO5 Switchover Threshold
LDO3 Voltage Regulation Range
LDO3IN > VLDO3+dropout
LDO3 Short-Circuit Current (Note 6)
LDO3 = GND
LDO3EN Input Voltage
Rising edge
1.1
-
2.5
V
Falling edge
0.94
-
1.06
V
1
µA
LDO3EN Input Leakage Current
LDO3EN = GND or VCC1
-1
LDO3 Discharge ON-Resistance
LDO3EN = GND
-
36
60

VCC Input Bias Current (Note 6)
EN1 = EN2 = VCC1, FB1 = FB2 = 0.65V
-
2
-
mA
VCC1 Start-up Voltage
EN1 = EN2 = LDO3EN = GND
3.45
3.6
3.75
V
VCC
3
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
These specifications apply for TA = -10°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C,
VIN = 12V. Boldface limits apply over the operating temperature range, -10°C to +100°C. (Continued)
Electrical Specifications
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
Rising edge
4.35
4.45
4.55
V
Falling edge
4.10
4.20
4.30
V
-
0.6
-
V
PARAMETER
CONDITIONS
VCC2 POR Threshold
PWM
Reference Voltage (Note 6)
Regulation Accuracy
VOUT regulated to 0.6V
-1
-
1
%
FB Input Bias Current
FB = 0.6V
-10
-
30
nA
200
-
600
kHz
Frequency Range
Frequency Set Accuracy (Note 7)
FSW = 300kHz
-12
-
12
%
VOUT Voltage Regulation Range
VIN > 6V for VOUT = 5.5V
0.6
-
5.5
V
-
14
50

Soft-start, I_PGOOD = 5mA sinking
-
32
100

UVP, I_PGOOD = 5mA sinking
-
95
200

OVP, I_PGOOD = 5mA sinking
-
63
150

OCP, I_PGOOD = 5mA sinking
-
32
100

PGOOD = VCC1
-
0
1
µA
-
5
-
mA
From EN high to PGOOD high (for one SMPS
channel)
2.20
2.75
3.70
ms
EN2(1) = Floating, from EN1(2) high to
PGOOD2(1) high
4.50
5.60
7.50
ms
VOUT Soft-Discharge Resistance
POWER-GOOD
PGOOD Pull-Down Impedance
PGOOD Leakage Current
Maximum PGOOD Sink Current (Note 6)
PGOOD Soft-start Delay
GATE DRIVER
UGATE Pull-Up ON-Resistance (Note 6)
200mA source current
-
1.0
1.5

UGATE Source Current (Note 6)
UGATE-PHASE = 2.5V
-
2.0
-
A
UGATE Pull-Down ON-Resistance (Note 6)
250mA source current
-
1.0
1.5

UGATE Sink Current (Note 6)
UGATE-PHASE = 2.5V
-
2.0
-
A
LGATE Pull-Up ON-Resistance (Note6)
250mA source current
-
1.0
1.5

LGATE Source Current (Note 6)
LGATE-PGND = 2.5V
-
2.0
-
A
LGATE Pull-Down ON-Resistance (Note 6)
250mA source current
-
0.5
0.9

LGATE Sink Current (Note 6)
LGATE-PGND = 2.5V
-
4.0
-
A
UGATE to LGATE Deadtime (Note 6)
UG falling to LG rising, no load
-
21
-
ns
LGATE to UGATE Deadtime (Note 6)
LG falling to UG rising, no load
-
21
-
ns
Bootstrap Diode Forward Voltage (Note 6)
2mA forward diode current
-
0.58
-
V
Bootstrap Diode Reverse Leakage Current
VR = 25V
-
0.2
1
µA
Low level (DCM enabled)
-
-
0.8
V
Float level (DCM with audio filter)
1.9
-
2.1
V
High level (Forced CCM)
2.4
-
-
V
FCCM = GND or VCC1
-2
-
2
µA
CONTROL
FCCM Input Voltage
FCCM Input Leakage Current
4
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
These specifications apply for TA = -10°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C,
VIN = 12V. Boldface limits apply over the operating temperature range, -10°C to +100°C. (Continued)
Electrical Specifications
PARAMETER
CONDITIONS
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
Audio Filter Switching Frequency (Note 6)
FCCM floating
-
28
-
kHz
EN Input Voltage
Clear fault level/SMPS OFF level
-
-
0.8
V
Delay start level
1.9
-
2.1
V
SMPS ON level
2.4
-
-
V
EN Input Leakage Current
EN = GND or VCC1
-3.5
-
3.5
µA
ISEN Input Impedance (Note 6)
EN = VCC1
-
600
-
k
ISEN Input Leakage Current (Note 6)
EN = GND
-
0.1
-
µA
OCSET Input Impedance (Note 6)
EN = VCC1
-
600
-
k
OCSET Input Leakage Current (Note 6)
EN = GND
-
0.1
-
µA
OCSET Current Source
EN = VCC1
9
10.0
10.5
µA
-1.75
0.0
1.75
mV
PROTECTION
OCP (VOCSET-VISEN) Threshold
UVP Threshold
Falling edge, referenced to FB
81
84
87
%
OVP Threshold
Rising edge, referenced to FB
113
116
120
%
Falling edge, referenced to FB
99.5
103
106
%
Rising edge
-
150
-
°C
Falling edge
-
135
-
°C
OTP Threshold (Note 6)
NOTES:
6. Limits established by characterization and are not production tested.
7. FSW accuracy reflects IC tolerance only; it does not include frequency variation due to VIN, VOUT, LOUT, ESRCOUT, or other application specific
parameters.
8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
5
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Typical Application Circuits
The below typical application circuits generate the 5V/8A and 3.3V/8A main supplies in a notebook
computer. The input supply (VBAT) range is 5.5V to 25V
VBAT
4x10µF
V IN
BO O T1
BO O T2
0.22µF
0.22µF
4.7µH
3 .3 V
330µF
IRF7821
UGATE1
UGATE2
PHASE1
PHASE2
LG ATE1
LGATE2
IRF7821
0.022µF 14k
4.7µH
5V
0.022µF
330µF
14k
IRF7832
14k
ISL62381
ISL62382
ISL62383
ISL62381C
ISL62382C
ISL62383C
750
OCSET1
45.3k
IS E N 1
1200pF
VOUT1
FB1
IRF7832
14k
750
OCSET2
68.1k
IS E N 2
FB2
3 .3 V
10k
LDO 3*
9.09k
L D O 3 IN *
100k
17.4k
4.7µF
PGOOD1
LDO 3FB*
5V
PGOOD2
10k
4.7µF
LDO5
LDO 3EN *
FCCM
FSET1
FSET2
VCC1
VCC2
1µF
LDO5
100k
EN1
EN2
10
1µF
1200pF
VOUT2
PGND
0.01µF
24.3k
GND
19.6k
0.01µF
*ISL62381, ISL62382, ISL62381C,
AND ISL62382C ONLY
FIGURE 1. TYPICAL APPLICATION CIRCUIT WITH INDUCTOR DCR CURRENT SENSE
VBAT
BOOT1
4x10µF
0.22µF
0.001
330µF
1k
IRF7821
UGATE1
UGATE2
PHASE1
PHASE2
LG ATE1
LG ATE2
IRF7832
1k
750
1200pF
BOOT2
IRF7821
4.7µH
3 .3 V
V IN
4.7µH
0.001
IS E N 1
VOUT1
FB1
1k
1k
ISL62381
ISL62382 O C S E T 2
ISL62383
IS E N 2
ISL62381C
VOUT2
ISL62382C
ISL62383C
750
68.1k
LDO 3*
4.7µF
5V
100k
PGOOD1
LDO 3FB*
PGOOD2
10k
LDO 5
EN1
EN2
10
FCCM
FSET1
FSET2
VCC2
1µF
PGND
GND
24.3k
19.6k
*ISL62381, ISL62382, ISL62381C,
AND ISL62382C ONLY
LDO5
100k
LD O 3EN*
VCC1
1µF
9.09K
L D O 3 IN *
17.4k
4.7µF
1200pF
FB2
3 .3 V
10k
5V
330µF
IRF7832
OCSET1
45.3k
0.22µF
0.01µF
0.01µF
FIGURE 2. TYPICAL APPLICATION CIRCUIT WITH RESISTOR CURRENT SENSE
6
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Typical Application Circuits
The below typical application circuits generate the 1.05V/15A and 1.5V/15A main supplies in a
notebook computer. The input supply (VBAT) range is 5.5V to 25V
VBAT
6x10µF
V IN
BO O T1
BO O T2
0.22µF
0.22µF
IRF7821x2
2.2µH
1 .0 5 V
330µFx2
UGATE1
UGATE2
PHASE1
PHASE2
LG ATE1
LGATE2
IRF7821x2
IRF7832x2
36.5k
1800pF
FB1
36.5k
VBAT
L D O 3 IN *
PGOOD1
LDO 3FB*
PGOOD2
10k
LDO5
LDO 3EN *
FCCM
FSET1
FSET2
VCC1
VCC2
1µF
LDO5
100k
EN1
EN2
10
1µF
24.3k
100k
17.4k
4.7µF
1800pF
FB2
LDO 3*
5V
590
16.2k
3 .3 V
4.7µF
330µFx2
IRF7832x2
ISL62381
OCSET2
O C S E T 1 ISL62382
ISL62383
IS E N 2
IS E N 1
ISL62381C
VOUT1
VOUT2
ISL62382C
ISL62383C
16.2k
48.7k
1 .5 V
16.2k 0.022µF
0.022µF 16.2k
590
2.2µH
PGND
0.01µF
17.4k
GND
14k
*ISL62381, ISL62382, ISL62381C,
AND ISL62382C ONLY
0.01µF
FIGURE 3. TYPICAL APPLICATION CIRCUIT WITH INDUCTOR DCR CURRENT SENSE
VBAT
IRF7821x2
BO O T1
6x10µF
V IN
IRF7821x2
BOOT2
0.22µF
0.22µF
2.2µH
1 .0 5 V
0.001
330µFx2
UGATE1
UG ATE2
PHASE1
PHASE2
LG ATE1
LG ATE2
2.2µH
IRF7832x2
2k
0.001
330µFx2
IRF7832x2
2k
1 .5 V
2k
2k
ISL62381
590
1800pF
O C S E T 1 ISL62382
IS E N 1
36.5k
VOUT1
FB1
ISL62383
ISL62381C
ISL62382C
ISL62383C
590
OCSET2
36.5k
IS E N 2
FB2
3 .3 V
LDO 3*
48.7k
4.7µF
5V
PGOOD1
LDO 3FB*
PGOOD2
10k
LDO5
100k
EN1
EN2
10
LDO 3EN *
FCCM
FSET1
FSET2
VCC1
VCC2
1µF
24.3k
100k
LDO5
1µF
VBAT
L D O 3 IN *
17.4k
4.7µF
1800pF
VOUT2
PGND
GND
17.4k
14k
0.01µF
0.01µF
*ISL62381, ISL62382, ISL62381C
AND ISL62382C ONLY
FIGURE 4. TYPICAL APPLICATION CIRCUIT WITH RESISTOR CURRENT SENSE
7
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Pin Descriptions
PIN NUMBER
28 LD
32 LD
NAME
FUNCTION
1
1
2
2
FSET2
Frequency control input for SMPS2. Connect a resistor to ground to program the switching frequency. A small
ceramic capacitor such as 10nF is necessary to parallel with this resistor to smooth the voltage.
3
3
FCCM
Logic input to control efficiency mode. Logic high forces continuous conduction mode (CCM). Logic low allows full
discontinuous conduction mode (DCM). Float this pin for ultrasonic DCM operation.
4
4
VCC2
SMPS2 analog power supply input for reference voltages and currents. Connect to VCC1 with a 10 resistor.
Bypass to ground with a 1µF ceramic capacitor near the IC.
5
5
VCC1
SMPS1 analog power supply input for reference voltages and currents. It is internally connected to the LDO5
output. Bypass to ground with a 1µF ceramic capacitor near the IC.
-
6
6
7
7
8
8
9
FB1
9
10
VOUT1
SMPS1 output voltage sense input. Used for soft-discharge.
10
11
ISEN1
SMPS1 current sense input. Used for overcurrent protection and R3 regulation.
11
12
12
13
13
14
PHASE1 SMPS1 switching node for high-side gate drive return and synthetic ripple modulation. Connect to the switching
NMOS source, the synchronous NMOS drain, and the output inductor for SMPS1.
14
15
UGATE1 High-side NMOS gate drive output for SMPS1. Connect to the gate of the SMPS1 switching FET.
15
16
BOOT1
SMPS1 bootstrap input for the switching NMOS gate drivers. Connect to PHASE1 with a 0.22µF ceramic capacitor.
16
17
LGATE1
Low-side NMOS gate drive output for SMPS1. Connect to the gate of the SMPS1 synchronous FET.
-
18
LDO3FB LDO3 linear regulator feedback input used for output voltage programming and regulation.
-
19
LDO3
-
20
LDO3IN
17
21
VIN
18
22
LDO5
5V linear regulator output, providing up to 100mA before switchover to SMPS2. Bypass to ground with a 4.7µF
ceramic capacitor.
19
23
PGND
Power ground for SMPS1 and SMPS2. This provides a return path for synchronous FET switching currents.
20
24
LGATE2
Low-side NMOS gate drive output for SMPS2. Connect to the gate of the SMPS2 synchronous FET.
21
25
BOOT2
SMPS2 bootstrap input for the switching NMOS gate drivers. Connect to PHASE2 with a 0.22µF ceramic capacitor.
22
26
UGATE2 High-side NMOS gate drive output for SMPS2. Connect to the gate of the SMPS2 switching FET.
23
27
PHASE2 SMPS2 switching node for high-side gate drive return and synthetic ripple modulation. Connect to the switching
NMOS source, the synchronous NMOS drain, and the output inductor for SMPS2.
24
28
25
29
26
30
ISEN2
SMPS2 current sense input. Used for overcurrent protection and R3 regulation.
27
31
VOUT2
SMPS2 output voltage sense input. Used for soft-discharge and switchover to LDO5 output.
PGOOD2 SMPS2 open-drain power-good status output. Connect to LDO5 through a 100kΩ resistor. Output will be high when
the SMPS2 output is within the regulation window with no faults detected.
LDO3EN Logic input for enabling and disabling the LDO3 linear regulator. Positive logic input.
FSET1
Frequency control input for SMPS1. Connect a resistor to ground to program the switching frequency. A small
ceramic capacitor such as 10nF is necessary to parallel with this resistor to smooth the voltage.
PGOOD1 SMPS1 open-drain power-good status output. Connect to LDO5 through a 100kΩ resistor. Output will be high when
the SMPS1 output is within the regulation window with no faults detected.
SMPS1 feedback input used for output voltage programming and regulation.
OCSET1 Input from current-sensing network used to program the overcurrent shutdown threshold for SMPS1.
EN1
EN2
Logic input to enable and disable SMPS1. A logic high will enable SMPS1 immediately. A logic low will disable
SMPS1. Floating this input will delay SMPS1 start-up until after SMPS2 achieves regulation.
LDO3 linear regulator output, providing up to 100mA. Bypass to ground with a 4.7µF ceramic capacitor.
Power input for LDO3. Must be connected to a voltage greater than the LDO3 set point plus the dropout voltage.
Feed-forward input for line voltage transient compensation. Connect to the power train input voltage.
Logic input to enable and disable SMPS2. A logic high will enable SMPS2 immediately. A logic low will disable
SMPS2. Floating this input will delay SMPS2 start-up until after SMPS1 achieves regulation.
OCSET2 Input from current-sensing network used to program the over-current shutdown threshold for SMPS2.
8
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Pin Descriptions (Continued)
PIN NUMBER
28 LD
32 LD
NAME
FUNCTION
28
32
FB2
SMPS2 feedback input used for output voltage programming and regulation.
Bottom
Pad
Bottom
Pad
GND
Analog ground of the IC. Unless otherwise stated, signals are reference to this GND.
Typical Performance
100
95
100
VIN = 7V
95
90
90
VIN = 12V
85
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 7V
VIN = 19V
80
75
70
65
80
70
65
60
55
55
1.00
10.00
IOUT (A)
FIGURE 5. CHANNEL 1 EFFICIENCY AT VO = 3.3V, DEM
OPERATION. HIGH-SIDE 1xIRF7821,
rDS(ON) = 9.1m; LOW-SIDE 1xIRF7832,
rDS(ON) = 4m; L = 4.7µH, DCR = 14.3m; CCM
FSW = 270kHz
VIN = 19V
75
60
50
0.10
VIN = 12V
85
50
0.01
0.10
1.00
10.00
IOUT (A)
FIGURE 6. CHANNEL 2 EFFICIENCY AT VO = 5V, DEM
OPERATION. HIGH-SIDE 1xIRF7821,
rDS(ON) = 9.1m; LOW-SIDE 1xIRF7832,
rDS(ON) = 4m; L = 4.7µH, DCR = 14.3m; CCM
FSW = 330kHz
VO1
VO1
FB1
FB1
PGOOD1
PGOOD1
PHASE1
FIGURE 7. POWER-ON, VIN = 12V, LOAD = 5A, VO = 3.3V
9
PHASE1
FIGURE 8. POWER-OFF, VIN = 12V, IO = 5A, VO = 3.3V
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Typical Performance (Continued)
VO1
VO1
FB1
FB1
PGOOD1
PGOOD1
EN1
EN1
FIGURE 9. ENABLE CONTROL, EN1 = HIGH, VIN = 12V,
VO = 3.3V, IO = 5A
FIGURE 10. ENABLE CONTROL, EN1 = LOW, VIN = 12V,
VO = 3.3V, IO = 5A
VO1
VO1
PHASE1
PHASE1
VO2
PHASE2
FIGURE 11. CCM STEADY-STATE OPERATION,VIN = 12V,
VO1 = 3.3V, IO1 = 5A, VO2 = 5V, IO2 = 5A
VO1
VO2
PHASE2
FIGURE 12. DCM STEADY-STATE OPERATION,VIN = 12V,
VO1 = 3.3V, IO1 = 0. 2A, VO2 = 5V, IO2 = 0. 2A
VO1
PHASE1
PHASE1
VO2
PHASE2
FIGURE 13. AUDIO FILTER OPERATION, VIN = 12V,
VO1 = 3.3V, VO2 = 5V, NO LOAD
10
IO1
FIGURE 14. TRANSIENT RESPONSE, VIN = 12V, VO = 3.3V,
IO = 0.1A/8.1A @ 2.5A/µs
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Typical Performance (Continued)
VO1
VO1
PHASE1
IO1
FIGURE 15. LOAD INSERTION RESPONSE, VIN = 12V,
VO = 3.3V, IO = 0.1A/8.1A @ 2.5A/µs
PHASE1
IO1
FIGURE 16. LOAD RELEASE RESPONSE, VIN = 12V,
VO = 3.3V, IO = 0.1A/8.1A @ 2.5A/µs
EN1
EN2
VO1
VO1
VO2
FIGURE 17. DELAYED START, VIN = 12V, VO1 = 3.3V, VO2 = 5V,
EN2 = FLOAT, NO LOAD
VO1
PGOOD1
VO2
FIGURE 18. DELAYED START, VIN = 12V, VO1 = 3.3V, VO2 = 5V,
EN1 = FLOAT, NO LOAD
VO1
PGOOD1
IO1
VO2
PGOOD2
FIGURE 19. DELAYED START, VIN = 12V, VO1 = 3.3V, VO2 = 5V,
EN1 = 1, EN2 = FLOAT, NO LOAD
11
FIGURE 20. OVERCURRENT PROTECTION, VIN = 12V,
VO = 3.3V
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Typical Performance (Continued)
VO1
VO1
UGATE1-PHASE1
UGATE1-PHASE1
LGATE1
LGATE1
PGOOD1
PGOOD1
FIGURE 21. CROWBAR OVERVOLTAGE PROTECTION,
VIN = 12V, VO = 3.3V, NO LOAD
12
FIGURE 22. TRI-STATE OVERVOLTAGE PROTECTION,
VIN = 12V, VO = 3.3V, NO LOAD
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Block Diagram
VIN
VOUT2
FSET1/2
4.8V
5V
LDO
FB1/2
R3
MODULATOR
VREF
LDO5
0.6V
BOOT1/2
FCCM
PWM
VOUT1/2
UGATE
DRIVER
UGATE1/2
PHASE1/2
SOFT DISCHARGE
LGATE
DRIVER
LGATE1/2
PGND
EN1
PGOOD1/2
START-UP
AND
SHUTDOWN
LOGIC
LDO3EN*
VCC1/2
BIAS AND
REFERENCE
10µA
OCSET1/2
OCP
T-PAD
PROTECTION LOGIC
OVP/UVP/OCP/OTP
ISEN1/2
LDO3IN*
VREF + 16%
LDO3FB*
UVP
3.3V
LDO
FB1/2
LDO3*
OVP
VREF - 16%
THERMAL
MONITOR
SOFT DISCHARGE
* ISL62381, ISL2382, ISL62381C AND ISL62382C ONLY
13
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Theory of Operation
Four Output Controller
The ISL62381, ISL62382, ISL62381C and ISL62382C
generate four regulated output voltages, including two PWM
controllers and two LDOs. The two PWM channels are
identical and almost entirely independent, with the exception
of sharing the GND pin. Unless otherwise stated, only one
individual channel is discussed, and the conclusion applies
to both channels.
PWM Modulator
The ISL62381, ISL62382, ISL62383, ISL62381C,
ISL62382C and ISL62383C modulator features Intersil’s R3
technology, a hybrid of fixed frequency PWM control and
variable frequency hysteretic control. Intersil’s R3 technology
can simultaneously affect the PWM switching frequency and
PWM duty cycle in response to input voltage and output load
transients. The R3 modulator synthesizes an AC signal VR,
which is an analog representation of the output inductor
ripple current. The duty-cycle of VR is the result of charge
and discharge current through a ripple capacitor CR. The
current through CR is provided by a transconductance
amplifier gm that measures the VIN and VO pin voltages.
The positive slope of VR can be written as Equation 1:
(EQ. 1)
V RPOS = g m   V IN – V OUT   C R
The negative slope of VR can be written as Equation 2:
V RNEG = g m  V OUT  C R
(EQ. 2)
Where gm is the gain of the transconductance amplifier.
WINDOW VOLTAGE VW
(WRT VCOMP)
RIPPLE CAPACITOR VOLTAGE VR
ERROR AMPLIFIER
VOLTAGE VCOMP
PWM
FIGURE 23. MODULATOR WAVEFORMS DURING LOAD
TRANSIENT
A window voltage VW is referenced with respect to the error
amplifier output voltage VCOMP, creating an envelope into
which the ripple voltage VR is compared. The amplitude of
VW is set by a resistor connected across the FSET and GND
pins. The VR, VCOMP, and VW signals feed into a window
comparator in which VCOMP is the lower threshold voltage
and VCOMP + VW is the higher threshold voltage. Figure 23
14
shows PWM pulses being generated as VR traverses the
VCOMP and VCOMP + VW thresholds. The PWM switching
frequency is proportional to the slew rates of the positive and
negative slopes of VR; it is inversely proportional to the
voltage between VW and VCOMP. Equation 3 illustrates how
to calculate the window size based on output voltage and
frequency set resistor RW.
(EQ. 3)
V W = g m  V OUT   1 – D   R W
Programming the PWM Switching Frequency
These controllers do not use a clock signal to produce
PWMs. The PWM switching frequency FSW is programmed
by the resistor RW that is connected from the FSET pin to
the GND pin. The approximate PWM switching frequency
can be expressed as written in Equation 4:
1
F SW = --------------------------------10  C R  R W
(EQ. 4)
For a desired FSW, the RW can be selected by Equation 5.
1
R W = -----------------------------------10  C R  F SW
(EQ. 5)
where CR = 17pF with ±20% error range. To smooth the
FSET pin voltage, a ceramic capacitor such as 10nF is
necessary to parallel with RW.
It is recommended that whenever the control loop
compensation network is modified, FSW should be checked
for the correct frequency and if necessary, adjust RW .
Power-On Reset
These controllers are disabled until the voltage at the VIN
pin has increased above the rising power-on reset (POR)
threshold voltage. The controller will be disabled when the
voltage at the VIN pin decreases below the falling POR
threshold.
In addition to VIN POR, the LDO5 pin is also monitored. If its
voltage falls below 4.2V, the SMPS outputs will be shut
down. This ensures that there is sufficient BOOT voltage to
enhance the upper MOSFET.
EN, Soft-Start and PGOOD
These controllers use a digital soft-start circuit to ramp the
output voltage of each SMPS to the programmed regulation
setpoint at a predictable slew rate. The slew rate of the
soft-start sequence has been selected to limit the in-rush
current through the output capacitors as they charge to the
desired regulation voltage. When the EN pins are pulled
above their rising thresholds, the PGOOD Soft-Start Delay,
tSS, starts and the output voltage begins to rise. The FB pin
ramps to 0.6V in approximately 1.5ms and the PGOOD pin
goes to high impedance approximately 1.25ms after the FB
pin voltage reaches 0.6V.
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
1.5ms
VOUT
tSOFTSTART
VCC and LDO5
EN
FB
PGOOD
2.75ms
PGOOD Delay
FIGURE 24. SOFT-START SEQUENCE FOR ONE SMPS
The PGOOD pin indicates when the converter is capable of
supplying regulated voltage. It is an undefined impedance if VIN
is not above the rising POR threshold or below the POR falling
threshold. When a fault is detected, these controllers will turn
on the open-drain NMOS, which will pull PGOOD low with a
nominal impedance of 63 or 95 This will flag the system
that one of the output voltages is out of regulation.
Separate enable pins allow for full soft-start sequencing.
Because low shutdown quiescent current is necessary to
prolong battery life in notebook applications, the LDO5 5V LDO
is held off until any of the three enable signals (EN1, EN2 or
LDO3EN) is pulled high. Soft-start of all outputs will only start
until after LDO5 is above the 4.2V POR threshold. In addition to
user-programmable sequencing, these controllers include a
pre-programmed sequential SMPS soft-start feature. Table 1
shows the SMPS enable truth table.
After VIN is applied, the VCC1 start-up 3.6V voltage can be
used as the logic high signal of any of EN1, EN2 and LDO3EN
to enable PVCC if there is no other power supply on the board.
MOSFET Gate-Drive Outputs LGATE and UGATE
These controllers have internal gate-drivers for the high-side
and low-side N-Channel MOSFETs. The low-side gate-drivers
are optimized for low duty-cycle applications where the low-side
MOSFET conduction losses are dominant, requiring a low
r DS(ON) MOSFET. The LGATE pull-down resistance is small in
order to clamp the gate of the MOSFET below the VGS(th) at
turn-off. The current transient through the gate at turn-off can
be considerable because the gate charge of a low r DS(ON)
MOSFET can be large. Adaptive shoot-through protection
prevents a gate-driver output from turning on until the opposite
gate-driver output has fallen below approximately 1V. The
dead-time shown in Figure 25 is extended by the additional
period that the falling gate voltage stays above the 1V
threshold. The typical dead-time is 21ns. The high-side
gate-driver output voltage is measured across the UGATE and
PHASE pins while the low-side gate-driver output voltage is
measured across the LGATE and PGND pins. The power for
the LGATE gate-driver is sourced directly from the LDO5 pin.
The power for the UGATE gate-driver is sourced from a “boot”
capacitor connected across the BOOT and PHASE pins. The
boot capacitor is charged from the 5V LDO5 supply through a
“boot diode” each time the low-side MOSFET turns on, pulling
the PHASE pin low. These controllers have integrated boot
diodes connected from the LDO5 pins to BOOT pins.
TABLE 1. SMPS ENABLE SEQUENCE LOGIC
tLGFUGR
EN1
EN2
START-UP SEQUENCE
0
0
Both SMPS outputs OFF simultaneously
0
Float
Both SMPS outputs OFF simultaneously
Float
0
Both SMPS outputs OFF simultaneously
UGATE
Float
Float
Both SMPS outputs OFF simultaneously
LGATE
0
1
SMPS1 OFF, SMPS2 ON
1
0
SMPS1 ON, SMPS2 OFF
1
1
Both SMPS outputs ON simultaneously
Float
1
SMPS1 enables after SMPS2 is in regulation
1
Float
SMPS2 enables after SMPS1 is in regulation
tUGFLGR
50%
50%
FIGURE 25. LGATE AND UGATE DEAD-TIME
VCC1
The VCC1 nominal operation voltage is 5V. If EN1, EN2 and
LDO3EN are all logic low, the VCC1 start-up voltage is 3.6V
when VIN is applied on these controllers. LDO5 is held off
until any of the three enable signals (EN1, EN2 or LDO3EN)
is pulled high. When LDO5 is above the 4.2V VCC1 POR
threshold, VCC1 will switchover to LDO5 internally.
15
Diode Emulation
FCCM is a logic input that controls the power state of these
controllers. If forced high, these controllers will operate in
forced continuous-conduction-mode (CCM) over the entire
load range. This will produce the best transient response to
all load conditions, but will have increased light-load power
loss. If FCCM is forced low, these controllers will
automatically operate in diode-emulation-mode (DEM) at
light load to optimize efficiency in the entire load range. The
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
transition is automatically achieved by detecting the load
current and turning off LGATE when the inductor current
reaches 0A.
Positive-going inductor current flows from either the source
of the high-side MOSFET, or the drain of the low-side
MOSFET. Negative-going inductor current flows into the
drain of the low-side MOSFET. When the low-side MOSFET
conducts positive inductor current, the phase voltage will be
negative with respect to the GND and PGND pins.
Conversely, when the low-side MOSFET conducts negative
inductor current, the phase voltage will be positive with
respect to the GND and PGND pins. These controllers
monitor the phase voltage when the low-side MOSFET is
conducting inductor current to determine its direction.
When the output load current is greater than or equal to ½
the inductor ripple current, the inductor current is always
positive, and the converter is always in CCM. These
controllers minimize the conduction loss in this condition by
forcing the low-side MOSFET to operate as a synchronous
rectifier.
When the output load current is less than ½ the inductor
ripple current, negative inductor current occurs. Sinking
negative inductor current through the low-side MOSFET
lowers efficiency through unnecessary conduction losses.
These controllers automatically enter DEM after the PHASE
pin has detected positive voltage and LGATE was allowed to
go high for eight consecutive PWM switching cycles. These
controllers will turn off the low-side MOSFET once the phase
voltage turns positive, indicating negative inductor current.
These controllers will return to CCM on the following cycle
after the PHASE pin detects negative voltage, indicating that
the body diode of the low-side MOSFET is conducting
positive inductor current.
Efficiency can be further improved with a reduction of
unnecessary switching losses by reducing the PWM
frequency. It is characteristic of the R3 architecture for the
PWM frequency to decrease while in diode emulation. The
extent of the frequency reduction is proportional to the
reduction of load current. Upon entering DEM, the PWM
frequency makes an initial step-reduction because of a 33%
step-increase of the window voltage V W.
Because the switching frequency in DEM is a function of
load current, very light load conditions can produce
frequencies well into the audio band. This can be
problematic if audible noise is coupled into audio amplifier
circuits. To prevent this from occurring, these controllers
allow the user to float the FCCM input. This will allow DEM at
light loads, but will prevent the switching frequency from
going below ~28kHz to prevent noise injection into the audio
band. A timer is reset each PWM pulse. If the timer exceeds
30µs, LGATE is turned on, causing the ramp voltage to
reduce until another UGATE is commanded by the voltage
loop.
16
Overcurrent Protection
The overcurrent protection (OCP) setpoint is programmed
with resistor, ROCSET, that is connected across the OCSET
and PHASE pins.
L
DCR
IL
PHASE1
+
ROCSET
ISL62381
10µA
OCSET1
+ VROCSET
VDCR
CSEN
VO
_
CO
_
RO
ISEN1
FIGURE 26. OVERCURRENT-SET CIRCUIT
Figure 26 shows the overcurrent-set circuit for SMPS1. The
inductor consists of inductance L and the DC resistance
(DCR). The inductor DC current IL creates a voltage drop
across DCR, given by Equation 6:
V DCR = I L  DCR
(EQ. 6)
Theses controllers sink a 10µA current into the OCSET1 pin,
creating a DC voltage drop across the resistor ROCSET,
given by Equation 7:
V ROCSET = 10A  R OCSET
(EQ. 7)
Resistor RO is connected between the ISEN1 pin and the
actual output of the converter. During normal operation, the
ISEN1 pin is a high impedance path, therefore there is no
voltage drop across RO. The DC voltage difference between
the OCSET1 pin and the ISEN1 pin can be established using
Equation 8:
V OCSET1 – V ISEN1 = I L  DCR – 10A  R OCSET
(EQ. 8)
These controllers monitor the OCSET1 pin and the ISEN1 pin
voltages. Once the OCSET1 pin voltage is higher than the
ISEN1 pin voltage for more than 10µs, these controllers
declare an OCP fault. The value of ROCSET is then written as
Equation 9:
I OC  DCR
R OCSET = --------------------------10A
(EQ. 9)
Where:
- ROCSET () is the resistor used to program the
overcurrent setpoint
- IOC is the output current threshold that will activate the
OCP circuit
- DCR is the inductor DC resistance
For example, if IOC is 20A and DCR is 4.5m, the choice of
ROCSET is ROCSET = 20A x 4.5m/10µA = 9k
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Resistor ROCSET and capacitor CSEN form an RC network
to sense the inductor current. To sense the inductor current
correctly, not only in DC operation but also during dynamic
operation, the RC network time constant ROCSETCSEN
needs to match the inductor time constant L/DCR. The value
of CSEN is then written as Equation 10:
L
C SEN = ----------------------------------------R OCSET  DCR
(EQ. 10)
For example, if L is 1.5µH, DCR is 4.5m, and ROCSET is
9kthe choice of CSEN = 1.5µH/(9kx 4.5m) = 0.037µF
Upon converter start-up, the CSEN capacitor bias is 0V. To
prevent false OCP during this time, a 10µA current source
flows out of the ISEN1 pin, generating a voltage drop on the
RO resistor, which should be chosen to have the same
resistance as ROCSET. When PGOOD pin goes high, the
ISEN1 pin current source will be removed.
When an OCP fault is declared, the PGOOD pin will
pull-down to 32and latch off the converter. The fault will
remain latched until the EN pin has been pulled below the
falling EN threshold voltage, or until VIN has decayed below
the falling POR threshold.
When using a discrete current sense resistor, inductor
time-constant matching is not required. Equation 7 remains
unchanged, but Equation 8 is modified in Equation 11:
V OCSET1 – V ISEN1 = I L  R SENSE – 10A  R OCSET
(EQ. 11)
Furthermore, Equation 9 is changed in Equation 12:
I OC  R SENSE
R OCSET = ------------------------------------10A
(EQ. 12)
Where RSENSE is the series power resistor for sensing
inductor current. For example, with an RSENSE = 1m and
an OCP target of 10A, ROCSET = 1k
Overvoltage Protection
The OVP fault detection circuit triggers after the FB pin
voltage is above the rising overvoltage threshold for more
than 2µs. The FB pin voltage is 0.6V in normal operation.
The rising over voltage threshold is typically 116% of that
value, or 1.16*0.6V = 0.696V.
When an OVP fault is declared, the PGOOD pin will pull
down with 65and latch-off the converter. The OVP fault will
remain latched until the EN pin has been pulled below the
falling EN threshold voltage, or until VIN has decayed below
the falling POR threshold.
For ISL62381, ISL62381C, ISL62383 and ISL62383C,
although the converter has latched-off in response to an
OVP fault, the LGATE gate-driver output will retain the ability
to toggle the low-side MOSFET on and off in response to the
output voltage transversing the OVP rising and falling
thresholds. The LGATE gate-driver will turn on the low-side
MOSFET to discharge the output voltage, thus protecting the
17
load from potentially damaging voltage levels. The LGATE
gate-driver will turn off the low-side MOSFET once the FB
pin voltage is lower than the falling overvoltage threshold for
more than 2µs. The falling overvoltage threshold is typically
106% of the reference voltage, or 1.06*0.6V = 0.636V. This
process repeats as long as the output voltage fault is
present, allowing the ISL62381, ISL62381C, ISL62383 and
ISL62383C to protect against persistent overvoltage
conditions.
For ISL62382 and ISL62382C, if OVP is detected, it simply tristates the PHASE node by turning UGATE and LGATE off.
Undervoltage Protection
The UVP fault detection circuit triggers after the FB pin
voltage is below the undervoltage threshold for more than
2µs. The undervoltage threshold is typically 86% of the
reference voltage, or 0.86*0.6V = 0.516V. If a UVP fault is
declared, and the PGOOD pin will pull-down with 93and
latch-off the converter. The fault will remain latched until the
EN pin has been pulled below the falling enable threshold, or
if VIN has decayed below the falling POR threshold.
Programming the Output Voltage
When the converter is in regulation, there will be 0.6V
between the FB and GND pins. Connect a two-resistor
voltage divider across the OUT and GND pins with the
output node connected to the FB pin, as shown in Figure 27.
Scale the voltage-divider network such that the FB pin is
0.6V with respect to the GND pin when the converter is
regulating at the desired output voltage. The output voltage
can be programmed from 0.6V to 5.5V.
Programming the output voltage is written as Equation 13:
R TOP 

V OUT = V REF   1 + -----------------------------
R

BOTTOM
(EQ. 13)
Where:
- VOUT is the desired output voltage of the converter
- The voltage to which the converter regulates the FB pin
is the VREF (0.6V)
- RTOP is the voltage-programming resistor that connects
from the FB pin to the converter output. In addition to
setting the output voltage, this resistor is part of the loop
compensation network
- RBOTTOM is the voltage-programming resistor that
connects from the FB pin to the GND pin
Choose RTOP first when compensating the control loop, and
then calculate RBOTTOM according to Equation 14:
V REF  R
TOP
R BOTTOM = ------------------------------------V OUT – V REF
(EQ. 14)
Compensation Design
Figure 27 shows the recommended Type-II compensation
circuit. The FB pin is the inverting input of the error amplifier.
The COMP signal, the output of the error amplifier, is inside the
chip and unavailable to users. CINT is a 100pF capacitor
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
integrated inside the IC that connects across the FB pin and the
COMP signal. RTOP, RFB, CFB and CINT form the Type-II
compensator. The frequency domain transfer function is given
by Equation 15:
1 + s   R TOP + R FB   C
FB
G COMP  s  = ------------------------------------------------------------------------------------------s  R TOP  C INT   1 + s  R FB  C 
(EQ. 15)
FB
CINT = 100pF
CFB
RFB
RTOP
-
VO
FB
LDO3IN pin, which must be connected to a voltage greater than
the LDO3 output voltage plus the dropout voltage.
Currents in excess of the limit will cause the LDO3 voltage to
drop dramatically, limiting the power dissipation.
Thermal Monitor and Protection
LDO3 and LDO5 can dissipate non-trivial power inside these
controllers at high input-to-output voltage ratios and full load
conditions. To protect the silicon, these controllers continually
monitor the die temperature. If the temperature exceeds
+150°C, all outputs will be turned off to sharply curtail power
dissipation. The outputs will remain off until the junction
temperature has fallen below +135°C.
EA
General Application Design Guide
RBOTTOM
COMP
+
REF
ISL6238
FIGURE 27. COMPENSATION REFERENCE CIRCUIT
The LC output filter has a double pole at its resonant frequency
that causes rapid phase change. The R3 modulator used in
these controllers make the LC output filter resemble a first order
system in which the closed loop stability can be achieved with
the recommended Type-II compensation network. Intersil
provides a PC-based tool (example page is shown later) that
can be used to calculate compensation network component
values and help simulate the loop frequency response.
LDO5 Linear Regulator
In addition to the two SMPS outputs, these controllers also
provide two linear regulator outputs. LDO5 is fixed 5V LDO
output capable of sourcing 100mA continuous current.
When the output of SMPS2 is programmed to 5V, SMPS2 will
automatically take over the load of LDO5. This provides a
large power savings and boosts the efficiency. After
switchover to SMPS2, the LDO5 output current plus the
MOSFET drive current should not exceed 100mA in order to
guarantee the LDO5 output voltage in the range of 5V ±5%.
The total MOSFET drive current can be estimated by
Equation 16.
(EQ. 16)
I DRIVE = Q g  F SW
where Qg is the total gate charge of all the power MOSFET
in two SMPS regulators. Then the LDO5 output load current
should be less than 100mA-IDRIVE.
LDO3 Linear Regulator
ISL62381, ISL62381C, ISL62382 and ISL62382C include
LDO3 linear regulator whose output is adjustable from 1.2V to
5V through LDO3FB pin with a 1.2V reference voltage. It can
be independently enabled from both SMPS channels. Logic
high of LDO3EN will enable LDO3. LDO3 is capable of
sourcing 100mA continuous current and draws its power from
This design guide is intended to provide a high-level
explanation of the steps necessary to design a single-phase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced in the
following section. In addition to this guide, Intersil provides
complete reference designs that include schematics, bills of
materials, and example board layouts.
Selecting the LC Output Filter
The duty cycle of an ideal buck converter is a function of the
input and the output voltage. This relationship is written as
Equation 17:
V OUT
D = --------------V IN
(EQ. 17)
The output inductor peak-to-peak ripple current is written as
Equation 18:
V OUT   1 – D 
I PP = -------------------------------------F SW  L
(EQ. 18)
A typical step-down DC/DC converter will have an IP-P of
20% to 40% of the maximum DC output load current. The
value of IP-P is selected based upon several criteria such as
MOSFET switching loss, inductor core loss, and the resistive
loss of the inductor winding. The DC copper loss of the
inductor can be estimated by Equation 19:
P COPPER = I LOAD
2

DCR
(EQ. 19)
Where ILOAD is the converter output DC current.
The copper loss can be significant so attention has to be given
to the DCR selection. Another factor to consider when choosing
the inductor is its saturation characteristics at elevated
temperatures. A saturated inductor could cause destruction of
circuit components, as well as nuisance OCP faults.
A DC/DC buck regulator must have output capacitance CO
into which ripple current IP-P can flow. Current IP-P develops
out of the capacitor. These two voltages are written as
Equation 20:
V ESR = I PP  E SR
(EQ. 20)
and Equation 21:
18
FN6665.5
May 13, 2011
I PP
V C = ------------------------------8  CO  F
(EQ. 21)
SW
If the output of the converter has to support a load with high
pulsating current, several capacitors will need to be paralleled
to reduce the total ESR until the required VP-P is achieved.
The inductance of the capacitor can cause a brief voltage dip
if the load transient has an extremely high slew rate. Low
inductance capacitors should be considered in this scenario.
A capacitor dissipates heat as a function of RMS current and
frequency. Be sure that IP-P is shared by a sufficient quantity
of paralleled capacitors so that they operate below the
maximum rated RMS current at FSW. Take into account that
the rated value of a capacitor can fade as much as 50% as
the DC voltage across it increases.
0.6
0.48
k=1
k = 0.75
k = 0.5
k = 0.25
k=0
0.36
0.24
0.12
0
0
0.1
0.3
0.2
0.4
0.5
0.6
0.7
0.8
0.9
1.0
DUTY CYCLE
Selection of the Input Capacitor
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and capable of
supplying the RMS current required by the switching circuit.
Their voltage rating should be at least 1.25 times greater
than the maximum input voltage, while a voltage rating of 1.5
times is a preferred rating. Figure 28 is a graph of the input
capacitor RMS ripple current, normalized relative to output
load current, as a function of duty cycle and is adjusted for
converter efficiency. The normalized RMS ripple current
calculation is written as Equation 22:
2
Dk
I MAX  D   1 – D  + -------------12
I C  RMS ,NORMALIZED  = ----------------------------------------------------------------------I MAX
IN
(EQ. 22)
Where:
- IMAX is the maximum continuous ILOAD of the converter
- k is a multiplier (0 to 1) corresponding to the inductor
peak-to-peak ripple amplitude expressed as a
percentage of IMAX (0% to 100%)
- D is the duty cycle that is adjusted to take into account
the efficiency of the converter which is written as:
V OUT
D = -------------------------V IN  EFF
NORMALIZED INPUT RMS RIPPLE CURRENT
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
(EQ. 23)
In addition to the bulk capacitance, some low ESL ceramic
capacitance is recommended to decouple between the drain
of the high-side MOSFET and the source of the low-side
MOSFET.
FIGURE 28. NORMALIZED RMS INPUT CURRENT @ EFF = 1
MOSFET Selection and Considerations
Typically, a MOSFET cannot tolerate even brief excursions
beyond their maximum drain to source voltage rating. The
MOSFETs used in the power stage of the converter should
have a maximum VDS rating that exceeds the sum of the
upper voltage tolerance of the input power source and the
voltage spike that occurs when the MOSFET switches off.
There are several power MOSFETs readily available that are
optimized for DC/DC converter applications. The preferred
high-side MOSFET emphasizes low gate charge so that the
device spends the least amount of time dissipating power in
the linear region. Unlike the low-side MOSFET which has the
drain-source voltage clamped by its body diode during turn
off, the high-side MOSFET turns off with a VDS of
approximately VIN - VOUT, plus the spike across it. The
preferred low-side MOSFET emphasizes low r DS(ON) when
fully saturated to minimize conduction loss. It should be
noted that this is an optimal configuration of MOSFET
selection for low duty cycle applications (D < 50%). For
higher output, low input voltage solutions, a more balanced
MOSFET selection for high- and low-side devices may be
warranted.
For the low-side (LS) MOSFET, the power loss can be
assumed to be conductive only and is written as Equation 24:
2
P CON_LS  I LOAD  r DS  ON _LS   1 – D 
(EQ. 24)
For the high-side (HS) MOSFET, the its conduction loss is
written as Equation 25:
P CON_HS = I LOAD
2

r DS  ON _HS  D
(EQ. 25)
For the high-side MOSFET, the switching loss is written as
Equation 26:
V IN  I PEAK  t OFF  f
V IN  I VALLEY  t ON  f
SW
SW
P SW_HS = ----------------------------------------------------------------- + ------------------------------------------------------------2
2
(EQ. 26)
19
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Where:
- IVALLEY is the difference of the DC component of the
inductor current minus 1/2 of the inductor ripple current
- IPEAK is the sum of the DC component of the inductor
current plus 1/2 of the inductor ripple current
- tON is the time required to drive the device into
saturation
- tOFF is the time required to drive the device into cut-off
Co
PIN 4 (VCC2)
L2
PIN 21 (VIN)
L2
ISL62381
AND ISL62382
Ci
LINE OF SYMMETRY
Selecting The Bootstrap Capacitor
Ci
The selection of the bootstrap capacitor is written as
Equation 27:
Qg
C BOOT = -----------------------V BOOT
L1
(EQ. 27)
Where:
- Qg is the total gate charge required to turn on the
high-side MOSFET
- VBOOT, is the maximum allowed voltage decay across
the boot capacitor each time the high-side MOSFET is
switched on
As an example, suppose the high-side MOSFET has a total
gate charge Qg, of 25nC at VGS = 5V, and a VBOOT of
200mV. The calculated bootstrap capacitance is 0.125µF; for
a comfortable margin, select a capacitor that is double the
calculated capacitance. In this example, 0.22µF will suffice.
Use an X7R or X5R ceramic capacitor.
Layout Considerations
As a general rule, power should be on the bottom layer of
the PCB and weak analog or logic signals are on the top
layer of the PCB. The ground-plane layer should be adjacent
to the top layer to provide shielding. The ground plane layer
should have an island located under the IC, the
compensation components, and the FSET components. The
island should be connected to the rest of the ground plane
layer at one point.
VIAS TO
VIAS
TO
GROUND
GROUND
PLANE
PLANE
GND
VOUT
INDUCTOR
INDUCTOR
HIGH-SIDE
HIGH-SIDE
MOSFETS
MOSFETS
PHASE
NODE
OUTPUT
OUTPUT
CAPACITORS
CAPACITORS
SCHOTTKY
SCHOTTKY
DIODE
DIODE
LOW-SIDE
LOW-SIDE
MOSFETS
MOSFETS
INPUT
INPUT
CAPACITORS
CAPACITORS
VIN
U2
PGND PLANE
PHASE PLANES
VOUT PLANES
VIN PLANE
U1
L1
Co
FIGURE 30. SYMMETRIC LAYOUT GUIDE
Signal Ground and Power Ground
The bottom of these controllers TQFN package is the signal
ground (GND) terminal for analog and logic signals of the IC.
Connect the GND pad of these controllers to the island of
ground plane under the top layer using several vias for a
robust thermal and electrical conduction path. Connect the
input capacitors, the output capacitors, and the source of the
lower MOSFETs to the power ground (PGND) plane.
The following pin descriptions use ISL62381 as an example.
PGND (Pin 23)
This is the return path for the pull-down of the LGATE
low-side MOSFET gate driver. Ideally, PGND should be
connected to the source of the low-side MOSFET with a
low-resistance, low-inductance path.
VIN (Pin 21)
The VIN pin should be connected close to the drain of the
high-side MOSFET, using a low resistance and low
inductance path.
VCC (Pins 4 and 5)
For best performance, place the decoupling capacitor very
close to the VCC and GND pins.
LDO5 (Pin 22)
For best performance, place the decoupling capacitor very
close to the LDO5 and respective PGND pin, preferably on
the same side of the PCB as the ISL62381 IC.
FIGURE 29. TYPICAL POWER COMPONENT PLACEMENT
EN (Pins 13 and 28) and PGOOD (Pins 1 and 8)
Because there are two SMPS outputs and only one PGND
pin, the power train of both channels should be laid out
symmetrically. The line of bilateral symmetry should be
drawn through pins 4 and 21 (pins 4 and 18 for ISL62383).
This layout approach ensures that the controller does not
favor one channel over another during critical switching
decisions. Figure 30 illustrates one example of how to
achieve proper bilateral symmetry.
20
These are logic signals that are referenced to the GND pin.
Treat as a typical logic signal.
OCSET (Pins 12 and 29) and ISEN (Pins 11 and 30)
For DCR current sensing, current-sense network, consisting
of ROCSET and CSEN, needs to be connected to the
inductor pads for accurate measurement. Connect ROCSET
to the phase-node side pad of the inductor, and connect
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
CSEN to the output side pad of the inductor. The ISEN
resistor should also be connected to the output pad of the
inductor with a separate trace. Connect the OCSET pin to
the common node of node of ROCSET and CSEN.
For resistive current sensing, connect ROCSET from the
OCSET pin to the inductor side of the resistor pad. The ISEN
resistor should be connected to the VOUT side of the resistor
pad.
In both current-sense configurations, the resistor and
capacitor sensing elements, with the exclusion of the current
sense power resistor, should be placed near the
corresponding IC pin. The trace connections to the inductor
or sensing resistor should be treated as Kelvin connections.
FB (Pins 9 and 32), and VOUT (Pins 10 and 31)
The VOUT pin is used to generate the R3 synthetic ramp
voltage and for soft-discharge of the output voltage during
shutdown events. This signal should be routed as close to
the regulation point as possible. The input impedance of the
FB pin is high, so place the voltage programming and loop
compensation components close to the VOUT, FB, and GND
pins keeping the high impedance trace short.
FSET (Pins 2 and 7)
These pins require a quiet environment. The resistor RFSET
and capacitor CFSET should be placed directly adjacent to
these pins. Keep fast moving nodes away from these pins.
LGATE (Pins 17 and 24)
The signal going through these traces are both high dv/dt
and high di/dt, with high peak charging and discharging
current. Route these traces in parallel with the trace from the
PGND pin. These two traces should be short, wide, and
away from other traces. There should be no other weak
signal traces in proximity with these traces on any layer.
BOOT (Pins 16 and 25), UGATE (Pins 15 and 26), and
PHASE (Pins 14 and 27)
The signals going through these traces are both high dv/dt
and high di/dt, with high peak charging and discharging
current. Route the UGATE and PHASE pins in parallel with
short and wide traces. There should be no other weak signal
traces in proximity with these traces on any layer.
Copper Size for the Phase Node
The parasitic capacitance and parasitic inductance of the
phase node should be kept very low to minimize ringing. It is
best to limit the size of the PHASE node copper in strict
accordance with the current and thermal management of the
application. An MLCC should be connected directly across
the drain of the upper MOSFET and the source of the lower
MOSFET to suppress the turn-off voltage spike.
21
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Package Outline Drawing
L28.4x4
28 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 9/06
A
4 . 00
2 . 50
PIN #1 INDEX AREA
CHAMFER 0 . 400 X 45°
0 . 40
22
28
1
0 . 40
15
3 . 20
2 . 50
4 . 00
21
0 . 4 x 6 = 2.40 REF
B
PIN 1
INDEX AREA
7
0 . 10
2X
14
8
0 . 20 ±0 . 05
0 . 10 M C A B
0 . 4 x 6 = 2 . 40 REF
TOP VIEW
3 . 20
BOTTOM VIEW
SEE DETAIL X''
0 . 10 C
(3 . 20)
C
PACKAGE BOUNDARY
MAX. 0 . 80
(28X 0 . 20)
SEATING PLANE
0 . 00 - 0 . 05
0 . 08 C
0 . 20 REF
(3 . 20)
(2 . 50)
SIDE VIEW
(0 . 40)
(0 . 40)
C
0 . 20 REF
5
0 ~ 0 . 05
(2 . 50)
(28X 0 . 60)
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Controlling dimensions are in mm.
Dimensions in ( ) for reference only.
2. Unless otherwise specified, tolerance : Decimal ±0.05
Angular ±2°
3. Dimensioning and tolerancing conform to AMSE Y14.5M-1994.
4. Bottom side Pin#1 ID is diepad chamfer as shown.
5. Tiebar shown (if present) is a non-functional feature.
22
FN6665.5
May 13, 2011
ISL62381, ISL62382, ISL62383, ISL62381C, ISL62382C, ISL62383C
Thin Quad Flat No-Lead Plastic Package (TQFN)
Thin Micro Lead Frame Plastic Package (TMLFP)
L32.5x5A
2X
0.15 C A
D
A
32 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220WJJD-1 ISSUE C)
D/2
MILLIMETERS
2X
6
INDEX
AREA
N
0.15 C B
1
2
3
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.70
0.75
0.80
-
A1
-
-
0.05
-
0.30
5, 8
3.55
7, 8
A3
E/2
b
E
D
D2
B
TOP VIEW
0.20 REF
0.18
5.00 BSC
3.30
C
0.08 C
SEATING PLANE
A3
SIDE VIEW
A1
3.45
-
E
5.00 BSC
-
5.75 BSC
9
3.30
e
/ / 0.10 C
-
E1
E2
A
0.25
3.45
3.55
0.50 BSC
7, 8
-
k
0.20
-
-
-
L
0.30
0.40
0.50
8
N
32
2
Nd
8
3
Ne
8
3
Rev. 2 05/06
NX b
5
0.10 M C A B
D2
NX k
D2
2
(DATUM B)
8
7
N
(DATUM A)
6
INDEX
AREA
E2
E2/2
3
2
1
NX L
N
7
(Ne-1)Xe
REF.
8
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5m-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
e
8
(Nd-1)Xe
REF.
BOTTOM VIEW
A1
NX b
5
SECTION "C-C"
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
23
FN6665.5
May 13, 2011