ISL95870, ISL95870A, ISL95870B The ISL95870, ISL95870A, ISL95870B ICs are Single-Phase Synchronous-Buck PWM regulators featuring Intersil’s proprietary R4 Technology™. The wide 3.3V to 25V input voltage range is ideal for systems that run on battery or AC-adapter power sources. The ISL95870A and ISL95870B are low-cost solutions for applications requiring dynamically selected slew-rate controlled output voltages. The soft-start and dynamic setpoint slew-rates are capacitor programmed. Voltage identification logic-inputs select four (ISL95870A, ISL95870B) resistor-programmed setpoint reference voltages that directly set the output voltage of the converter between 0.5V and 1.5V, and up to 5V with a feedback voltage divider. Compared with R3 modulator, the R4 modulator has equivalent light-load efficiency, faster transient performance, accurately regulated frequency control and all internal compensation. These updates, together with integrated MOSFET drivers and schottky bootstrap diode, allow for a high-performance regulator that is highly compact and needs few external components. The differential remote sensing for output voltage and selectable switching frequency are another two new functions. For maximum efficiency, the converter automatically enters diode-emulation mode (DEM) during light-load conditions such as system standby. Features • Input Voltage Range: 3.3V to 25V • Output Voltage Range: 0.5V to 5V • Precision Regulation - Proprietary R4™ Frequency Control Loop - ±0.5% System Accuracy Over -10°C to +100°C • Optimal Transient Response - Intersil’s R4™ Modulator Technology • Output Remote Sense • Extremely Flexible Output Voltage Programmability - 2-Bit VID Selects Four Independent Setpoint Voltages for ISL95870B - 2-Bit VID Selects Four Dependent or Three Independent Setpoint Voltages for ISL95870A - Simple Resistor Programming of Setpoint Voltages • Selectable 300kHz, 500kHz, 600kHz or 1MHz PWM Frequency in Continuous Conduction • Automatic Diode Emulation Mode for Highest Efficiency • Power-Good Monitor for Soft-Start and Fault Detection Applications*(see page 26) • Mobile PC Graphical Processing Unit VCC Rail • Mobile PC I/O Controller Hub (ICH) VCC Rail • Mobile PC Memory Controller Hub (GMCH) VCC Rail RVCC +5V CVCC VCC PGOOD QHS 12 11 LO 10 VOUT 0.5V TO 5V QLS 9 VO CBOOT 8 ROFS CIN ROCSET PHASE SREF VIN 3.3V TO 25V RPGOOD 13 14 PVCC LGATE EN 5 CSOFT ROFS1 4 UGATE OCSET GPIO RTN 7 3 BOOT FB 2 GND 6 1 RFB1 FSEL RTN1 PGND 16 15 CPVCC CO CSEN RTN1 RO RFB 0 FIGURE 1. ISL95870 APPLICATION SCHEMATIC WITH ONE OUTPUT VOLTAGE SETPOINT AND DCR CURRENT SENSE December 22, 2009 FN6899.0 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2009. All Rights Reserved All other trademarks mentioned are the property of their respective owners. ISL95870, ISL95870A, ISL95870B PWM DC/DC Controller with VID Inputs for Portable GPU Core-Voltage Regulator ISL95870, ISL95870A, ISL95870B Applications Schematics: ISL95870 RVCC +5V CVCC CPVCC 9 FSEL UGATE VOUT 0.5V TO 5V LO PHASE QLS PGOOD CO ROCSET 4 QHS BOOT CBOOT 8 10 RPGOOD VCC 13 PVCC 14 LGATE 3 5 CSOFT ROFS1 SREF 11 FB GPIO 2 CIN CSEN VO EN 12 7 RTN 1 6 GND RFB1 OCSET RTN1 15 16 PGND VIN 3.3V TO 25V RTN1 RO RFB 0 ROFS FIGURE 2. ISL95870 APPLICATION SCHEMATIC WITH ONE OUTPUT VOLTAGE SETPOINT AND DCR CURRENT SENSE RVCC +5V CVCC CPVCC 9 QHS UGATE LO RSEN PHASE VOUT 0.5V TO 5V QLS PGOOD CBOOT VO ROCSET 4 CIN BOOT 8 10 RPGOOD VCC 13 14 PVCC LGATE 3 7 CSOFT ROFS1 SREF 11 6 GPIO 2 FB EN 12 OCSET RTN 1 5 GND RFB1 FSEL RTN1 15 16 PGND VIN 3.3V TO 25V CO CSEN RTN1 RO RFB ROFS 0 FIGURE 3. ISL95870 APPLICATION SCHEMATIC WITH ONE OUTPUT VOLTAGE SETPOINT AND RESISTOR CURRENT SENSE 2 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Applications Schematics: ISL95870A RVCC RFB1 PVCC 19 3 18 4 17 RTN VID1 GPIO 5 16 6 15 VID0 14 8 13 9 FB SET1 7 RSET2 RSET3 12 CIN QHS BOOT UGATE LO PHASE VOUT 0.5V TO 5V QLS EN PGOOD CBOOT FSEL VO CO CSEN RTN1 RO RFB CSOFT RSET1 11 SET0 10 SREF CVCC OCSET ROFS1 RTN1 2 VIN 3.3V TO 25V ROCSET GND VCC RPGOOD PGND 20 CPVCC 1 LGATE +5V ROFS 0 FIGURE 4. ISL95870A APPLICATION SCHEMATIC WITH FOUR OUTPUT VOLTAGE SETPOINTS AND DCR CURRENT SENSE RVCC RFB1 PVCC 19 3 18 4 17 RTN VID1 5 16 6 15 VID0 SET0 14 8 9 13 FB SET1 7 RSET2 RSET3 CSOFT RSET1 11 SREF 10 GPIO 12 CVCC CIN QHS BOOT UGATE VOUT 0.5V TO 5V RSEN LO PHASE EN QLS PGOOD FSEL CBOOT VO CO CSEN RTN1 RO OCSET ROFS1 RTN1 2 VIN 3.3V TO 25V ROCSET GND VCC RPGOOD PGND 20 CPVCC 1 LGATE +5V RFB ROFS 0 FIGURE 5. ISL95870A APPLICATION SCHEMATIC WITH FOUR OUTPUT VOLTAGE SETPOINTS AND RESISTOR CURRENT SENSE 3 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Applications Schematics: ISL95870B RPGOOD VCC 13 5 12 6 RSET3 UGATE VOUT 0.5V TO 5V LO PHASE QLS EN PGOOD CO CBOOT FSEL CSEN RTN1 10 7 11 QHS BOOT ROCSET 18 GND 4 CIN RO RFB ROFS CSOFT 17 LGATE 14 RSET4 RSET2 SET1 3 CVCC VO SET0 15 FB RSET1 2 9 SREF 16 OCSET VID0 VIN 3.3V TO 25V 1 8 VID1 GPIO 19 20 ROFS1 RTN SET2 RFB1 RTN1 PGND CPVCC PVCC RVCC +5V 0 FIGURE 6. ISL95870B APPLICATION SCHEMATIC WITH FOUR OUTPUT VOLTAGE SETPOINTS AND DCR CURRENT SENSE RPGOOD VCC 17 12 6 RSET3 FB QHS BOOT LO UGATE RSEN PHASE QLS EN PGOOD CBOOT FSEL VOUT 0.5V TO 5V CO CSEN RTN1 10 7 11 CIN ROCSET 18 13 5 RSET4 CSOFT RSET2 SET1 14 GND 4 CVCC VO SET0 3 SET2 RSET1 15 9 SREF 2 OCSET VID0 16 8 VID1 GPIO VIN 3.3V TO 25V 1 ROFS ROFS1 RTN 19 RFB1 RTN1 LGATE 20 PGND CPVCC PVCC RVCC +5V RFB RO 0 FIGURE 7. ISL95870B APPLICATION SCHEMATIC WITH FOUR OUTPUT VOLTAGE SETPOINTS AND RESISTOR CURRENT SENSE 4 FN6899.0 December 22, 2009 Block Diagram VCC POR SOFT-START CIRCUITRY BOOT EN 5 DRIVER PHASE DEAD-TIME GENERATION FB INTERNAL COMPENSATION AMPLIFIER SREF PVCC OVERVOLTAGE/ UNDERVOLTAGE DRIVER + LGATE PGND *SET 0 R4 *SET 1 MODULATOR VO **SET2 *VID1 REFERENCE VOLTAGE CIRCUITRY REMOTE SENSE CIRCUITRY OVERCURRENT *VID0 Fs SELECTION CIRCUITRY GND *ISL95870A, ISL95870B ONLY FN6899.0 December 22, 2009 RTN FSEL **ISL95870B ONLY FIGURE 8. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM OF ISL95870, ISL95870A, ISL95870B OCSET ISL95870, ISL95870A, ISL95870B PGOOD CIRCUITRY PGOOD UGATE ISL95870, ISL95870A, ISL95870B Pin Configurations 20 PVCC 13 VCC 14 PVCC 15 LGATE 16 PGND PGND 2 1 LGATE ISL95870A (20 LD 3.2X1.8 ΜTQFN) TOP VIEW ISL95870 (16 LD 2.6X1.8 ΜTQFN) TOP VIEW 19 VCC GND 3 18 BOOT RTN 2 11 UGATE RTN 4 17 UGATE EN 3 10 PHASE VID1 5 16 PHASE VID0 6 15 EN SREF 7 14 PGOOD SET0 8 13 FSEL SET1 9 12 VO VO 8 OCSET 7 9 PGOOD FB 6 17 VCC 18 PVCC 19 LGATE 20 PGND ISL95870B (20 LD 3X4 QFN) TOP VIEW RTN 1 16 BOOT VID1 2 15 UGATE VID0 3 6 14 PHASE GND 12 PGOOD SET1 6 11 FSEL VO 10 SET0 5 OCSET 9 13 EN FB 8 SREF 4 SET2 7 FSEL 5 SREF 4 OCSET 11 12 BOOT FB 10 GND 1 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B ISL95870 Functional Pin Descriptions PIN NUMBER SYMBOL DESCRIPTION 1 GND IC ground for bias supply and signal reference. 2 RTN Negative remote sense input of VOUT. If resistor divider consisting of RFB and ROFS is used at FB pin, the same resistor divider should be used at RTN pin, i.e. keep RFB1=RFB, and ROFS1=ROFS. 3 EN Enable input for the IC. Pulling EN above the rising threshold voltage initializes the soft-start sequence. 4 SREF Soft-start and voltage slew-rate programming capacitor input. Connects internally to the inverting input of the VSET voltage setpoint amplifier. 5 FSEL Input for programming the regulator switching frequency. Pull this pin to VCC for 1MHz switching. Pull this pin to GND with a 100kΩ resistor for 600kHz switching. Leave this pin floating for 500kHz switching. Pull this pin directly to GND for 300kHz switching. 6 FB 7 OCSET Input for the overcurrent detection circuit. The overcurrent setpoint programming resistor ROCSET connects from this pin to the sense node. 8 VO Output voltage sense input for the R4 modulator. The VO pin also serves as the reference input for the overcurrent detection circuit. 9 PGOOD Power-good open-drain indicator output. This pin changes to high impedance when the converter is able to supply regulated voltage. 10 PHASE Return current path for the UGATE high-side MOSFET driver, VIN sense input for the R4 modulator, and inductor current polarity detector input. 11 UGATE High-side MOSFET gate driver output. Connect to the gate terminal of the high-side MOSFET of the converter. 12 BOOT Positive input supply for the UGATE high-side MOSFET gate driver. The BOOT pin is internally connected to the cathode of the Schottky boot-strap diode. Connect an MLCC between the BOOT pin and the PHASE pin. 13 VCC Input for the IC bias voltage. Connect +5V to the VCC pin and decouple with at least a MLCC to the GND pin. 14 PVCC Input for the LGATE and UGATE MOSFET driver circuits. The PVCC pin is internally connected to the anode of the Schottky boot-strap diode. Connect +5V to the PVCC pin and decouple with a MLCC to the PGND pin. 15 LGATE Low-side MOSFET gate driver output. Connect to the gate terminal of the low-side MOSFET of the converter. 16 PGND Return current path for the LGATE MOSFET driver. Connect to the source of the low-side MOSFET. Voltage feedback sense input. Connects internally to the inverting input of the control-loop error amplifier. The converter is in regulation when the voltage at the FB pin equals the voltage on the SREF pin. 7 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B ISL95870A Functional Pin Descriptions PIN NUMBER SYMBOL DESCRIPTION 1 LGATE Low-side MOSFET gate driver output. Connect to the gate terminal of the low-side MOSFET of the converter. 2 PGND Return current path for the LGATE MOSFET driver. Connect to the source of the low-side MOSFET. 3 GND IC ground for bias supply and signal reference. 4 RTN Negative remote sense input of VOUT. If resistor divider consisting of RFB and ROFS is used at FB pin, the same resistor divider should be used at RTN pin, i.e. keep RFB1=RFB, and ROFS1=ROFS. 5 VID1 Logic input for setpoint voltage selector. Use in conjunction with the VID0 pin to select among four setpoint reference voltages. 6 VID0 Logic input for setpoint voltage selector. Use in conjunction with the VID1 pin to select among four setpoint reference voltages. 7 SREF Soft-start and voltage slew-rate programming capacitor input and setpoint reference voltage programming resistor input. Connects internally to the inverting input of the VSET voltage setpoint amplifier. 8 SET0 Voltage set-point programming resistor input. 9 SET1 Voltage set-point programming resistor input. 10 FB Voltage feedback sense input. Connects internally to the inverting input of the control-loop error transconductance amplifier. The converter is in regulation when the voltage at the FB pin equals the voltage on the SREF pin. 11 OCSET Input for the overcurrent detection circuit. The overcurrent setpoint programming resistor ROCSET connects from this pin to the sense node. 12 VO Output voltage sense input for the R4 modulator. The VO pin also serves as the reference input for the overcurrent detection circuit. 13 FSEL Input for programming the regulator switching frequency. Pull this pin to VCC for 1MHz switching. Pull this pin to GND with a 100kΩ resistor for 600kHz switching. Leave this pin floating for 500kHz switching. Pull this pin directly to GND for 300kHz switching. 14 PGOOD Power-good open-drain indicator output. This pin changes to high impedance when the converter is able to supply regulated voltage. 15 EN 16 PHASE Return current path for the UGATE high-side MOSFET driver, VIN sense input for the R4 modulator, and inductor current polarity detector input. 17 UGATE High-side MOSFET gate driver output. Connect to the gate terminal of the high-side MOSFET of the converter. 18 BOOT Positive input supply for the UGATE high-side MOSFET gate driver. The BOOT pin is internally connected to the cathode of the Schottky boot-strap diode. Connect an MLCC between the BOOT pin and the PHASE pin. 19 VCC Input for the IC bias voltage. Connect +5V to the VCC pin and decouple with at least a MLCC to the GND pin. 20 PVCC Input for the LGATE and UGATE MOSFET driver circuits. The PVCC pin is internally connected to the anode of the Schottky boot-strap diode. Connect +5V to the PVCC pin and decouple with a MLCC to the PGND pin. Enable input for the IC. Pulling EN above the rising threshold voltage initializes the soft-start sequence. 8 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B ISL95870B Functional Pin Descriptions PIN NUMBER SYMBOL DESCRIPTION 1 RTN Negative remote sense input of VOUT. If resistor divider consisting of RFB and ROFS is used at FB pin, the same resistor divider should be used at RTN pin, i.e. keep RFB1=RFB, and ROFS1=ROFS. 2 VID1 Logic input for setpoint voltage selector. Use in conjunction with the VID0 pin to select among four setpoint reference voltages. 3 VID0 Logic input for setpoint voltage selector. Use in conjunction with the VID1 pin to select among four setpoint reference voltages. 4 SREF Soft-start and voltage slew-rate programming capacitor input and setpoint reference voltage programming resistor input. Connects internally to the inverting input of the VSET voltage setpoint amplifier. 5 SET0 Voltage set-point programming resistor input. 6 SET1 Voltage set-point programming resistor input. 7 SET2 Voltage set-point programming resistor input. 8 FB Voltage feedback sense input. Connects internally to the inverting input of the control-loop error transconductance amplifier. The converter is in regulation when the voltage at the FB pin equals the voltage on the SREF pin. 9 OCSET Input for the overcurrent detection circuit. The overcurrent setpoint programming resistor ROCSET connects from this pin to the sense node. 10 VO Output voltage sense input for the R4 modulator. The VO pin also serves as the reference input for the overcurrent detection circuit. 11 FSEL Input for programming the regulator switching frequency. Pull this pin to VCC for 1MHz switching. Pull this pin to GND with a 100kΩ resistor for 600kHz switching. Leave this pin floating for 500kHz switching. Pull this pin directly to GND for 300kHz switching. 12 PGOOD Power-good open-drain indicator output. This pin changes to high impedance when the converter is able to supply regulated voltage. 13 EN Enable input for the IC. Pulling EN above the rising threshold voltage initializes the soft-start sequence. 14 PHASE Return current path for the UGATE high-side MOSFET driver, VIN sense input for the R4 modulator, and inductor current polarity detector input. 15 UGATE High-side MOSFET gate driver output. Connect to the gate terminal of the high-side MOSFET of the converter. 16 BOOT Positive input supply for the UGATE high-side MOSFET gate driver. The BOOT pin is internally connected to the cathode of the Schottky boot-strap diode. Connect an MLCC between the BOOT pin and the PHASE pin. 17 VCC Input for the IC bias voltage. Connect +5V to the VCC pin and decouple with at least a MLCC to the GND pin. 18 PVCC Input for the LGATE and UGATE MOSFET driver circuits. The PVCC pin is internally connected to the anode of the Schottky boot-strap diode. Connect +5V to the PVCC pin and decouple with a MLCC to the PGND pin. 19 LGATE Low-side MOSFET gate driver output. Connect to the gate terminal of the low-side MOSFET of the converter. 20 PGND Return current path for the LGATE MOSFET driver. Connect to the source of the low-side MOSFET. Bottom Pad GND IC ground for bias supply and signal reference. 9 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Ordering Information PART NUMBER (Note 2) PART MARKING ISL95870HRUZ-T (Notes 1, 4) GAV TEMP RANGE (°C) PACKAGE (Pb-Free) PKG. DWG. # -10 to +100 16 Ld 2.6x1.8 µTQFN L16.2.6x1.8A ISL95870AHRUZ-T (Notes 1, 4) GAW -10 to +100 20 Ld 3.2x1.8 µTQFN L20.3.2x1.8 ISL95870BHRZ (Note 3) 870B -10 to +100 20 Ld 3x4 QFN L20.3x4 ISL95870BHRZ-T (Notes 1, 3) 870B -10 to +100 20 Ld 3x4 QFN L20.3x4 ISL95870IRUZ-T (Notes 1, 4) GAZ -40 to +100 16 Ld 2.6x1.8 µTQFN L16.2.6x1.8A ISL95870AIRUZ-T (Notes 1, 4) GAX -40 to +100 20 Ld 3.2x1.8 µTQFN L20.3.2x1.8 ISL95870BIRZ (Note 3) 870I -40 to +100 20 Ld 3x4 QFN L20.3x4 ISL95870BIRZ-T (Notes 1, 3) 870I -40 to +100 20 Ld 3x4 QFN L20.3x4 NOTES: 1. Please refer to TB347 for details on reel specifications. 2. For Moisture Sensitivity Level (MSL), please see device information page for ISL95870, ISL95870A, ISL95870B. For more information on MSL please see techbrief TB363. 3. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 4. These Intersil Pb-free plastic packaged products employ special Pb-free material sets; molding compounds/die attach materials and NiPdAu plate - e4 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 10 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Table of Contents Applications Schematics: ISL95870 .................................................................................................... 2 Applications Schematics: ISL95870A.................................................................................................. 3 Applications Schematics: ISL95870B.................................................................................................. 4 Block Diagram .................................................................................................................................... 5 ISL95870 Functional Pin Descriptions ................................................................................................ 7 ISL95870A Functional Pin Descriptions .............................................................................................. 8 ISL95870B Functional Pin Descriptions .............................................................................................. 9 Absolute Maximum Ratings .............................................................................................................. 12 Thermal Information ........................................................................................................................ 12 Recommended Operating Conditions ................................................................................................ 12 Electrical Specifications ..................................................................................................................... 12 Theory of Operation.......................................................................................................................... 15 Power-On Reset .............................................................................................................................. Start-Up Timing .............................................................................................................................. Start-Up and Voltage-Step Operation for ISL95870.............................................................................. Start-Up and Voltage-Step Operation for ISL95870A, ISL95870B........................................................... Output Voltage Programming for ISL95870......................................................................................... Output Voltage Programming for ISL95870A ....................................................................................... Output Voltage Programming for ISL95870B ....................................................................................... High Output Voltage Programming ..................................................................................................... R4 Modulator.................................................................................................................................. Stability ......................................................................................................................................... Transient Response ......................................................................................................................... Diode Emulation.............................................................................................................................. Overcurrent.................................................................................................................................... Overvoltage ................................................................................................................................... Undervoltage.................................................................................................................................. Over-Temperature........................................................................................................................... PGOOD Monitor............................................................................................................................... Integrated MOSFET Gate-Drivers ....................................................................................................... Adaptive Shoot-Through Protection.................................................................................................... 15 15 15 15 16 16 17 19 19 19 20 20 20 21 21 21 22 22 22 General Application Design Guide ..................................................................................................... 22 Selecting the LC Output Filter ........................................................................................................... Selecting the Input Capacitor ............................................................................................................ Selecting the Bootstrap Capacitor ...................................................................................................... Driver Power Dissipation .................................................................................................................. MOSFET Selection and Considerations ................................................................................................ Layout Considerations ...................................................................................................................... 22 23 23 23 24 24 Revision History ............................................................................................................................... 26 Products ........................................................................................................................................... 26 L16.2.6x1.8A ..................................................................................................................................... 27 L20.3.2x1.8 ........................................................................................................................................ 28 L20.3x4.............................................................................................................................................. 29 11 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Absolute Maximum Ratings Thermal Information VCC, PVCC, PGOOD, FSEL to GND . . . . . . . . -0.3V to +7.0V VCC, PVCC to PGND . . . . . . . . . . . . . . . . . . -0.3V to +7.0V GND to PGND . . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V EN, SET0, SET1, SET2, VO, VID0, VID1, FB, RTN, OCSET, SREF-0.3V to GND, VCC + 0.3V BOOT Voltage (VBOOT-GND) . . . . . . . . . . . . . . . -0.3V to 33V BOOT To PHASE Voltage (VBOOT-PHASE) . . . -0.3V to 7V (DC) -0.3V to 9V (<10ns) PHASE Voltage . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 28V GND -8V (<20ns Pulse Width, 10µJ) UGATE Voltage . . . . . . . . . . . . VPHASE - 0.3V (DC) to VBOOT VPHASE - 5V (<20ns Pulse Width, 10µJ) to VBOOT LGATE Voltage . . . . . . . . . . GND - 0.3V (DC) to VCC + 0.3V . . . . . . GND - 2.5V (<20ns Pulse Width, 5µJ) to VCC + 0.3V ESD Rating Human Body Model . . . . . . . . . . . . . . . . . . . . . . . . . 2kV Machine Model . . . . . . . . . . . . . . . . . . . . . . . . . . . 200V Charged Device Model . . . . . . . . . . . . . . . . . . . . . . . 1kV Latch Up . . . . . . . . . . . . . . . . JEDEC Class II Level A at +125°C Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W) 16 Ld µTQFN (Note 5). . . . . . . . . . 90 N/A 20 Ld µTQFN (Note 5). . . . . . . . . . 88 N/A 20 Ld QFN (Notes 6, 7) . . . . . . . . . 44 5 Junction Temperature Range . . . . . . . . . . . -55°C to +150°C Operating Temperature Range: For “H” Version Parts . . . . . . . . . . . . . . . . -10°C to +100°C For “I” Version Parts . . . . . . . . . . . . . . . . -40°C to +100°C Storage Temperature . . . . . . . . . . . . . . . . -65°C to +150°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . .see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Ambient Temperature Range: For “H” Version Parts . . . . . . . . For “I” Version Parts . . . . . . . . Converter Input Voltage to GND VCC, PVCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -10°C to +100°C -40°C to +100°C . . . . 3.3V to 25V . . . . . . . 5V ±5% CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 5. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 6. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 7. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications PARAMETER All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply over the operating temperature range, -40°C to +100°C, unless otherwise stated. MIN MAX (Note 11) TYP (Note 11) UNIT SYMBOL TEST CONDITIONS VCC Input Bias Current IVCC EN = 5V, VCC = 5V, FB = 0.55V, SREF < FB - 1.2 1.9 mA VCC Shutdown Current IVCCoff EN = GND, VCC = 5V - 0 1.0 µA PVCC Shutdown Current IPVCCoff EN = GND, PVCC = 5V - 0 1.0 µA VCC and PVCC VCC POR THRESHOLD Rising VCC POR Threshold Voltage VVCC_THR 4.40 4.52 4.60 V Falling VCC POR Threshold Voltage V 4.10 4.22 4.35 V VID0 = VID1 = VCC, PWM Mode = CCM (For “H” Version Parts, TA = -10°C to +100°C) -0.5 - +0.5 % VID0 = VID1 = VCC, PWM Mode = CCM -0.75 +0.5 % VCC_THF REGULATION System Accuracy PWM Switching Frequency Accuracy FSW PWM Mode = CCM (For “H” Version Parts, TA = -10°C to +100°C) -15 - +15 % PWM Mode = CCM -22 - +15 % EN = 5V - 600 - kΩ VENTHR < EN, SREF = Soft-Start Mode - 8.5 - µA VO VO Input Impedance RVO VO Reference Offset Current IVOSS 12 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Electrical Specifications All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply over the operating temperature range, -40°C to +100°C, unless otherwise stated. (Continued) PARAMETER SYMBOL VO Input Leakage Current IVOoff TEST CONDITIONS EN = GND, VO = 3.6V MIN MAX (Note 11) TYP (Note 11) UNIT - 0 - µA ERROR AMPLIFIER FB Input Bias Current IFB EN = 5V, FB = 0.50V -20 - +50 nA ISS SREF = Soft-Start Mode 8.5 17 25.5 µA ±51 85 ±119 µA IVS SREF = Setpoint-Stepping Mode (For “H” Version Parts, TA = -10°C to +100°C) SREF = Setpoint-Stepping Mode ±46 ±85 ±127 µA SREF (Note 8) Soft-Start Current Voltage Step Current POWER GOOD PGOOD Pull-down Impedance RPG PGOOD = 5mA Sink - 50 150 Ω PGOOD Leakage Current IPG PGOOD = 5V - 0.1 1.0 µA GATE DRIVER UGATE Pull-Up Resistance (Note 9) RUGPU 200mA Source Current - 1.1 1.7 Ω UGATE Source Current (Note 9) IUGSRC UGATE - PHASE = 2.5V - 1.8 - A UGATE Sink Resistance (Note 9) RUGPD 250mA Sink Current - 1.1 1.7 Ω UGATE Sink Current (Note 9) IUGSNK UGATE - PHASE = 2.5V - 1.8 - A LGATE Pull-Up Resistance (Note 9) RLGPU 250mA Source Current - 1.1 1.7 Ω LGATE Source Current (Note 9) ILGSRC LGATE - GND = 2.5V - 1.8 - A LGATE Sink Resistance (Note 9) RLGPD 250mA Sink Current - 0.55 1.0 Ω LGATE Sink Current (Note 9) ILGSNK LGATE - PGND = 2.5V - 3.6 - A UGATE to LGATE Deadtime tUGFLGR UGATE falling to LGATE rising, no load - 21 - ns LGATE to UGATE Deadtime tLGFUGR LGATE falling to UGATE rising, no load - 21 - ns RPHASE - 33 - kΩ PHASE PHASE Input Impedance BOOTSTRAP DIODE Forward Voltage VF PVCC = 5V, IF = 2mA - 0.58 - V Reverse Leakage IR VR = 25V - 0 - µA CONTROL INPUTS EN High Threshold Voltage VENTHR 2.0 - - V EN Low Threshold Voltage VENTHF - - 1.0 V 0.85 1.7 2.55 µA - 0 1.0 µA EN Input Bias Current IEN EN Leakage Current IENoff EN = 5V EN = GND VID<0,1> High Threshold Voltage (Note 10) VVIDTHR 0.65 - - V VID<0,1> Low Threshold Voltage (Note 10) VVIDTHF - - 0.5 V EN = 5V - 0.5 - µA EN=0V - 0 - µA VID<0,1> Input Bias Current (Note 10) IVID VID<0,1> Leakage Current (Note 10) IVIDoff PROTECTION 13 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Electrical Specifications PARAMETER All typical specifications TA = +25°C, VCC = 5V. Boldface limits apply over the operating temperature range, -40°C to +100°C, unless otherwise stated. (Continued) SYMBOL OCP Threshold Voltage VOCPTH OCP Reference Current IOCP TEST CONDITIONS MIN MAX (Note 11) TYP (Note 11) UNIT VOCSET - VO -1.75 - 1.75 mV EN = 5.0V (For “H” Version Parts, TA = -10°C to +100°C) 7.65 8.5 9.35 µA EN = 5.0V 7.05 8.5 9.35 µA OCSET Input Resistance ROCSET EN = 5.0V - 600 - kΩ OCSET Leakage Current IOCSET EN = GND - 0 - µA UVP Threshold Voltage VUVTH VFB = %VSREF 81 84 87 % 113 116 120 % VFB = %VSREF 112.5 116 120 % VFB = %VSREF 98 102 106 % OVP Rising Threshold Voltage VOVRTH VFB = %VSREF (For “H” Version Parts, TA = -10°C to +100°C) OVP Falling Threshold Voltage VOVFTH OTP Rising Threshold Temperature (Note 9) TOTRTH - 150 - °C OTP Hysteresis (Note 9) TOTHYS - 25 - °C NOTES: 8. For ISL95870,there is one internal reference 0.5V. For ISL95870A, ISL95870B, there are four resistor-programmed reference voltages. 9. Limits established by characterization and are not production tested. 10. VID function is only for ISL95870A, ISL95870B. 11. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. 14 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Theory of Operation Where: The following sections will provide a detailed description of the inner workings of the ISL95870, ISL95870A, ISL95870B. Power-On Reset The IC is disabled until the voltage at the VCC pin has increased above the rising power-on reset (POR) threshold voltage VVCC_THR. The controller will become disabled when the voltage at the VCC pin decreases below the falling POR threshold voltage VVCC_THF. The POR detector has a noise filter of approximately 1µs. Start-Up Timing Once VCC has ramped above VVCC_THR, the controller can be enabled by pulling the EN pin voltage above the input-high threshold VENTHR. Approximately 20µs later, the voltage at the SREF pin begins slewing to the designated VID set-point. The converter output voltage at the FB feedback pin follows the voltage at the SREF pin. During soft-start, The regulator always operates in CCM until the soft-start sequence is complete. Start-Up and Voltage-Step Operation for ISL95870 When the voltage on the VCC pin has ramped above the rising power-on reset voltage VVCC_THR, and the voltage on the EN pin has increased above the rising enable threshold voltage VENTHR, the SREF pin releases its discharge clamp, and enables the reference amplifier VSET. The soft-start current ISS is limited to 17µA and is sourced out of the SREF pin and charges capacitor CSOFT until VSREF equals VREF. The regulator controls the PWM such that the voltage on the FB pin tracks the rising voltage on the SREF pin. The elapsed time from when the EN pin is asserted to when VSREF has charged CSOFT to VREF is called the soft-start delay tSS which is given by Equation 1: V SREF ⋅ C SOFT t SS = ------------------------------------------I SS (EQ. 1) Where: - ISS is the soft-start current source at the 17µA limit - VSREF is the buffered VREF reference voltage The end of soft-start is detected by ISS tapering off when capacitor CSOFT charges to VREF. The internal SSOK flag is set, the PGOOD pin goes high, and diode emulation mode (DEM) is enabled. Choosing the CSOFT capacitor to meet the requirements of a particular soft-start delay tSS is calculated using Equation 2, which is written as follows: t SS ⋅ I SS C SOFT = ----------------------V SREF (EQ. 2) - tSS is the soft-start delay ISS is the soft-start current source at the 17µA limit - VSREF is the buffered VREF reference voltage Start-Up and Voltage-Step Operation for ISL95870A, ISL95870B When the voltage on the VCC pin has ramped above the rising power-on reset voltage VVCC_THR, and the voltage on the EN pin has increased above the rising enable threshold voltage VENTHR, the SREF pin releases its discharge clamp and enables the reference amplifier VSET. The soft-start current ISS is limited to 17µA and is sourced out of the SREF pin into the parallel RC network of capacitor CSOFT and resistance RT. The resistance RT is the sum of all the series connected RSET programming resistors and is written as Equation 3: R T = R SET1 + R SET2 + …R SET ( n ) The voltage on the SREF pin rises as ISS charges CSOFT to the voltage reference setpoint selected by the state of the VID inputs at the time the EN pin is asserted. The regulator controls the PWM such that the voltage on the FB pin tracks the rising voltage on the SREF pin. Once CSOFT charges to the selected setpoint voltage, the ISS current source comes out of the 17µA current limit and decays to the static value set by VSREF/RT. The elapsed time from when the EN pin is asserted to when VSREF has reached the voltage reference setpoint is the soft-start delay tSS which is given by Equation 4: V START-UP t SS = – ( R T ⋅ C SOFT ) ⋅ LN(1 – ------------------------------) I SS ⋅ R T (EQ. 4) Where: - ISS is the soft-start current source at the 17µA limit - VSTART-UP is the setpoint reference voltage selected by the state of the VID inputs at the time EN is asserted - RT is the sum of the RSET programming resistors The end of soft-start is detected by ISS tapering off when capacitor CSOFT charges to the designated VSET voltage reference setpoint. The SSOK flag is set, and the PGOOD pin goes high. The ISS current source changes over to the voltage-step current source IVS which has a current limit of ±85µA. Whenever the VID inputs or the external setpoint reference programs a different setpoint reference voltage, the IVS current source charges or discharges capacitor CSOFT to that new level at ±85µA. Once CSOFT charges to the selected setpoint voltage, the IVS current source comes out of the 85µA current limit and decays to the static value set by VSREF/RT. The elapsed time to charge CSOFT to the new voltage is called the voltagestep delay tVS and is given by Equation 5: ( V NEW – V OLD ) t VS = – ( R T ⋅ C SOFT ) ⋅ LN(1 – -------------------------------------------) I VS ⋅ R T 15 (EQ. 3) (EQ. 5) FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Choosing the CSOFT capacitor to meet the requirements of a particular soft-start delay tSS is calculated with Equation 6, which is written as: – t SS C SOFT = --------------------------------------------------------------------V START-UP ⎞ ⎛ ⎜ R T ⋅ LN(1 – ------------------------------)⎟ I SS ⋅ R T ⎠ ⎝ (EQ. 6) - tSS is the soft-start delay - ISS is the soft-start current source at the 17µA limit - VSTART-UP is the setpoint reference voltage selected by the state of the VID inputs at the time EN is asserted - RT is the sum of the RSET programming resistors Choosing the CSOFT capacitor to meet the requirements of a particular voltage-step delay tVS is calculated with Equation 7, which is written as: (EQ. 7) Where: tVS is the voltage-step delay VNEW is the new setpoint voltage VOLD is the setpoint voltage that VNEW is changing from - IVS is the ±85µA setpoint voltage-step current; positive when VNEW > VOLD, negative when VNEW < VOLD - RT is the sum of the RSET programming resistors - Output Voltage Programming for ISL95870 The ISL95870 has a fixed 0.5V reference voltage (VSREF). As shown in Figure 9, the output voltage is the reference voltage if RFB is shorted and ROFS is open. A resistor divider consisting of ROFS and RFB allows the user to scale the output voltage between 0.5V and 5V. The relation between the output voltage and the reference voltage is given in Equation 8: R FB + R OFS V OUT = V SREF ⋅ ---------------------------------R OFS 16 RFB FB VCOMP − EA + VREF + VSET − SREF FIGURE 9. ISL95870 VOLTAGE PROGRAMMING CIRCUIT Where: – t VS C SOFT = -----------------------------------------------------------------------------V NEW – V OLD ⎞ ⎛ ⎜ R T ⋅ LN(1 – ---------------------------------------)⎟ I VS ⋅ R T ⎝ ⎠ VOUT ROFS - IVS is the ±85µA setpoint voltage-step current; positive when VNEW > VOLD, negative when VNEW < VOLD - VNEW is the new setpoint voltage selected by the VID inputs - VOLD is the setpoint voltage that VNEW is changing from - RT is the sum of the RSET programming resistors CSOFT Where: (EQ. 8) Output Voltage Programming for ISL95870A The ISL95870A allows the user to select four different reference voltages, thus four different output voltages, by voltage identification pins VID1 and VID0. The maximum reference voltage cannot be designed higher than 1.5V. The implementation scheme is shown in Figure 10. The setpoint reference voltages are programmed with resistors that use the naming convention RSET(x) where (x) is the first, second, or third programming resistor connected in series starting at the SREF pin and ending at the GND pin. As shown in Table 1, different combinations of VID1 and VID0 closes different switches and leaves other switches open. For example, for the case of VID1 = 1 and VID0 = 0, switch SW1 closes and all the other three switches SW0, SW2 and SW3 are open. For one combination of VID1 and VID0, the internal switch connects the inverting input of the VSET amplifier to a specific node among the string of RSET programming resistors. All the resistors between that node and the SREF pin serve as the feedback impedance RF of the VSET amplifier. Likewise, all the resistors between that node and the GND pin serve as the input impedance RIN of the VSET amplifier. Equation 9 gives the general form of the gain equation for the VSET amplifier: RF ⎞ ⎛ V SETX = V REF ⋅ ⎜ 1 + ----------⎟ R IN⎠ ⎝ (EQ. 9) Where: - VREF is the 0.5V internal reference of the IC - VSETx is the resulting setpoint reference voltage that appears at the SREF pin TABLE 1. ISL95870A VID TRUTH TABLE VID STATE RESULT VID1 VID0 CLOSE VSREF VOUT 1 1 SW0 VSET1 VOUT1 1 0 SW1 VSET2 VOUT2 0 1 SW2 VSET3 VOUT3 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B TABLE 1. ISL95870A VID TRUTH TABLE (Continued) VOUT RESULT VID1 VID0 CLOSE VSREF VOUT 0 0 SW1, SW3 VSET4 VOUT4 RFB FB VCOMP − EA ROFS VID STATE + + VSET − Equations 10, 11, 12 and 13 give the specific VSET equations for the ISL95870A setpoint reference voltages. VREF 0.5V The ISL95870A VSET1 setpoint is written as Equation 10: (EQ. 10) The ISL95870A VSET3 setpoint is written as Equation 12: R SET1 + R SET2⎞ ⎛ V SET3 = V REF ⋅ ⎜ 1 + --------------------------------------------⎟ R SET3 ⎝ ⎠ - VSET1 < VSET2 < VSET3 < VSET4 Thus, - VOUT1 < VOUT2 < VOUT3 < VOUT4 For given four user selected reference voltages VSETx, the following equation needs to be satisfied in order to have non-zero solution for RSETx. SET3 SW0 SW1 SW2 SW3 (EQ. 13) The VSET1 is fixed at 0.5V because it corresponds to the closure of internal switch SW0 that configures the VSET amplifier as a unity-gain voltage follower for the 0.5V voltage reference VREF. Theoretically, VSET3 can be higher or lower or equal to VSET4 depending on the selection of RSET1, RSET2 and RSET3. However, it is recommended to design the four reference voltages in the following order: V SET1 ⋅ V SET2 + V SET0 SET1 (EQ. 12) The ISL95870A VSET4 setpoint is written as Equation 13: R SET1⎞ ⎛ V SET4 = V REF ⋅ ⎜ 1 + ------------------⎟ R SET2⎠ ⎝ RSET1 (EQ. 11) RSET3 R SET1 ⎛ ⎞ V SET2 = V REF ⋅ ⎜ 1 + --------------------------------------------⎟ R + R ⎝ SET2 SET3⎠ RSET2 The ISL95870A VSET2 setpoint is written as Equation 11: SREF CSOFT V SET1 = V REF ⋅ V SET4 – V SET2 ⋅ V SET3 – V SET2 ⋅ V SET4 = 0 (EQ. 14) The programmed resistors RSET1, RSET2 and RSET3 are designed in the following way. First, assign an initial value to RSET3 of approximately 100kΩ then calculate RSET1 and RSET2 using Equations 15 and 16 respectively. R SET3 ⋅ ( V SET4 – V REF ) ⋅ ( V SET2 – V REF ) R SET1 = --------------------------------------------------------------------------------------------------------------------V REF ⋅ ( V SET4 – V SET2 ) (EQ. 15) R SET3 ⋅ ( V SET2 – V REF ) R SET2 = -------------------------------------------------------------------V SET4 – V SET2 (EQ. 16) The sum of all the programming resistors should be approximately 300kΩ, as shown in Equation 17, otherwise adjust the value of RSET3 and repeat the calculations. R SET1 + R SET2 + R SET3 ≅ 300kΩ 17 (EQ. 17) FIGURE 10. ISL95870A VOLTAGE PROGRAMMING CIRCUIT If the output voltage is in the range of 0.5V to 1.5V, the external resistor-divider is not necessary. The output voltage is equal to one of the reference voltages depending on the status of VID1 and VID0. The external resistor divider consisting of RFB and ROFS allows the user to program the output voltage in the range of 1.5V to 5V. The relation between the output voltage and the reference voltage is given in Equation 18: R FB + R OFS V OUT = V SREF ⋅ ---------------------------------- = V SREF ⋅ k R (EQ. 18) OFS In this case, the four output voltages are equal to each of the corresponding reference voltages multiplying the factor k. V OUTx = V SETx ⋅ k (EQ. 19) Output Voltage Programming for ISL95870B The ISL95870B allows the user to select four different reference voltages, thus four different output voltages, by voltage identification pins VID1 and VID0. The maximum reference voltage cannot be designed higher than 1.5V. The implementation scheme is shown in Figure 11. The setpoint reference voltages are programmed with resistors that use the naming convention RSET(x) where (x) is the first, second, third, or fourth programming resistor connected in series starting at the SREF pin and ending at the GND pin. As shown in Table 2, different combinations of VID1 and VID0 close different switches and leave other switches open. For example, for the case of VID1 = 1 and VID0 = 0, switch SW1 closes and all the other three switches SW0, SW2 and SW3 are open. For one combination of VID1 and VID0, the internal switch FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B connects the inverting input of the VSET amplifier to a specific node among the string of RSET programming resistors. All the resistors between that node and the SREF pin serve as the feedback impedance RF of the VSET amplifier. Likewise, all the resistors between that node and the GND pin serve as the input impedance RIN of the VSET amplifier. Equation 20 gives the general form of the gain equation for the VSET amplifier: RF ⎞ ⎛ V SETX = V REF ⋅ ⎜ 1 + ----------⎟ R IN⎠ ⎝ (EQ. 20) Where: - VREF is the 0.5V internal reference of the IC - VSETx is the resulting setpoint reference voltage that appears at the SREF pin calculate RSET1, RSET2 and RSET3 using Equations 25, 26, and 27 respectively. R SET4 ⋅ V SET4 ⋅ ( V SET2 – V REF ) R SET1 = ------------------------------------------------------------------------------------------V REF ⋅ V SET2 (EQ. 25) R SET4 ⋅ V SET4 ⋅ ( V SET3 – V SET2 ) R SET2 = ---------------------------------------------------------------------------------------------V SET2 ⋅ V SET3 (EQ. 26) R SET4 ⋅ ( V SET4 – V SET3 ) R SET3 = ----------------------------------------------------------------------V SET3 (EQ. 27) The sum of all the programming resistors should be approximately 300kΩ, as shown in Equation 28, otherwise adjust the value of RSET4 and repeat the calculations. R SET1 + R SET2 + R SET3 + R SET4 ≅ 300kΩ (EQ. 28) TABLE 2. ISL95870B VID TRUTH TABLE VID1 VID0 RESULT CLOSE VSREF VOUT 1 1 SW0 VSET1 VOUT1 1 0 SW1 VSET2 VOUT2 0 1 SW2 VSET3 VOUT3 0 0 SW3 VSET4 VOUT4 VOUT RFB FB + + VSET − Equations 21, 22, 23 and 24 give the specific VSET equations for the ISL95870B setpoint reference voltages. (EQ. 22) The ISL95870B VSET3 setpoint is written as Equation 23: R SET1 + R SET2⎞ ⎛ V SET3 = V REF ⋅ ⎜ 1 + --------------------------------------------⎟ R SET3 + R SET4⎠ ⎝ (EQ. 23) The ISL95870B VSET4 setpoint is written as Equation 24: R SET1 + R SET2 + R ⎛ SET3⎞ V SET4 = V REF ⋅ ⎜ 1 + ---------------------------------------------------------------------⎟ R ⎝ ⎠ SET4 (EQ. 24) The VSET1 is fixed at 0.5V because it corresponds to the closure of internal switch SW0 that configures the VSET amplifier as a unity-gain voltage follower for the 0.5V voltage reference VREF. The setpoint reference voltages use the naming convention VSET(x) where (x) is the first, second, third, or fourth setpoint reference voltage where: - VSET1 < VSET2 < VSET3 < VSET4 Thus, - VOUT1 < VOUT2 < VOUT3 < VOUT4 For given four user selected reference voltages VSETx, the programmed resistors RSET1, RSET2, RSET3 and RSET4 are designed in the following way. First, assign an initial value to RSET4 of approximately 100kΩ then RSET1 SET1 SET2 SW0 SW1 SW2 SW3 FIGURE 11. ISL95870B VOLTAGE PROGRAMMING CIRCUIT If the output voltage is in the range of 0.5V to 1.5V, the external resistor-divider is not necessary. The output voltage is equal to one of the reference voltages depending on the status of VID1 and VID0. The external resistor divider consisting of RFB and ROFS allows the user to program the output voltage in the range of 1.5V to 5V. The relation between the output voltage and the reference is given in Equation 29: R FB + R OFS V OUT = V SREF ⋅ ---------------------------------- = V SREF ⋅ k R (EQ. 29) OFS In this case, the four output voltages are equal to each of the corresponding reference voltages multiplying the factor k. V OUTx = V SETx ⋅ k 18 SET0 RSET2 R SET1 ⎛ ⎞ V SET2 = V REF ⋅ ⎜ 1 + ---------------------------------------------------------------------⎟ R + R + R ⎝ SET2 SET3 SET4⎠ RSET3 The ISL95870B VSET2 setpoint is written as Equation 22: VREF 0.5V RSET4 (EQ. 21) V SET1 = V REF SREF CSOFT The ISL95870B VSET1 setpoint is written as Equation 21: VCOMP − EA ROFS VID STATE (EQ. 30) FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B High Output Voltage Programming The ISL95870 has a fixed 0.5V reference voltage (VSREF). For high output voltage application, the resistor divider consisting of RFB and ROFS requires large ratio (RFB :ROFS = 9:1 for 5V output). The FB pin with large ratio resistor divider is noise sensitive and the PCB layout should be carefully routed. It is recommended to use small value resistor divider such as RFB=1kΩ. In general the ISL95870A and ISL95870B have much better jitter performance than the ISL95870 when the output voltage is in the range of 3.3V to 5V, particularly in DCM. This is because VSREF voltage can be set to 1.5V and a smaller ratio resistor divider can be used. This makes the singal to noise ratio at FB pin much better. So for 3.3V to 5V output, the ISL95870A and ISL95870B are recommended with VSREF set to 1.5V. COMPENSATION TO COUNTER INTEGRATOR POLE V V OUT V COMP VDAC FIGURE 12. INTEGRATOR ERROR-AMPLIFIER CONFIGURATION R3 LOOP GAIN (dB) INTEGRATOR POLE p1 L/C DOUBLE-POLE R4 Modulator Stability p2 p3 CURRENT-MODE ZERO z1 -20dB CROSSOVER REQUIRED FOR STABILITY COMPENSATOR TO ADD z2 IS NEEDED ec /d dB 0 -2 c de B/ 0d -4 / dec -60dB The R4 modulator is an evolutionary step in R3 technology. Like R3, the R4 modulator allows variable frequency in response to load transients and maintains the benefits of current-mode hysteretic controllers. However, in addition, the R4 modulator reduces regulator output impedance and uses accurate referencing to eliminate the need for a high-gain voltage amplifier in the compensation loop. The result is a topology that can be tuned to voltage-mode hysteretic transient speed while maintaining a linear control model and removes the need for any compensation. This greatly simplifies the regulator design for customers and reduces external component cost. INTEGRATOR FOR HIGH DC GAIN f (Hz) FIGURE 13. UNCOMPENSATED INTEGRATOR OPEN-LOOP RESPONSE The removal of compensation derives from the R4 modulator’s lack of need for high DC gain. In traditional architectures, high DC gain is achieved with an integrator in the voltage loop. The integrator introduces a pole in the open-loop transfer function at low frequencies. That, combined with the double-pole from the output L/C filter, creates a three pole system that must be compensated to maintain stability. Figure 12 illustrates the classic integrator configuration for a voltage loop error-amplifier. While the integrator provides the high DC gain required for accurate regulation in traditional technologies, it also introduces a low-frequency pole into the control loop. Figure 13 shows the open-loop response that results from the addition of an integrating capacitor in the voltage loop. The compensation components found in Figure 12 are necessary to achieve stability. Classic control theory requires a single-pole transition through unity gain to ensure a stable system. Current-mode architectures (includes peak, peak-valley, current-mode hysteretic, R3 and R4) generate a zero at or near the L/C resonant point, effectively canceling one of the system’s poles. The system still contains two poles, one of which must be canceled with a zero before unity gain crossover to achieve stability. Compensation components are added to introduce the necessary zero. Because R4 does not require a high-gain voltage loop, the integrator can be removed, reducing the number of inherent poles in the loop to two. The current-mode zero continues to cancel one of the poles, ensuring a single-pole crossover for a wide range of output filter choices. The result is a stable system with no need for compensation components or complex equations to properly tune the stability. R2 VOUT VCOMP R1 VDAC FIGURE 14. NON-INTEGRATED R4 ERROR-AMPLIFIER CONFIGURATION 19 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Figure 14 shows the R4 error-amplifier that does not require an integrator for high DC gain to achieve accurate regulation. The result to the open loop response can be seen in Figure 15. R4 LOOP GAIN (dB) L/C DOUBLE-POLE p1 SYSTEM HAS 2 POLES AND 1 ZERO p2 NO COMPENSATOR IS NEEDED ec /d dB dec 0 -2 B/ c 0d de -2 B/ 0d -4 CURRENT-MODE ZERO z1 f (Hz) FIGURE 15. UNCOMPENSATED R4 OPEN-LOOP RESPONSE Transient Response In addition to requiring a compensation zero, the integrator in traditional architectures also slows system response to transient conditions. The change in COMP voltage is slow in response to a rapid change in output voltage. If the integrating capacitor is removed, COMP moves as quickly as VOUT, and the modulator immediately increases or decreases switching frequency to recover the output voltage. IOUT t R4 R3 current, can be either positive or negative. Should the sum of the AC and DC components of the inductor current remain positive for the entire switching period, the converter is in continuous-conduction-mode (CCM). However, if the inductor current becomes negative or zero, the converter is in discontinuous-conduction-mode (DCM). Unlike the standard DC/DC buck regulator, the synchronous rectifier can sink current from the output filter inductor during DCM, reducing the light-load efficiency with unnecessary conduction loss as the low-side MOSFET sinks the inductor current. The ISL95870, ISL95870A, ISL95870B controllers avoid the DCM conduction loss by making the low-side MOSFET emulate the current-blocking behavior of a diode. This smart-diode operation called diode-emulation-mode (DEM) is triggered when the negative inductor current produces a positive voltage drop across the rDS(ON) of the low-side MOSFET for eight consecutive PWM cycles while the LGATE pin is high. The converter will exit DEM on the next PWM pulse after detecting a negative voltage across the rDS(ON) of the low-side MOSFET. It is characteristic of the R4 architecture for the PWM switching frequency to decrease while in DCM, increasing efficiency by reducing unnecessary gate-driver switching losses. The extent of the frequency reduction is proportional to the reduction of load current. Upon entering DEM, the PWM frequency is forced to fall approximately 30% by forcing a similar increase of the window voltage V W. This measure is taken to prevent oscillating between modes at the boundary between CCM and DCM. The 30% increase of VW is removed upon exit of DEM, forcing the PWM switching frequency to jump back to the nominal CCM value. Overcurrent VCOMP t VOUT t The overcurrent protection (OCP) setpoint is programmed with resistor ROCSET, which is connected across the OCSET and PHASE pins. Resistor RO is connected between the VO pin and the actual output voltage of the converter. During normal operation, the VO pin is a high impedance path, therefore there is no voltage drop across RO. The value of resistor RO should always match the value of resistor ROCSET. FIGURE 16. R3 vs R4 IDEALIZED TRANSIENT RESPONSE The dotted red and blue lines in Figure 16 represent the time delayed behavior of VOUT and VCOMP in response to a load transient when an integrator is used. The solid red and blue lines illustrate the increased response of R4 in the absence of the integrator capacitor. L DCR PHASE + ROCSET 8.5µA OCSET Diode Emulation + VROCSET IL VDCR CSEN _ VO CO _ RO The polarity of the output inductor current is defined as positive when conducting away from the phase node, and defined as negative when conducting towards the phase node. The DC component of the inductor current is positive, but the AC component known as the ripple 20 VO FIGURE 17. OVERCURRENT PROGRAMMING CIRCUIT FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Figure 17 shows the overcurrent set circuit. The inductor consists of inductance L and the DC resistance DCR. The inductor DC current IL creates a voltage drop across DCR, which is given by Equation 31: (EQ. 31) V DCR = I L ⋅ DCR The IOCSET current source sinks 8.5µA into the OCSET pin, creating a DC voltage drop across the resistor ROCSET, which is given by Equation 32: (EQ. 32) V ROCSET = 8.5μA ⋅ R OCSET The DC voltage difference between the OCSET pin and the VO pin, which is given by Equation 33: V OCSET – V VO = V DCR – V ROCSET = I L ⋅ DCR – I OCSET ⋅ R OCSET (EQ. 33) The IC monitors the voltage of the OCSET pin and the VO pin. When the voltage of the OCSET pin is higher than the voltage of the VO pin for more than 10µs, an OCP fault latches the converter off. The value of ROCSET is calculated with Equation 34, which is written as: I OC ⋅ DCR R OCSET = ---------------------------I OCSET (EQ. 34) Where: - ROCSET (Ω) is the resistor used to program the overcurrent setpoint - IOC is the output DC load current that will activate the OCP fault detection circuit - DCR is the inductor DC resistance For example, if IOC is 20A and DCR is 4.5mΩ, the choice of ROCSET is equal to 20A x 4.5mΩ/8.5µA = 10.5kΩ. Resistor ROCSET and capacitor CSEN form an R-C network to sense the inductor current. To sense the inductor current correctly not only in DC operation, but also during dynamic operation, the R-C network time constant ROCSET CSEN needs to match the inductor time constant L/DCR. The value of CSEN is then written as Equation 35: L C SEN = -----------------------------------------R OCSET ⋅ DCR (EQ. 35) For example, if L is 1.5µH, DCR is 4.5mΩ, and ROCSET is 9kΩ, the choice of CSEN = 1.5µH/(9kΩ x 4.5mΩ) = 0.037µF. When an OCP fault is declared, the converter will be latched off and the PGOOD pin will be asserted low. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage V VCC_THF. Overvoltage The OVP fault detection circuit triggers after the FB pin voltage is above the rising overvoltage threshold VOVRTH for more than 2µs. For example, if the converter is programmed to regulate 1.0V at the FB pin, that voltage would have to rise above the typical VOVRTH threshold of 21 116% for more than 2µs in order to trip the OVP fault latch. In numerical terms, that would be 116% x 1.0V = 1.16V. When an OVP fault is declared, the converter will be latched off and the PGOOD pin will be asserted low. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. Although the converter has latched-off in response to an OVP fault, the LGATE gate-driver output will retain the ability to toggle the low-side MOSFET on and off, in response to the output voltage transversing the VOVRTH and VOVFTH thresholds. The LGATE gate-driver will turnon the low-side MOSFET to discharge the output voltage, protecting the load. The LGATE gate-driver will turn-off the low-side MOSFET once the FB pin voltage is lower than the falling overvoltage threshold VOVRTH for more than 2µs. The falling overvoltage threshold VOVFTH is typically 102%. That means if the FB pin voltage falls below 102% x 1.0V = 1.02V for more than 2µs, the LGATE gate-driver will turn off the low-side MOSFET. If the output voltage rises again, the LGATE driver will again turn on the low-side MOSFET when the FB pin voltage is above the rising overvoltage threshold VOVRTH for more than 2µs. By doing so, the IC protects the load when there is a consistent overvoltage condition. Undervoltage The UVP fault detection circuit triggers after the FB pin voltage is below the undervoltage threshold VUVTH for more than 2µs. For example if the converter is programmed to regulate 1.0V at the FB pin, that voltage would have to fall below the typical VUVTH threshold of 84% for more than 2µs in order to trip the UVP fault latch. In numerical terms, that would be 84% x 1.0V = 0.84V. When a UVP fault is declared, the converter will be latched off and the PGOOD pin will be asserted low. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. Over-Temperature When the temperature of the IC increases above the rising threshold temperature TOTRTH, it will enter the OTP state that suspends the PWM, forcing the LGATE and UGATE gate-driver outputs low. The status of the PGOOD pin does not change nor does the converter latch-off. The PWM remains suspended until the IC temperature falls below the hysteresis temperature TOTHYS at which time normal PWM operation resumes. The OTP state can be reset if the EN pin is pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. All other protection circuits remain functional while the IC is in the OTP state. It is likely that the IC will detect an UVP fault because in the absence of PWM, the output voltage decays below the undervoltage threshold VUVTH. FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B PGOOD Monitor The PGOOD pin indicates when the converter is capable of supplying regulated voltage. The PGOOD pin is an undefined impedance if the VCC pin has not reached the rising POR threshold VVCC_THR, or if the VCC pin is below the falling POR threshold VVCC_THF. If there is a fault condition of output overcurrent, overvoltage or undervoltage, PGOOD is asserted low. The PGOOD pull-down impedance is 50Ω. UGATE 1V 1V 1V 1V Integrated MOSFET Gate-Drivers The LGATE pin and UGATE pins are MOSFET driver outputs. The LGATE pin drives the low-side MOSFET of the converter while the UGATE pin drives the high-side MOSFET of the converter. The LGATE driver is optimized for low duty-cycle applications where the low-side MOSFET experiences long conduction times. In this environment, the low-side MOSFETs require exceptionally low rDS(ON) and tend to have large parasitic charges that conduct transient currents within the devices in response to high dv/dt switching present at the phase node. The drain-gate charge in particular can conduct sufficient current through the driver pull-down resistance that the VGS(th) of the device can be exceeded and turned on. For this reason, the LGATE driver has been designed with low pull-down resistance and high sink current capability to ensure clamping the MOSFETs gate voltage below VGS(th). Adaptive Shoot-Through Protection Adaptive shoot-through protection prevents a gate-driver output from turning on until the opposite gate-driver output has fallen below approximately 1V. The dead-time shown in Figure 18 is extended by the additional period that the falling gate voltage remains above the 1V threshold. The high-side gate-driver output voltage is measured across the UGATE and PHASE pins while the low-side gate-driver output voltage is measured across the LGATE and PGND pins. The power for the LGATE gate-driver is sourced directly from the PVCC pin. The-power for the UGATE gate-driver is supplied by a boot-strap capacitor connected across the BOOT and PHASE pins. The capacitor is charged each time the phase node voltage falls a diode drop below PVCC such as when the low-side MOSFET is turned on. LGATE FIGURE 18. GATE DRIVE ADAPTIVE SHOOT-THROUGH PROTECTION General Application Design Guide This design guide is intended to provide a high-level explanation of the steps necessary to design a singlephase buck converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced in the following. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts. Selecting the LC Output Filter The duty cycle of an ideal buck converter is a function of the input and the output voltage. This relationship is expressed in Equation 36: VO D = --------V IN (EQ. 36) The output inductor peak-to-peak ripple current is expressed in Equation 37: VO ⋅ ( 1 – D ) I P-P = ------------------------------F SW ⋅ L (EQ. 37) A typical step-down DC/DC converter will have an IPP of 20% to 40% of the maximum DC output load current. The value of IP-P is selected based upon several criteria such as MOSFET switching loss, inductor core loss, and the resistive loss of the inductor winding. The DC copper loss of the inductor can be estimated using Equation 38: 2 P COPPER = I LOAD ⋅ DCR (EQ. 38) Where, ILOAD is the converter output DC current. The copper loss can be significant so attention has to be given to the DCR of the inductor. Another factor to consider when choosing the inductor is its saturation characteristics at elevated temperature. A saturated inductor could cause destruction of circuit components, as well as nuisance OCP faults. 22 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B ΔV ESR = I P-P ⋅ E SR (EQ. 39) I P-P ΔV C = --------------------------------8 ⋅ CO ⋅ F (EQ. 40) SW If the output of the converter has to support a load with high pulsating current, several capacitors will need to be paralleled to reduce the total ESR until the required VP-P is achieved. The inductance of the capacitor can significantly impact the output voltage ripple and cause a brief voltage spike if the load transient has an extremely high slew rate. Low inductance capacitors should be considered. A capacitor dissipates heat as a function of RMS current and frequency. Be sure that IP-P is shared by a sufficient quantity of paralleled capacitors so that they operate below the maximum rated RMS current at FSW. Take into account that the rated value of a capacitor can fade as much as 50% as the DC voltage across it increases. Selecting the Input Capacitor The important parameters for the bulk input capacitors are the voltage rating and the RMS current rating. For reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and capable of supplying the RMS current required by the switching circuit. Their voltage rating should be at least 1.25x greater than the maximum input voltage, while a voltage rating of 1.5x is a preferred rating. Figure 19 is a graph of the input RMS ripple current, normalized relative to output load current, as a function of duty cycle that is adjusted for converter efficiency. The ripple current calculation is written as Equation 41: 2 2 2 2 D ( I MAX ⋅ ( D – D ) ) + ⎛ x ⋅ I MAX ⋅ ------ ⎞ ⎝ 12 ⎠ I IN_RMS = -------------------------------------------------------------------------------------------------------I MAX (EQ. 41) Where: - IMAX is the maximum continuous ILOAD of the converter - x is a multiplier (0 to 1) corresponding to the inductor peak-to-peak ripple amplitude expressed as a percentage of IMAX (0% to 100%) - D is the duty cycle that is adjusted to take into account the efficiency of the converter Duty cycle is written as Equation 42: VO D = -------------------------V IN ⋅ EFF (EQ. 42) In addition to the bulk capacitors, some low ESL ceramic capacitors are recommended to decouple between the 23 drain of the high-side MOSFET and the source of the low-side MOSFET. 0.6 NORMALIZED INPUT RMS RIPPLE CURRENT A DC/DC buck regulator must have output capacitance CO into which ripple current IP-P can flow. Current IP-P develops a corresponding ripple voltage VP-P across CO, which is the sum of the voltage drop across the capacitor ESR and of the voltage change stemming from charge moved in and out of the capacitor. These two voltages are expressed in Equations 39 and 40: 0.5 x=0 0.4 x = 0.5 0.3 0.2 x=1 0.1 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 DUTY CYCLE FIGURE 19. NORMALIZED INPUT RMS CURRENT FOR EFF = 1 Selecting the Bootstrap Capacitor The integrated driver features an internal bootstrap schottky diode. Simply adding an external capacitor across the BOOT and PHASE pins completes the bootstrap circuit. The bootstrap capacitor voltage rating is selected to be at least 10V. Although the theoretical maximum voltage of the capacitor is PVCC-VDIODE (voltage drop across the boot diode), large excursions below ground by the phase node requires at least a 10V rating for the bootstrap capacitor. The bootstrap capacitor can be chosen from Equation 43: Q GATE C BOOT ≥ -----------------------ΔV BOOT (EQ. 43) Where: - QGATE is the amount of gate charge required to fully charge the gate of the upper MOSFET - ΔVBOOT is the maximum decay across the BOOT capacitor As an example, suppose the high-side MOSFET has a total gate charge Qg, of 25nC at VGS = 5V, and a ΔVBOOT of 200mV. The calculated bootstrap capacitance is 0.125µF; for a comfortable margin, select a capacitor that is double the calculated capacitance. In this example, 0.22µF will suffice. Use a low temperature-coefficient ceramic capacitor. Driver Power Dissipation Switching power dissipation in the driver is mainly a function of the switching frequency and total gate charge of the selected MOSFETs. Calculating the power dissipation in the driver for a desired application is critical to ensuring safe operation. Exceeding the maximum allowable power dissipation level will push the IC beyond the maximum recommended operating junction temperature of +125°C. When designing the application, it is recommended that the following calculation be performed to ensure safe operation at the desired frequency for the selected MOSFETs. The power FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B dissipated by the drivers is approximated as Equation 44: (EQ. 44) P = F sw ( 1.5V U Q + V L Q ) + P L + P U U L 2 P CON_LS ≈ I LOAD ⋅ r DS ( ON )_LS ⋅ ( 1 – D ) Where: Fsw is the switching frequency of the PWM signal VU is the upper gate driver bias supply voltage VL is the lower gate driver bias supply voltage QU is the charge to be delivered by the upper driver into the gate of the MOSFET and discrete capacitors - QL is the charge to be delivered by the lower driver into the gate of the MOSFET and discrete capacitors - PL is the quiescent power consumption of the lower driver - PU is the quiescent power consumption of the upper driver - 1000 QU =100nC QL =200nC 900 QU =50nC QL =100nC POWER (mW) 800 QU =50nC QL=50nC 700 600 QU =20nC QL=50nC 500 For the low-side MOSFET, (LS), the power loss can be assumed to be conductive only and is written as Equation 45: 400 (EQ. 45) For the high-side MOSFET, (HS), its conduction loss is written as Equation 46: 2 P CON_HS = I LOAD ⋅ r DS ( ON )_HS ⋅ D (EQ. 46) For the high-side MOSFET, its switching loss is written as Equation 47: V IN ⋅ I VALLEY ⋅ t ON ⋅ F V IN ⋅ I PEAK ⋅ t OFF ⋅ F SW SW P SW_HS = ---------------------------------------------------------------------- + -----------------------------------------------------------------2 2 (EQ. 47) Where: - IVALLEY is the difference of the DC component of the inductor current minus 1/2 of the inductor ripple current - IPEAK is the sum of the DC component of the inductor current plus 1/2 of the inductor ripple current - tON is the time required to drive the device into saturation - tOFF is the time required to drive the device into cut-off 300 Layout Considerations 200 As a general rule, power layers should be close together, either on the top or bottom of the board, with the weak analog or logic signal layers on the opposite side of the board. The ground-plane layer should be adjacent to the signal layer to provide shielding. The ground plane layer should have an island located under the IC, the components connected to analog or logic signals. The island should be connected to the rest of the ground plane layer at one quiet point. 100 0 0 200 400 600 800 1k 1.2k 1.4k 1.6k 1.8k 2k FREQUENCY (Hz) FIGURE 20. POWER DISSIPATION vs FREQUENCY MOSFET Selection and Considerations The choice of MOSFETs depends on the current each MOSFET will be required to conduct, the switching frequency, the capability of the MOSFETs to dissipate heat, and the availability and nature of heat sinking and air flow. Typically, a MOSFET cannot tolerate even brief excursions beyond their maximum drain to source voltage rating. The MOSFETs used in the power stage of the converter should have a maximum VDS rating that exceeds the sum of the upper voltage tolerance of the input power source and the voltage spike that occurs when the MOSFETs switch. There are several power MOSFETs readily available that are optimized for DC/DC converter applications. The preferred high-side MOSFET emphasizes low gate charge so that the device spends the least amount of time dissipating power in the linear region. The preferred lowside MOSFET emphasizes low r DS(on) when fully saturated to minimize conduction loss. 24 There are two sets of components in a DC/DC converter, the power components and the small signal components. The power components are the most critical because they switch large amount of energy. The small signal components connect to sensitive nodes or supply critical bypassing current and signal coupling. The power components should be placed first and these include MOSFETs, input and output capacitors, and the inductor. Keeping the distance between the power train and the control IC short helps keep the gate drive traces short. These drive signals include the LGATE, UGATE, PGND, PHASE and BOOT. When placing MOSFETs, try to keep the source of the upper MOSFETs and the drain of the lower MOSFETs as close as thermally possible. See Figure 21. Input high frequency capacitors should be placed close to the drain of the upper MOSFETs and the source of the lower MOSFETs. Place the output inductor and output FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B VIAS TO GROUND PLANE VOUT INDUCTOR OCSET AND VO PINS GND PHASE NODE HIGH-SIDE MOSFETS VIN OUTPUT CAPACITORS SCHOTTKY DIODE LOW-SIDE MOSFETS INPUT CAPACITORS FIGURE 21. TYPICAL POWER COMPONENT PLACEMENT capacitors between the MOSFETs and the load. High frequency output decoupling capacitors (ceramic) should be placed as close as possible to the decoupling target, making use of the shortest connection paths to any internal planes. Place the components in such a way that the area under the IC has less noise traces with high dV/dt and di/dt, such as gate signals and phase node signals. VCC AND PVCC PINS Place the decoupling capacitors as close as practical to the IC. In particular, the PVCC decoupling capacitor should have a very short and wide connection to the PGND pin. The VCC decoupling capacitor should be referenced to GND pin. EN, PGOOD, VID0, VID1, AND FSEL PINS These are logic signals that are referenced to the GND pin. Treat as a typical logic signal. 25 The current-sensing network consisting of ROCSET, RO, and CSEN needs to be connected to the inductor pads for accurate measurement of the DCR voltage drop. These components however, should be located physically close to the OCSET and VO pins with traces leading back to the inductor. It is critical that the traces are shielded by the ground plane layer all the way to the inductor pads. The procedure is the same for resistive current sense. FB, SREF, SET0, SET1, SET2, AND RTN PINS The input impedance of these pins is high, making it critical to place the components connected to these pins as close as possible to the IC. LGATE, PGND, UGATE, BOOT, AND PHASE PINS The signals going through these traces are high dv/dt and high di/dt, with high peak charging and discharging current. The PGND pin can only flow current from the gate-source charge of the low-side MOSFETs when LGATE goes low. Ideally, route the trace from the LGATE pin in parallel with the trace from the PGND pin, route the trace from the UGATE pin in parallel with the trace from the PHASE pin. In order to have more accurate zero-crossing detection of inductor current, it is recommended to connect Phase pin to the drain of the low-side MOSFETs with Kelvin connection. These pairs of traces should be short, wide, and away from other traces with high input impedance; weak signal traces should not be in proximity with these traces on any layer. FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest Rev. DATE REVISION December 22, 2009 CHANGE FN6899.0 Initial Release Products Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks. Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a complete list of Intersil product families. *For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on intersil.com: ISL95870, ISL95870A, ISL95870B To report errors or suggestions for this datasheet, please go to www.intersil.com/askourstaff FITs are available from our website at http://rel.intersil.com/reports/search.php For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 26 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Ultra Thin Quad Flat No-Lead Plastic Package (UTQFN) D L16.2.6x1.8A B 16 LEAD ULTRA THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE MILLIMETERS 6 INDEX AREA 2X A N SYMBOL E 0.10 C 1 2 2X MIN NOMINAL MAX NOTES A 0.45 0.50 0.55 - A1 - - 0.05 - 0.10 C A3 TOP VIEW 0.10 C C A 0.05 C 0.127 REF - b 0.15 0.20 0.25 5 D 2.55 2.60 2.65 - E 1.75 1.80 1.85 - e 0.40 BSC - SEATING PLANE A1 SIDE VIEW K 0.15 - - - L 0.35 0.40 0.45 - L1 0.45 0.50 0.55 - N 16 2 Nd 4 3 Ne 4 3 e PIN #1 ID K 1 2 NX L L1 θ NX b 5 16X 0.10 M C A B 0.05 M C (DATUM B) (DATUM A) BOTTOM VIEW 0 - 12 4 Rev. 5 2/09 NOTES: 1. Dimensioning and tolerancing conform to ASME Y14.5-1994. 2. N is the number of terminals. 3. Nd and Ne refer to the number of terminals on D and E side, respectively. 4. All dimensions are in millimeters. Angles are in degrees. 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. CL (A1) NX (b) L 5 e SECTION "C-C" 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 7. Maximum package warpage is 0.05mm. TERMINAL TIP C C 8. Maximum allowable burrs is 0.076mm in all directions. 9. JEDEC Reference MO-255. 10. For additional information, to assist with the PCB Land Pattern Design effort, see Intersil Technical Brief TB389. 3.00 1.80 1.40 1.40 2.20 0.90 0.40 0.20 0.50 0.20 0.40 10 LAND PATTERN 27 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Package Outline Drawing L20.3.2x1.8 20 LEAD ULTRA THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE (UTQFN) Rev 0, 5/08 1.80 A 6 PIN #1 ID 16X 0.40 B 20 6 PIN 1 ID# 1 19 2 3.20 0.50±0.10 (4X) 0.10 9 12 11 10 VIEW “A-A” TOP VIEW 0.10 M C A B 0.05 M C 4 20X 0.20 19X 0.40 ± 0.10 BOTTOM VIEW ( 1.0 ) (1 x 0.70) SEE DETAIL "X" 0.10 C MAX 0.55 C BASE PLANE ( 2. 30 ) SEATING PLANE 0.05 C SIDE VIEW ( 16 X 0 . 40 ) C 0 . 2 REF 5 ( 20X 0 . 20 ) 0 . 00 MIN. 0 . 05 MAX. ( 19X 0 . 60 ) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 28 FN6899.0 December 22, 2009 ISL95870, ISL95870A, ISL95870B Package Outline Drawing L20.3x4 20 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 0, 6/07 3.00 0.10 M C A B 0.05 M C A B 4 20X 0.25 16X 0.50 +0.05 -0.07 17 A PIN 1 INDEX AREA (C 0.40) 20 16 1 PIN 1 INDEX AREA 4.00 2.65 11 +0.10 -0.15 6 0.15 (4X) A 10 7 VIEW "A-A" 1.65 TOP VIEW +0.10 -0.15 20x 0.40±0.10 BOTTOM VIEW SEE DETAIL "X" 0.10 C C 0.9± 0.10 SEATING PLANE 0.08 C SIDE VIEW (16 x 0.50) (2.65) (3.80) (20 x 0.25) C 0.2 REF (20 x 0.60) 5 0.00 MIN. 0.05 MAX. (1.65) (2.80) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature. 29 FN6899.0 December 22, 2009