LTC3868-1 Low IQ, Dual 2-Phase Synchronous Step-Down Controller Description Features n n n n n n n n n n n n n n n n n n The LTC®3868-1 is a high performance dual step-down switching regulator controller that drives all N-channel synchronous power MOSFET stages. A constant frequency current mode architecture allows a phase-lockable frequency of up to 850kHz. Power loss and noise due to the input capacitor ESR are minimized by operating the two controller outputs out of phase. Low Operating IQ: 170µA (One Channel On) Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 14V Wide VIN Range: 4V to 24V RSENSE or DCR Current Sensing Out-of-Phase Controllers Reduce Required Input Capacitance and Power Supply Induced Noise OPTI-LOOP® Compensation Minimizes COUT Phase-Lockable Frequency (75kHz to 850kHz) Programmable Fixed Frequency (50kHz to 900kHz) Selectable Continuous, Pulse-Skipping or Burst Mode® Operation at Light Loads Very Low Dropout Operation: 99% Duty Cycle Adjustable Output Voltage Soft-Start Power Good Output Voltage Monitor Output Overvoltage Protection Output Latchoff Protection During Short Circuit Low Shutdown IQ: 8µA Internal LDO Powers Gate Drive from VIN or EXTVCC No Current Foldback During Start-Up Small 4mm × 5mm QFN and Narrow SSOP Packages The 170μA no-load quiescent current extends operating life in battery powered systems. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The LTC3868-1 features a precision 0.8V reference and a power good output indicator. A wide 4V to 24V input supply range encompasses a wide range of intermediate bus voltages and battery chemistries. Independent soft-start pins for each controller ramp the output voltages during start-up. Current foldback limits MOSFET heat dissipation during short-circuit conditions. The output short-circuit latchoff feature further protects the circuit in short-circuit conditions. Applications n n n n For a leadless 32-pin QFN package with the additional features of adjustable current limit, clock out, phase modulation and two PGOOD outputs, see the LTC3868 data sheet. Notebook and Palmtop Computers Portable Instruments Battery Operated Digital Devices Distributed DC Power Systems L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, µModule, Linear Technology and the Linear logo are registered trademarks and No RSENSE and UltraFast are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5705919, 5929620, 6100678, 6144194, 6177787, 6304066, 6580258. Typical Application High Efficiency Dual 8.5V/3.3V Step-Down Converter 22µF 50V 4.7µF 0.1µF 3.3µH VIN INTVCC BOOST1 BOOST2 LTC3868-1 0.1µF 80 7.2µH BG2 PGND SENSE2+ 0.01Ω 0.007Ω 62.5k 150µF 680pF 20k 15k SENSE2– VFB2 ITH2 SENSE1– VFB1 ITH1 SS1 0.1µF SGND SS2 0.1µF 193k 680pF 15k 10000 90 SW2 SENSE1+ VOUT1 3.3V 5A 100 20k VOUT2 8.5V 3.5A 150µF 1000 70 EFFICIENCY 60 50 100 POWER LOSS 40 10 30 20 VIN = 12V VOUT = 3.3V FIGURE 12 CIRCUIT 10 0 0.0001 0.001 0.01 0.1 1 OUTPUT CURRENT (A) 10 POWER LOSS (mW) SW1 BG1 TG2 EFFICIENCY (%) TG1 Efficiency and Power Loss vs Load Current VIN 9V TO 24V 1 0.1 38681 TA01b 38681 TA01 38681fb LTC3868-1 Absolute Maximum Ratings (Note 1) Input Supply Voltage (VIN).......................... –0.3V to 28V Topside Driver Voltages BOOST1, BOOST2 ................................. –0.3V to 34V Switch Voltage (SW1, SW2) ......................... –5V to 28V (BOOST1-SW1), (BOOST2-SW2) . ............... –0.3V to 6V RUN1, RUN2 . .............................................. –0.3V to 8V Maximum Current Sourced into Pin from Source >8V .......................................................100µA SENSE1+, SENSE2+, SENSE1– SENSE2– Voltages....................................... –0.3V to 16V PLLIN/MODE, FREQ Voltages ............... –0.3V to INTVCC EXTVCC . ..................................................... –0.3V to 14V ITH1, ITH2, VFB1, VFB2 Voltages...................... –0.3V to 6V PGOOD1 Voltage . ........................................ –0.3V to 6V SS1, SS2, INTVCC Voltages .......................... –0.3V to 6V Operating Temperature Range (Note 2).... –40°C to 85°C Junction Temperature (Note 3).............................. 125°C Storage Temperature Range.................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) SSOP................................................................. 300°C Pin Configuration TOP VIEW 28 27 26 25 24 23 3 26 TG1 SENSE1– 4 25 SW1 FREQ 5 24 BOOST1 PLLIN/MODE 6 23 BG1 SGND 7 22 VIN RUN1 8 21 PGND RUN2 9 20 EXTVCC SENSE2– 10 19 INTVCC SENSE2+ 11 18 BG2 TG1 27 PGOOD1 SENSE1+ SS1 VFB1 ITH1 28 SS1 2 VFB1 ITH1 1 SW1 PGOOD1 TOP VIEW SENSE1+ 1 22 BOOST1 SENSE1– 2 21 BG1 FREQ 3 20 VIN PLLIN/MODE 4 19 PGND 29 SGND SGND 5 18 EXTVCC RUN1 6 17 INTVCC RUN2 7 16 BG2 SENSE2– 8 15 BOOST2 SW2 TG2 SS2 ITH2 VFB2 SENSE2+ 9 10 11 12 13 14 UFD PACKAGE 28-LEAD (4mm s 5mm) PLASTIC QFN VFB2 12 17 BOOST2 ITH2 13 16 SW2 SS2 14 15 TG2 GN PACKAGE 28-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 43°C/W EXPOSED PAD (PIN 29) IS SGND, MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 90°C/W order information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3868EUFD-1#PBF LTC3868EUFD-1#TRPBF 38681 28-Lead (4mm × 5mm) Plastic QFN –40°C to 85°C LTC3868IUFD-1#PBF LTC3868IUFD-1#TRPBF 38681 28-Lead (4mm × 5mm) Plastic QFN –40°C to 85°C LTC3868EGN-1#PBF LTC3868EGN-1#TRPBF LTC3868GN-1 28-Lead Plastic SSOP –40°C to 85°C LTC3868IGN-1#PBF LTC3868IGN-1#TRPBF LTC3868GN-1 28-Lead Plastic SSOP –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 38681fb LTC3868-1 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. SYMBOL PARAMETER VIN Input Supply Operating Voltage Range CONDITIONS VFB1,2 Regulated Feedback Voltage (Note 4) ITH1,2 Voltage = 1.2V IFB1,2 Feedback Current (Note 4) VREFLNREG Reference Voltage Line Regulation (Note 4) VIN = 4.5V to 24V VLOADREG Output Voltage Load Regulation (Note4) Measured in Servo Loop, ∆ITH Voltage = 1.2V to 0.7V (Note4) Measured in Servo Loop, ∆ITH Voltage = 1.2V to 2V MIN TYP 4 24 UNITS V 0.8 0.812 V ±5 ±50 nA 0.002 0.02 %/V l 0.01 0.1 % l –0.01 –0.1 % l 0.788 MAX gm1,2 Transconductance Amplifier gm (Note 4) ITH1,2 = 1.2V, Sink/Source = 5µA IQ Input DC Supply Current (Note 5) Pulse-Skipping or Forced Continuous Mode (One Channel On) RUN1 = 5V and RUN2 = 0V or RUN1 = 0V and RUN2 = 5V, VFB1 = 0.83V (No Load) Pulse-Skipping or Forced Continuous Mode (Both Channels On) RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) Sleep Mode (One Channel On) RUN1 = 5V and RUN2 = 0V or RUN1 = 0V and RUN2 = 5V, VFB1 = 0.83V (No Load) 170 250 µA Sleep Mode (Both Channels On) RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) 300 450 µA µA 2 1.3 mA 2 mA Shutdown RUN1,2 = 0V UVLO Undervoltage Lockout INTVCC Ramping Up INTVCC Ramping Down VOVL Feedback Overvoltage Protection Measured at VFB1,2, Relative to Regulated VFB1,2 SENSE+ Pin Current Each Channel SENSE – Pins Current Each Channel VOUT1,2 < INTVCC – 0.5V VOUT1,2 > INTVCC + 0.5V DFMAX Maximum Duty Factor In Dropout, FREQ = 0V 98 99.4 ISS1,2 Soft-Start Charge Current VSS1,2 = 0V 0.7 VRUN1,2 On RUN Pin On Threshold VRUN1, VRUN2 Rising 1.21 ISENSE+ ISENSE – l l 8 25 3.6 4 3.8 4.2 4 V V 7 10 13 % ±1 µA ±1 700 µA µA 1 1.4 µA 1.26 1.31 540 l VRUN1,2 Hyst RUN Pin Hysteresis VSS1,2 LA SS Pin Latchoff Arming Threshold VSS1,2 LT IDSC1,2 LT mmho % 50 V mV VSS1, VSS2 Rising from 1V 1.9 2 2.1 SS Pin Latchoff Threshold VSS1, VSS2 Rising from 2V 1.3 1.5 1.7 V SS Discharge Current Short-Circuit Condition VFB1,2 = 0.5V VSS1,2 = 4.5V 7 10 13 µA VFB1,2 = 0.7V, VSENSE1–,2– = 3.3V 43 50 57 mV VSENSE(MAX) Maximum Current Sense Threshold V Gate Driver TG1,2 Pull-Up On-Resistance Pull-Down On-Resistance 2.5 1.5 Ω Ω BG1,2 Pull-Up On-Resistance Pull-Down On-Resistance 2.4 1.1 Ω Ω 38681fb LTC3868-1 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS TG1,2 tr TG1,2 t f TG Transition Time: Rise Time Fall Time (Note 6) CLOAD = 3300pF CLOAD = 3300pF 25 16 ns ns BG1,2 tr BG1,2 t f BG Transition Time: Rise Time Fall Time (Note 6) CLOAD = 3300pF CLOAD = 3300pF 28 13 ns ns TG/BG t1D Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time CLOAD = 3300pF Each Driver 30 ns BG/TG t1D Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time CLOAD = 3300pF Each Driver 30 ns tON(MIN) Minimum On-Time (Note 7) 95 ns INTVCC Linear Regulator VINTVCCVIN Internal VCC Voltage 6V < VIN < 24V, VEXTVCC = 0V VLDOVIN INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 0V VINTVCCEXT Internal VCC Voltage 6V < VEXTVCC < 13V 4.85 4.85 VLDOEXT INTVCC Load Regulation ICC = 0mA to 50mA, VEXTVCC = 8.5V VEXTVCC EXTVCC Switchover Voltage EXTVCC Ramping Positive VLDOHYS EXTVCC Hysteresis 4.5 5.1 5.35 V 0.7 1.1 % 5.1 5.35 V 0.6 1.1 % 4.7 4.9 V 250 mV 105 kHz Oscillator and Phase-Locked Loop f 25kΩ Programmable Frequency RFREQ = 25k, PLLIN/MODE = DC Voltage f 65kΩ Programmable Frequency RFREQ = 65k, PLLIN/MODE = DC Voltage f105kΩ Programmable Frequency RFREQ = 105k, PLLIN/MODE = DC Voltage fLOW Low Fixed Frequency VFREQ = 0V, PLLIN/MODE = DC Voltage fHIGH High Fixed Frequency VFREQ = INTVCC, PLLIN/MODE = DC Voltage fSYNC Synchronizable Frequency PLLIN/MODE = External Clock 375 440 505 835 l kHz kHz 320 350 380 kHz 485 535 585 kHz 850 kHz 0.4 V ±1 µA 75 PGOOD1 Output VPGL PGOOD1 Voltage Low IPGOOD = 2mA IPGOOD PGOOD1 Leakage Current VPGOOD = 5V VPG PGOOD1 Trip Level VFB with Respect to Set Regulated Voltage VFB Ramping Negative Hysteresis –13 –10 2.5 –7 % % VFB with Respect to Set Regulated Voltage VFB Ramping Positive Hysteresis 7 10 2.5 13 % % tPG Delay for Reporting a Fault (PGOOD Low) Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Ratings for extended periods may affect device reliability and lifetime. Note 2: The LTC3868E-1 is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3868I-1 is guaranteed over the full –40°C to 85°C operating temperature range. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA) 0.2 25 µs where θJA = 43°C for the QFN package and θJA = 90°C for the SSOP package. Note 4: The LTC3868-1 is tested in a feedback loop that servos VITH1,2 to a specified voltage and measures the resultant VFB1,2. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications information. Note 6: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 7: The minimum on-time condition is specified for an inductor peak-to-peak ripple current ≥ 40% of IMAX (See Minimum On-Time Considerations in the Applications Information section). 38681fb LTC3868-1 Typical Performance Characteristics Efficiency and Power Loss vs Output Current Efficiency vs Load Current 100 90 60 100 50 10 40 30 Burst Mode OPERATION PULSESKIPPING FCM 20 10 0.001 0.01 0.1 1 OUTPUT CURRENT (A) EFFICIENCY (%) 70 VIN = 5V 80 1000 POWER LOSS (mW) EFFICIENCY (%) FIGURE 12 CIRCUIT 90 VIN = 12V VOUT = 3.3V 80 0 0.0001 100 10000 70 VIN = 12V 60 50 40 30 20 1 10 10 0.1 0 0.0001 VOUT = 3.3V FIGURE 12 CIRCUIT 0.001 0.01 0.1 1 OUTPUT CURRENT (A) 38681 G02 38681 G01 98 FIGURE 12 CIRCUIT VOUT = 3.3V IOUT = 4A 96 EFFICIENCY (%) 94 Load Step (Forced Continuous Mode) Load Step (Burst Mode Operation) Efficiency vs Input Voltage VOUT 100mV/DIV ACCOUPLED 92 10 VOUT 100mV/DIV ACCOUPLED 90 88 IL 2A/DIV 86 IL 2A/DIV 84 82 80 0 5 20 10 15 INPUT VOLTAGE (V) 25 28 VOUT = 3.3V 20µs/DIV FIGURE 12 CIRCUIT 38681 G04 38681 G05 20µs/DIV VOUT = 3.3V FIGURE 12 CIRCUIT 38681 G03 Inductor Current at Light Load Load Step (Pulse-Skipping Mode) VOUT 100mV/DIV ACCOUPLED IL 2A/DIV Soft Start-Up FORCED CONTINUOUS MODE VOUT2 2V/DIV Burst Mode OPERATION 2A/DIV VOUT1 2V/DIV PULSESKIPPING MODE VOUT = 3.3V 20µs/DIV FIGURE 12 CIRCUIT 38681 G06 VOUT = 3.3V 2µs/DIV ILOAD = 200µA FIGURE 12 CIRCUIT 38681 G07 20ms/DIV FIGURE 12 CIRCUIT 38681 G08 38681fb LTC3868-1 Typical Performance Characteristics Total Input Supply Current vs Input Voltage 300µA LOAD 250 200 NO LOAD 150 100 50 0 10 5 15 20 INPUT VOLTAGE (V) 25 5.4 5.2 INTVCC 5.0 EXTVCC RISING 4.8 EXTVCC FALLING 4.6 4.4 4.0 –45 –20 28 80 55 30 TEMPERATURE (°C) 5 38681 G10 Maximum Current Sense Voltage vs ITH Voltage –50 –20 –150 –200 –250 –300 –350 –400 –450 –500 5% DUTY CYCLE 0.2 0.4 0.6 0.8 1.0 ITH PIN VOLTAGE 1.2 1.4 –550 –600 0 10 5 VSENSE COMMON MODE VOLTAGE (V) 80 50 40 30 20 10 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 FEEDBACK VOLTAGE (V) 38681 G16 10 15 20 INPUT VOLTAGE (V) 240 230 PLLIN/MODE = 0 V = 12V 220 VIN = 3.3V OUT 210 ONE CHANNEL ON 200 190 180 170 160 150 140 130 120 110 80 55 5 –45 –20 30 TEMPERATURE (°C) 25 28 38681 G12 80 60 40 20 0 15 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 38681 G15 Shutdown Current vs Temperature Quiescent Current vs Temperature QUIESCENT CURRENT (µA) MAXIMUM CURRENT SENSE VOLTAGE (mV) Foldback Current Limit 90 60 5 38681 G14 38681 G13 70 0 Maximum Current Sense Threshold vs Duty Cycle –100 0 0 5.0 130 0 20 –40 105 MAXIMUM CURRENT SENSE VOLTAGE (mV) 40 5.1 SENSE– Pins Input Bias Current PULSE-SKIPPING FORCED CONTINUOUS Burst Mode OPERATION (FALLING) Burst Mode OPERATION (RISING) 60 5.1 38681 G11 SENSE– CURRENT (µA) CURRENT SENSE THRESHOLD (mV) 80 5.2 4.2 10 SHUTDOWN CURRENT (µA) SUPPLY CURRENT (µA) 300 5.2 INTVCC VOLTAGE (V) FIGURE 12 CIRCUIT VOUT = 3.3V ONE CHANNEL ON 350 INTVCC Line Regulation 5.6 EXTVCC AND INTVCC VOLTAGE (V) 400 EXTVCC Switchover and INTVCC Voltages vs Temperature 9 8 7 6 5 105 130 38681 G17 4 –45 –20 55 30 80 5 TEMPERATURE (°C) 105 130 38681 G18 38681fb LTC3868-1 Typical Performance Characteristics Soft-Start Pull-Up Current vs Temperature 1.40 1.20 808 REGULATED FEEDBACK VOLTAGE (mV) 1.35 1.15 1.30 1.10 RUN PIN VOLTAGE (V) SS PULL-UP CURRENT (µA) Regulated Feedback Voltage vs Temperature Shutdown (RUN) Threshold vs Temperature 1.05 1.00 0.95 0.90 1.25 1.20 1.15 1.10 1.05 1.00 0.85 0.95 0.80 –45 –20 80 55 30 TEMPERATURE (°C) 5 105 0.90 –45 130 55 30 5 80 TEMPERATURE (°C) –20 105 800 798 796 794 792 –45 –20 130 130 14 800 12 700 600 10 8 6 4 0 105 130 38681 G21 FREQUENCY (kHz) INPUT CURRENT (µA) 105 80 55 30 TEMPERATURE (°C) 5 Oscillator Frequency vs Temperature FREQ = INTVCC 500 FREQ = GND 400 300 200 2 VOUT = 28V 80 55 5 30 TEMPERATURE (°C) 802 Shutdown Current vs Input Voltage VOUT = 3.3V 100 5 10 25 20 15 INPUT VOLTAGE (V) 38681 G22 28 0 –45 –20 80 55 30 TEMPERATURE (°C) 5 38681 G23 105 130 38681 G24 Undervoltage Lockout Threshold vs Temperature Oscillator Frequency vs Input Voltage 4.4 356 4.3 354 4.2 INTVCC VOLTAGE (V) OSCILLATOR FREQUENCY (kHz) SENSE– CURRENT (µA) SENSE– Pin Input Current vs Temperature –20 804 38681 G20 38681 G19 50 0 –50 –100 –150 –200 –250 –300 –350 –400 –450 –500 –550 –600 –45 806 352 350 348 4.0 3.9 3.8 3.7 3.6 346 344 4.1 3.5 5 10 20 15 INPUT VOLTAGE (V) 25 28 38681 G28 3.4 –45 –20 55 30 5 80 TEMPERATURE (°C) 105 130 38681 G25 38681fb LTC3868-1 Typical Performance Characteristics Latchoff Thresholds vs Temperature INTVCC vs Load Current 5.20 2.3 VIN = 12V 2.2 2.1 INTVCC VOLTAGE (V) INTVCC VOLTAGE (V) 5.15 5.10 EXTVCC = 0V 5.05 EXTVCC = 8V 5.00 ARMING THRESHOLD 2.0 1.9 1.8 1.7 1.6 LATCH-OFF THRESHOLD 1.5 1.4 1.3 4.95 0 20 40 60 80 100 120 140 160 180 200 LOAD CURRENT (mA) 38681 G26 Pin Functions 1.2 –45 –20 55 30 5 80 TEMPERATURE (°C) 105 130 38681 G27 (QFN/SSOP) SENSE1–, SENSE2– (Pin 2, Pin 4/Pin 8, Pin 10): The (–) Input to the Differential Current Comparators. When greater than INTVCC – 0.5V, the SENSE– pin supplies current to the current comparator. FREQ (Pin 3/Pin 5): The Frequency Control Pin for the Internal VCO. Connecting this pin to GND forces the VCO to a fixed low frequency of 350kHz. Connecting this pin to INTVCC forces the VCO to a fixed high frequency of 535kHz. Other frequencies between 50kHz and 900kHz can be programmed using a resistor between FREQ and GND. An internal 20µA pull-up current develops the voltage to be used by the VCO to control the frequency PLLIN/MODE (Pin 4/Pin 6): External Synchronization Input to Phase Detector and Forced Continuous Mode Input. When an external clock is applied to this pin, the phase-locked loop will force the rising TG1 signal to be synchronized with the rising edge of the external clock. When not synchronizing to an external clock, this input, which acts on both controllers, determines how the LTC3868-1 operates at light loads. Pulling this pin to ground selects Burst Mode operation. An internal 100k resistor to ground also invokes Burst Mode operation when the pin is floated. Tying this pin to INTVCC forces continuous inductor current operation. Tying this pin to a voltage greater than 1.2V and less than INTVCC – 1.3V selects pulse-skipping operation. SGND (Pin 5, Exposed Pad Pin 29/Pin 7): Small-signal ground common to both controllers, must be routed separately from high current grounds to the common (–) terminals of the CIN capacitors. The exposed pad (QFN only) must be soldered to the PCB for rated thermal performance. RUN1, RUN2 (Pin 6, Pin 8/Pin 7, Pin 9): Digital Run Control Inputs for Each Controller. Forcing either of these pins below 1.26V shuts down that controller. Forcing both of these pins below 0.7V shuts down the entire LTC3868‑1, reducing quiescent current to approximately 8µA. Do not float these pins. 38681fb LTC3868-1 Pin Functions (QFN/SSOP) INTVCC (Pin 17/Pin 19): Output of the Internal Linear Low Dropout Regulator. The driver and control circuits are powered from this voltage source. Must be decoupled to power ground with a minimum of 4.7µF ceramic or other low ESR capacitor. Do not use the INTVCC pin for any other purpose. EXTVCC (Pin 18/Pin 20): External Power Input to an Internal LDO Connected to INTVCC. This LDO supplies INTVCC power, bypassing the internal LDO powered from VIN whenever EXTVCC is higher than 4.7V. See EXTVCC Connection in the Applications Information section. Do not exceed 14V on this pin. PGND (Pin 19/Pin 21): Driver Power Ground. Connects to the sources of bottom (synchronous) N-channel MOSFETs and the (–) terminal(s) of CIN. VIN (Pin 20/Pin 22): Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. TG1, TG2 (Pin 24, Pin 26/Pin 13, Pin 15): High Current Gate Drives for Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V superimposed on the switch node voltage SW. PGOOD1 (Pin 25/Pin 27): Open-Drain Logic Output. PGOOD1 is pulled to ground when the voltage on the VFB1 pin is not within ±10% of its set point. SS1, SS2 (Pin 26, Pin 28/Pin 12, Pin 14): External SoftStart Input. The LTC3868-1 regulates the VFB1,2 voltage to the smaller of 0.8V or the voltage on the SS1,2 pin. An internal 1µA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the ramp time to final regulated output voltage. This pin is also used as the short-circuit latchoff timer. ITH1, ITH2 (Pin 27, Pin 1/Pin 11, Pin 13): Error Amplifier Outputs and Switching Regulator Compensation Points. Each associated channel’s current comparator trip point increases with this control voltage. BG1, BG2 (Pin 21, Pin 23/Pin 16, Pin 18): High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is from ground to INTVCC. VFB1, VFB2 (Pin 28, Pin 2/Pin 10, Pin 12): Receives the remotely sensed feedback voltage for each controller from an external resistive divider across the output. BOOST1, BOOST2 (Pin 22, Pin 24/Pin 15, Pin 17): Bootstrapped Supplies to the Topside Floating Drivers. Capacitors are connected between the BOOST and SW pins and Schottky diodes are tied between the BOOST and INTVCC pins. Voltage swing at the BOOST pins is from INTVCC to (VIN + INTVCC). SENSE1+, SENSE2+ (Pin 1, Pin 3/Pin 9, Pin 11): The (+) input to the differential current comparators are normally connected to DCR sensing networks or current sensing resistors. The ITH pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold. SW1, SW2 (Pin 23, Pin 25/Pin 14, Pin 16): Switch Node Connections to Inductors. 38681fb LTC3868-1 FUNCTIONAL Diagram INTVCC DUPLICATE FOR SECOND CONTROLLER CHANNEL PGOOD1 BOOST DROP OUT DET 0.88V VFB1 + – + 0.72V S Q R Q TOP ON SWITCH LOGIC BOT INTVCC BG VOUT CLK2 0.425V – CLK1 + SLEEP – ICMP PFD + + –+ +– – SYNC DET PLLIN/MODE IR – SENSE+ 2(VFB) 0.45V 100k SENSE– SLOPE COMP VFB VIN + EA – OV – 5.1V LDO EN LDO EN 0.5µA + – SHDN RST 2(VFB) 0.80V TRACK/SS 0.88V SGND INTVCC RUN RA CC ITH CC2 FOLDBACK 1µA 11V RB + EXTVCC 5.1V RSENSE L 3mV 4.7V COUT PGND VCO CIN SW 20µA FREQ CB D BOT SHDN DB TG TOP VIN SHORT CKT LATCH-OFF RC SS CSS SHDN 10µA 38681 FD 38681fb 10 LTC3868-1 Operation (Refer to the Functional Diagram) Main Control Loop The LTC3868-1 uses a constant frequency, current mode step-down architecture with the two controller channels operating 180 degrees out of phase. During normal operation, each external top MOSFET is turned on when the clock for that channel sets the RS latch, and is turned off when the main current comparator, ICMP, resets the RS latch. The peak inductor current at which ICMP trips and resets the latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier, EA. The error amplifier compares the output voltage feedback signal at the VFB pin (which is generated with an external resistor divider connected across the output voltage, VOUT , to ground) to the internal 0.800V reference voltage. When the load current increases, it causes a slight decrease in VFB relative to the reference, which causes the EA to increase the ITH voltage until the average inductor current matches the new load current. After the top MOSFET is turned off each cycle, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current comparator IR, or the beginning of the next clock cycle. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open or tied to a voltage less than 4.7V, the VIN LDO (low dropout linear regulator) supplies 5.1V from VIN to INTVCC. If EXTVCC is taken above 4.7V, the VIN LDO is turned off and an EXTVCC LDO is turned on. Once enabled, the EXTVCC LDO supplies 5.1V from EXTVCC to INTVCC. Using the EXTVCC pin allows the INTVCC power to be derived from a high efficiency external source such as one of the LTC3868-1 switching regulator outputs. Each top MOSFET driver is biased from the floating bootstrap capacitor, CB, which normally recharges during each cycle through an external diode when the top MOSFET turns off. If the input voltage VIN decreases to a voltage close to VOUT , the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about one-twelfth of the clock period every tenth cycle to allow CB to recharge. Shutdown and Start-Up (RUN1, RUN2 and SS1, SS2 Pins) The two channels of the LTC3868-1 can be independently shut down using the RUN1 and RUN2 pins. Pulling either of these pins below 1.26V shuts down the main control loop for that controller. Pulling both pins below 0.7V disables both controllers and most internal circuits, including the INTVCC LDOs. In this state, the LTC3868-1 draws only 8µA of quiescent current. The RUN pin may be externally pulled up or driven directly by logic. When driving the RUN pin with a low impedance source, do not exceed the absolute maximum rating of 8V. The RUN pin has an internal 11V voltage clamp that allows the RUN pin to be connected through a resistor to a higher voltage (for example, VIN), so long as the maximum current into the RUN pin does not exceed 100µA. The start-up of each controller’s output voltage VOUT is controlled by the voltage on the SS pin for that channel. When the voltage on the SS pin is less than the 0.8V internal reference, the LTC3868-1 regulates the VFB voltage to the SS pin voltage instead of the 0.8V reference. This allows the SS pin to be used to program a soft-start by connecting an external capacitor from the SS pin to SGND. An internal 1µA pull-up current charges this capacitor creating a voltage ramp on the SS pin. As the SS voltage rises linearly from 0V to 0.8V (and beyond up to the absolute maximum rating of 6V), the output voltage VOUT rises smoothly from zero to its final value. Short-Circuit Latchoff After the controller has been started and been given adequate time to ramp up the output voltage, the SS capacitor is used in a short-circuit timeout circuit. Specifically, once the voltage on the SS pin rises above 2V (the arming threshold), the short-circuit timeout circuit is enabled (see Figure 1). If the output voltage falls below 70% of its nominal regulated voltage, the SS capacitor begins discharging with a net 9µA pulldown current on the assumption that the output is in an overcurrent and/or short-circuit condition. If the condition lasts long enough to allow the SS pin voltage to fall below 1.5V (the latchoff threshold), the controller will shut down (latch off) until the RUN pin voltage or the VIN voltage is recycled. 38681fb 11 LTC3868-1 Operation (Refer to the Functional Diagram) current is provided due to internal current foldback and actual power wasted is low due to the efficient nature of the current mode switching regulator. Foldback current limiting is disabled during the soft-start interval (as long as the VFB voltage is keeping up with the SS voltage). INTVCC SS VOLTAGE 2V 1.5V 0.8V LATCHOFF COMMAND SS PIN CURRENT Light Load Current Operation (Burst Mode Operation, Pulse-Skipping or Forced Continuous Mode) (PLLIN/MODE Pin) 0V 1µA 1µA –9µA OUTPUT VOLTAGE LATCHOFF ENABLE ARMING tLATCH 38681 F01 SOFT-START INTERVAL Figure 1. Latchoff Timing Diagram The delay time from when a short-circuit occurs until the controller latches off can be calculated using the following equation tLATCH ~ CSS (VSS – 1.5V)/9µA where VSS is the initial voltage (must be greater than 2V) on the SS pin at the time the short-circuit occurs. Normally the SS pin voltage will have been pulled up to the INTVCC voltage (5.1V) by the internal 1µA pull-up current. Note that the two controllers on the LTC3868-1 have separate, independent short-circuit latchoff circuits. Latchoff can be overridden/defeated by connecting a resistor 150k or less from the SS pin to INTVCC. This resistor provides enough pull-up current to overcome the 9µA pull-down current present during a short-circuit. Note that this resistor also shortens the soft-start period. Foldback Current On the other hand, when the output voltage falls to less than 70% of its nominal level, foldback current limiting is also activated, progressively lowering the peak current limit in proportion to the severity of the overcurrent or short-circuit condition. Even if a short-circuit is present and the short-circuit latchoff is not yet enabled (when SS voltage has not yet reached 2V), a safe, low output The LTC3868-1 can be enabled to enter high efficiency Burst Mode operation, constant frequency pulse-skipping mode, or forced continuous conduction mode at low load currents. To select Burst Mode operation, tie the PLLIN/ MODE pin to ground. To select forced continuous operation, tie the PLLIN/MODE pin to INTVCC. To select pulseskipping mode, tie the PLLIN/MODE pin to a DC voltage greater than 1.2V and less than INTVCC – 1.3V. When a controller is enabled for Burst Mode operation, the minimum peak current in the inductor is set to approximately 30% of the maximum sense voltage even though the voltage on the ITH pin indicates a lower value. If the average inductor current is higher than the load current, the error amplifier, EA, will decrease the voltage on the ITH pin. When the ITH voltage drops below 0.425V, the internal sleep signal goes high (enabling sleep mode) and both external MOSFETs are turned off. In sleep mode, much of the internal circuitry is turned off, reducing the quiescent current. If one channel is shut down and the other channel is in sleep mode, the LTC3868-1 draws only 170µA of quiescent current. If both channels are in sleep mode, the LTC3868-1 draws only 300µA of quiescent current. In sleep mode, the load current is supplied by the output capacitor. As the output voltage decreases, the EA’s output begins to rise. When the output voltage drops enough, the ITH pin is reconnected to the output of the EA, the sleep signal goes low, and the controller resumes normal operation by turning on the top external MOSFET on the next cycle of the internal oscillator. When a controller is enabled for Burst Mode operation, the inductor current is not allowed to reverse. The reverse 38681fb 12 LTC3868-1 Operation (Refer to the Functional Diagram) current comparator, IR, turns off the bottom external MOSFET just before the inductor current reaches zero, preventing it from reversing and going negative. Thus, the controller is in discontinuous operation. In forced continuous operation or when clocked by an external clock source to use the phase-locked loop (see Frequency Selection and Phase-Locked Loop section), the inductor current is allowed to reverse at light loads or under large transient conditions. The peak inductor current is determined by the voltage on the ITH pin, just as in normal operation. In this mode, the efficiency at light loads is lower than in Burst Mode operation. However, continuous operation has the advantages of lower output voltage ripple and less interference to audio circuitry. In forced continuous mode, the output ripple is independent of load current. When the PLLIN/MODE pin is connected for pulse-skipping mode, the LTC3868-1 operates in PWM pulse-skipping mode at light loads. In this mode, constant frequency operation is maintained down to approximately 1% of designed maximum output current. At very light loads, the current comparator, ICMP, may remain tripped for several cycles and force the external top MOSFET to stay off for the same number of cycles (i.e., skipping pulses). The inductor current is not allowed to reverse (discontinuous operation). This mode, like forced continuous operation, exhibits low output ripple as well as low audio noise and reduced RF interference when compared to Burst Mode operation. It provides higher light load efficiency than forced continuous mode, but not nearly as high as Burst Mode operation. Frequency Selection and Phase-Locked Loop (FREQ and PLLIN/MODE Pins) The selection of switching frequency is a trade off between efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching losses, but requires larger inductance and/or capacitance to maintain low output ripple voltage. The switching frequency of the LTC3868-1’s controllers can be selected using the FREQ pin. If the PLLIN/MODE pin is not being driven by an external clock source, the FREQ pin can be tied to SGND, tied to INTVCC or programmed through an external resistor. Tying FREQ to SGND selects 350kHz while tying FREQ to INTVCC selects 535kHz. Placing a resistor between FREQ and SGND allows the frequency to be programmed between 50kHz and 900kHz. A phase-locked loop (PLL) is available on the LTC3868-1 to synchronize the internal oscillator to an external clock source that is connected to the PLLIN/MODE pin. The phase detector adjusts the voltage (through an internal lowpass filter) of the VCO input to align the turn-on of controller 1’s external top MOSFET to the rising edge of the synchronizing signal. Thus, the turn-on of controller 2’s external top MOSFET is 180 degrees out of phase to the rising edge of the external clock source. The VCO input voltage is prebiased to the operating frequency set by the FREQ pin before the external clock is applied. If prebiased near the external clock frequency, the PLL loop only needs to make slight changes to the VCO input in order to synchronize the rising edge of the external clock’s to the rising edge of TG1. The ability to prebias the loop filter allows the PLL to lock-in rapidly without deviating far from the desired frequency. The typical capture range of the phase-locked loop is from approximately 55kHz to 1MHz, with a guarantee over all manufacturing variations to be between 75kHz and 850kHz. In other words, the LTC3868-1’s PLL is guaranteed to lock to an external clock source whose frequency is between 75kHz and 850kHz. The typical input clock thresholds on the PLLIN/MODE pin are 1.6V (rising) and 1.1V (falling). Output Overvoltage Protection An overvoltage comparator guards against transient overshoots as well as other more serious conditions that may overvoltage the output. When the VFB pin rises by more than 10% above its regulation point of 0.800V, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. 38681fb 13 LTC3868-1 Operation (Refer to the Functional Diagram) Power Good (PGOOD) Pin The PGOOD1 pin is connected to an open drain of an internal N-channel MOSFET. The MOSFET turns on and pulls the PGOOD1 pin low when the corresponding VFB1 pin voltage is not within ±10% of the 0.8V reference voltage. The PGOOD1 pin is also pulled low when the RUN1 pin is low (shut down). When the VFB1 pin voltage is within the ±10% requirement, the MOSFET is turned off and the pin is allowed to be pulled up by an external resistor to a source no greater than 6V. pulses increased the total RMS current flowing from the input capacitor, requiring the use of more expensive input capacitors and increasing both EMI and losses in the input capacitor and battery. Theory and Benefits of 2-Phase Operation With 2-phase operation, the two channels of the dual switching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency. Why the need for 2-phase operation? Up until the 2-phase family, constant frequency dual switching regulators operated both channels in phase (i.e., single phase operation). This means that both switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor and battery. These large amplitude current Figure 2 compares the input waveforms for a representative single phase dual switching regulator to the LTC3868-1 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2-phase operation dropped the input current from 2.53ARMS to 1.55ARMS. While this is an impressive reduction in itself, remember that the power losses are 5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV IIN(MEAS) = 2.53ARMS IIN(MEAS) = 1.55ARMS 38681 F01 Figure 2. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency 38681fb 14 LTC3868-1 Operation (Refer to the Functional Diagram) proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative duty cycles which, in turn, are dependent upon the input voltage VIN (Duty Cycle = VOUT/VIN). Figure 3 shows how the RMS input current varies for single-phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, for most applications is that 2-phase operation will reduce the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle. 3.0 SINGLE PHASE DUAL CONTROLLER INPUT RMS CURRENT (A) 2.5 2.0 1.5 2-PHASE DUAL CONTROLLER 1.0 0.5 0 VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40 38681 F03 Figure 3. RMS Input Current Comparison 38681fb 15 LTC3868-1 Applications Information The Typical Application on the first page is a basic LTC3868‑1 application circuit. LTC3868-1 can be configured to use either DCR (inductor resistance) sensing or low value resistor sensing. The choice between the two current sensing schemes is largely a design trade off between cost, power consumption and accuracy. DCR sensing is becoming popular because it saves expensive current sensing resistors and is more power efficient, especially in high current applications. However, current sensing resistors provide the most accurate current limits for the controller. Other external component selection is driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is used) and inductor value. Next, the power MOSFETs and Schottky diodes are selected. Finally, input and output capacitors are selected. SENSE+ and SENSE– Pins The SENSE+ and SENSE– pins are the inputs to the current comparators. The common mode voltage range on these pins is 0V to 16V (Absolute Maximum), enabling the LTC3868-1 to regulate output voltages up to a nominal 14V (allowing margin for tolerances and transients). programmed current limit unpredictable. If inductor DCR sensing is used (Figure 5b), resistor R1 should be placed close to the switching node, to prevent noise from coupling into sensitive small-signal nodes. TO SENSE FILTER, NEXT TO THE CONTROLLER COUT 38681 F04 INDUCTOR OR RSENSE Figure 4. Sense Lines Placement with Inductor or Sense Resistor VIN INTVCC BOOST TG SW LTC3868-1 Filter components mutual to the sense lines should be placed close to the LTC3868-1, and the sense lines should run close together to a Kelvin connection underneath the current sense element (shown in Figure 4). Sensing current elsewhere can effectively add parasitic inductance and capacitance to the current sense element, degrading the information at the sense terminals and making the VOUT BG SENSE+ PLACE CAPACITOR NEAR SENSE PINS SENSE– The SENSE+ pin is high impedance over the full common mode range, drawing at most ±1µA. This high impedance allows the current comparators to be used in inductor DCR sensing. The impedance of the SENSE– pin changes depending on the common mode voltage. When SENSE– is less than INTVCC – 0.5V, a small current of less than 1µA flows out of the pin. When SENSE– is above INTVCC + 0.5V, a higher current (~550µA) flows into the pin. Between INTVCC – 0.5V and INTVCC + 0.5V, the current transitions from the smaller current to the higher current. VIN SGND 38681 F05a (5a) Using a Resistor to Sense Current VIN INTVCC VIN BOOST INDUCTOR TG L SW LTC3868-1 DCR VOUT BG R1 SENSE+ C1* R2 SENSE– SGND *PLACE C1 NEAR SENSE PINS (R1||R2) • C1 = L DCR RSENSE(EQ) = DCR R2 R1 + R2 38681 F05b (5b) Using the Inductor DCR to Sense Current Figure 5. Current Sensing Methods 38681fb 16 LTC3868-1 Applications Information Low Value Resistor Current Sensing A typical sensing circuit using a discrete resistor is shown in Figure 5a. RSENSE is chosen based on the required output current. The current comparator has a maximum threshold VSENSE(MAX) of 50mV. The current comparator threshold voltage sets the peak of the inductor current, yielding a maximum average output current, IMAX, equal to the peak value less half the peak-to-peak ripple current, ∆IL. To calculate the sense resistor value, use the equation: RSENSE = VSENSE(MAX) IMAX + ∆IL 2 When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability criterion for buck regulators operating at greater than 50% duty factor. A curve is provided in the Typical Performance Characteristics section to estimate this reduction in peak output current depending upon the operating duty factor. using a good RLC meter, but the DCR tolerance is not always the same and varies with temperature; consult the manufacturers’ data sheets for detailed information. Using the inductor ripple current value from the Inductor Value Calculation section, the target sense resistor value is: RSENSE(EQUIV) = If the external R1||R2 • C1 time constant is chosen to be exactly equal to the L/DCR time constant, the voltage drop across the external capacitor is equal to the drop across the inductor DCR multiplied by R2/(R1 + R2). R2 scales the voltage across the sense terminals for applications where the DCR is greater than the target sense resistor value. To properly dimension the external filter components, the DCR of the inductor must be known. It can be measured IMAX + ∆IL 2 To ensure that the application will deliver full load current over the full operating temperature range, choose the minimum value for the maximum current sense threshold voltage (VSENSE(MAX)) in the Electrical Characteristics table. Next, determine the DCR of the inductor. When provided, use the manufacturer’s maximum value, usually given at 20°C. Increase this value to account for the temperature coefficient of copper resistance, which is approximately 0.4%/°C. A conservative value for TL(MAX) is 100°C. To scale the maximum inductor DCR to the desired sense resistor (RD) value, use the divider ratio: Inductor DCR Sensing For applications requiring the highest possible efficiency at high load currents, the LTC3850 is capable of sensing the voltage drop across the inductor DCR, as shown in Figure 5b. The DCR of the inductor represents the small amount of DC resistance of the copper wire, which can be less than 1mΩ for today’s low value, high current inductors. In a high current application requiring such an inductor, power loss through a sense resistor would cost several points of efficiency compared to inductor DCR sensing. VSENSE(MAX) RD = RSENSE(EQUIV ) DCRMAX at TL(MAX ) C1 is usually selected to be in the range of 0.1µF to 0.47µF. This forces R1||R2 to around 2k, reducing error that might have been caused by the SENSE+ pin’s ±1µA current. The equivalent resistance R1||R2 is scaled to the room temperature inductance and maximum DCR: R1|| R2 = L DCR at 20°C • C1 ( ) The sense resistor values are: R1 = R1 • RD R1|| R2 ; R2 = RD 1 – RD 38681fb 17 LTC3868-1 Applications Information The maximum power loss in R1 is related to duty cycle, and will occur in continuous mode at the maximum input voltage: PLOSS R1 = ( VIN(MAX) – VOUT ) • VOUT R1 Ensure that R1 has a power rating higher than this value. If high efficiency is necessary at light loads, consider this power loss when deciding whether to use DCR sensing or sense resistors. Light load power loss can be modestly higher with a DCR network than with a sense resistor, due to the extra switching losses incurred through R1. However, DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads. Peak efficiency is about the same with either method. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current ∆IL decreases with higher inductance or higher frequency and increases with higher VIN: ΔIL = V 1 VOUT 1– OUT ( f) (L) VIN Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ∆IL = 0.3(IMAX). The maximum ∆IL occurs at the maximum input voltage. The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average inductor current required results in a peak current below 30% of the current limit determined by RSENSE. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite or molypermalloy cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance value selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred for high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Power MOSFET and Schottky Diode (Optional) Selection Two external power MOSFETs must be selected for each controller in the LTC3868-1: one N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5.1V during start-up (see EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. The only exception is if low input voltage is expected (VIN < 4V); then, sub-logic level threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic-level MOSFETs are limited to 30V or less. 38681fb 18 LTC3868-1 Applications Information Selection criteria for the power MOSFETs include the on-resistance, RDS(ON), Miller capacitance, CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in VDS. This result is then multiplied by the ratio of the application applied VDS to the gate charge curve specified VDS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT VIN Synchronous Switch Duty Cycle = VIN − VOUT VIN The MOSFET power dissipations at maximum output current are given by: V 2 PMAIN = OUT (IMAX ) (1+ δ) RDS(ON) + VIN 2 I (R ) (C ( VIN) MAX )• 2 DR MILLER 1 1 + ( f) VINTVCC – VTHMIN VTHMIN PSYNC = VIN – VOUT 2 IMAX ) (1+ δ) RDS(ON) ( VIN where δ is the temperature dependency of RDS(ON) and RDR (approximately 2Ω) is the effective driver resistance at the MOSFET’s Miller threshold voltage. VTHMIN is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1+ δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. The optional Schottky diodes D1 and D2 shown in Figure 10 conduct during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead-time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. CIN and COUT Selection The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst-case capacitor RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used in the formula shown in Equation 1 to determine the maximum RMS capacitor current requirement. Increasing the output current drawn from the other controller will actually decrease the input RMS ripple current from its maximum value. The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor of 30% to 70% when compared to a single phase power supply solution. In continuous mode, the source current of the top MOSFET is a square wave of duty cycle (VOUT)/(VIN). To prevent large voltage transients, a low ESR capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX 1/2 (1) V V – V ( ) ( ) VIN OUT IN OUT 38681fb 19 LTC3868-1 Applications Information Equation 1 has a maximum at VIN = 2VOUT , where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturers’ ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet size or height requirements in the design. Due to the high operating frequency of the LTC3868-1, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The benefit of the LTC3868-1 2-phase operation can be calculated by using Equation 1 for the higher power controller and then calculating the loss that would have resulted if both controller channels switched on at the same time. The total RMS power lost is lower when both controllers are operating due to the reduced overlap of current pulses required through the input capacitor’s ESR. This is why the input capacitor’s requirement calculated above for the worst-case controller is adequate for the dual controller design. Also, the input protection fuse resistance, battery resistance, and PC board trace resistance losses are also reduced due to the reduced peak currents in a 2-phase system. The overall benefit of a multiphase design will only be fully realized when the source impedance of the power supply/battery is included in the efficiency testing. The sources of the top MOSFETs should be placed within 1cm of each other and share a common CIN(s). Separating the sources and CIN may produce undesirable voltage and current resonances at VIN. A small (0.1µF to 1µF) bypass capacitor between the chip VIN pin and ground, placed close to the LTC3868-1, is also suggested. A 10Ω resistor placed between CIN (C1) and the VIN pin provides further isolation between the two channels. The selection of COUT is driven by the effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: ΔVOUT 1 ≈ ΔIL ESR + 8 • f • COUT where f is the operating frequency, COUT is the output capacitance and ∆IL is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Setting Output Voltage The LTC3868-1 output voltages are each set by an external feedback resistor divider carefully placed across the output, as shown in Figure 6. The regulated output voltage is determined by: R VOUT = 0.8V 1+ B RA To improve the frequency response, a feedforward capacitor, CFF , may be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. VOUT RB 1/2 LTC3868-1 CFF VFB RA 38681 F05 Figure 6. Setting Output Voltage Soft-Start (SS Pins) The start-up of each VOUT is controlled by the voltage on the respective SS pin. When the voltage on the SS pin is less than the internal 0.8V reference, the LTC3868-1 regulates the VFB pin voltage to the voltage on the SS pin instead of 0.8V. The SS pin can be used to program an external soft-start function. Soft-start is enabled by simply connecting a capacitor from the SS pin to ground, as shown in Figure 7. An internal 1µA current source charges the capacitor, providing a 1/2 LTC3868-1 SS CSS SGND 38681 F06 Figure 7. Using the SS Pin to Program Soft-Start 38681fb 20 LTC3868-1 Applications Information linear ramping voltage at the SS pin. The LTC3868-1 will regulate the VFB pin (and hence VOUT) according to the voltage on the SS pin, allowing VOUT to rise smoothly from 0V to its final regulated value. The total soft-start time will be approximately: 0.8 V tSS = CSS • 1µA INTVCC Regulators The LTC3868-1 features two separate internal P-channel low dropout linear regulators (LDO) that supply power at the INTVCC pin from either the VIN supply pin or the EXTVCC pin depending on the connection of the EXTVCC pin. INTVCC powers the gate drivers and much of the LTC3868-1’s internal circuitry. The VIN LDO and the EXTVCC LDO regulate INTVCC to 5.1V. Each of these can supply a peak current of 50mA and must be bypassed to ground with a minimum of 4.7µF low ESR capacitor. No matter what type of bulk capacitor is used, an additional 1µF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is needed to supply the high transient currents required by the MOSFET gate drivers and to prevent interaction between the channels. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3868-1 to be exceeded. The INTVCC current, which is dominated by the gate charge current, may be supplied by either the VIN LDO or the EXTVCC LDO. When the voltage on the EXTVCC pin is less than 4.7V, the VIN LDO is enabled. Power dissipation for the IC in this case is highest and is equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, the LTC3868-1 INTVCC current is limited to less than 22mA from a 28V supply when not using the EXTVCC supply at 70°C ambient temperature in the SSOP package: To prevent the maximum junction temperature from being exceeded, the input supply current must be checked while operating in forced continuous mode (PLLIN/MODE = INTVCC) at maximum VIN. When the voltage applied to EXTVCC rises above 4.7V, the VIN LDO is turned off and the EXTVCC LDO is enabled. The EXTVCC LDO remains on as long as the voltage applied to EXTVCC remains above 4.5V. The EXTVCC LDO attempts to regulate the INTVCC voltage to 5.1V, so while EXTVCC is less than 5.1V, the LDO is in dropout and the INTVCC voltage is approximately equal to EXTVCC. When EXTVCC is greater than 5.1V, up to an absolute maximum of 14V, INTVCC is regulated to 5.1V. Using the EXTVCC LDO allows the MOSFET driver and control power to be derived from one of the LTC3868-1’s switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during normal operation and from the VIN LDO when the output is out of regulation (e.g., start-up, short-circuit). If more current is required through the EXTVCC LDO than is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. In this case, do not apply more than 6V to the EXTVCC pin and make sure that EXTVCC ≤ VIN. Significant efficiency and thermal gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Switcher Efficiency). For 5V to 14V regulator outputs, this means connecting the EXTVCC pin directly to VOUT . Tying the EXTVCC pin to a 8.5V supply reduces the junction temperature in the previous example from 125°C to: TJ = 70°C + (45mA)(8.5V)(90°C/W) = 87°C However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from the output. TJ = 70°C + (22mA)(28V)(90°C/W) = 125°C 38681fb 21 LTC3868-1 Applications Information The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC Left Open (or Grounded). This will cause INTVCC to be powered from the internal 5.1V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC Connected Directly to VOUT . This is the normal connection for a 5V to 14V regulator and provides the highest efficiency. 3. EXTVCC Connected to an External Supply. If an external supply is available in the 5V to 14V range, it may be used to power EXTVCC. Ensure that EXTVCC < VIN. 4. EXTVCC Connected to an Output-Derived Boost Network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage that has been boosted to greater than 4.7V. This can be done with the capacitive charge pump shown in Figure 8. Ensure that EXTVCC < VIN. CIN BAT85 VIN BAT85 MTOP VN2222LL TG1 1/2 LTC3868-1 EXTVCC L SW RSENSE BAT85 VOUT MBOT BG1 D PGND COUT 38681 F08 Figure 8. Capacitive Charge Pump for EXTVCC Topside MOSFET Driver Supply (CB, DB) External bootstrap capacitors, CB, connected to the BOOST pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the Functional Diagram is charged though external diode DB from INTVCC when the SW pin is low. When one of the topside MOSFETs is to be turned on, the driver places the CB voltage across the gate-source of the desired MOSFET. This enhances the top MOSFET switch and turns it on. The switch node voltage, SW, rises to VIN and the BOOST pin follows. With the topside MOSFET on, the boost voltage is above the input supply: VBOOST = VIN + VINTVCC. The value of the boost capacitor, CB, needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than VIN(MAX). When adjusting the gate drive level, the final arbiter is the total input current for the regulator. If a change is made and the input current decreases, then the efficiency has improved. If there is no change in input current, then there is no change in efficiency. Fault Conditions: Current Limit and Current Foldback When the output current hits the current limit, the output voltage begins to drop. If the output voltage falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered to about one-half of its maximum selected value. Under short-circuit conditions with very low duty cycles, the LTC3868-1 will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The shortcircuit ripple current is determined by the minimum ontime, tON(MIN), of the LTC3868-1 (≈90ns), the input voltage and inductor value: V ΔIL(SC) = tON(MIN) IN L The resulting average short-circuit current is: ISC = 50% • ILIM(MAX) RSENSE 1 – ∆IL(SC) 2 Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the controller is operating. A comparator monitors the output for overvoltage conditions. The comparator detects faults greater than 10% 38681fb 22 LTC3868-1 Applications Information above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The bottom MOSFET remains on continuously for as long as the overvoltage condition persists; if VOUT returns to a safe level, normal operation automatically resumes. 1000 900 FREQUENCY (kHz) 800 A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. Phase-Locked Loop and Frequency Synchronization The LTC3868-1 has an internal phase-locked loop (PLL) comprised of a phase frequency detector, a lowpass filter, and a voltage-controlled oscillator (VCO). This allows the turn-on of the top MOSFET of controller 1 to be locked to the rising edge of an external clock signal applied to the PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET is thus 180 degrees out of phase with the external clock. The phase detector is an edge sensitive digital type that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector does not exhibit false lock to harmonics of the external clock. If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced continuously from the phase detector output, pulling up the VCO input. When the external clock frequency is less than fOSC, current is sunk continuously, pulling down the VCO input. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. The voltage at the VCO input is adjusted until the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the phase detector output is high impedance and the internal filter capacitor, CLP , holds the voltage at the VCO input. Typically, the external clock (on the PLLIN/MODE pin) input high threshold is 1.6V, while the input low threshold is 1.1V. 700 600 500 400 300 200 100 0 15 25 35 45 55 65 75 85 95 105 115 125 FREQ PIN RESISTOR (kΩ) 38681 F09 Figure 9. Relationship Between Oscillator Frequency and Resistor Value at the FREQ Pin Rapid phase locking can be achieved by using the FREQ pin to set a free-running frequency near the desired synchronization frequency. The VCO’s input voltage is prebiased at a frequency corresponding to the frequency set by the FREQ pin. Once prebiased, the PLL only needs to adjust the frequency slightly to achieve phase lock and synchronization. Although it is not required that the free-running frequency be near external clock frequency, doing so will prevent the operating frequency from passing through a large range of frequencies as the PLL locks. Note that the LTC3868-1 can only be synchronized to an external clock whose frequency is within range of the LTC3868-1’s internal VCO, which is nominally 55kHz to 1MHz. This is guaranteed to be between 75kHz and 850kHz. Table 2 summarizes the different states in which the FREQ pin can be used. Table 2 FREQ PIN PLLIN/MODE PIN FREQUENCY 0V DC Voltage 350kHz INTVCC DC Voltage 535kHz Resistor DC Voltage 50kHz–900kHz Any of the Above External Clock Phase–Locked to External Clock 38681fb 23 LTC3868-1 Applications Information Minimum On-Time Considerations Minimum on-time, tON(MIN), is the smallest time duration that the LTC3868-1 is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: tON(MIN) < VOUT VIN f () If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. The minimum on-time for the LTC3868-1 is approximately 95ns. However, as the peak sense voltage decreases the minimum on-time gradually increases up to about 130ns. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3868-1 circuits: 1) IC VIN current, 2) INTVCC regulator current, 3) I2R losses, 4) topside MOSFET transition losses. 1. The VIN current is the DC input supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents. VIN current typically results in a small (<0.1%) loss. 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the control circuit current. In continuous mode, IGATECHG = f(QT + QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. Supplying INTVCC from an output-derived power source through EXTVCC will scale the VIN current required for the driver and control circuits by a factor of (Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 2.5mA of VIN current. This reduces the midcurrent loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous mode the average output current flows through L and RSENSE, but is chopped between the topside MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L, RSENSE and ESR to obtain I2R losses. For example, if each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and RESR = 40mΩ (sum of both input and output capacitance losses), then the total resistance is 130mΩ. This results in losses ranging from 3% to 13% as the output current increases from 1A to 5A for a 5V output, or a 4% to 20% loss for a 3.3V output. Efficiency varies as the inverse square of VOUT for the same external components and output power level. The combined effects of increasingly lower output voltages and higher currents required by high performance digital systems is not doubling but quadrupling the importance of loss terms in the switching regulator system! 38681fb 24 LTC3868-1 Applications Information 4. Transition losses apply only to the topside MOSFET(s), and become significant only when operating at high input voltages (typically 15V or greater). Transition losses can be estimated from: can also be estimated by examining the rise time at the pin. The ITH external components shown in Figure 12 circuit will provide an adequate starting point for most applications. Transition Loss = (1.7) • VIN • 2 • IO(MAX) • CRSS • f The ITH series RC-CC filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop gain and phase. An output current pulse of 20% to 80% of full-load current having a rise time of 1µs to 10µs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Other hidden losses such as copper trace and internal battery resistances can account for an additional 5% to 10% efficiency degradation in portable systems. It is very important to include these system level losses during the design phase. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a minimum of 20µF to 40µF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. The LTC3868-1 2-phase architecture typically halves this input capacitance requirement over competing solutions. Other losses including Schottky conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load current transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT . ∆ILOAD also begins to charge or discharge COUT generating the feedback error signal that forces the regulator to adapt to the current change and return VOUT to its steady-state value. During this recovery time VOUT can be monitored for excessive overshoot or ringing, which would indicate a stability problem. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The availability of the ITH pin not only allows optimization of control loop behavior, but it also provides a DC coupled and AC filtered closed-loop response test point. The DC step, rise time and settling at this test point truly reflects the closed-loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth Placing a resistive load and a power MOSFET directly across the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC is decreased, the zero frequency will be kept the same, thereby keeping the phase shift the same in the most critical frequency range of the feedback loop. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of 38681fb 25 LTC3868-1 Applications Information CLOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. Design Example The power dissipation on the topside MOSFET can be easily estimated. Choosing a Fairchild FDS6982S dual MOSFET results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At maximum input voltage with T(estimated) = 50°C: PMAIN = ΔIL(NOM) = VOUT V 1– OUT ( f) (L) VIN(NOM) A 3.9µH inductor will produce 29% ripple current. The peak inductor current will be the maximum DC value plus one half the ripple current, or 6.88A. Increasing the ripple current will also help ensure that the minimum on-time of 95ns is not violated. The minimum on-time occurs at maximum VIN: VOUT 3.3V tON(MIN) = = = 4299ns VIN(MAX ) ( f) 22V (350kHz ) The equivalent RSENSE resistor value can be calculated by using the minimum value for the maximum current sense threshold (43mV): 43mV RSENSE ≤ = 0.006Ω 6.88A ( ) ( As a design example for one channel, assume VIN = 12V(nominal), VIN = 22V (max), VOUT = 3.3V, IMAX = 6A, VSENSE(MAX) = 50mV and f = 350kHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the FREQ pin to GND, generating 350kHz operation. The minimum inductance for 30% ripple current is: 2 3.3V 6A 1+ 0.005 50°C – 25°C 22V 2 6A 2.5Ω 215pF • 0.035Ω + 22V 2 1 1 5V – 2.3V + 2.3V 350kHz = 433mW ) ( ( )( ) ( ( ) )( ) ) A short-circuit to ground will result in a folded back current of: ( ) 25mV 1 95ns 22V ISC = – = 3.9A 0.006Ω 2 3.9µH with a typical value of RDS(ON) and δ = (0.005/°C)(25°C) = 0.125. The resulting power dissipated in the bottom MOSFET is: ( P = 3.9A SYNC )2 (1.125)(0.022Ω) = 376mW which is less than under full-load conditions. CIN is chosen for an RMS current rating of at least 3A at temperature assuming only this channel is on. COUT is chosen with an ESR of 0.02Ω for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(∆IL) = 0.02Ω(1.75A) = 35mVP-P Choosing 1% resistors: RA = 25k and RB = 78.1k yields an output voltage of 3.299V. 38681fb 26 LTC3868-1 Applications Information PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. These items are also illustrated graphically in the layout diagram of Figure 10. Figure 11 illustrates the current waveforms present in the various branches of the 2-phase synchronous regulators operating in the continuous mode. Check the following in your layout: 1. Are the top N-channel MOSFETs MTOP1 and MTOP2 located within 1cm of each other with a common drain connection at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large resonant loop. 2. Are the signal and power grounds kept separate? The combined IC signal ground pin and the ground return of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET, Schottky diode and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. 3. Do the LTC3868-1 VFB pins’ resistive dividers connect to the (+) terminals of COUT? The resistive divider must be connected between the (+) terminal of COUT and signal ground. The feedback resistor connections should not be along the high current input feeds from the input capacitor(s). v 4. Are the SENSE– and SENSE+ leads routed together with minimum PC trace spacing? The filter capacitor between SENSE+ and SENSE– should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. 5. Is the INTVCC decoupling capacitor connected close to the IC, between the INTVCC and the power ground pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1µF ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 6. Keep the switching nodes (SW1, SW2), top gate nodes (TG1, TG2), and boost nodes (BOOST1, BOOST2) away from sensitive small-signal nodes, especially from the opposites channel’s voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the output side of the LTC3868-1 and occupy minimum PC trace area. 7. Use a modified star ground technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTVCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the SGND pin of the IC. PC Board Layout Debugging Start with one controller on at a time. It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be maintained over the input voltage range down to dropout and until the output load drops below the low current operation threshold—typically 10% of the maximum designed current level in Burst Mode operation. The duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after each controller is checked for its individual performance should both controllers be turned on at the same time. A particularly difficult region of operation is when one controller channel is nearing its current comparator trip point when the other channel is turning on its top MOSFET. This occurs around 50% duty cycle on either channel due to the phasing of the internal clocks and may cause minor duty cycle jitter. 38681fb 27 LTC3868-1 Applications Information Reduce VIN from its nominal level to verify operation of the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while monitoring the outputs to verify operation. Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems coincide with high input voltavges and low output currents, look for capacitive coupling between the BOOST, SW, TG, and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between CIN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC. An embarrassing problem, which can be missed in an otherwise properly working switching regulator, results when the current sensing leads are hooked up backwards. The output voltage under this improper hookup will still be maintained but the advantages of current mode control will not be realized. Compensation of the voltage loop will be much more sensitive to component selection. This behavior can be investigated by temporarily shorting out the current sensing resistor—don’t worry, the regulator will still maintain control of the output voltage. SS1 LTC3868-1 ITH1 VFB1 RPU1 PGOOD1 PGOOD1 VPULL-UP (<6V) L1 SENSE1+ TG1 SENSE1– SW1 CB1 FREQ M1 BOOST1 M2 RSENSE VOUT1 D1 BG1 RUN1 RUN2 EXTVCC INTVCC SENSE2+ BG2 ITH2 SS2 RIN 1µF CERAMIC COUT1 PGND SENSE2– VFB2 CVIN VOUT1 CINTVCC VIN CIN COUT2 1µF CERAMIC M3 BOOST2 SW2 GND + SGND VIN + PLLIN/MODE + fIN M4 D2 CB2 RSENSE TG2 VOUT2 L2 38681 F10 Figure 10. Recommended Printed Circuit Layout Diagram 38681fb 28 LTC3868-1 Applications Information SW1 L1 D1 RSENSE1 VOUT1 COUT1 RL1 VIN RIN CIN SW2 BOLD LINES INDICATE HIGH SWITCHING CURRENT. KEEP LINES TO A MINIMUM LENGTH. D2 L2 RSENSE2 VOUT2 COUT2 RL2 38681 F11 Figure 11. Branch Current Waveforms 38681fb 29 LTC3868-1 Typical ApplicationS RB1 215k CF1 15pF LTC3868-1 SENSE1+ C1 1nF RA1 68.1k SENSE1– VFB1 CITH1A 150pF INTVCC PGOOD1 100k MBOT1 BG1 SW1 CITH1 820pF ITH1 CSS1 0.1µF D1 INTVCC PLLIN/MODE SGND TG2 EXTVCC RUN1 BOOST2 RUN2 FREQ RITH2 27k CITH2A 100pF SW2 ITH2 BG2 VIN 9V TO 24V D2 MTOP2 CB2 0.47µF L2 7.2µH RSENSE2 10mΩ MBOT2 VOUT2 8.5V 3A COUT2 150µF VFB2 SENSE2– C2 1nF CF2 39pF SS2 CIN 22µF CINT 4.7µF PGND CSS2 0.1µF COUT1 150µF VOUT1 3.3V 5A MTOP1 TG1 VIN SS1 RA2 44.2k RSENSE1 7mΩ CB1 0.47µF BOOST1 RITH1 15k CITH2 680pF L1 3.3µH SENSE2+ RB2 442k 38681 F12 COUT1, COUT2: SANYO 10TPD150M L1: SUMIDA CDEP105-3R2M L2: SUMIDA CDEP105-7R2M MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP Start-Up Efficiency vs Output Current SW Node Waveforms 100 90 EFFICIENCY (%) 80 70 VOUT = 8.5V VOUT2 2V/DIV VOUT = 3.3V SW1 5V/DIV 60 50 VOUT1 2V/DIV 40 SW2 5V/DIV 30 20 10 VIN = 12V Burst Mode OPERATION 0 0.00001 0.0001 0.001 0.01 0.1 OUTPUT CURRENT (A) 1 10 20ms/DIV 38681 F12c 1µs/DIV 38681 F12d 38681 F12b Figure 12. High Efficiency Dual 8.5V/3.3V Step-Down Converter 38681fb 30 LTC3868-1 Typical ApplicationS High Efficiency Dual 2.5V/3.3V Step-Down Converter RB1 143k CF1 22pF RA1 68.1k C1 1nF LTC3868-1 SENSE1+ SENSE1– VFB1 CITH1A 100pF INTVCC PGOOD1 100k MBOT1 BG1 SW1 RITH1 22k CITH1 820pF ITH1 CSS1 0.01µF RITH2 15k CITH2A 150pF RA2 68.1k CF2 15pF C2 1nF COUT1 150µF VOUT1 2.5V 5A MTOP1 TG1 D1 VIN INTVCC PLLIN/MODE SGND TG2 EXTVCC RUN1 BOOST2 RUN2 FREQ SS2 SW2 ITH2 BG2 CIN 22µF CINT 4.7µF PGND CSS2 0.01µF RSENSE1 7mΩ CB1 0.47µF BOOST1 SS1 CITH2 820pF L1 2.4µH VIN 4V TO 24V D2 MTOP2 CB2 0.47µF L2 3.2µH RSENSE2 7mΩ MBOT2 VOUT2 3.3V COUT2 5A 150µF VFB2 SENSE2– SENSE2+ RB2 215k COUT1, COUT2: SANYO 10TPD150M L1: SUMIDA CDEP105-2R5 L2: SUMIDA CDEP105-3R2M MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP 38681 F13 38681fb 31 LTC3868-1 Typical ApplicationS High Efficiency Dual 12V/5V Step-Down Converter RB1 422k CF1 33pF RA1 34k C1 1nF SENSE1+ SENSE1– INTVCC PGOOD1 100k MBOT1 BG1 VFB1 CITH1A 100pF SW1 RITH1 33k CITH1 680pF CSS1 0.01µF D1 LTC3868-1 VIN INTVCC RFREQ 60k CSS2 0.01µF RITH2 17k CITH2A 100pF RA2 75k CF2 15pF C2 1nF PGND PLLIN/MODE SGND TG2 EXTVCC RUN1 BOOST2 RUN2 FREQ SS2 SW2 ITH2 BG2 VFB2 SENSE2– SENSE2+ COUT1 47µF VOUT1 12V 3A MTOP1 TG1 ITH1 RSENSE1 10mΩ CB1 0.47µF BOOST1 SS1 CITH2 680pF L1 8.8µH CIN 22µF CINT 4.7µF VIN 12.5V TO 24V D2 MTOP2 CB2 0.47µF L2 4.3µH RSENSE2 7mΩ MBOT2 VOUT2 5V COUT2 5.5A 150µF COUT1: KEMET T525D476M016E035 COUT2: SANYO 10TPD150M L1: SUMIDA CDEP105-8R8M L2: SUMIDA CDEP105-4R3M MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP RB2 393k 38681 TA02a 38681fb 32 LTC3868-1 Typical ApplicationS High Efficiency Dual 1V/1.2V Step-Down Converter RB1 28.7k CF1 56pF RA1 115k C1 1nF SENSE1+ SENSE1– INTVCC PGOOD1 L1 MBOT1 0.47µH BG1 VFB1 CITH1A 220pF SW1 RITH1 3.93k CITH1 1000pF RFREQ 60k CSS2 0.01µF RITH2 3.43k CITH2A 220pF RA2 115k VIN CF2 56pF RB2 57.6k C2 1nF INTVCC PGND PLLIN/MODE SGND TG2 EXTVCC RUN1 BOOST2 RUN2 FREQ SS2 SW2 ITH2 BG2 VFB2 SENSE2– SENSE2+ COUT1 220µF ×2 VOUT1 1V 8A D1 LTC3868-1 CSS1 0.01µF MTOP1 TG1 ITH1 RSENSE1 4mΩ CB1 0.47µF BOOST1 SS1 CITH2 1000pF 100k CIN 22µF CINT 4.7µF VIN 12V D2 MTOP2 CB2 0.47µF L2 0.47µH MBOT2 RSENSE2 4mΩ VOUT2 1.2V COUT2 8A 220µF ×2 COUT1, COUT2: SANYO 2RSTPE220M L1: SUMIDA CDEP105-3R2M L2: SUMIDA CDEP105-7R2M MTOP1, MTOP2: RENESAS RJK0305 MBOT1, MBOT2: RENESAS RJK0328 38681 TA03a 38681fb 33 LTC3868-1 Typical ApplicationS High Efficiency Dual 1V/1.2V Step-Down Converter with Inductor DCR Current Sensing RB1 28.7k CF1 56pF RA1 115k RS1 1.18k C1 0.1µF SENSE1+ SENSE1– INTVCC PGOOD1 CITH1A 200pF L1 0.47µH MBOT1 BG1 VFB1 SW1 RITH1 3.93k CITH1 1000pF RFREQ 65k CSS2 0.01µF RITH2 3.93k CITH2A 220pF RA2 115k CF2 56pF RB2 57.6k C2 0.1µF VOUT1 1V 8A D1 LTC3868-1 CSS1 0.01µF MTOP1 TG1 ITH1 COUT1 220µF ×2 CB1 0.47µF BOOST1 VIN SS1 CITH2 1000pF 100k INTVCC PGND PLLIN/MODE SGND TG2 EXTVCC RUN1 BOOST2 RUN2 FREQ SS2 SW2 ITH2 BG2 CIN 22µF CINT 4.7µF VIN 12V D2 MTOP2 CB2 0.47µF L2 0.47µH VOUT2 1.2V COUT2 8A 220µF ×2 MBOT2 VFB2 SENSE2– SENSE2+ COUT1, COUT2: SANYO 2R5TPE220M L1, L2: SUMIDA IHL P2525CZERR47M06 MTOP1, MTOP2: RENESAS RJK0305 MBOT1, MBOT2: RENESAS RJK0328 RS2 1.18k 38681 TA05 38681fb 34 LTC3868-1 Package Description UFD Package 28-Lead Plastic QFN (4mm × 5mm) (Reference LTC DWG # 05-08-1712 Rev B) 0.70 p0.05 4.50 p 0.05 3.10 p 0.05 2.50 REF 2.65 p 0.05 3.65 p 0.05 PACKAGE OUTLINE 0.25 p0.05 0.50 BSC 3.50 REF 4.10 p 0.05 5.50 p 0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 p 0.10 (2 SIDES) 0.75 p 0.05 R = 0.05 TYP PIN 1 NOTCH R = 0.20 OR 0.35 s 45o CHAMFER 2.50 REF R = 0.115 TYP 27 28 0.40 p 0.10 PIN 1 TOP MARK (NOTE 6) 1 2 5.00 p 0.10 (2 SIDES) 3.50 REF 3.65 p 0.10 2.65 p 0.10 (UFD28) QFN 0506 REV B 0.200 REF 0.00 – 0.05 0.25 p 0.05 0.50 BSC BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WXXX-X). 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 38681fb 35 LTC3868-1 Package Description GN Package 28-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .386 – .393* (9.804 – 9.982) .045 p.005 28 27 26 25 24 23 22 21 20 19 18 17 1615 .254 MIN .033 (0.838) REF .150 – .165 .229 – .244 (5.817 – 6.198) .0165 p.0015 .150 – .157** (3.810 – 3.988) .0250 BSC 1 RECOMMENDED SOLDER PAD LAYOUT .015 p .004 s 45o (0.38 p 0.10) .0075 – .0098 (0.19 – 0.25) 2 3 4 5 6 7 8 .0532 – .0688 (1.35 – 1.75) 9 10 11 12 13 14 .004 – .0098 (0.102 – 0.249) 0o – 8o TYP .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) .008 – .012 (0.203 – 0.305) TYP .0250 (0.635) BSC GN28 (SSOP) 0204 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 38681fb 36 LTC3868-1 Revision History (Revision history begins at Rev B) REV DATE DESCRIPTION PAGE NUMBER B 12/09 Change to Absolute Maximum Ratings Change to Electrical Characteristics Change to Typical Performance Characteristics Change to Pin Functions Text Changes to Operation Section Text Changes to Applications Information Section 2 3, 4 6 8, 9 11, 12, 13 21, 22, 23, 24, 26 Change to Figure 10 28 Changes to Related Parts 38 38681fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 37 LTC3868-1 Related Parts PART NUMBER DESCRIPTION COMMENTS LTC3857/LTC3857-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle Phase-Lockable Fixed Operating Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 24V, IQ = 50µA, LTC3858/LTC3858-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle Phase-Lockable Fixed Operating Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 24V, 0.8V ≤ VOUT ≤ 14V, IQ = 170µA, LTC3834/LTC3834-1 Low IQ, Synchronous Step-Down DC/DC Controllers Phase-Lockable Fixed Operating Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30µA, LTC3835/LTC3835-1 Low IQ, Synchronous Step-Down DC/DC Controllers Phase-Lockable Fixed Operating Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 80µA, LT3845 Low IQ, High Voltage Synchronous Step-Down DC/DC Controller Adjustable Fixed Operating Frequency 100kHz to 500kHz, 4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 120µA, TSSOP-16 LT3800 Low IQ, High Voltage Synchronous Step-Down DC/DC Controller Fixed 200kHz Operating Frequency, 4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 100µA, TSSOP-16 LTC3824 Low IQ, High Voltage DC/DC Controller, 100% Duty Cycle Selectable Fixed 200kHz to 600kHz Operating Frequency, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-10E LTC3850/LTC3850-1 LTC3850-2 Dual 2-Phase, High Efficiency Synchronous Step-Down DC/DC Controllers, RSENSE or DCR Current Sensing and Tracking Phase-Lockable Fixed Operating Frequency 250kHz to 780kHz, 4V ≤ VIN ≤ 30V, 0.8V ≤ VOUT ≤ 5.25V LTC3855 Dual, Multiphase, Synchronous DC/DC Step-Down Controller with Diffamp and DCR Temperature Compensation Phase-Lockable Fixed Frequency 250kHz to 770kHz, 4.5V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 12.5V LTC3853 Triple Output, Multiphase Synchronous Step-Down DC/DC Phase-Lockable Fixed Operating Frequency 250kHz to 750kHz, Controller, RSENSE or DCR Current Sensing and Tracking 4V ≤ VIN ≤ 24V, VOUT Up to 13.5V Small Footprint Wide VIN Range Synchronous Step-Down Fixed 400kHz Operating Frequency, 4.5V ≤ VIN ≤ 38V, DC/DC Controller 0.8V ≤ VOUT ≤ 5.25V, 2mm × 3mm QFN-12, MSOP-12 LTC3854 LTC3775 High Frequency Synchronous Voltage Mode Step-Down DC/DC Controller Fast Transient Response, tON(MIN) = 30ns, 4V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 0.8VIN, MSOP-16E, 3mm × 3mm QFN-16 LTC3851A/ LTC3851A-1 No RSENSE ™ Wide VIN Range Synchronous Step-Down DC/DC Controllers Phase-Lockable Fixed Operating Frequency 250kHz to 750kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.25V, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16 LTC3878/LTC3879 No RSENSE Constant On-Time Synchronous Step-Down DC/DC Controllers Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, VOUT Up 90% of VIN, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16 38681fb 38 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LT 0110 REV B • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2009