SIPEX SP6127EK1L/TR

SP6127
High-Voltage, Step-Down Controller in TSOT6
FEATURES
Wide 4.5V – 29V Input Voltage Range
Internal compensation
Built-in High-Current PMOS Driver
Adjustable Overcurrent Protection
Internal soft start
900kHz Constant Frequency Operation
0.6V Reference Voltage
1% output setpoint accuracy
Lead Free, RoHS Compliant Package:
Small 6-pin TSOT
LX
GND
FB
6
5
4
SP6127
6 PinTSOT
1
VIN
2
GATE
3
VDR
__________________________________________________________ DESCRIPTION
The SP6127 is a PWM controlled step down (buck) voltage mode regulator with VIN feedforward and
internal Type-II compensation. It operates from 4.5V to 29V VIN, making it suitable for 5V, 12V and 24V
applications. By using a PMOS driver, this device is capable of operating at 100% duty cycle. The highside driver is designed to drive the gate 5V below VIN. The programmable overcurrent protection is
based on the high-side MOSFET’s ON resistance sensing and allows setting the overcurrent protection
value up to 300mV threshold (measured between VIN-LX). The SP6127 is available in a space-saving 6pin TSOT package making it the smallest controller available capable of operating from 24VDC supplies.
_______________________________________ TYPICAL APPLICATION CIRCUIT
VIN
C1
4.7uF
Q1
Si2343DS
12V
2
1
Gate
GND
Vin
L1, IHLP-2525CZ
3.3uH, 30mOhm, 6A
Rs=1k
VOUT
LX
C7
0.1uF
6
SP6127
Ds
MBRA340T3G
3
C4
22uF
RZ
2K
VDR
1.8V
0-2.0A
R1
200k, 1%
CZ
33pF
4
GND
VFB
GND
R2
100k, 1%
5
D1 1N4148
SHDN
High=Of f
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 1
 2007 Sipex Corporation
ABSOLUTE MAXIMUM RATINGS
Input Voltage….................................................-0.3V to 30V
Lx………………………………………………….…-2V to 30V
FB……………..................................................-0.3V to 5.5V
Storage Temperature..………………...……-65 °C to 150 °C
Junction Temperature...................................-40°C to 125°C
Lead Temperature (Soldering, 10 sec)…..…………..300 °C
ESD Rating………..….…1kV LX, 2kV all other nodes, HBM
These are stress ratings only, and functional operation of the
device at these ratings or any other above those indicated in
the operation sections of the specifications below is not
implied. Exposure to absolute maximum rating conditions for
extended periods of time may affect reliability.
____________________________________________ ELECTRICAL SPECIFICATIONS
Specifications are for TAMB=TJ=25°C, and those denoted by ♦ apply over the full operating range, -40°C< Tj <125°C. Unless
otherwise specified: VIN =4.5V to 29V, CIN = 4.7µF.
PARAMETER
TYP
MAX
UVLO Turn-On Threshold
4.2
4.35
4.5
V
0°C< Tj <125°C
UVLO Turn-Off Threshold
4.0
4.15
4.4
V
0°C< Tj <125°C
UVLO Hysteresis
0.2
UNITS
♦
MIN
V
Operating Input Voltage Range
4.5
29
V
Operating Input Voltage Range
7
29
V
Operating VCC Current
Reference Voltage Accuracy
0.3
0.5
3
1
mA
%
Reference Voltage Accuracy
0.5
2
%
Reference Voltage
0.594
0.6
0.606
V
Reference Voltage
0.588
0.6
0.612
V
750
900
VIN/5
1050
kHz
V
40
100
ns
0
%
%
Switching Frequency
Peak-to-peak ramp Voltage
Minimum ON-Pulse Duration
Minimum Duty Cycle
Maximum Duty Cycle
100
Gate Driver Turn-Off Resistance
50
60
kΩ
Gate Driver Pull-Down
Resistance
4
8
Ω
Gate Driver Pull-up Resistance
3
6
Ω
5.5
V
VIN - VDR voltage difference
4.5
Overcurrent Threshold
LX pin Input Current
OFF interval during hiccup
270
25
300
30
70
330
35
mV
uA
ms
3
5
9
ms
1.0
1.2
Soft start time
SHDN Threshold
SHDN Threshold Hysteresis
June 26, 2007
0.8
100
CONDITIONS
V
0°C< Tj <125°C
♦
VFB=1.2V
♦
♦
♦
Internal resistor between GATE and
VIN
VIN=12V, VFB=0.5V, Measure
resistance between GATE and VDR
VIN=12V, VFB=0.7V, Measure
resistance between GATE and VIN
♦
Measure VIN – VDR, VIN>7V
Measure VIN - LX
VLX = VIN
VFB=0.58V, measure between
VIN=4.5V and first GATE pulse
♦
Apply voltage to FB
mV
SP6127 TSOT-6 PFET Buck Controller
Page 2
 2007 Sipex Corporation
_______________________________________________________ PIN DESCRIPTION
PIN #
PIN
NAME
1
VIN
2
GATE
3
VDR
4
FB
5
GND
6
LX
DESCRIPTION
Input power supply for the controller. Place input decoupling capacitor as close
as possible to this pin.
Connect to the gate terminal of the external P-channel MOSFET.
Power supply for the internal driver. This voltage is internally regulated to
about 5V below VIN. Place a 0.1µF decoupling capacitor between VDR and
VIN as close as possible to the IC.
Regulator feedback input. Connect to a resistive voltage-divider network to set
the output voltage. This pin can be also used for ON/OFF control. If this pin is
pulled above 1V the P-channel driver is disabled and controller resets internal
soft start circuit.
Ground pin.
This pin is used as a current limit input for the internal current limit comparator.
Connect to the drain pin of the external MOSFET through an optional resistor.
Internal threshold is pre-set to 300mV nominal and can be decreased by
changing the external resistor based on the following formula: VTRSHLD =
300mV – 30uA * R
______________________________________________________ BLOCK DIAGRAM
VIN
5V
VDR
Oscillator
Vin - 5V LDO
VIN
5V Internal LDO
I = k x VIN
FAULT
PWM Latch
Reset Dominant
VREF
GATE
S
+
FB
R
+
-
PWM Comparator
Error Amplifier
VDR
FAULT
FAULT
ENBL
LX
-
70ms delay
+
-
1V
June 26, 2007
FAULT
Register
S
R
R
+
UVLO
4-Bit counter
Overcurrent
Comparator
30uA
VIN - 0.3V
GND
POR
Set Dominant
SP6127 TSOT-6 PFET Buck Controller
Page 3
 2007 Sipex Corporation
____________________________________________________ GENERAL OVERVIEW
The SP6127 is a fixed frequency, Voltage-mode,
non-synchronous PWM controller optimized for
minimum component, small form factor and cost
effectiveness. It has been designed for singlesupply operation ranging from 4.5V to 29V.
SP6127 has Type-II internal compensation for
use with Electrolytic/Tantalum output capacitors.
For ceramic capacitors Type-III compensation
can be implemented by simply adding an R and
C between output and Feedback. A precision
0.6V reference, present on the positive terminal
of the Error Amplifier, permits programming of the
output voltage down to 0.6V via the FB pin. The
output of the Error Amplifier is internally
compared to a feed-forward (VIN/5 peak-to-peak)
ramp and generates the PWM control. Timing is
governed by an internal oscillator that sets the
PWM frequency at 900kHz.
SP6127 contains useful protection features.
Overcurrent protection is based on the high-side
MOSFET’s RDS(ON) and is programmable via a
resistor placed at LX node. Under-Voltage LockOut (UVLO) ensures that the controller starts
functioning only when sufficient voltage exists for
powering IC’s internal circuitry.
SP6127 Loop Compensation
where:
f ESRZERO =
f DBPOLE =
1
2.π .R ESR .C OUT
1
2.π . L ⋅ C OUT
……….. (2)
………… (3)
Creating a Type-III compensation Network
The above condition requires the ESR zero to be
at a lower frequency than the double-pole from
the LC filter. If this condition is not met, Type-III
compensation should be used and can be
accomplished by placing a series RC
combination in parallel with R1 as shown below.
The value of CZ can be calculated as follows and
RZ selected from table 1.
CZ =
The SP6127 includes Type-II internal compensation
components for loop compensation.
External
compensation components are not required for
systems with tantalum or aluminum electrolytic
output capacitors with sufficiently high ESR. Use
the condition below as a guideline to determine
whether or not the internal compensation is
sufficient for your design.
Type-II internal compensation is sufficient if the
following condition is met:
L ⋅C
………….. (4)
1.25 × R1
fESRZERO/
fDBPOLE
1X
2X
3X
5X
>= 10X
RZ
50KΩ
40KΩ
30KΩ
10KΩ
2KΩ
f ESRZERO < f DBPOLE ………………. (1)
Table1- Selection of RZ
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 4
 2007 Sipex Corporation
______________________________________________________ GENERAL OVERVIEW
Vout
SP6127
CP1
2pF
RZ
CZ2
130pF
RZ2
200k
CZ
R1
200k, 1%
VFB
+
Vref =0.6V
R2
Error Amplif ier
Figure 1- RZ and CZ in conjunction with internal compensation components form a TypeIII compensation network
Loop Compensation Example 1- A converter
utilizing a SP6127 has a 3.3µH inductor and a
22µF/5mΩ ceramic capacitor. Determine
whether Type-III compensation is needed.
From equation (2)
fESRZERO
= 1.45MHz. From
equation (3) fDBPOLE = 18.4 kHz. Since the
condition specified in (1) is not met, Type-III
compensation must be used by adding
external components RZ and CZ. Using
equation (4) CZ is calculated to be 34pF (use
33pF). Following the guideline given in table 1,
a 2kΩ RZ should be used.
The steps followed in example 1 were used to
compensate the typical application circuit
shown on page 1. Satisfactory frequency
response of the circuit, seen in figure 2,
validates the above procedure.
Loop Compensation Example 2- A converter
utilizing the SP6127 has a 3.3µH inductor and
a 220µF, 82mΩ Aluminum Electrolytic
capacitor.
Determine
whether
Type-III
compensation is needed.
From equation (2)
fESRZERO
= 8.8kHz. From
equation (3) fDBPOLE = 5.9kHz. Since the
condition specified in (1) is not met, Type-III
compensation needs to be used by adding
external components RZ and CZ. Using
equation (4) CZ is calculated 108pF (use 100
pF). Since fESRZERO /fDBPOLE is approximately 2,
RZ must be set at 40kΩ.
Figure 2- Satisfactory frequency response of typical application circuit shown on page 1.
Crossover frequency fc is 100kHz with a corresponding phase margin of 60 degrees.
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 5
 2007 Sipex Corporation
______________________________________________________ GENERAL OVERVIEW
for Rs in the range of 0.5kΩ to 3kΩ. For Rs larger
than 3kΩ, test results are lower than those
predicted by (5), due to circuit parasitics.
Therefore the maximum value of Rs should be
limited to 3kΩ.
Overcurrent Protection (OCP)
Vin
SP6127
Gate
Q1
Ov er-Current Comparator
LX
Rs
+
Ds
30uA
Vin - 0.3V
Note that in order to safeguard against false
overcurrent trigger due to transients, there is a
150ns delay between the turn on of the MOSFET
and when OCP circuit is activated. As a
consequence at very high Vo/VIN ratio, where
MOSFET on-time is less than 150ns, the OCP
circuit will not detect overcurrent.
Using the ON/OFF Function
Figure 3 Overcurrent protection circuit
The overcurrent protection (OCP) circuit
functions by monitoring the voltage across the
high-side FET Q1. When this voltage exceeds
0.3V, the overcurrent comparator triggers and
the controller enters hiccup mode. For example
if Q1 has RDS(ON)=0.1Ω, then the overcurrent
will trigger at I = 0.3V/0.1Ω=3A. To program a
lower overcurrent use a resistor Rs as shown in
figure 1. Calculate Rs from:
Rs =
0.3 − (1.15 × Iout × Rds (on) × Kt )
...(5)
30uA
The Feedback pin serves a dual role of ON/OFF
control. The MOSFET driver is disabled when a
voltage greater than 1V is applied at the FB pin.
Maximum voltage rating of this pin is 5.5V. The
controlling signal should be applied through a
small signal diode as shown on page 1.
Please note that an optional 10kΩ bleeding
resistor across the output helps keep the output
capacitor discharged under no load condition.
Programming the Output Voltage
To program the output voltage, calculate R2
using the following equation:
Where:
R2 =
1.15 is used to calculate peak inductor current
which is nominally 15% higher than average
output current
RDS(ON) is MOSFET ON-resistance rating
Kt is a multiplier that accounts for increase in
RDS(ON) due to temperature
Example: A switching MOSFET used with
SP6127 has RDS(ON) of 0.08Ω and Kt is 1.5.
Program the over-current circuit so that
maximum output is 2A.
Rs =
0.3 − (1.15 × 2 A × 0.08Ohm × 1.5)
30uA
Rs=800Ω
Using the above equation there is good
agreement between calculated and test results
June 26, 2007
R1
 Vout 

− 1
 Vref

Where:
VREF=0.6 is the reference voltage of the SP6127
R1=200kΩ is a fixed-value resistor that, in
addition to being a voltage divider, it is part of the
compensation network. In order to simplify
compensation calculations, R1 is fixed at 200kΩ.
Soft Start
Soft Start is preset internally to 5ms (nominal).
Internal Soft Start eliminates the need for the
external capacitor CSS that is commonly used to
program this function.
SP6127 TSOT-6 PFET Buck Controller
Page 6
 2007 Sipex Corporation
______________________________________________________ GENERAL OVERVIEW
MOSFET Gate Drive
The P-channel drive is derived through an
internal regulator that generates VIN-5V. This pin
(VDR) must be connected to VIN with a 0.1µF
decoupling capacitor. The gate drive circuit
swings between VIN and VIN-5 and employs
powerful drivers for efficient switching of the Pchannel MOSFET.
voltage selection. For a low duty cycle application
such as the circuit shown on first page, the
Schottky diode is conducting most of the time and
its conduction losses are the largest component
of losses in the converter. Conduction losses can
be estimated from:
 Vout 
Pc = Vf × Iout × 1 −

Vin 

where:
VF is diode forward voltage at IOUT
Power MOSFET Selection
Select the Power MOSFET for Voltage rating
BVDSS, On resistance RDS(ON), and thermal
resistance RTHJA. BVDSS should be about twice
as high as VIN in order to guard against switching
transients. The recommended MOSFET voltage
rating for VIN of 5V, 12V and 24V is 12V, 30V and
40V respectively. RDS(ON) must be selected such
that when operating at peak current and junction
temperature, the Overcurrent threshold of the
SP6127 is not exceeded. Allowing 50% for
temperature coefficient of RDS(ON) and 15% for
inductor current ripple, the following expression
can be used:
0.3V


RDS (ON ) ≤ 

 1.5 × 1.15 × Iout 
Within this constraint, selecting MOSFETs with
lower RDS(ON) will reduce conduction losses at
the expense of increased switching losses. As a
rule of thumb, select the highest RDS(ON)
MOSFET that meets the above criteria. Switching
losses can be assumed to roughly equal to the
conduction losses. A simplified expression for
conduction losses is given by:
 Vout 
Pcond = Iout 2 × RDS (ON ) × 

 Vin 
The MOSFET’s junction temperature can be
estimated from:
T = (2 × Pc × Rthja ) + Tambient
Schottky Rectifier selection
The Schottky diode’s AC losses due to its
switching capacitance are negligible.
Inductor Selection
Select the Inductor for inductance L and
saturation current ISAT. Select an inductor with
ISAT higher than the programmed overcurrent.
Calculate inductance from:
 Vout   1   1 

L = (Vin − Vout ) × 
 ×   × 
 Vin   f   Irip 
where:
VIN is converter input voltage
VOUT is converter output voltage
f is switching frequency
IRIP is inductor peak-to-peak current ripple
(nominally set to 30% of IOUT)
Keep in mind that a higher IRIP results in a
smaller inductor which has the advantages of
small size, low DC equivalent resistance DCR,
high saturation current ISAT and allows the use of
a lower output capacitance to meet a given step
load transient. A higher IRIP, however, increases
the output voltage ripple and increases the
current at which converter enters Discontinuous
Conduction Mode. The output current at which
converter enters DCM is ½ of IRIP. Note that a
negative current step load that drives the
converter into DCM will result in a large output
voltage transient. Therefore the lowest current for
a step load should be larger than ½ of IRIP.
Select the Schottky Diode for Voltage rating VR,
Forward voltage Vf, and thermal resistance
RTHJA. The Voltage rating should be selected
using the same guidelines outlined for MOSFET
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 7
 2007 Sipex Corporation
______________________________________________________ GENERAL OVERVIEW
Output Capacitor Selection
Note that a smaller inductor results in a higher
Select the output capacitor for voltage rating,
capacitance and Equivalent Series Resistance
(ESR). Nominally the voltage rating is selected to
be twice as large as the output voltage. Select
the capacitance to satisfy the specification for
output voltage overshoot/undershoot caused by
current step load. A steady-state output current
IOUT corresponds to inductor stored energy of ½ L
IOUT2. A sudden decrease in IOUT forces the
energy surplus in L to be absorbed by COUT. This
causes an overshoot in output voltage that is
corrected by the reduced duty cycle of the power
switch. Use the following equation to calculate
COUT:
IRIP, therefore requiring a larger COUT and/or
lower ESR in order to meet VRIP.
 I 2 − I1 
Cout = L × 
2
2 
 Vos - Vout 
In general, total input voltage ripple should be
kept below 1.5% of VIN (not to exceed 180mV).
Input voltage ripple has three components: ESR
and ESL cause a step voltage drop upon turn on
of the MOSFET. During on time, the capacitor
discharges linearly as it supplies IOUT-IIN. The
contribution to Input voltage ripple by each term
can be calculated from:
Where:
L is the output inductance
I2 is the step load high current
I1 is the step load low current
Vos is output voltage including overshoot
VOUT is steady state output voltage
Select ESR such that output voltage ripple (VRIP)
specification is met. There are two components to
VRIP: The first component arises from charge
transferred to and from COUT during each cycle.
The second component of VRIP is due to inductor
ripple current flowing through the output
capacitor’s ESR. It can be calculated from:


1

Vrip = Irip × ESR + 
 8 × Cout × fs 
Where:
IRIP is inductor ripple current
fs is switching frequency
COUT is output capacitor calculated above
June 26, 2007
Select the input capacitor for Voltage,
Capacitance, ripple current, ESR and ESL.
Voltage rating is nominally selected to be twice
the input voltage. The RMS value of the input
capacitor current, assuming a low inductor ripple
current (IRIP), can be calculated from:
Icin = Iout × D(1 − D )
∆V , Cin =
Output voltage undershoot calculation is more
complicated. Test results for SP6127 buck
circuits show that undershoot is approximately
equal to overshoot. Therefore the above equation
provides a satisfactory method for calculating
COUT.
2
Input Capacitor Selection
2
Iout × Vout × (Vin − Vout )
fs × Cin × Vin 2
∆V , ESR = ESR(Iout − 0.5Irip )
∆V , ESL = ESL
(Iout − 0.5Irip )
Trise
Where TRISE is the rise time of current through
capacitor
Total input voltage ripple is sum of the above:
∆V , Tot = ∆V , Cin + ∆V , ESR + ∆V , ESL
In circuits where converter input voltage is
applied via a mechanical switch, excessive
ringing may be present at turn-on that may
interfere with smooth startup of the SP6127. The
addition of an inexpensive 100µF Aluminum
Electrolytic capacitor at the input will help reduce
ringing and restore a smooth startup.
SP6127 TSOT-6 PFET Buck Controller
Page 8
 2007 Sipex Corporation
_____________________________________ TYPICAL PERFORMANCE CHARACTERISTICS
VIN
C1
4.7uF
Q1
Si2343DS
12V
2
1
Gate
GND
Vin
L1, IHLP-2525CZ
3.3uH, 30mOhm, 6A
Rs=1k
VOUT
LX
C7
0.1uF
6
SP6127
Ds
MBRA340T3G
3
C4
22uF
RZ
2K
VDR
1.8V
0-2.0A
R1
200k, 1%
CZ
33pF
4
GND
VFB
GND
R2
100k, 1%
5
D1 1N4148
SHDN
High=Of f
Figure 4- Application circuit
SP6127 Efficiency versus Iout,
Vin=12V,Ta=25C
SP6127 Load Regulation
Vin=12V
90
1.810
80
1.805
Vout (V)
Efficiency (%)
Vout=1.8V
70
60
1.800
1.795
50
1.790
0.0
0.5
1.0
1.5
2.0
0.0
0.5
Iout (A)
1.5
2.0
Iout (A)
Figure 5- Efficiency, natural convection
June 26, 2007
1.0
Figure 6- Load regulation
SP6127 TSOT-6 PFET Buck Controller
Page 9
 2007 Sipex Corporation
_____________________________________ TYPICAL PERFORMANCE CHARACTERISTICS
Figure 7- Step load 1-2A
ch1: VIN ch2: VOUT, ch3: IOUT
Figure 8- Overcurrent shutdown
ch1: VIN, ch2: VOUT, ch3: Inductor current, ch4: IOUT
Figure 9- Startup no load
ch1: VIN, ch2: VOUT, ch3: IOUT
Figure 10- Start up 2A
ch1: VIN ch2: VOUT, ch3: IOUT
Figure 11- Output ripple at 0A is 32mV
ch1: VIN, ch2: VOUT, ch3: IOUT
June 26, 2007
Figure 12- Output ripple at 2A is 12mV
ch1: VIN, ch2: VOUT, ch3: IOUT
SP6127 TSOT-6 PFET Buck Controller
Page 10
 2007 Sipex Corporation
______________________________________________________ PACKAGE: 6PIN TSOT
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 11
 2007 Sipex Corporation
Ordering Information:
Part Number
Temperature Range
Package
SP6127EK1-L………………………...……….-40°C to +125°C……………….………………..6 Pin TSOT
SP6127EK1-L/TR………………………….....-40°C to +125°C…….…………….………….....6 Pin TSOT
For further assistance:
Email:
WWW Support page:
Sipex Application Notes:
[email protected]
http://www.sipex.com/content.aspx?p=support
http://www.sipex.com/applicationNotes.aspx
Sipex Corporation
Solved by
TM
Headquarters and
Sales Office
233 South Hillview Drive
Milpitas, CA95035
tel: (408) 934-7500
FAX: (408) 935-7600
Sipex Corporation reserves the right to make changes to any products described herein. Sipex does not assume any liability arising out of the application or use of any
product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others.
June 26, 2007
SP6127 TSOT-6 PFET Buck Controller
Page 12
 2007 Sipex Corporation