ADS5421 ADS 542 1 SBAS237D – DECEMBER 2001 – REVISED JULY 2004 14-Bit, 40MHz Sampling ANALOG-TO-DIGITAL CONVERTER FEATURES DESCRIPTION ● HIGH DYNAMIC RANGE: High SFDR: 83dB at 10MHz fIN High SNR: 75dB at 10MHz fIN ● PREMIUM TRACK-AND-HOLD: Differential Inputs Selectable Full-Scale Input Range ● LOW POWER: 850mW ● FLEXIBLE CLOCKING: Differential or Single-Ended Accepts Sine or Square Wave Clocking Down to 0.5Vp-p Variable Threshold Level The ADS5421 is a high-dynamic range 14-bit, 40MHz, pipelined Analog-to-Digital Converter (ADC). It includes a high-bandwidth linear track-and-hold amplifier that gives excellent spurious performance up to and beyond the Nyquist rate. The clock input can accept a low-level differential sine wave or square wave signal down to 0.5Vp-p, further improving the Signal-to-Noise Ratio (SNR) performance. The ADS5421 has a 4Vp-p differential input range (2Vp-p • 2 inputs) for optimum Spurious-Free Dynamic Range (SFDR). The differential operation gives the lowest even-order harmonic components. A lower input voltage can also be selected using the internal references, further optimizing SFDR. The ADS5421 is available in a small LQFP-64 package. APPLICATIONS ● COMMUNICATIONS RECEIVERS ● TEST INSTRUMENTATION ● PROFESSIONAL CCD IMAGING +VS DV CLK ADS5421 Timing Circuitry CLK 1Vp-p 14-Bit Pipelined ADC Core IN T&H 1Vp-p IN Error Correction Logic 3-State Outputs D0 • • • D13 CM (+2.5V) Reference Ladder and Driver Reference and Mode Select REFT VREF SEL1 SEL2 REFB OE VDRV Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. Copyright © 2001-2004 Texas Instruments, Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com ELECTROSTATIC DISCHARGE SENSITIVITY ABSOLUTE MAXIMUM RATINGS(1) +VSA, +VSD, VDRV ............................................................................... +6V Analog Input .......................................................... (–0.3V) to (+VS + 0.3V) Logic Input ............................................................ (–0.3V) to (+VS + 0.3V) Case Temperature ......................................................................... +100°C Junction Temperature .................................................................... +150°C Storage Temperature ..................................................................... +150°C This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. NOTE: (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. EVALUATION BOARD PRODUCT ADS5421EVM DESCRIPTION USER’S GUIDE Populated Evaluation Board SBAU084 PACKAGE/ORDERING INFORMATION(1) SPECIFIED TEMPERATURE RANGE PACKAGE MARKING PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR ADS5421Y LQFP-64 PM –40°C to +85°C ADS5421Y ADS5421Y/T Tape and Reel, 250 " " " " ADS5421Y/R Tape and Reel, 1500 " ORDERING NUMBER TRANSPORT MEDIA, QUANTITY NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet. ELECTRICAL CHARACTERISTICS TA = specified temperature range, typical at +25°C, +VSA = +VSD = +5V, differential input range = 1.5V to 3.5V each input (4Vp-p), sampling rate = 40MHz, internal reference, VDRV = +3V, and –1dBFS, unless otherwise noted. ADS5421Y PARAMETER CONDITIONS MIN RESOLUTION SPECIFIED TEMPERATURE RANGE ANALOG INPUT Standard Differential Input Range Common-Mode Voltage Optional Input Ranges Analog Input Bias Current Analog Input Bandwidth Input Capacitance Ambient Air Full-Scale = 4Vp-p DYNAMIC CHARACTERISTICS Differential Linearity Error (largest code error) f = 1MHz f = 10MHz No Missing Codes Integral Nonlinearity Error, f = 1MHz Spurious-Free Dynamic Range(1) f = 1MHz f = 10MHz f = 30MHz 2-Tone Intermodulation Distortion(3) f = 14.5MHz and 15.5MHz (–7dB each tone) Signal-to-Noise Ratio (SNR) f = 1MHz f = 10MHz f = 30MHz Signal-to-(Noise + Distortion) (SINAD) f = 1MHz f = 10MHz f = 30MHz Effective Number of Bits(4) Output Noise Aperture Delay Time Aperture Jitter Over-Voltage Recovery Time Full-Scale Step Acquisition Time 2 MAX Bits –40 to +85 °C 3.5 V V V µA MHz pF 40M Samples/sec Clk Cyc 2.5 2Vp-p or 3Vp-p 1 500 9 1M 10 ±0.5 ±0.5 Tested ±2.5 78 72 72 f = 1MHz IN and IN tied to CM UNITS 14 Tested 1.5 Selectable CONVERSION CHARACTERISTICS Sample Rate Data Latency TYP ±1.0 LSB LSB LSB 88 85 82 dBFS(2) dBFS dBFS –90 dBc 76 75 75 dBFS dBFS dBFS 75 74 74 12.2 0.4 3 1 5 5 dB dB dBFS Bits LSB rms ns ps rms ns ns ADS5421 www.ti.com SBAS237D ELECTRICAL CHARACTERISTICS (Cont.) TA = specified temperature range, typical at +25°C, +VSA = +VSD = +5V, differential input range = 1.5V to 3.5V each input (4Vp-p), sampling rate = 40MHz, internal reference, VDRV = +3V, and –1dBFS, unless otherwise noted. ADS5421Y PARAMETER DIGITAL INPUTS Clock Input Logic Family (other than clock inputs) High Level Input Current(5) (VIN = 5V) Low Level Input Current (VIN = 0V) High Level Input Voltage Low Level Input Voltage Input Capacitance DIGITAL OUTPUTS(6) Logic Family Logic Coding Low Output Voltage (IOL = 50µA to 0.5mA) High Output Voltage (IOH = 50µA to 0.5mA) Low Output Voltage (IOL = 50µA to 1.6mA) High Output Voltage (IOH = 50µA to 1.6mA) 3-State Enable Time 3-State Disable Time Output Capacitance ACCURACY Zero Error (Referred to –FS) Zero Error Drift (Referred to –FS) Gain Error(7) Gain Error Drift(7) Power-Supply Rejection of Gain Internal REF Tolerance (VREFT, VREFB) External REF Voltage Range Reference Input Resistance POWER-SUPPLY REQUIREMENTS Supply Voltage: +VSA, +VSD Supply Current: +IS Output Driver Supply Current (VDRV) Power Dissipation: VDRV = 5V VDRV = 3V Power Down Thermal Resistance, θJA LQFP-64 CONDITIONS MIN Rising Edge of Convert Clock +0.5 TYP MAX UNITS +VSD Vp-p 100 10 µA µA V V pF +3V/+5V Compatible CMOS +2.0 +1.0 5 +3V/+5V Compatible CMOS Straight Offset Binary VDRV = 3V +0.2 +2.5 VDRV = 5V +0.2 +2.5 OE = LOW OE = HIGH 20 2 5 40 10 at +25°C ±0.5 15 ±0.2 35 68 ±10 2 1.0 ±1.0 at +25°C ∆VS = ±5% Deviation from Ideal 0.9 Operating, fIN = 10MHz Operating, fIN = 10MHz Operating +4.75 +5.0 170 12 900 850 40 48 ±1.0 ±50 2.025 +5.25 925 V V V V ns ns pF %FS ppm/°C %FS ppm/°C dB mV V kΩ V mA mA mW mW mW °C/W NOTES: (1) Spurious-Free Dynamic Range refers to the magnitude of the largest harmonic. (2) dBFS means dB relative to Full-Scale. (3) 2-tone intermodulation distortion is referred to the largest fundamental tone. This number will be 6dB higher if it is referred to the magnitude of the 2-tone fundamental envelope. (4) Effective Number of Bits (ENOB) is defined by (SINAD – 1.76)/6.02. (5) A 50kΩ pull-down resistor is inserted internally. (6) Recommended maximum capacitance loading, 15pF. (7) Includes internal reference. ADS5421 SBAS237D www.ti.com 3 PIN CONFIGURATION REFBY GND IN GND IN GND GND GND GND REFT CM REFB GND 63 GND 64 +VSA TQFP +VSA Top View 62 61 60 59 58 57 56 55 54 53 52 51 50 49 +VSA 1 48 GND +VSA 2 47 GND +VSD 3 46 VREF +VSD 4 45 SEL1 +VSD 5 44 SEL2 +VSD 6 43 GND GND 7 42 GND GND 8 CLK 41 BTC ADS5421Y 9 40 PD CLK 10 39 OE GND 11 38 GNDRV 25 26 27 28 29 30 31 32 NC 24 NC 23 B14 (LSB) 22 B13 21 B12 20 B11 19 B10 18 B9 17 B8 33 VDRV B7 34 VDRV DV 16 B6 DNC 15 B5 35 VDRV B4 GNDRV 14 B3 36 GNDRV B2 37 GNDRV B1 (MSB) GND 12 GNDRV 13 PIN DESCRIPTIONS PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 4 I/O I I O O O O O O O O O O O O O O DESIGNATOR +VSA +VSA +VSD +VSD +VSD +VSD GND GND CLK CLK GND GND GNDRV GNDRV DNC DV B1 B2 B3 B4 B5 B6 B7 B8 B9 B10 B11 B12 B13 B14 NC NC DESCRIPTION Analog Supply Voltage Analog Supply Voltage Digital Supply Voltage Digital Supply Voltage Digital Supply Voltage Digital Supply Voltage Ground Ground Clock Input Complementary Clock Input Ground Ground Ground Ground Do Not Connect Data Valid Pulse: HI = Data Valid Data Bit 1 (D13) (MSB) Data Bit 2 (D12) Data Bit 3 (D11) Data Bit 4 (D10) Data Bit 5 (D9) Data Bit 6 (D8) Data Bit 7 (D7) Data Bit 8 (D6) Data Bit 9 (D5) Data Bit 10 (D4) Data Bit 11 (D3) Data Bit 12 (D2) Data Bit 13 (D1) Data Bit 14 (D0) (LSB) No Internal Connection No Internal Connection PIN 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 48 49 50 51 52 53 54 55 56 57 58 59 60 61 62 63 64 I/O I I I I DESIGNATOR VDRV VDRV VDRV GNDRV GNDRV GNDRV OE PD BTC GND GND SEL2 SEL1 VREF GND GND GND REFB CM REFT GND GND GND GND IN GND IN GND REFBY GND +VSA +VSA DESCRIPTION Output Driver Supply Voltage Output Driver Supply Voltage Output Driver Supply Voltage Ground Ground Ground Output Enable: HI = High Impedance Power Down: HI = Power Down; LO = Normal HI = Binary Two’s Complement Ground Ground Reference Select 2: See Table on Page 5 Reference Select 1: See Table on Page 5 Internal Reference Voltage Ground Ground Ground Bottom Reference Voltage Bypass Common-Mode Voltage (Midscale) Top Reference Voltage Bypass Ground Ground Ground Ground Complementary Analog Input Ground Analog Input Ground Reference Bypass Ground Analog Supply Voltage Analog Supply Voltage ADS5421 www.ti.com SBAS237D TIMING DIAGRAM N+9 N+8 N+2 N+1 Analog In N+4 N+3 N tD N+5 tL tCONV N + 10 N+7 N+6 tH Clock 10 Clock Cycles t2 Data Out N – 10 N–9 N–8 N–7 N–6 N–5 N–4 N–3 N–2 N–1 Data Invalid N t1 Data Valid Output tDV SYMBOL t CONV tL tH tD t1 t2 tDV DESCRIPTION MIN Convert Clock Period Clock Pulse LOW Clock Pulse HIGH Aperture Delay Data Hold Time, CL = 0pF New Data Delay Time, CL = 15pF max 25 11.5 11.5 Data Valid Output, CL = 15pF 3.9 TYP MAX UNITS 1µs t CONV /2 t CONV /2 3 7.2 12.7 ns ns ns ns ns ns 4.4 ns REFERENCE AND FULL-SCALE RANGE SELECT TABLE DESIRED FULL-SCALE RANGE SEL1 SEL2 INTERNAL VREF 4Vp-p 3Vp-p 2Vp-p GND GND VREF GND +VSA GND 2V 1.5V 1V NOTE: For external reference operation, tie VREF to +VSA. The full-scale range will be 2x the reference value. For example, selecting a 2V external reference will set the full-scale values of 1.5V to 3.5V for both IN and IN inputs. ADS5421 SBAS237D www.ti.com 5 TYPICAL CHARACTERISTICS TA = 25°C, +VSA = +VSD = +5V, differential input range = 1.5V to 3.5V each input (4Vp-p), sampling rate = 40MSPS, internal reference, and VDRV = 3V, unless otherwise noted. SPECTRAL PERFORMANCE SPECTRAL PERFORMANCE 0 0 FIN = 1MHz, –1dBFS FIN = 10MHz, –1dBFS SFDR = 88.4dBFS –20 SFDR = 85dBFS –20 SNR = 74.5dBFS –40 Amplitude (dB) Amplitude (dB) SNR = 76.2dBFS –60 –80 –100 –40 –60 –80 –100 –120 –120 0 4 8 12 16 20 0 4 8 Frequency (MHz) SPECTRAL PERFORMANCE FIN = 15MHz, –1dBFS FIN = 15MHz, –3dBFS SFDR = 84.9dBFS SFDR = 87.9dBFS –20 SNR = 72.7dBFS 16 20 16 20 SNR = 73.6dBFS –40 Amplitude (dB) Amplitude (dB) 20 0 –20 –60 –80 –100 –40 –60 –80 –100 –120 –120 0 4 8 12 16 20 0 4 8 Frequency (MHz) 12 Frequency (MHz) SPECTRAL PERFORMANCE 2-TONE INTERMODULATION 0 0 FIN = 15MHz, –6dBFS F1 (–7dBc) = 14.5MHz SFDR = 85.3dBFS –20 F2 (–7dBc) = 15.5MHz –20 SNR = 74.6dBFS SFDR = –90dB –40 Amplitude (dB) Amplitude (dB) 16 SPECTRAL PERFORMANCE 0 –60 –80 –100 –40 –60 –80 –100 –120 –120 0 4 8 12 16 20 0 Frequency (MHz) 6 12 Frequency (MHz) 4 8 12 Frequency (MHz) ADS5421 www.ti.com SBAS237D TYPICAL CHARACTERISTICS (Cont.) TA = 25°C, +VSA = +VSD = +5V, differential input range = 1.5V to 3.5V each input (4Vp-p), sampling rate = 40MSPS, internal reference, and VDRV = 3V, unless otherwise noted. DIFFERENTIAL LINEARITY ERROR 0.5 INTEGRAL LINEARITY ERROR 4 FIN = 1MHz 0.4 FIN = 1MHz 3 0.3 2 0.1 ILE (LSB) DLE (LSB) 0.2 0.0 –0.1 1 0 –1 –0.2 –2 –0.3 –3 –0.4 –0.5 –4 0 2048 4096 6144 8192 10240 12288 14336 16384 0 2048 4096 6144 8192 10240 12288 14336 16384 Code Code SFDR AND SNR vs INPUT FREQUENCY (FCLK = 40MHz) SFDR AND SNR vs CLOCK FREQUENCY (FIN = 15MHz) 100 90 SFDR SFDR 85 SFDR, SNR (dBFS) SFDR, SNR (dBFS) 90 80 70 SNR 60 50 75 SNR 70 65 40 60 1.0 100 10 10 15 20 25 30 FIN (MHz) FCLK (MHz) SFDR AND SNR vs CLOCK FREQUENCY (FIN = 10MHz) SWEPT POWER (SFDR) (FIN = 10MHz) 35 40 –10 0 120 90 SFDR 110 85 dBFS 100 SFDR (dBFS, dBc) SFDR, SNR (dBFS) 80 80 75 SNR 70 65 90 80 70 60 50 dBc 40 30 20 10 60 0 10 15 20 25 30 35 40 –60 ADS5421 SBAS237D –50 –40 –30 –20 Input Amplitude (dBFS) FCLK (MHz) www.ti.com 7 TYPICAL CHARACTERISTICS (Cont.) TA = 25°C, +VSA = +VSD = +5V, differential input range = 1.5V to 3.5V each input (4Vp-p), sampling rate = 40MSPS, internal reference, and VDRV = 3V, unless otherwise noted. SWEPT POWER (SNR) (FIN = 10MHz) OUTPUT NOISE HISTOGRAM (DC Common-Mode Input) 700000 90 dBc 80 600000 500000 60 50 Count SNR (dBFS, dBc) 70 dBFS 40 30 400000 300000 200000 20 100000 10 0 0 –60 –50 –40 –30 –20 –10 N–3 0 N–2 N–1 Input Amplitude (dBFS) APPLICATION INFORMATION THEORY OF OPERATION The ADS5421 is a high-speed, high-performance, CMOS ADC build with a fully differential pipeline architecture. Each stage contains a low-resolution quantizer and digital error correction logic ensuring good differential linearity. The conversion process is initiated by a rising edge of the external convert clock. Once the signal is captured by the input trackand-hold amplifier, the bits are sequentially encoded starting with the Most Significant Bit (MSB). This process results in a data latency of 10 clock cycles after which the output data is available as a 14-bit parallel word either coded in a Straight Offset Binary or Binary Two’s Complement format. The analog input of the ADS5421 consists of a differential track-and-hold circuit, as shown in Figure 1. The differential topology produces a high level of AC performance at high sampling rates. It also results in a very high usable input bandwidth—especially important for Intermediate Frequency (IF) or undersampling applications. Both inputs (IN, IN) require external biasing up to a common-mode voltage that is typically at the mid-supply level (+VS/2). This is because the on-resistance of the CMOS switches is lowest at this voltage, minimizing the effects of the signal-dependent, 8 N N+1 N+2 N+3 Code nonlinearity of RON. For ease of use, the ADS5421 incorporates a selectable voltage reference, a versatile clock input, and a logic output driver designed to interface to 3V or 5V logic. S5 ADS5421 S3 VBIAS S1 CIN S2 CIN IN T&H IN S4 S6 VBIAS Tracking Phase: S1, S2, S3, S4 closed; S5, S6 open Hold Phase: S1, S2, S3, S4 open; S5, S6 closed FIGURE 1. Simplified Circuit of Input Track-and-Hold Amplifier. ADS5421 www.ti.com SBAS237D ANALOG INPUTS TYPES OF APPLICATIONS The analog input of the ADS5421 can be configured in various ways and driven with different circuits, depending on the application and the desired level of performance. Offering an extremely high dynamic range at high input frequencies, the ADS5421 is particularly well suited for communication systems that digitize wideband signals. Features on the ADS5421, like the input range selector, or the option of an external reference, provide the needed flexibility to accommodate a wide range of applications. In any case, the analog interface/driver requirements should be carefully examined before selecting the appropriate circuit configuration. The circuit definition should include considerations on the input frequency spectrum and amplitude, as well as the available power supplies. DIFFERENTIAL INPUTS The ADS5421 input structure is designed to accept the applied signal in differential format. Differential operation of the ADS5421 requires an input signal that consists of an in-phase and a 180° out-of-phase component simultaneously applied to the inputs (IN, IN). Differential signals offer a number of advantages, which in many applications will be instrumental in achieving the best harmonic performance of the ADS5421: • The signal amplitude is half of that required for the singleended operation and is, therefore, less demanding to achieve while maintaining good linearity performance from the signal source. • The reduced signal swing allows for more headroom of the interface circuitry and, therefore, a wider selection of the best suitable driver amplifier. • Even-order harmonics are minimized. • Improves the noise immunity based on the commonmode input rejection of the converter. Both inputs are identical in terms of their impedance and performance with the exception that by applying the signal to the complementary input (IN) instead of the IN input will invert the orientation of the input signal relative to the output code. 3dB down compared to the 4Vp-p range, while an improvement in the distortion performance of the driver amplifier may be realized due to the reduced output power level required. The third option, 2Vp-p full-scale range, may be considered mainly for applications requiring DC-coupling and/or singlesupply operation of the driver and the converter. INPUT BIASING (VCM) The ADS5421 operates from a single +5V supply, and requires each of the analog inputs to be externally biased to a common-mode voltage of typically +2.5V. This allows a symmetrical signal swing while maintaining sufficient headroom to either supply rail. Communication systems are usually AC-coupled in between signal processing stages, making it convenient to set individual common-mode voltages and allow optimizing the DC operating point for each stage. Other applications, such as imaging, process mainly unipolar or DC-restored signals. In this case, the common-mode voltage may be shifted such that the full input range of the converter is utilized. It should be noted that the CM pin is not internally buffered, but ties directly to the reference ladder. Therefore, it is recommended to keep loading of this pin to a minimum (< 100µA) to avoid an increase in the nonlinearity of the converter. Additionally, the DC voltage at the CM pin is not precisely +2.5V, but is subject to the tolerance of the top and bottom references, as well as the resistor ladder. Furthermore, the common-mode voltage typically declines with an increase in sampling frequency. This, however, does not affect the performance. INPUT IMPEDANCE The input of the ADS5421 is capacitive, and the driving source needs to provide the slew current to charge or discharge the input sampling capacitor while the track-and-hold amplifier is in track mode (see Figure 1). This effectively results in a dynamic input impedance that is a function of the sampling frequency. Figure 2 depicts the differential input impedance of the ADS5421 as a function of the input frequency. INPUT FULL-SCALE RANGE VERSUS PERFORMANCE 1000 100 ZIN (kΩ) Employing dual-supply amplifiers and AC-coupling will usually yield the best results. DC-coupling and/or single-supply amplifiers impose additional design constraints due to their headroom requirements, especially when selecting the 4Vp-p input range. The full-scale input range of the ADS5421 is defined either by the settings of the reference select pins (SEL1, SEL2) or by an external reference voltage (see Table I). By choosing between the different signal input ranges, trade-offs can be made between noise and distortion performance. For maximizing the SNR—important for timedomain applications—the 4Vp-p range may be selected. This range may also be used with low-level (–6dBFS to –40dBFS) but high-frequency inputs (multi-tone). The 3Vp-p range may be considered for achieving a combination of both low-noise and distortion performance. Here, the SNR number is typically 1 0.1 0.01 0.1 1 10 100 1000 fIN (MHz) FIGURE 2. Differential Input Impedance vs Input Frequency. ADS5421 SBAS237D 10 www.ti.com 9 For applications that use op amps to drive the ADC, it is recommended that a series resistor be added between the amplifier output and the converter inputs. This will isolate the capacitive input of the converter from the driving source and avoid gain peaking, or instability; furthermore, it will create a 1st-order, low-pass filter in conjunction with the specified input capacitance of the ADS5421. Its cutoff frequency can be adjusted further by adding an external shunt capacitor from each signal input to ground. The optimum values of this RC network, however, depend on a variety of factors, including the ADS5421 sampling rate, the selected op amp, the interface configuration, and the particular application (time domain versus frequency domain). Generally, increasing the size of the series resistor and/or capacitor will improve the SNR, however, depending on the signal source, large resistor values may be detrimental to the harmonic distortion performance. In any case, the use of the RC network is optional but optimizing the values to adapt to the specific application is encouraged. ANALOG INPUT DRIVER CONFIGURATIONS The following section provides some principal circuit suggestions on how to interface the analog input signal to the ADS5421. Applications that have a requirement for DCcoupling a new differential amplifier, such as the THS4502, can be used to drive the ADS5421, as shown in Figure 3. The THS4502 amplifier allows a single-ended to differential conversion to be performed easily, which reduces component cost. In addition, the VCM pin on the THS4502 can be directly tied to the common-mode pin (CM) of the ADS5421 in order to set up the necessary bias voltage for the converter inputs. As shown in Figure 3, the THS4502 is configured for unity gain. If required, higher gain can easily be configured, and a low-pass filter can be created by adding small capacitors (e.g., 10pF) in parallel to the feedback resistors. Due to the THS4502 driving a capacitive load, small series resistors in the output ensure stable operation. Further details of this and other functions of the THS4502 may be found in its product datasheet located at the Texas Instruments web site (www.ti.com). In general, differential amplifiers provide for a high-performance driver solution for baseband applications, and different differential amplifier models can be selected depending on the system requirements. TRANSFORMER-COUPLED INTERFACE CIRCUITS If the application allows for AC-coupling but requires a signal conversion from a single-ended source to drive the ADS5421 differentially, using a transformer offers a number of advantages. As a passive component, it does not add to the total noise, and by using a step-up transformer, further signal amplification can be realized. As a result, the signal swing of the amplifier driving the transformer can be reduced, leading to an increased headroom for the amplifier and improved distortion performance. A transformer interface solution is given in Figure 4. The input signal is assumed to be an IF and bandpass filtered prior to the IF amplifier. Dedicated IF amplifiers are commonly fixed-gain blocks and feature a very high bandwidth, low-noise figure, and a high intercept point, but at the expense of high quiescent currents, which are often around 100mA. The IF amplifier may be AC-coupled, or directly connected to the primary side of the transformer. A variety of miniature RF transformers are readily available from different manufacturers, (e.g., Mini-Circuits, Coilcraft, or Trak). For selection, it is important to carefully examine the application requirements and determine the correct model, the desired impedance ratio, and frequency characteristics. Furthermore, the appropriate model must support the targeted distortion level and should not exhibit any core saturation at full-scale voltage levels. The transformer center tap can be directly tied to the CM pin of the converter because it does not appreciably load the ADC reference (see Figure 4). The value of termination resistor RT must be chosen to satisfy the termination requirements of the source impedance (RS). It can be calculated using the equation RT = n2 • RS to ensure proper impedance matching. +5V 10pF(1) +5V 392Ω RS 392Ω 25Ω IN 56.2Ω VCM THS4502 25Ω ADS5421 22pF IN 0.1µF 392Ω 412Ω CM 10pF(1) –5V NOTES: Supply bypassing not shown. (1) Optional. FIGURE 3. Using the THS4502 Differential Amplifier (Gain = 1) to Drive the ADS5421 in a DC-Coupled Configuration. 10 ADS5421 www.ti.com SBAS237D +5V VIN (IF) Optional Bandpass Filter RS IF Amplifier XFR 1:n RIN IN RT RIN CIN ADS5421 IN CM NOTE: Supply bypassing not shown. + 0.1µF 2.2µF FIGURE 4. Driving the ADS5421 with a Low-Distortion IF Amplifier and a Transformer Suited for IF Sampling Applications. TRANSFORMER-COUPLED, SINGLE-ENDED-TODIFFERENTIAL CONFIGURATION For applications in which the input frequency is limited to approximately 10MHz (e.g., baseband), a high-speed operational amplifier may be used. The OPA847 is configured for the noninverting mode; this amplifies the single-ended input signal and drives the primary of a RF transformer (see Figure 5). To maintain the very low distortion performance of the OPA847, it may be advantageous to set the full-scale input range of the ADS5421 to 3Vp-p or 2Vp-p (refer to the Reference section for details on selecting the converter’s full-scale range). The circuit also shows the use of an additional RC low-pass filter placed in series with each converter input. This optional filter can be used to set a defined corner frequency and attenuate some of the wideband noise. The actual component values would need to be tuned for individual application requirements. As a guideline, resistor values are typically in the range of 10Ω to 50Ω, and capacitors in the range of 10pF to 100pF. In any case, the RIN and CIN values should have a low tolerance. This will ensure that the ADS5421 sees closely matched source impedances. +5V –5V +5V RG RS VIN OPA847 0.1µF RIN 1:n IN RT R1 RIN CIN ADS5421 IN VCM ≈ +2.5V CM R2 + 2.2µF 0.1µF FIGURE 5. Converting a Single-Ended Input Signal into a Differential Signal Using a RF Transformer. ADS5421 SBAS237D www.ti.com 11 The interface circuit example presented in Figure 6 employs two OPA687s, (decompensated voltage-feedback op amps), optimized for gains of 12V/V or higher. Implementing a new compensation technique allows the OPA847s to operate with a reduced signal gain of 8.5V/V, while maintaining the high loop gain and the associated excellent distortion performance offered by the decompensated architecture. For a detailed discussion on this circuit and the compensation scheme, refer to the OPA847 data sheet (SBOS251) located at www.ti.com. Input transformer, T1, converts the singleended input signal to a differential signal required at the inverting inputs of the amplifier, which are tuned to provide a 50Ω impedance match to an assumed 50Ω source. To achieve the 50Ω input match at the primary of the 1:2 transformer, the secondary must see a 200Ω load impedance. Both amplifiers are configured for the inverting mode resulting in close gain and phase matching of the differential signal. This technique, along with a highly symmetrical layout, is instrumental in achieving a substantial reduction of the 2nd-harmonic, while retaining excellent 3rd-order performance. A common-mode voltage, VCM, is applied to the noninverting inputs of the OPA847. Additional series 20Ω resistors isolate the output of the op amps from the capacitive load presented by the 40pF capacitors and the input capacitance of the ADS5421. This 20Ω/47pF combination sets a pole at approximately 85MHz and rolls off some of the wideband noise resulting in a reduction of the noise floor. For the measured 2-tone, 3rd-order distortion for the amplifier portion of the circuit of Figure 6, see Figure 7. The upper curve is for a total 2-tone envelope of 4Vp-p, requiring two tones, each 2Vp-p across the OPA847 outputs. The lower curve is for a 2Vp-p envelope, resulting in a 1Vp-p amplitude per tone. The basic measurement dynamic range for the two close-in spurious tones is approximately 85dBc. The 4Vp-p test does not show measurable 3rd-order spurious until 25MHz, while the 2Vp-p is unmeasureable up to 40MHz center frequency. 2-tone, 2nd-order intermodulation distortion was unmeasureable for this circuit. –60 3rd-Order Spurious (dBc) AC-COUPLED, DIFFERENTIAL INTERFACE WITH GAIN –65 4Vp-p –70 –75 2Vp-p –80 –85 0 5 10 15 20 25 30 35 40 45 50 Center Frequency (MHz) FIGURE 7. Measured 2-Tone, 3rd-Order Distortion for a Differential ADC Driver. +5V VCM 100Ω 20Ω OPA847 –5V +5V 1.7pF T1 50Ω Source 1:2 39pF 850Ω IN < 6dB Noise Figure 39pF ADS5421 47pF 850Ω IN CM 1.7pF 100Ω VCM +5V 20Ω 0.1µF OPA847 VCM –5V FIGURE 6. High Dynamic Range Interface Circuit with the OPA847 Set for a Gain of +8.5V/V. 12 ADS5421 www.ti.com SBAS237D REFERENCE REFERENCE OPERATION Integrated into the ADS5421 is a bandgap reference circuit, including logic that provides a +1V, +1.5V, or +2V reference output by selecting the corresponding pin-strap configuration. Table I lists a complete overview of the possible reference options and pin configurations. Figure 8 shows the basic model of the internal reference circuit. The functional blocks are a 1V bandgap voltage reference, a selectable gain amplifier, the drivers for the top and bottom reference (REFT, REFB), and the resistive reference ladder. The ladder resistance measures approximately 1kΩ between the REFT and REFB pins. The ladder is split into two equal segments establishing a common-mode voltage at the ladder midpoint, labeled CM. The ADS5421 requires solid bypassing for all reference pins to keep the effects of clock feedthrough to a minimum and to achieve the specified level of performance. Figure 8 shows the recommended decoupling scheme. All 0.1µF capacitors must be located as close to the pins as possible. In addition, pins REFT, CM, and REFB must be decoupled with tantalum surface-mount capacitors (2.2µF or 4.7µF). When operating the ADS5421 with the internal reference, the effective full-scale input span for each of the inputs, IN and IN, is determined by the voltage at the VREF pin, given to: The top and bottom reference outputs can be used to provide up to 1mA of current (sink or source) to external circuits. Degradation of the differential linearity (DNL) and, consequently, the dynamic performance, of the ADS5421 may occur if this limit is exceeded. USING EXTERNAL REFERENCES For even more design flexibility, the ADS5421 can be operated with external references. The utilization of an external reference voltage may be considered for applications requiring higher accuracy, improved temperature stability, or a continuous adjustment of the converter full-scale range. Especially in multichannel applications, the use of a common external reference offers the benefit of improving the gain matching between converters. Selection between internal or external reference operation is controlled through the VREF pin. The internal reference will become disabled if the voltage applied to the VREF pin exceeds +3.5VDC. Once selected, the ADS5421 requires two reference voltages: a top reference voltage applied to the REFT pin and a bottom reference voltage applied to the REFB pin (see Table I). As illustrated in Figure 9, a micropower reference (REF1004) and a dual, single-supply amplifier (OPA2234) can be used to generate a precision external reference. Note that the function of the range select pins, SEL1 and SEL2, are disabled while the converter is operating in external reference mode. (1) Input Span (differential, each input) = VREF = (REFT – REFB) in Vp-p DESIRED FULL-SCALE RANGE (FSR) (DIFFERENTIAL) CONNECT SEL1 (PIN 45) TO: CONNECT SEL2 (PIN 44) TO: 4Vp-p (+16dBm) GND GND +2.0V +3.5V +1.5V 3Vp-p (+13dBm) GND +VSA +1.5V +3.25V +1.75V 2Vp-p (+10dBm) VREF GND +1.0V +3.0V +2.0V — — > +3.5V +2.75V to +4.5V +0.5V to +2.25V External Reference VOLTAGE AT VREF (PIN 46) VOLTAGE AT REFT (PIN 52) VOLTAGE AT REFB (PIN 50) TABLE I. Reference Pin Configurations and Corresponding Voltages on the Reference Pins. SEL1 SEL2 45 REFBY 61 0.1µF 44 Range Select and Gain Amplifier Top Reference Driver REFT 52 0.1µF + 2.2µF 500Ω CM +1VDC Bandgap Reference 51 0.1µF + 2.2µF 500Ω Bottom Reference Driver ADS5421 REFB 50 0.1µF + 2.2µF 46 VREF 0.1µF FIGURE 8. Internal Reference Circuit of the ADS5421 and Recommended Bypass Scheme. ADS5421 SBAS237D www.ti.com 13 +5V +5V 1/2 OPA2234 4.7kΩ REFT + R3 + 0.1µF ADS5421 R4 R1 REF1004 +2.5V 2.2µF 10µF 1/2 OPA2234 R2 REFB + 0.1µF 2.2µF 0.1µF FIGURE 9. Example for an External Reference Circuit Using a Dual, Single-Supply Op Amp. DIGITAL INPUTS AND OUTPUTS CLOCK INPUT CLK Unlike most ADCs, the ADS5421 contains internal clock conditioning circuitry. This enables the converter to adapt to a variety of application requirements and different clock sources. With no input signal connected to either clock pin, the threshold level is set to approximately +1.6V by the onchip resistive voltage divider, as shown in Figure 10. The parallel combination of R1 || R2 and R3 || R4 sets the input impedance of the clock inputs (CLK, CLK) to approximately 2.7kΩ single-ended, or 5.5kΩ differentially. The associated ground referenced input capacitance is approximately 5pF for each input. If a logic voltage other than the nominal +1.6V is desired, the clock inputs can be externally driven to establish an alternate threshold voltage. +5V R1 8.5kΩ ADS5421 R3 8.5kΩ CLK CLK R2 4kΩ R4 4kΩ FIGURE 10. The Differential Clock Inputs are Internally Biased. The ADS5421 can be interfaced to standard TTL or CMOS logic and accepts 3V or 5V compliant logic levels. In this case, the clock signal should be applied to the CLK input, whereas the complementary clock input (CLK) should be bypassed to ground by a low-inductance ceramic chip capacitor, as shown in Figure 11. Depending on the quality of the signal, inserting a series, damping resistor can be beneficial to reduce ringing. When digitizing at high sampling rates the clock should have a 50% duty cycle (tH = tL) to maintain good distortion performance. TTL/CMOS Clock Source (3V/5V) ADS5421 CLK 47nF FIGURE 11. Single-Ended TTL/CMOS Clock Source. Applying a single-ended clock signal will provide satisfactory results in many applications. However, unbalanced high-speed logic signals can introduce a high amount of disturbances, such as ringing or ground bouncing. In addition, a high amplitude can cause the clock signal to have unsymmetrical rise-and-fall times, potentially affecting the converter distortion performance. Proper termination practice and a clean PC board layout will help to keep those effects to a minimum. To take full advantage of the excellent distortion performance of the ADS5421, it is recommended to drive the clock inputs differentially. A differential clock improves the digital feedthrough immunity and minimizes the effect of modulation between the signal and the clock. Figure 12 illustrates a simple method of converting a square wave clock from single-ended to differential using an RF transformer. Small surface-mount transformers are readily available from several manufacturers (e.g., model ADT1-1 by Mini-Circuits). A capacitor in series with the primary side may be inserted to block any DC voltage present in the signal. The secondary side connects directly to the two clock inputs of the converter because the clock inputs are self-biased. Square Wave or Sine Wave Clock Source RS XFR 1:1 0.1µF RT CLK ADS5421 CLK FIGURE 12. Connecting a Ground-Referenced Clock Source to the ADS5421 Using an RF Transformer. 14 ADS5421 www.ti.com SBAS237D MINIMUM SAMPLING RATE The clock inputs of the ADS5421 can be connected in a number of ways. However, the best performance is obtained when the clock input pins are driven differentially. Operating in this mode, the clock inputs accommodate signal swings ranging from 2.5Vp-p down to 0.5Vp-p differentially. This allows direct interfacing of clock sources such as voltage-controlled crystal oscillators (VCXO) to the ADS5421. The advantage here is the elimination of external logic, usually necessary to convert the clock signal into a suitable logic (TTL or CMOS) signal that otherwise would create an additional source of jitter. In any case, a very low-jitter clock is fundamental to preserving the excellent AC performance of the ADS5421. The converter itself is specified for a low jitter, characterizing the outstanding capability of the internal clock and track-andhold circuitry. Generally, as the input frequency increases, the clock jitter becomes more dominant for maintaining a good signal-to-noise ratio. This is particularly critical in IF sampling applications where the sampling frequency is lower than input frequency (undersampling). The following equation can be used to calculate the achievable SNR for a given input frequency and clock jitter (tJA in ps rms): SNR = 20 log10 1 π 2 f ( INt JA ) The pipeline architecture of the ADS5421 uses a switchedcapacitor technique in its internal track-and-hold stages. With each clock cycle, charges representing the captured signal level are moved within the ADC pipeline core. The high sampling speed necessitates the use of very small capacitor values. In order to hold the droop errors low, the capacitors require a minimum refresh rate. To maintain accuracy of the acquired sample charge, the sampling clock on the ADS5421 must not drop below the specified minimum of 1MHz. DATA OUTPUT FORMAT (BTC) The ADS5421 makes two data output formats available, either the Straight Offset Binary (SOB) code or the Binary Two’s Complement (BTC) code. The selection of the output coding is controlled through the BTC pin. Applying a logic HIGH will enable the BTC coding, whereas a logic LOW will enable the SOB code. The BTC output format is widely used to interface to microprocessors, for example. The two code structures are identical with the exception that the MSB is inverted for the BTC format, as shown in Table II. (2) If the input signal exceeds the full-scale range, the output code will remain at all 1s or all 0s. Depending on the nature of the clock source output impedance, impedance matching might become necessary. For this, a termination resistor, RT, can be installed (see Figure 12). To calculate the correct value for this resistor, consider the impedance ratio of the selected transformer and the differential clock input impedance of the ADS5421, which is approximately 5.5kΩ. Shown in Figure 13 is one preferred method for clocking the ADS5421. Here, the single-ended clock source can be either a square wave or a sine wave. Using the high-speed differential translator SN65LVDS100 from Texas Instruments, a low-jitter clock can be generated to drive the clock inputs of the ADS5421 differentially. DIFFERENTIAL INPUT STRAIGHT OFFSET BINARY (SOB) BINARY TWO’S COMPLEMENT (BTC) +FS – 1LSB (IN = +3.5V, IN = +1.5V) 11 1111 1111 1111 01 1111 1111 1111 +1/2 FS 11 0000 0000 0000 01 0000 0000 0000 Bipolar Zero (IN = IN = VCM) 10 0000 0000 0000 00 0000 0000 0000 –1/2 FS 01 0000 0000 0000 11 0000 0000 0000 –FS (IN = +1.5V, IN = +3.5V) 00 0000 0000 0000 10 0000 0000 0000 TABLE II. Coding Table for Differential Input Configuration and 4Vp-p Full-Scale Input Range. +5V 0.01µF Square Wave Or Sine Wave Clock Input SN65LVDS100 0.01µF A RT(1) 100Ω 0.01µF Y CLK B 0.01µF Z VBB ADS5421 CLK 50Ω 50Ω 0.01µF NOTE: (1) Additional termination resistor RT may be necessary depending on the source requirements FIGURE 13. Differential Clock Driver Using an LVDS Translator. ADS5421 SBAS237D www.ti.com 15 OUTPUT ENABLE (OE ) POWER DISSIPATION The digital outputs of the ADS5421 can be set to high impedance (tri-state), exercising the output enable pin (OE). For normal operation, this pin must be at a logic LOW potential, whereas a logic HIGH voltage disables the outputs. Even though this function affects the output driver stage, the threshold voltages for the OE pin do not depend on the output driver supply (VDRV), but are fixed (see the Electrical Characteristics Table and the Digital Inputs Sections). Operating the OE function dynamically (e.g., high-speed multiplexing) should be avoided as it will corrupt the conversion process. A majority of the ADS5421 total power consumption is used for biasing, therefore; it is independent of the applied clock frequency. Figure 14 shows the typical variation in power consumption versus the clock speed. The current on the VDRV supply is directly related to the capacitive loading of the data output pins and care must be taken to minimize such loading. 45 FIN = 10MHz 40 Sample Rate (MSPS) POWER DOWN (PD) A power-down pin is provided; when taken HIGH, this pin shuts down portions within the ADS5421 and reduces the power dissipation to less than 40mW. The remaining active blocks include the internal reference, ensuring a fast reactivation time. During power-down, data in the converter pipeline will be lost and new valid data will be subject to the specified pipeline delay. If the PD pin is not used, it should be tied to ground or a logic LOW level. 35 30 25 20 15 700 720 740 760 780 800 820 840 880 Power Dissipation (mW) OUTPUT LOADING It is recommended to keep the capacitive loading on the data output lines as low as possible, preferably below 15pF. Higher capacitive loading will cause larger dynamic currents as the digital outputs are changing. For example, with a typical output slew rate of 0.8V/ns and a total capacitive loading of 10pF (including 4pF output capacitance, 5pF input capacitance of external logic buffer, and 1pF PC board parasitics), a bit transition can cause a dynamic current of (10pF • 0.8V/1ns = 8mA). These high current surges can feed back to the analog portion of the ADS5421 and adversely affect the performance. If necessary, external buffers or latches close to the converter’s output pins can be used to minimize the capacitive loading. They also provide the added benefit of isolating the ADS5421 from any digital activities on the bus coupling back high-frequency noise. POWER SUPPLIES When defining the power supplies for the ADS5421, it is highly recommended to consider linear supplies instead of switching types. Even with good filtering, switching supplies can radiate noise that could interfere with any highfrequency input signal and cause unwanted modulation products. At its full conversion rate of 40MHz, the ADS5421 typically requires 170mA of supply current on the +5V supplies. Note that this supply voltage should stay within a 5% tolerance. 16 FIGURE 14. Power Dissipation vs Clock Frequency. DIGITAL OUTPUT DRIVER SUPPLY (VDRV) A dedicated supply pin, VDRV, provides power to the logic output drivers of the ADS5421 and can be operated with a supply voltage in the range of +3.0V to +5.0V. This can simplify interfacing to various logic families, in particular lowvoltage CMOS. It is recommended to operate the ADS5421 with a +3.3V supply voltage on VDRV. This will lower the power dissipation in the output stages due to the lower output swing and reduce current glitches on the supply line that may affect the AC performance of the converter. The analog supply (+VSA) and digital supply (+VSD) may be tied together, with a ferrite bead or inductor between the supply pins. Each of the these supply pins must be bypassed separately with at least one 0.1µF ceramic chip capacitor, forming a pi-filter (see Figure 15). The recommended operation for the ADS5421 is +5V for the +VS pins and +3.3V on the output driver pin (VDRV). The configuration of the supplies requires that a specific power-up sequence be followed for the ADS5421. Analog voltage must be applied to the analog supply pin (+VSA) before applying a voltage to the driver supply (VDRV) or before bringing both the digital supply (+VSD) and VDRV up simultaneously. Powering up +VSD and VDRV prior to +VSA will cause a large current on +VSA and result in the ADS5421 not functioning properly. ADS5421 www.ti.com SBAS237D VIN 50Ω ADT2-1 4.7µF + +VA (5V) 0.1µF 0.1µF 22Ω 22Ω 4.7µF + 4.7µF + 0.1µF 0.1µF 22pF 56 55 54 53 52 51 50 49 GND GND IN GND IN GND GND GND GND REFT CM REFB GND +VSA 57 GND 48 GND 47 +VSD VREF 46 +VSD SEL1 45 5 +VSD SEL2 44 6 +VSD GND 43 7 GND GND 42 8 GND BTC 41 9 CLK PD 40 10 CLK OE 39 11 GND GNDRV 38 12 GND GNDRV 37 13 GNDRV GNDRV 36 14 GNDRV VDRV 35 15 DNC VDRV 34 16 DV VDRV 33 B8 B9 B10 B11 B12 B13 B14 NC NC ADS5421 B7 50Ω 4 58 B6 0.1µF ADT2-1 3 59 B5 RS CLKIN +VSA 60 B4 0.1µF 10µF 2 61 B3 +VD (5V) +VSA 62 B2 0.01µF 1 63 B1 0.1µF 64 +VSA 10µF + 0.1µF REFBY 0.1µF 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 0.1µF 0.01µF NC NC D13 D12 D11 D9 D10 D8 D7 D6 D5 D4 D3 D2 D1 DV DO 0.1µF 10µF + 0.1µF +VDR (3.3V) FIGURE 15. Basic Application Circuit of the ADS5421 Includes Recommended Supply and Reference Bypassing. ADS5421 SBAS237D www.ti.com 17 LAYOUT AND DECOUPLING CONSIDERATIONS Proper grounding and bypassing, short lead length, and the use of ground planes are particularly important for highfrequency designs. Achieving optimum performance with a fast sampling converter like the ADS5421 requires careful attention to the PC board layout to minimize the effect of board parasitics and optimize component placement. A multilayer board usually ensures best results and allows convenient component placement. The ADS5421 must be treated as an analog component and the +VSA pins connected to a clean analog supply. This ensures the most consistent results, because digital supplies often carry a high level of switching noise that could couple into the converter and degrade the performance. As mentioned previously, the driver supply pins (VDRV) must also be connected to a low-noise supply. Supplies of adjacent digital circuits can carry substantial current transients. The supply voltage must be thoroughly filtered before connecting to the VDRV supply of the converter. All ground connections on the ADS5421 are internally bonded to the metal flag (bottom of package) that forms a large ground plane. All ground pins must directly connect to an analog ground plane that covers the PC board area under the converter. Due to its high sampling frequency, the ADS5421 generates high-frequency current transients and noise (clock feedthrough) that are fed back into the supply and reference lines. If not sufficiently bypassed, this adds noise to the conversion process. See Figure 15 for the recommended supply decoupling scheme for the ADS5421. All +VS pins should be bypassed with a combination of 10nF, 0.1µF ceramic chip capacitors (0805, low ESR) and a 10µF tantalum tank capacitor. A similar approach may be used on the driver supply pins, VDRV. In order to minimize the lead and trace inductance, the capacitors must be located as close to 18 the supply pins as possible. They are best placed directly under the package where double-sided component mounting is allowed. In addition, larger bipolar decoupling capacitors (2.2µF to 10µF), effective at lower frequencies, must also be used on the main supply pins. They can be placed on the PC board in proximity (< 0.5") of the ADC. If the analog inputs to the ADS5421 are driven differentially, it is especially important to optimize towards a highly symmetrical layout. Small trace length differences can create phase shifts compromising a good distortion performance. For this reason, the use of two single op amps rather than one dual amplifier enables a more symmetrical layout and a better match of parasitic capacitances. The pin orientation of the ADS5421 package follows a flow-through design with the analog inputs located on one side of the package whereas the digital outputs are located on the opposite side of the quad-flat package. This provides a good physical isolation between the analog and digital connections. While designing the layout, it is important to keep the analog signal traces separated from any digital lines to prevent noise coupling onto the analog portion. Try to match trace length for the differential clock signal (if used) to avoid mismatches in propagation delays. Singleended clock lines must be short and should not cross any other signal traces. Short circuit traces on the digital outputs will minimize capacitive loading. Trace length must be kept short to the receiving gate (< 2") with only one CMOS gate connected to one digital output. If possible, the digital data outputs must be buffered (with the TI SN74AVC16244, for example). Dynamic performance can also be improved with the insertion of series resistors at each data output line. This sets a defined time constant and reduces the slew rate that would otherwise flow due to the fast edge rate. The resistor value may be chosen to result in a time constant of 15% to 25% of the used data rate. ADS5421 www.ti.com SBAS237D PACKAGE OPTION ADDENDUM www.ti.com 9-Dec-2004 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty ADS5421Y/R ACTIVE LQFP PM 64 1500 None CU SNPB Level-3-220C-168 HR ADS5421Y/T ACTIVE LQFP PM 64 250 None CU SNPB Level-3-220C-168 HR Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. None: Not yet available Lead (Pb-Free). Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens, including bromine (Br) or antimony (Sb) above 0.1% of total product weight. (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 1 MECHANICAL DATA MTQF008A – JANUARY 1995 – REVISED DECEMBER 1996 PM (S-PQFP-G64) PLASTIC QUAD FLATPACK 0,27 0,17 0,50 0,08 M 33 48 49 32 64 17 0,13 NOM 1 16 7,50 TYP Gage Plane 10,20 SQ 9,80 12,20 SQ 11,80 0,25 0,05 MIN 0°– 7° 0,75 0,45 1,45 1,35 Seating Plane 0,08 1,60 MAX 4040152 / C 11/96 NOTES: A. B. C. D. All linear dimensions are in millimeters. This drawing is subject to change without notice. Falls within JEDEC MS-026 May also be thermally enhanced plastic with leads connected to the die pads. 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