® ADS822 ADS825 ADS 825 ADS 822 For most current data sheet and other product information, visit www.burr-brown.com 10-Bit, 40MHz Sampling ANALOG-TO-DIGITAL CONVERTERS TM ● +3V OR +5V LOGIC I/O COMPATIBLE (ADS825) ● POWER DOWN: 20mW ● 28-LEAD SSOP PACKAGE FEATURES ● ● ● ● ● HIGH SNR: 60dB HIGH SFDR: 72dBFS LOW POWER: 190mW INTERNAL/EXTERNAL REFERENCE OPTION SINGLE-ENDED OR FULLY DIFFERENTIAL ANALOG INPUT ● PROGRAMMABLE INPUT RANGE ● LOW DNL: 0.5LSB ● SINGLE +5V SUPPLY OPERATION APPLICATIONS ● MEDICAL IMAGING ● TEST EQUIPMENT ● COMPUTER SCANNERS ● COMMUNICATIONS ● VIDEO DIGITIZING DESCRIPTION The ADS822 and ADS825 employ digital error correction techniques to provide excellent differential linearity for demanding imaging applications. Its low distortion and high SNR give the extra margin needed for medical imaging, communications, video, and test instrumentation. The ADS822 and ADS825 offer power dissipation of 190mW and also provide a power-down mode, thus reducing power dissipation to only 20mW. The ADS825 is +3V or +5V Logic I/O compatible. The ADS822 and ADS825 are pipeline, CMOS analog-to-digital converters that operate from a single +5V power supply. These converters provide excellent performance with a single-ended input and can be operated with a differential input for added spurious performance. These high-performance converters include a 10-bit quantizer, high-bandwidth track-and-hold, and a high-accuracy internal reference. They also allow for the user to disable the internal reference and utilize external references. This external reference option provides excellent gain and offset matching when used in multi-channel applications or in applications where full-scale range adjustment is required. +VS CLK ADS822 ADS825 VIN IN IN T/H The ADS822 and ADS825 are specified at a maximum sampling frequency of 40MHz and a single-ended input range of 1.5V to 3.5V. The ADS822 and ADS825 are available in a 28-lead SSOP package and are pin-for-pin compatible with the 10-bit, 60MHz ADS823 and ADS826, and the 10-bit, 70MHz ADS824, providing an upgrade path to higher sampling frequencies. VDRV Timing Circuitry 10-Bit Pipelined A/D Core Error Correction Logic 3-State Outputs D0 • • • D9 Internal Reference CM Optional External Reference Int/Ext PD OE International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 © 1997 Burr-Brown Corporation PDS-1385E Printed in U.S.A. October, 1999 SPECIFICATIONS At TA = full specified temperature range, VS = +5V, single-ended input range = 1.5V to 3.5V, and sampling rate = 40MHz, external reference, unless otherwise noted. ADS825E(1) ADS822E PARAMETER CONDITIONS MIN RESOLUTION SPECIFIED TEMPERATURE RANGE ANALOG INPUT Standard Single-Ended Input Range Optional Single-Ended Input Range Common-Mode Range Optional Differential Input Range Analog Input Bias Current Input Impedance Track-Mode Input Bandwidth Ambient Air 2Vp-p 1Vp-p 1.5 2 2Vp-p 2 DIGITAL INPUTS Logic Family Convert Command High Level Input Current(6) (VIN = 5VDD) Low Level Input Current (VIN = 0V) High Level Input Voltage Low Level Input Voltage Input Capacitance MAX MIN MAX UNITS Bits –40 to +85 –40 to +85 °C 3.5 3 ✻ ✻ 3 ✻ 10k ✻ ✻ ✻ ✻ 40M ✻ ±1.0 ✻ ✻ Guaranteed ✻ ±2.0 V V V V µA MΩ || pF MHz ✻ Samples/s Clk Cyc ✻ LSB LSB ✻ LSBs ✻ 5 ±0.25 ±0.5 Guaranteed ±0.5 ✻ ✻ ✻ 1 1.25 || 5 300 –3dBFS Input TYP 10 Guaranteed 2.5 CONVERSION CHARACTERISTICS Sample Rate Data Latency DYNAMIC CHARACTERISTICS Differential Linearity Error (largest code error) f = 1MHz f = 10MHz No Missing Codes Integral Nonlinearity Error, f = 1MHz Spurious Free Dynamic Range(2) f = 1MHz f = 10MHz Two-Tone Intermodulation Distortion(4) f = 9.5MHz and 9.9MHz (–7dB each tone) Signal-to-Noise Ratio (SNR) f = 1MHz f = 10MHz Signal-to-(Noise + Distortion) (SINAD) f = 1MHz f = 10MHz Effective Number of Bits(5), f = 1MHz Output Noise Aperture Delay Time Aperture Jitter Overvoltage Recovery Time Full-Scale Step Acquisition Time TYP 10 Guaranteed Referred to Full Scale 71 65 dBFS(3) dBFS ✻ dBc ✻ ✻ dB dB ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ dB dB Bits LSBs rms ns ps rms ns ns CMOS-Compatible Rising Edge of Convert Clock 100 10 +3.5 +1.0 5 TTL, +3V/+5V CMOS-Compatible Rising Edge of Convert Clock ✻ ✻ +2.0 +0.8 ✻ µA µA V V pF CMOS-Compatible Straight Offset Binary +0.1 +4.9 +0.1 +2.8 2 40 2 10 5 CMOS-Compatible Straight Offset Binary 63 72 66 60 –67 Referred to Full Scale 57 60 60 ✻ Referred to Full Scale 56 Input Tied to Common-Mode Start Conversion DIGITAL OUTPUTS Logic Family Logic Coding Low Output Voltage (IOL = 50µA to 1.6mA) High Output Voltage, (IOH = 50µA to 0.5mA) Low Output Voltage, (IOL = 50µA to 1.6mA) High Output Voltage, (IOH = 50µA to 0.5mA) 3-State Enable Time 3-State Disable Time Output Capacitance VDRV = 5V VDRV = 3V OE = H to L OE = L to H ACCURACY (Internal Reference, 2Vp-p, Unless Otherwise Noted) Zero Error (referred to –FS) at 25°C Zero Error Drift (referred to –FS) Midscale Offset Error at 25°C Gain Error(7) at 25°C Gain Error Drift(7) Gain Error(8) at 25°C Gain Error Drift(8) Power Supply Rejection of Gain ∆ VS = ±5% REFT Tolerance Deviation From Ideal 3.5V REFB Tolerance Deviation From Ideal 1.5V External REFT Voltage Range External REFB Voltage Range Reference Input Resistance REFT to REFB REFB + 0.8 1.25 ® ADS822, ADS825 2 59 58 9.5 0.2 3 1.2 2 5 ✻ ±1.0 5 ±3.0 ±1.5 38 ±0.75 25 70 ±10 ±10 3.5 1.5 1.6 ±2.5 ✻ ✻ ✻ ✻ ±1.5 ±25 ±25 VS – 1.25 REFT – 0.8 ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ±0.29 ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ V V V V ns ns pF % FS ppm/°C % FS % FS ppm/°C % FS ppm/°C dB mV mV V V kΩ SPECIFICATIONS (Cont.) At TA = full specified temperature range, VS = +5V, single-ended input range = 1.5V to 3.5V, and sampling rate = 40MHz, external reference, unless otherwise noted. ADS825E(1) ADS822E PARAMETER POWER SUPPLY REQUIREMENTS Supply Voltage: +VS Supply Current: +IS Power Dissipation: VDRV = 5V VDRV = 3V VDRV = 5V VDRV = 3V Power Down Thermal Resistance, θJA 28-Lead SSOP CONDITIONS MIN TYP MAX MIN TYP MAX UNITS Operating Operating (External Reference) External Reference External Reference Internal Reference Internal Reference Operating +4.75 +5.0 40 200 190 250 240 20 +5.25 ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ V mA mW mW mW mW mW 230 ✻ ✻ 89 °C/W ✻ Indicates the same specifications as the ADS822E. NOTES: (1) ADS825E accepts a +3V clock input. (2) Spurious Free Dynamic Range refers to the magnitude of the largest harmonic. (3) dBFS means dB relative to Full Scale. (4) Two-tone intermodulation distortion is referred to the largest fundamental tone. This number will be 6dB higher if it is referred to the magnitude of the two-tone fundamental envelope. (5) Effective number of bits (ENOB) is defined by (SINAD – 1.76)/6.02. (6) A 50kΩ pull-down resistor is inserted internally on OE pin. (7) Includes internal reference. (8) Excludes internal reference. PIN DESCRIPTIONS PIN CONFIGURATION Top View SSOP GND 1 28 VDRV Bit 1 (MSB) 2 27 +VS Bit 2 3 26 GND Bit 3 4 25 IN Bit 4 5 24 IN Bit 5 6 23 CM Bit 6 7 22 REFT 21 ByT ADS822 ADS825 Bit 7 8 Bit 8 9 20 ByB Bit 9 10 19 REFB Bit 10 (LSB) 11 18 INT/EXT OE 12 17 RSEL PD 13 16 GND CLK 14 15 +VS PIN DESIGNATOR 1 2 3 4 5 6 7 8 9 10 11 12 GND Bit 1 Bit 2 Bit 3 Bit 4 Bit 5 Bit 6 Bit 7 Bit 8 Bit 9 Bit 10 OE 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 PD CLK +VS GND RSEL INT/EXT REFB ByB ByT REFT CM IN IN GND +VS VDRV DESCRIPTION Ground Data Bit 1 (D9) (MSB) Data Bit 2 (D8) Data Bit 3 (D7) Data Bit 4 (D6) Data Bit 5 (D5) Data Bit 6 (D4) Data Bit 7 (D3) Data Bit 8 (D2) Data Bit 9 (D1) Data Bit 10 (D0) (LSB) Output Enable. HI = high impedance state LO = normal operation (internal pull-down resistor) Power Down. HI = enable; LO = disable Convert Clock Input +5V Supply Ground Input Range Select. HI = 2V; LO = 1V Reference Select. HI = external, LO = internal Bottom Reference Bottom Ladder Bypass Top Ladder Bypass Top Reference Common-Mode Voltage Output Complementary Input (–) Analog Input (+) Analog Ground +5V Supply Output Logic Driver Supply Voltage ® 3 ADS822, ADS825 TIMING DIAGRAM N+2 N+1 Analog In N+4 N+3 N tD N+5 tL tCONV N+7 N+6 tH Clock 5 Clock Cycles t2 Data Out N–5 N–4 N–3 N–2 N–1 Data Invalid SYMBOL tCONV tL tH tD t1 t2 N N+1 N+2 t1 DESCRIPTION MIN Convert Clock Period Clock Pulse Low Clock Pulse High Aperture Delay Data Hold Time, CL = 0pF New Data Delay Time, CL = 15pF max 25 11.5 11.5 TYP MAX UNITS 100µs ns ns ns ns ns ns 12.5 12.5 3 3.9 12 PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER ADS822E " ADS825E " SSOP-28 " SSOP-28 " 324 " 324 " SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER(1) TRANSPORT MEDIA –40°C to +85°C " –40°C to +85°C " ADS822E " ADS825E " ADS822E ADS822E/1K ADS825E ADS825E/1K Rails Tape and Reel Rails Tape and Reel NOTE: (1) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /1K indicates 1000 devices per reel). Ordering 1000 pieces of ADS822E/1K” will get a single 1000-piece Tape and Reel. ELECTROSTATIC DISCHARGE SENSITIVITY DEMO BOARD ORDERING INFORMATION PRODUCT DEMO BOARD ADS822E DEM-ADS822E This integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ABSOLUTE MAXIMUM RATINGS +VS ....................................................................................................... +6V Analog Input ............................................................. –0.3V to (+VS + 0.3V) Logic Input ............................................................... –0.3V to (+VS + 0.3V) Case Temperature ......................................................................... +100°C Junction Temperature .................................................................... +150°C Storage Temperature ..................................................................... +150°C ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® ADS822, ADS825 4 TYPICAL PERFORMANCE CURVES At TA = full specified temperature range, VS = +5V, single-ended input range = 1.5V to 3.5V, and sampling rate = 40MHz, external reference, unless otherwise noted. SPECTRAL PERFORMANCE SPECTRAL PERFORMANCE 0 0 fIN = 1MHz fIN = 10MHz –20 Magnitude (dB) Magnitude (dB) –20 –40 –60 –80 –100 0 5 10 15 20 0 5 10 15 Frequency (MHz) Frequency (MHz) SPECTRAL PERFORMANCE (Differential Input, 1Vp-p) SPECTRAL PERFORMANCE (Single-Ended, 1Vp-p) 0 20 0 fIN = 10MHz SNR = 58dBFS SFDR = 74dBFS fIN = 10MHz SNR = 57dBFS SFDR = 71dBFS –20 Magnitude (dB) –20 Magnitude (dB) –60 –80 –100 –40 –60 –80 –40 –60 –80 –100 –100 0 5 10 15 20 0 5 10 15 Frequency (MHz) Frequency (MHz) SPECTRAL PERFORMANCE (Single-Ended, 1Vp-p) UNDERSAMPLING (Differential Input, 2Vp-p) 0 20 0 fIN = 20MHz SNR = 57dBFS SFDR = 70dBFS fS = 40MHz fIN = 45MHz SNR = 60dBFS SFDR = 74dBFS –20 Magnitude (dB) –20 Magnitude (dB) –40 –40 –60 –80 –40 –60 –80 –100 –100 0 5 10 15 20 0 Frequency (MHz) 5 10 15 20 Frequency (MHz) ® 5 ADS822, ADS825 TYPICAL PERFORMANCE CURVES (Cont.) At TA = full specified temperature range, VS = +5V, single-ended input range = 1.5V to 3.5V, and sampling rate = 40MHz, external reference, unless otherwise noted. UNDERSAMPLING (Differential Input, 2Vp-p) TWO-TONE INTERMODULATION DISTORTION 0 0 fS = 40MHz fIN = 75MHz SNR = 59dBFS SFDR = 66dBFS –20 Magnitude (dB) Magnitude (dB) –20 f1 = 9.5MHz at –7dBFS f2 = 9.9MHz at –7dBFS IMD (3) = –67dB –40 –60 –40 –60 –80 –80 –100 –100 0 5 10 15 0 20 5 10 DIFFERENTIAL LINEARITY ERROR 1.0 fIN = 1MHz fIN = 10MHz 0.5 DLE (LSB) DLE (LSB) 20 DIFFERENTIAL LINEARITY ERROR 1.0 0.5 0 –0.5 0 –0.5 –1.0 –1.0 0 20 40 60 80 1024 0 20 40 Output Code 60 80 1024 Output Code INTEGRAL LINEARITY ERROR SWEPT POWER SFDR 2.0 100 80 SFDR (dBFS, dBc) 1.0 ILE (LSB) 15 Frequency (MHz) Frequency (MHz) 0 –1.0 dBFS 60 40 dBc 20 –2.0 0 0 256 512 768 1024 –60 Output Code –40 –30 –20 Input Amplitude (dBFS) ® ADS822, ADS825 –50 6 –10 0 TYPICAL PERFORMANCE CURVES (Cont.) At TA = full specified temperature range, VS = +5V, single-ended input range = 1.5V to 3.5V, and sampling rate = 40MHz, external reference, unless otherwise noted. DYNAMIC PERFORMANCE vs TEMPERATURE DYNAMIC PERFORMANCE vs INPUT FREQUENCY 75 75 SFDR SFDR, SNR (dBFS) SFDR, SNR (dBFS) 70 65 60 SNR 55 70 SFDR (fIN = 10MHz) 65 SFDR (fIN = 20MHz) SNR (fIN = 10MHz) 60 SNR (fIN = 20MHz) 50 55 0.1 1 10 –50 100 –25 0 25 50 75 Frequency (MHz) Temperature (°C) SIGNAL-TO-(NOISE + DISTORTION) vs TEMPERATURE DIFFERENTIAL LINEARITY ERROR vs TEMPERATURE 60 100 .60 fIN = 1MHz DLE (LSB) Sinad (dBFS) fIN = 20MHz 59 fIN = 10MHz 58 .50 fIN = 10MHz .40 fIN = 20MHz 57 .30 –50 –25 0 25 50 75 100 –50 –25 0 Temperature (°C) 25 50 75 100 Temperature (°C) POWER DISSIPATION vs TEMPERATURE OUTPUT NOISE (DC Input) 205 800k Counts Power (mW) 600k 200 400k 195 200k 0 190 –50 –25 0 25 50 75 N-2 100 N-1 N N+1 N+2 Code Temperature (°C) ® 7 ADS822, ADS825 APPLICATION INFORMATION The selection for the optimum interface configuration will depend on the individual application requirements and system structure. For example, communications applications often process a band of frequencies that do not include DC, whereas in imaging applications, the previously restored DC level must be maintained correctly up to the A/D converter. Features on the ADS822 and ADS825, such as the input range select (RSEL pin) or the option for an external reference, provide the needed flexibility to accommodate a wide range of applications. In any case, the ADS822 and ADS825 should be configured such that the application objectives are met while observing the headroom requirements of the driving amplifier in order to yield the best overall performance. THEORY OF OPERATION The ADS822 and ADS825 are high-speed CMOS analog-todigital converters which employ a pipelined converter architecture consisting of 9 internal stages. Each stage feeds its data into the digital error correction logic ensuring excellent differential linearity and no missing codes at the 10-bit level. The output data becomes valid on the rising clock edge (see Timing Diagram). The pipeline architecture results in a data latency of 5 clock cycles. The analog inputs of the ADS822 and ADS825 are differential track-and-hold (see Figure 1). The differential topology, along with tightly matched capacitors, produce a high level of AC performance while sampling at very high rates. INPUT CONFIGURATIONS AC-Coupled, Single-Supply Interface Figure 2 shows the typical circuit for an AC-coupled analog input configuration of the ADS822 and ADS825 while all components are powered from a single +5V supply. With the RSEL pin connected high, the full-scale input range is set to 2Vp-p. In this configuration, the top and bottom references (REFT, REFB) provide an output voltage of +3.5V and +1.5V, respectively. Two resistors ( 2x 1.62kΩ) are used to create a common-mode voltage (VCM) of approximately +2.5V to bias the inputs of the driving amplifier A1. Using the OPA680 on a single +5V supply, its ideal common-mode point is at +2.5V which coincides with the recommended common-mode input level for the ADS822 and ADS825. This obviates the need of a coupling capacitor between the amplifier and the converter. Even though the OPA680 has an AC gain of +2, the DC gain is only +1 due to the blocking capacitor at resistor RG. The addition of a small series resistor (RS) between the output of the op amp and the input of the ADS822 and ADS825 will be beneficial in almost all interface configurations. This will decouple the op amp’s output from the capacitive load and avoid gain peaking, which can result in increased noise. For best spurious and distortion performance, the resistor value should be kept below 100Ω. Furthermore, the series resistor in combination with the 10pF capacitor establishes a passive low-pass filter limiting the bandwidth for the wideband noise, thus helping improve the SNR performance. AC-Coupled, Dual Supply Interface The circuit provided in Figure 3 shows typical connections for the analog input in case the selected amplifier operates on dual supplies. This might be necessary to take full advantage of very low distortion operational amplifiers, like the OPA642. The advantage is that the driving amplifier can be operated with a ground referenced bipolar signal swing. This will keep the distortion performance at its lowest since the signal range stays within the linear region of the op amp and sufficient headroom to the supply rails can be maintained. By capacitively coupling the single-ended signal to the input of the ADS822 and ADS825, its common-mode requirements can easily be satisfied with two resistors connected between the top and bottom reference. The ADS822 and ADS825 allow their analog inputs to be driven either single-ended or differentially. The typical configuration for the ADS822 and ADS825 is the single-ended mode in which the input track-and-hold performs a singleended-to-differential conversion of the analog input signal. Both inputs (IN, IN) require external biasing using a common-mode voltage that is typically at the mid-supply level (+VS/2). The following application discussion focuses on the singleended configuration. Typically, its implementation is easier to achieve and the rated specifications for the ADS822 and ADS825 are characterized using the single-ended mode of operation. DRIVING THE ANALOG INPUT The ADS822 and ADS825 achieve excellent AC performance either in the single-ended or differential mode of operation. Op Amp Bias φ1 VCM φ1 CH φ2 CI IN IN φ1 φ2 OUT φ1 OUT φ1 CI φ2 CH φ1 φ1 Input Clock (50%) Op Amp Bias VCM Internal Non-overlapping Clock φ1 φ2 φ1 FIGURE 1. Simplified Circuit of Input Track-and-Hold with Timing Diagram. ® ADS822, ADS825 8 1.62kΩ +5V VCM +2.5V 1.62kΩ +5V 0.1µF 50Ω REFB +1.5V RS 50Ω VIN REFT +3.5V RSEL +VS IN OPA680 10pF +VIN 0V ADS822 ADS825 RF 402Ω –VIN CM IN RG 402Ω 0.1µF INT/EXT 0.1µF GND FIGURE 2. AC-Coupled Input Configuration for a 2Vp-p Full-Scale Range and a Common-Mode Voltage, VCM, at +2.5V Derived From the Internal Top (REFT) and Bottom Reference (REFB). +5V 1.62kΩ +5V RS 24.9Ω VIN REFT +3.5V 0.1µF RSEL +VS IN OPA642 100pF ADS822 ADS825 –5V RF 402Ω 1.62kΩ CM IN 0.1µF RG 402Ω REFB +1.5V INT/EXT GND FIGURE 3. AC-Coupling the Dual Supply Amplifier OPA642 to the ADS822 for a 2Vp-p Full-Scale Input Range. For applications requiring the driving amplifier to provide a signal amplification, with a gain ≥ 5, consider using decompensated voltage-feedback op amps, like the OPA643, or current-feedback op amps like the OPA681 and OPA658. input (IN) may be appropriately biased using the +2.5V common-mode voltage available at the CM pin. One half of amplifier A1 buffers the REFB pin and drives the voltage divider R1, R2. Due to the op amp’s noise gain of +2V/V, assuming RF = RIN , the common-mode voltage (VCM) has to be re-scaled to +1.25V. This results in the correct DC level of +2.5V for the signal input (IN). Any DC voltage differences between the IN and IN inputs of the ADS822 and ADS825 effectively produces an offset, which can be corrected for by adjusting the resistor values of the divider, R1and R2. The selection criteria for a suitable op amp should include the supply voltage, input bias current, output voltage swing, distortion, and noise specification. Note that in this example, the overall signal phase is inverted. To re-establish the original signal polarity, it is always possible to interchange the IN and IN connections. DC-coupled with Level Shift Several applications may require that the bandwidth of the signal path include DC, in which case, the signal has to be DC-coupled to the A/D converter. In order to accomplish this, the interface circuit has to provide a DC level shift to the analog input signal. The circuit shown in Figure 4 employs a dual op amp, A1, to drive the input of the ADS822 and ADS825, and level shifts the signal to be compatible with the selected input range. With the RSEL pin tied to the supply and the INT/EXT pin to ground, the ADS822 and ADS825 are configured for a 2Vp-p input range and use the internal references. The complementary ® 9 ADS822, ADS825 +5V RF 499Ω RIN 499Ω VIN RSEL RS 50Ω 1/2 OPA2681 +VS IN 2Vp-p 10pF ADS822 ADS825 NOTE: RF = RIN, G = –1 CM (+2.5) IN +5V 0.1µF REFB (+1.5V) REFT (+3.5V) INT/EXT 50Ω R2 200Ω 0.1µF 1/2 OPA2681 VCM = +1.25V 0.1µF R1 1kΩ RF 1kΩ FIGURE 4. DC-Coupled Interface Circuit with Dual Current-Feedback Amplifier OPA2681. SINGLE-ENDED-TO-DIFFERENTIAL CONFIGURATION (Transformer Coupled) If the application requires a signal conversion from a singleended source to feed the ADS822 and ADS825 differentially, a RF transformer might be a good solution. The selected transformer must have a center tap in order to apply the common-mode DC voltage necessary to bias the converter inputs. AC-grounding the center tap will generate the differential signal swing across the secondary winding. Consider a step-up transformer to take advantage of a signal amplification without the introduction of another noise source. Furthermore, the reduced signal swing from the source may lead to an improved distortion performance. circuit. The component values of the R-C low-pass may be optimized depending on the desired roll-off frequency. The resistor across the secondary side (RT) should be calculated using the equation RT = n2 • RG to match the source impedance (RG) for good power transfer and Voltage Standing Wave Ratio (VSWR). REFERENCE OPERATION Figure 6 depicts the simplified model of the internal reference circuit. The internal blocks are the bandgap voltage reference, the drivers for the top and bottom reference, and RSEL The differential input configuration may provide a noticeable advantage of achieving good SFDR performance over a wide range of input frequencies. In this mode, both inputs of the ADS822 and ADS825 see matched impedances, and the differential signal swing can be reduced to half of the swing required for single-ended drive. Figure 5 shows the schematic for the suggested transformer-coupled interface ADS822 50kΩ +VS INT/EXT 50kΩ Bandgap Reference and Logic VREF RG 0.1µF 1:n +1 22Ω VIN +1 IN 47pF ADS822 ADS825 RT 400Ω 400Ω 400Ω 400Ω 22Ω IN CM REFT RSEL INT/EXT ByT CM ByB REFB 47pF +5V + 10µF 0.1µF Bypass Capacitors: 0.1µF || 2.2µF each FIGURE 6. Equivalent Reference Circuit with Recommended Reference Bypassing. FIGURE 5. Transformer Coupled Input. ® ADS822, ADS825 10 The common-mode voltage available at the CM pin may be used as a bias voltage to provide the appropriate offset for the driving circuitry. However, care must be taken not to appreciably load this node, which is not buffered and has a high impedance. An alternative way of generating a common-mode voltage is given in Figure 7. Here, two external precision resistors (tolerance 1% or better) are located between the top and bottom reference pins. The commonmode voltage, CMV, will appear at the midpoint. the resistive reference ladder. The bandgap reference circuit includes logic functions that allows setting the analog input swing of the ADS822 and ADS825 to either a 1Vp-p or 2Vp-p full-scale range simply by tying the RSEL pin to a Low or High potential, respectively. While operating the ADS822 in the external reference mode, the buffer amplifiers for the REFT and REFB are disconnected from the reference ladder. As shown, the ADS822 and ADS825 have internal 50kΩ pull-up resistors at the range select pin (RSEL) and reference select pin (INT/EXT). Leaving these pins open configures the ADS822 and ADS825 for a 2Vp-p input range and external reference operation. Setting the ADS822 and ADS825 up for internal reference mode requires bringing the INT/EXT pin Low. The reference buffers can be utilized to supply up to 1mA (sink and source) to external circuitry. The resistor ladders of the ADS822 and ADS825 are divided into several segments and have two additional nodes, ByT and ByB, which are brought out for external bypassing only (see Figure 6). To ensure proper operation with any reference configurations, it is necessary to provide solid bypassing at all reference pins in order to keep the clock feedthrough to a minimum. All bypassing capacitors should be located as close to their respective pins as possible. EXTERNAL REFERENCE OPERATION For even more design flexibility, the internal reference can be disabled and an external reference voltage be used. The utilization of an external reference may be considered for applications requiring higher accuracy, improved temperature performance, or a wide adjustment range of the converter’s full-scale range. Especially in multichannel applications, the use of a common external reference has the benefit of obtaining better matching of the full-scale range between converters. The external references can vary as long as the value of the external top reference REFTEXT stays within the range of (VS – 1.25V) and (REFB + 0.8V), and the external bottom reference REFBEXT stays within 1.25V and (REFT – 0.8V) (See Figure 8). DIGITAL INPUTS AND OUTPUTS ADS822 ADS825 REFT +3.5V R1 1.6kΩ 0.1µF Clock Input Requirements REFB +1.5V Clock jitter is critical to the SNR performance of high-speed, high-resolution A/D converters. Clock jitter leads to aperture jitter (tA), which adds noise to the signal being converted. The ADS822 and ADS825 samples the input signal on the rising edge of the CLK input. Therefore, this edge should have the lowest possible jitter. The jitter noise contribution to total R2 1.6kΩ 0.1µF CMV +2.5V FIGURE 7. Alternative Circuit to Generate CM Voltage. +5V B A - Short for 1Vp-p Input Range B - Short for 2Vp-p Input Range (Default) +VS VIN A RSEL INT/EXT GND IN ADS822 ADS825 CMV +2.5VDC IN REFT External Top Reference REFT = REFB +0.8V to +3.75V ByT GND 4 x 0.1µF || 2.2µF ByB REFB External Bottom Reference REFB = REFT –0.8V to +1.25V FIGURE 8. Configuration Example for External Reference Operation. ® 11 ADS822, ADS825 Digital Output Driver (VDRV) The ADS822 features a dedicated supply pin for the output logic drivers, VDRV, which is not internally connected to the other supply pins. Setting the voltage at VDRV to +5V or +3V, the ADS822 and ADS825 produce corresponding logic levels and can directly interface to the selected logic family. The output stages are designed to supply sufficient current to drive a variety of logic families. However, it is recommended to use the ADS822 and ADS825 with +3V logic supply. This will lower the power dissipation in the output stages due to the lower output swing and reduce current glitches on the supply line which may affect the ACperformance of the converter. In some applications, it might be advantageous to decouple the VDRV pin with additional capacitors or a pi-filter. SNR is given by the following equation. If this value is near your system requirements, input clock jitter must be reduced. Jitter SNR = 20 log 1 rms signal to rms noise 2 π ƒ IN t A where: ƒIN is input signal frequency tA is rms clock jitter Particularly in undersampling applications, special consideration should be given to clock jitter. The clock input should be treated as an analog input in order to achieve the highest level of performance. Any overshoot or undershoot of the clock signal may cause degradation of the performance. When digitizing at high sampling rates, the clock should have 50% duty cycle (tH = tL), along with fast rise and fall times of 2ns or less. The clock input of the ADS825 can be driven with either 3V or 5V logic levels. Using low-voltage logic (3V) may lead to improved AC performance of the converter. GROUNDING AND DECOUPLING Proper grounding and bypassing, short lead length, and the use of ground planes are particularly important for high frequency designs. Multilayer PC boards are recommended for best performance since they offer distinct advantages like minimizing ground impedance, separation of signal layers by ground layers, etc. The ADS822 and ADS825 should be treated as analog components. Whenever possible, the supply pins should be powered by the analog supply. This will ensure the most consistent results since digital supply lines often carry high levels of noise which otherwise would be coupled into the converter and degrade the achievable performance. All ground connections on the ADS822 and ADS825 are internally joined together obviating the design of split ground planes. The ground pins (1, 16, 26) should directly connect to an analog ground plane which covers the PC board area around the converter. While designing the layout, it is important to keep the analog signal traces separated from any digital lines to prevent noise coupling onto the analog signal path. Due to their high sampling rates, the ADS822 and ADS825 generate high frequency current transients, and noise (clock feedthrough) that are fed back into the supply and reference lines. This requires that all supply and reference pins are sufficiently bypassed. Figure 9 shows the recommended decoupling scheme for the ADS822 and ADS825. In most cases, 0.1µF ceramic chip capacitors at each pin are adequate to keep the impedance low over a wide frequency range. Their effectiveness largely depends on the proximity to the individual supply pin. Therefore, they should be located as close to the supply pins as possible. In addition, a larger bipolar capacitor (1µF to 22µF) should be placed on the PC board in proximity of the converter circuit. Digital Outputs The output data format of the ADS822 and ADS825 are in positive Straight Offset Binary code (see Tables I and II). This format can easily be converted into the Binary Two’s Complement code by inverting the MSB. It is recommended to keep the capacitive loading on the data lines as low as possible (≤ 15pF). Higher capacitive loading will cause larger dynamic currents as the digital outputs are changing. Those high current surges can feed back to the analog portion of the ADS822 and ADS825 and affect the performance. If necessary, external buffers or latches close to the converter’s output pins may be used to minimize the capacitive loading. They also provide the added benefit of isolating the ADS822 and ADS825 from any digital noise activities on the bus coupling back high frequency noise. SINGLE-ENDED INPUT (IN = CMV) +FS –1LSB (IN = REFT) +1/2 Full Scale Bipolar Zero (IN = CMV) –1/2 Full Scale –FS (IN = REFB) STRAIGHT OFFSET BINARY (SOB) 11 11 10 01 00 1111 0000 0000 0000 0000 1111 0000 0000 0000 0000 TABLE I. Coding Table for Single-Ended Input Configuration with IN Tied to the Common-Mode Voltage (CMV). DIFFERENTIAL INPUT +FS –1LSB (IN = +3V, IN = +2V) +1/2 Full Scale Bipolar Zero (IN = IN = CMV) –1/2 Full Scale –FS (IN = +2V, IN = +3V) STRAIGHT OFFSET BINARY (SOB) 11 11 10 01 00 1111 0000 0000 0000 0000 1111 0000 0000 0000 0000 +VS 27 GND 26 ADS822 ADS825 +VS 15 0.1µF 0.1µF TABLE II. Coding Table for Differential Input Configuration and 2Vp-p Full-Scale Range. GND 16 VDRV 28 0.1µF 10µF + +5V +3/+5V Figure 9. Recommended Bypassing for the Supply Pins. ® ADS822, ADS825 12