CX74017
On the Direct Conversion Receiver
Abstract
Increased pressure for low power, small form factor, low cost,
and reduced bill of materials in such radio applications as mobile
communications has driven academia and industry to resurrect
the Direct Conversion Receiver (DCR). Long abandoned in favor
of the mature superheterodyne receiver, direct conversion has
emerged over the last decade or so thanks to improved
semiconductor process technologies and astute design
techniques. This paper describes the characteristics of the DCR
and the issues it raises.
Introduction
Very much like its well-established superheterodyne receiver
counterpart, introduced in 1918 by Armstrong [1], the origins of
the DCR date back to the first half of last century when a single
down-conversion receiver was first described by F.M. Colebrook
in 1924 [2], and the term homodyne was applied. Additional
developments led to the publication in 1947 of an article by
D. G. Tucker [3], which first coined the term synchrodyne, in a
receiver, which was designed as a precision demodulator for
measurement equipment rather than a radio. Another paper by
the latter in 1954 [4] reports the various single down-conversion
receivers published at the time, and clarifies the difference
between the homodyne (sometimes referred to as coherent
detector) and the synchrodyne receivers: the former obtains the
Local Oscillator (LO) directly, for example, from the transmitter,
whereas the latter synchronizes a free-running LO to the
incoming carrier.
brief description of alternative and well-established receiver
architectures, this paper presents the direct conversion
reception technique and highlights some of the system-level
issues associated with DCR.
Traditional Reception Techniques
The Superheterodyne
The superheterodyne, or more generally heterodyne1, receiver
is the most widely used reception technique. This technique
finds numerous applications from personal communication
devices to radio and TV tuners, and has been tried inside out
and is therefore well understood. It comes in a variety of
combinations [7-9], but essentially relies on the same idea: the
RF signal is first amplified in a frequency selective low-noise
stage, then translated to a lower intermediate frequency (IF),
with significant amplification and additional filtering, and finally
downconverted to baseband either with a phase discriminatory
or straight mixer, depending on the modulation format. This is
illustrated in the generic line-up of Figure 1.
RF band-select
filter
RF
Overcoming some of the problems associated with the
traditional superheterodyne and being more prone to integration,
DCR has nevertheless an array of inherent challenges. After a
Application Note
Channel-select
filter
IF
LNA
RF
Over the last decade or so, the drive of the wireless market and
enabling monolithic integration technology have triggered
research activities on DCRs, which integrated with the remaining
analog and digital sections of the transceiver, has the potential
to reach the “one-chip radio”. Besides, it favors multi-mode,
multi-standard applications and constitutes thereby another step
towards software radio.
The present article often refers to several recent publications
[5-6], providing a thorough survey and insight, and displaying
the renewed interest for DCRs.
Image-reject
filter
IF
f
f
101735A 1_071801
Figure 1. The Superheterodyne Receiver
Superheterodyning entails several trade-offs. Image rejection is
a prevailing concern in this architecture. During the first
Homo: Greek from “homos” - same; Hetero: Greek from
“heteros” – other; Synchro: Greek from “sunkhronos” – same
time; Dyne: Greek from “dunamis” – power.
1
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101735A
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CX74017
On the Direct Conversion Receiver
downconversion to IF, any unwanted activity at a frequency
spaced at fIF offset from fLO, on the opposite side of fLO from the
desired RF channel, produces a mixing product falling right into
the downconverted channel, at fIF. In practice, a RF bandpass
filter, usually a Surface Acoustic Wave (SAW) device, is utilized
to perform band selection ahead of the Low Noise Amplifier
(LNA), while a second filter follows the LNA to perform image
rejection. If these filters are identical, then, in fact, they share
the burden of the two functions. But some amount of image
rejection must particularly follow the LNA, for without it, the LNA
noise figure effectively doubles due to the mixing of amplified
image noise into the IF channel. Instead of the RF SAW filter,
other passive filtering technologies, such as dielectric or
ceramic, can also be featured. It can be seen from Figure 2 that
the higher the IF, the more relaxed the requirements on the
image reject filter cut-off frequency. Once at the IF, the
presence of an interfering signal in the vicinity of the channel
mandates sharp filtering around the channel; this is performed
after the first mixer by the channel select filter, which is also
often an IF SAW filter. Essentially, the exercise is that of a
carefully engineered balance among several variables:
•
•
•
Rejection provided by the various filters
Frequency planning
Linearity of the active stages
Dual IFs provide additional room to maneuver with filter
selectivity, but complicate the frequency planning somewhat.
The selectivity required of the two aforementioned filters, in
terms of fractional bandwidth, makes them unsuitable
candidates in the foreseeable future for integration. This is
because of low Qs of current silicon processes and the need to
be implemented by bulky, off-chip components. The IF channel
filter especially requires high-Q resonators for its
implementation: the higher the IF, the lesser the filter’s fractional
bandwidth, that is, its ratio of bandwidth to center frequency,
Channel
necessitating ever-higher Q. This high-Q requirement is most
commonly met by the use of a piezoelectric SAW and crystal
filters. This introduces additional constraints, as those filters
require often-inconvenient terminating impedances, and
matching may impinge on such issues as noise, gain, linearity,
and power dissipation of the adjoining active stages. The
narrower the fractional bandwidth, the more likely that the filter’s
passband shape will exhibit an extreme sensitivity to variations
in matching element values. Additionally, the specificity of the IF
filter to the signal bandwidth and hence the standard used,
makes superheterodyne receivers unsuitable for multi-standard
operation.
Nonetheless, superheterodyne is praised for its high selectivity
and sensitivity.
Image-Reject Receivers
Alternatively, by a smart use of trigonometric identities, the
image can be removed without the need of any post-LNA
image-reject filtering. This is the principle of image-reject
receivers [8] and [10]. The first is the Hartley architecture,
introduced in [11] in 1928, and shown in Figure 3. It uses two
mixers with their local oscillators in a quadrature phase
relationship. This separates the IF signal into in-phase (I) and
quadrature (Q) components. It then shifts the Q component by
90° before recombining the two paths. This is where the desired
signal, present in both paths with identical polarities, is
reinforced, while the image, present in both paths with opposite
polarities, is cancelled out. The dual of the Hartley architecture,
known as the Weaver image-reject receiver [12], achieves the
relative phase shift of one path by 90º by the use of a second
LO enroute to another IF or to baseband, see Figure 4. The
same result is achieved.
However, the reliability of these receivers depends heavily on
the accuracy of the I/Q paths, that is, the gain and phase
imbalance between the two branches.
Channel
Image
Interferer
Interferer
Channelselect BPF
Imagereject BPF
fRF
finter
fIF
fLO
fim
fIF
fIF
101735A 2_071901
Figure 2. Image-Rejection and Selectivity in a Superheterodyne Receiver (High-Side LO Injection)
2
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July 20, 2001
On the Direct Conversion Receiver
-fIF
CX74017
0
+fIF
RF cos(ωLOt)
fIF
Σ
sin(ωLOt)
-fIF
0
+fIF
90°
-fLO
0
+fLO
j
-fIF
-fIF
0
0 fIF
fIF
101735A 3_071801
IF bandpass filtering is large, making it possible to implement it
with low-Q components. The IF SAW, or crystal filter needed in
the high IF case, can be replaced with an active resistive
capacitor (RC) filter or other filter suitable for low frequency
operation that is also conducive to silicon integration. The low IF
signal may be translated to baseband through another mixer, or
preferably, in the digital domain following analog-to-digital (A/D)
conversion. Of course, this comes at the expense of faster and
higher resolution analog-to-digital converters (ADCs). If the IF
frequency is equal to only one or two channel widths, then it is
not possible to provide image-rejection at radio frequency (RF),
as the RF filter must be wide enough to pass all channels of the
system. In this case, all image rejection must come from the
quadrature downconversion to the low IF, which itself resembles
the Hartley architecture, once the baseband conversion is
added.
Figure 3. Hartley Image-Reject Architecture
BPF
Image-reject
filter
-f2
0
+f2
RF
0
RF cos(ω1t)
f2
-f1
0
+
cos(ω2t)
sin(ω1t)
Σ
sin(ω2t)
0
BPF
RF
-
+f1
LNA
Low IF
2×
Low IF
0
j
-f2
0
0 0
f2
f
101735A 4_071801
Figure 4. Weaver Image-Reject Architecture
Low IF Single Conversion Receiver
f
Figure 5. Low IF Single-Conversion Receiver
Wideband IF with double conversion
Low IF single conversion, see Figure 5, is an offspring of the
DCR, and is covered in the following paragraphs. Its main
purpose is to protect the receiver from all the direct current
(DC)-related obstacles that pertain to DCR, while retaining the
DCR’s benefit of the elimination of high-Q IF filters. As its name
indicates, instead of directly converting the signal to baseband,
the LO is slightly offset from the RF carrier, typically one to two
channels. The low IF means that the fractional bandwidth of the
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0
101735A 5_072001
This architecture, shown in Figure 6, is very similar to the
superheterodyne. In this case, the first mixer utilizes an LO that
is at a fixed frequency, and all RF band channels are translated
to IF, retaining their positions relative to one another. The
second mixer utilizes a tunable LO, thus selecting the desired
channel to be translated to baseband. A subsequent lowpass
filter suppresses adjacent channels.
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CX74017
On the Direct Conversion Receiver
RF band-select
filter
RF
LPF
Image-reject
filter
Wideband filter
IF
LNA
LPF
RF
Wideband IF
f
0
DC
f
0
f
101735A 6_071801
Figure 6. Wideband IF with Double Conversion.
Direct Conversion Receivers
Direct conversion reception, also referred to as homodyne, or
zero-IF, shown in Figure 7, is the most natural solution to
receiving information vehicled by a carrier. However, it has only
been a decade or so that this type of reception has found
applications other than pagers, for example [13]. For it has
several qualities which make it very suitable for integration as
well as multi-band, multi-standard operation, but severe inherent
obstacles that had for long kept it in the shadow of the
superheterodyne.
sideband signals, is the channel itself. Then, only one LO is
needed, which means only one phase noise contribution. The
need for the bulky, off-chip filters is consequently removed.
Filtering now only occurs at low, that is, baseband, frequencies
with some amplification. This means less current consumption
than at higher frequencies (to drive device parasitics), fewer
components, lower cost, etc. Practically, however, strong out-ofband interferers or blockers may need to be removed prior to
downconversion in order to avoid desensitizing the receiver by
saturating subsequent stages, as well as producing harmonics
and intermodulation terms, which then appear in baseband.
Such a filter may be placed after the LNA, for example.
DCR, however, brings its own set of issues. The following
paragraphs describe those in more detail.
LPF
RF band-select
filter
RF
DC offsets
LNA
RF
In direct conversion, as the signal of interest is converted to
baseband very early in the receive chain, without any filtering
other than RF band-selection, various phenomena contribute to
the creation of DC signals, which are directly appearing as
interfering signals in the band of interest.
DC
LPF
DC
f
0
f
101735A 7_071801
Figure 7. The Direct Conversion Receiver
First, the image problem is no longer present, since the IF is
zero and the image to the desired channel, for all but single4
The LO may be conducted or radiated through an unintended
path to the mixer’s RF input port, thus effectively mixing with
itself, producing an unwanted DC component at the mixer
output, see Figure 8.
Worse still, this LO leakage may reach the LNA input, producing
an even stronger result. This effect presents a high barrier
against the integration of LO, mixer, and LNA on a single silicon
substrate, where numerous mechanisms can contribute to poor
isolation. These include substrate coupling, ground bounce,
bond wire radiation, and capacitive and magnetic coupling.
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101735A
July 20, 2001
On the Direct Conversion Receiver
CX74017
Conversely, a strong in-band interferer, once amplified by the
LNA, may find a path to the LO input port of the mixer, see
Figure 9, thus producing self-mixing, once again.
R
e
r
a
d
i
a
t
e
Reflected,
Refracted
d
LO leakage
101735A 8_071801
Figure 8. LO Leakage
101735A 10_071801
Figure 10. Emission, Reflection, and Refraction
of LO Signal via Antenna
Whether or not the DC product desensitizes the receiver
depends on the system type. Obviously, it is preferable to
alternate current (AC)-couple at the mixer output to eliminate the
DC. Some modulation schemes used in paging applications,
such as Frequency-Shift Keying (FSK), show little degradation if
low frequency spectrum components are filtered out, see
Figure 11.
Strong
Interferer
101735A 9_071801
High pass
Figure 9. Strong In-Band Interferer, Amplified by the LNA
Some amount of LO power is conducted through the mixer and
LNA, due to their non-ideal reverse isolation, to the antenna.
The radiated power, appearing as an interferer to other
receivers in the corresponding band, may violate emissions
standards of the given system.
It is important to note that since the LO frequency is inside the
receive band, the front-end filters do nothing to suppress this LO
emission. Additionally, the radiated LO signal can then be
reflected by buildings or moving objects and re-captured by the
antenna, as shown in Figure 10. This effect, however, is not of
significant importance compared to the previously mentioned LO
self-mixing and blocker self-mixing.
LO or RF signal leakage to the opposite mixer port is not the
only way in which unwanted DC can be produced. Any stage
that exhibits even-order non-linearity also generates a DC
output. This is covered in more detail later.
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0f
c1
fc2
f
101735A 11_071801
Figure 11. High-Pass Filtering of the Modulation Spectrum
However, other modulation schemes present a peak at DC, and
capacitive AC-coupling will infer significant information loss,
hence considerably degrading the BER. In TDMA systems such
as GSM, although there is no significant low frequency spectral
peak, it still becomes impossible to AC couple. This is because
of the conflicting requirements on an AC-coupling capacitor in a
Time-Division Multiple Access (TDMA) system: the capacitor
must be large enough to avoid causing a wide notch at DC, but
it must be small enough that all transients settle out upon
receiver power-up (every frame) before data reception begins.
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CX74017
On the Direct Conversion Receiver
In TDMA receivers that cannot be AC coupled, the idle time slot,
that is, just before reception, can still be put to good use by
storing the value of the offset on a capacitor and then
subtracting it from the signal path during the burst. This is
exactly the same method which is normally used to correct DC
offsets occurring at the second mix of superheterodyne TDMA
receivers, where this mix goes to baseband. In that case the
only problem causing DC is LO self-mixing. In this method, the
DC value produced by the receiver is obtained in a premeasurement prior to the receive burst.
It is important when using this method, that the signal path prior
to the mixer be opened during the DC pre-measurement, to
prevent any large blocking signals from affecting the result.
Blocking signals, which can appear at any time, most often
induce variable or wandering offsets. The measurement-andsubtraction process cannot correct these offsets, because the
blocking signals may appear during the measurement and not
during the burst, or vice-versa. For blocking-induced DC, the
most effective measures are the elimination of self-mixing paths
and the maximizing of linearity to prevent the DC to start. Failing
these, there is still the possibility of DC-correction after-the-fact
in the digital signal processing (DSP) occurring at baseband.
DSP techniques can be used to remove the DC offset in TDMA
systems, in a way that cannot be duplicated in the analog
domain: a full timeslot of the received signal can be buffered,
the mean of which is determined and then removed from each
data point of the signal. The resulting signal has zero mean. For
systems such as Global System for Mobile communications®
(GSM®), an unwanted result of this is that any DC that is part of
the signal is lost, but the typical effect of this is minimal.
Figure 12 illustrates the use of such a method for a typical GSM
receiver. This technique can be further refined by tracking the
mean over portions of the burst, allowing the detection of
sudden interferers or blockers and canceling their DC product
only where it occurs.
Non-Linearities
As mentioned previously, another DCR problem is non-linearity.
Just as with the superheterodyne receiver, the DCR exhibits
spurious responses. For the superheterodyne, these occur at
RF input frequencies where
N (RF ) ± M (LO ) = IF ,
while for the DCR they occur where
N (RF ) − M (LO ) = IF
When a blocking signal’s carrier falls on one of these spurious
frequencies, the signal is translated to baseband with an
attendant shift in its bandwidth, dependent on the spurious
order.
However, more importantly, large blocking signals also cause
DC in the DCR, whether on a spurious frequency or not. The DC
is produced at the mixer output and amplified by the baseband
stages. It is due primarily to second order mixer non-linearity,
characterized by IP2 (second order intercept point), IM2 (second
order intermodulation.) It can be alleviated by extremely
well-balanced circuit design. However, only a short time ago, the
mixer and LNA used to require a single-ended design because
the antenna and a hypothetical preselect filter were usually
single-ended.
In most systems, third order intermodulation is important, as it
usually falls in-band, in the vicinity of the signals of interest, and
is characterized by IP3 (third order intercept point). In direct
conversion, the second order intermodulation becomes critical,
as it produces baseband signals, which now appear as
interfering signals in the down-converted desired signal. The
second order non-linearity is measured by the IP2. IP2 is
defined in the same manner as IP3 Figure 13.
Either a 2-tone, or 1-tone test can be performed, and the IP2 is
defined by extrapolating the low-frequency beat tone in the
former or the DC component in the latter, until it intercepts the
fundamental curve. To illustrate in the case of a single tone test,
the input signal is:
Careful layout can also improve isolation.
x(t ) = A cos(ωt )
Assuming a non-linearity modeled by a polynomial:
y ( x) = a1 x + a 2 x 2 + a 3 x 3 + ...
cos(2ωt ) + 1
y ( x ) = a1 A cos(ωt ) + a 2 A 2
+ ...
2
2
101735A 12_071801
Figure 12. BER Improvement with DSP-Based DC Offset
Cancellation
6
a2 A
a A2
= 2
+ a1 A cos(ωt ) +
cos(2ωt ) + ...
1
42
43
23
2
12
DC
fundamental
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101735A
July 20, 2001
On the Direct Conversion Receiver
CX74017
It can be seen from these equations and in Figure 13 that the
DC component due to the second order non-linearity is growing
with twice the slope of the fundamental on a logarithmic scale.
At the intercept point,
2a
a2 A2
= a1 A ⇔ A = 1 = IIP2
2
a2
complicates the use of MOS transistors for RF circuits, since the
main method of reducing it in MOS is to increase the transistor’s
size, which increases the device capacitance. This adversely
affects RF gain. For this reason, it is preferable to use bipolar
transistors for DCR mixer designs. In the first baseband stages
after the mixer, it becomes possible to use MOS devices, as the
transistor-size tradeoff is feasible at low frequencies.
Due to the doubled slope of the second-order product,
I/Q mismatches
IIP2 = Pin + ∆
Due to the high frequency of the LO, it is not possible to
implement the IQ demodulator digitally. An analog IQ
demodulator exhibits gain and phase imbalances between the
two branches, as well as the introduction of DC offsets. Such
imperfections distort the recovered constellation. Assuming
α ,ϕ being the amplitude and phase mismatch respectively
between the demodulator quadrature ports, and the complex
signal incident upon it having in-phase and quadrature
components I and Q:
with
∆ = Pout − IM2
Noise
Low frequency noise becomes a great concern in DCR [14], as
significant gain is allocated to baseband stages after the mixer.
Weak baseband signal levels of a few millivolts are still very
vulnerable to noise. This requires stronger RF stage gain to
alleviate the poor noise figure of baseband blocks, but of course
this must be traded against the linearity problems, just
described, that accompany higher RF gain.
Flicker noise, that is, 1/f noise, is the major baseband noise
contributor. Associated with DC flow, it has a spectral response
proportional to 1/f. In RF circuits, 1/f noise tends to be
modulated onto the RF signal. In the case of a mixer with
baseband output, 1/f noise sees especially high conversion gain.
In practice, flicker noise becomes an issue for Metal Oxide
Semiconductor (MOS) devices more than bipolar, and is
modeled as a voltage source in series with the gate. 1/f noise
IP3
Pout
OIP2
Iout = (I cos(ωt ) + Q sin(ωt ) ) ⋅ 2 cos(ωt )
Qout = (I cos(ωt ) + Q sin(ωt ) ) ⋅ 2 (1 + α ) sin(ωt + ϕ )
Filtering out the high frequency terms:
Iout = I
Qout = (1 + α ) (− I sin(ϕ ) + Q cos(ϕ ) )
Pout
OIP2
IP2
IP2
Pout
1
fu
1
1
1
r)
nd
l
ta
ord
e
∆
3
en
am
2 nd
2
3 rd o
rder
2 nd
ord
e
r)
al
1
IIP2
2
DC
(
1
t
en
DC
(
n
fu
m
da
Pin
1
Pin
∆
IIP2
Pin
+49dBm
101735A 13_071801
Figure 13. Second Order Intercept Point (IP2)
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CX74017
On the Direct Conversion Receiver
Figure 14 and Figure 15 sketch how this affects a given
constellation diagram. In DCR systems however, the IQ
matching is not as critical as in image-rejection architectures.
Rather, it is only important insofar as the accuracy of the
modulation is concerned.
Analog and digital (DSP based) calibration and adaptation
methods have been described so as to correct for these
imbalances, for example in [15].
Q
gain
imbalance
References
[1]
L. Lessing, “Man of High Fidelity: Edwin Howard
Armstrong, a Biography”, Bantam Books, New York,
1969.
[2]
F.M. Colebrook, “Homodyne”, Wireless World and Radio
Rev., 13, p. 774, 1924.
[3]
D.G. Tucker, “The Synchrodyne”, Electronic Engng, 19, p.
75-76, March 1947.
[4]
D.G. Tucker, “The History of the Homodyne and the
Synchrodyne”, Journal of the British Institution of Radio
Engineers, April 1954.
[5]
A.A. Abidi, “Direct-Conversion Radio transceivers for
Digital Communications”, IEEE Journal of Solid-State
Circuits, vol. 30, no. 12, December 1995.
[6]
B. Razavi, “Design Considerations for Direct-Conversion
Receivers”, IEEE Transactions on Circuits and Systems-II:
Analog and Digital Signal Processing, vol. 44, no. 6, June
1997.
[7]
S.J. Franke, “ECE 353 – Radio Communication Circuits”,
Department of Electrical and Computer Engineering,
University of Illinois, Urbana, 1994.
[8]
B. Razavi, “RF Microelectronics”, Prentice Hall, Upper
Saddle River, NJ, 1998.
[9]
J.C. Rudell et al., “Recent Developments in High
Integration Multi-Standard CMOS Transceivers for
Personal Communication Systems”, International
Symposium on Low Power Electronics and Design, 1998.
ideal
constellation
I
101735A 14_071801
Figure 14. IQ Demodulator Imperfections (Gain Imbalance)
Q
phase
imbalance
[10] J.C. Rudell, “Issues in RFIC Design”, lecture notes,
University of California Berkeley/National Technological
University, 1997.
I
[11] R. Hartley, “Single-Sideband Modulator”, U.S. Patent no.
1666206, April 1928.
101735A 15_071801
Figure 15. IQ Demodulator Imperfections (Phase Imbalance)
Conclusion
The direct conversion receiver is an attractive yet challenging
receiving technique. It has recently been applied successfully to
devices such as pagers, mobile phones, PC and internet
wireless connectivity cards, satellite receivers etc. in a variety of
process technologies and increasing integration levels. It is
poised to appear in many more applications in the near future.
8
[12] D.K. Weaver, “A Third Method of Generation and
Detection of Single Sideband Signals”, Proceedings of the
IRE, vol. 44, p. 1703-1705, December 1956.
[13] I.A.W. Vance, “Fully Integrated Radio Paging Receiver”,
IEE Proc., vol. 129, no. 1, p. 2-6, 1982.
[14] P.R. Gray, R.G. Meyer, “Analysis and Design of Analog
Integrated Circuits”, Third edition, John Wiley & Sons,
New York, 1993.
[15] J.K. Cavers, M. W. Liao, “Adaptive Compensation for
Imbalance and Offset Losses in Direct Conversion
Transceivers”, IEEE Transactions on Vehicular
Technology, vol. 42, p. 581-588, November 1993.
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The following are trademarks of Skyworks Solutions, Inc.: Skyworks™, the Skyworks symbol, and “Breakthrough Simplicity”™. Product names or services listed in this
publication are for identification purposes only, and may be trademarks of third parties. Third-party brands and names are the property of their respective owners.
Additional information, posted at www.skyworksinc.com, is incorporated by reference.
General Information:
Skyworks Solutions, Inc.
4311 Jamboree Rd.
Newport Beach, CA 92660-3007
www.skyworksinc.com