LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 LM2745 / LM2748 Synchronous Buck Controller with Pre-Bias Startup, and Optional Clock Synchronization Check for Samples: LM2745, LM2748 FEATURES DESCRIPTION • • The LM2745/8 are high-speed synchronous buck regulator controllers with an accurate feedback voltage accuracy of ±1.5%. It can provide simple down conversion to output voltages as low as 0.6V. Though the control sections of the IC are rated for 3 to 6V, the driver sections are designed to accept input supply rails as high as 14V. The use of adaptive non-overlapping MOSFET gate drivers helps avoid potential shoot-through problems while maintaining high efficiency. The IC is designed for the more costeffective option of driving only N-channel MOSFETs in both the high-side and low-side positions. It senses the low-side switch voltage drop for providing a simple, adjustable current limit. 1 2 • • • • • • • • • • • Switching Frequency from 50 kHz to 1 MHz Switching Frequency Synchronize Range 250 kHz to 1 MHz (LM2745 Only) Startup with a Pre-biased Output Load Power Stage Input Voltage from 1V to 14V Control Stage Input Voltage from 3V to 6V Output Voltage Adjustable Down to 0.6V Power Good Flag and Shutdown Output Overvoltage and Undervoltage Detection ±1.5% Feedback Voltage Accuracy Over Temperature Low-side Adjustable Current Sensing Adjustable Soft-start Tracking and Sequencing with Shutdown and Soft Start Pins TSSOP-14 Package The LM2745/8 features a fixed-frequency voltagemode PWM control architecture which is adjustable from 50 kHz to 1 MHz with one external resistor. In addition, the LM2745 also allows the switching frequency to be synchronized to an external clock signal over the range of 250 kHz to 1 MHz. This wide range of switching frequency gives the power supply designer the flexibility to make better tradeoffs between component size, cost and efficiency. APPLICATIONS • • • • • Down Conversion from 3.3V Cable Modem, DSL and ADSL Laser Jet and Ink Jet Printers Low Voltage Power Modules DSP, ASIC, Core and I/O Features include the ability to startup with a prebiased load on the output, soft-start, input undervoltage lockout (UVLO) and Power Good (based on both undervoltage and overvoltage detection). In addition, the shutdown pin of the IC can be used for providing startup delay, and the soft-start pin can be used for implementing precise tracking, for the purpose of sequencing with respect to an external rail. Typical Application VCC = 3.3V RCC VIN = 3.3V D1 CBOOT RPULL-UP + Q1 VCC LM2745 SD CCC BOOT FREQ/SYNC CClk RFADJ CSS RCS L1 VOUT = 1.2V@16A ISEN PWGD Clk CIN1,2 HG LG SS/TRACK PGND SGND PGND EAO FB Q2 RFB2 RC2 + CO1,2 CC3 RFB1 CC1 CC2 RC1 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2005–2013, Texas Instruments Incorporated LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 1 2 3 4 5 6 7 BOOT LG HG LM2745 PGND SD PGND SGND FREQ/SYNC VCC PWGD ISEN FB SS/TRACK EAO www.ti.com 14 1 13 2 12 3 11 4 10 5 9 6 8 7 Figure 1. 14-Lead Plastic TSSOP, θJA = 155°C/W Top View Package Number PW0014A BOOT LG HG LM2748 PGND PGND SD SGND FREQ VCC PWGD ISEN FB SS/TRACK EAO 14 13 12 11 10 9 8 Figure 2. 14-Lead Plastic TSSOP, θJA = 155°C/W Top View Package Number PW0014A Pin Descriptions BOOT (Pin 1) - Bootstrap pin. This is the supply rail for the high-side gate driver. When the high-side MOSFET turns on, the voltage on this pin should be at least one gate threshold above the regulator input voltage VIN to properly turn on the MOSFET. See MOSFET Gate Drivers in the Application Information section for more details on how to select MOSFETs. LG (Pin 2) - Low-gate drive pin. This is the gate drive for the low-side N-channel MOSFET. This signal is interlocked with the high-side gate drive HG (Pin 14), so as to avoid shoot-through. PGND (Pins 3, 13) - Power ground. This is also the ground for the low-side MOSFET driver. Both the pins must be connected together on the PCB and form a ground plane, which is usually also the system ground. SGND (Pin 4) - Signal ground. It should be connected appropriately to the ground plane with due regard to good layout practices in switching power regulator circuits. VCC (Pin 5) Supply rail for the control sections of the IC. PWGD (Pin 6) - Power Good pin. This is an open drain output, which is typically meant to be connected to VCC or any other low voltage source through a pull-up resistor. Choose the pull-up resistor so that the current going into this pin is kept below 1 mA. A recommended value for the pull-up resistor is 100 kΩ for most applications. The voltage on this pin is thus pulled low under output undervoltage or overvoltage fault conditions and also under input UVLO. ISEN (Pin 7) - Current limit threshold setting pin. This sources a fixed 40 µA current. A resistor of appropriate value should be connected between this pin and the drain of the low-side MOSFET (switch node). The minimum value for this resistor is 1 kΩ. EAO (Pin 8) - Output of the error amplifier. The voltage level on this pin is compared with an internally generated ramp signal to determine the duty cycle. This pin is necessary for compensating the control loop. SS/TRACK (Pin 9) - Soft-start and tracking pin. This pin is internally connected to the non-inverting input of the error amplifier during softstart, and in fact any time the SS/TRACK pin voltage happens to be below the internal reference voltage. For the basic soft-start function, a capacitor of minimum value 1 nF is connected from this pin to ground. To track the rising ramp of another power supply’s output, connect a resistor divider from the output of that supply to this pin as described in Application Information. FB (Pin 10) - Feedback pin. This is the inverting input of the error amplifier, which is used for sensing the output voltage and compensating the control loop. FREQ/SYNC (Pin 11) - Frequency adjust pin. The switching frequency is set by connecting a resistor of suitable value between this pin and ground. Some typical values (rounded up to the nearest standard values) are 150 kΩ for 200 kHz, 100 kΩ for 300 kHz, 51.1 kΩ for 500 kHz, 18.7 kΩ for 1 MHz. This pin is also used in the LM2745 to synchronize to an external clock. SD (Pin 12) - IC shutdown pin. Pull this pin to VCC to ensure the IC is enabled. Connect to ground to disable the IC. Under shutdown, both high-side and low-side drives are off. This pin also features a precision threshold for power supply sequencing purposes, as well as a low threshold to ensure minimal quiescent current. HG (Pin 14) - High-gate drive pin. This is the gate drive for the high-side N-channel MOSFET. This signal is interlocked with LG (Pin 2) to avoid shoot-through. These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 Absolute Maximum Ratings (1) (2) VCC -0.3 to 7V BOOT Voltage -0.3 to 18V ISEN -0.3 to 14V FREQ/SYNC Voltage -0.5 to VCC + 0.3V All other pins -0.3 to VCC + 0.3V Junction Temperature 150°C Storage Temperature −65°C to 150°C Soldering Information ESD Rating (1) (2) (3) Lead Temperature (soldering, 10sec) 260°C Infrared or Convection (20sec) 235°C (3) 2kV Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating ratings indicate conditions for which the device operates correctly. Operating Ratings do not imply specified performance limits. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. ESD using the human body model which is a 100pF capacitor discharged through a 1.5 kΩ resistor into each pin. Operating Ratings Supply Voltage Range, VCC (1) 3V to 6V BOOT Voltage Range 1V to 17V −40°C to +125°C Junction Temperature Range (TJ) Thermal Resistance (θJA) (1) 155°C/W The power MOSFETs can run on a separate 1V to 14V rail (Input voltage, VIN). Practical lower limit of VIN depends on selection of the external MOSFET. See the MOSFET GATE DRIVERS section under Application Information for further details. Electrical Characteristics VCC = 3.3V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA= TJ= 25°C. Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or statistical analysis. Symbol Parameter Conditions VFB FB Pin Voltage VCC = 3V to 6V VON UVLO Thresholds VCC Rising VCC Falling IQ_VCC Operating VCC Current Min Typ Max Units 0.591 0.6 0.609 V 2.79 2.42 V VCC = 3.3V, VSD = 3.3V fSW = 600 kHz LM2745 1.1 1.7 2.3 LM2748 1.0 1.5 2.1 VCC = 5V, VSD = 3.3V fSW = 600 kHz LM2745 1.3 2 2.6 LM2748 1.0 1.8 2.6 Shutdown VCC Current VCC = 3.3V, VSD = 0V 1 3 tPWGD1 PWGD Pin Response Time VFB Rising 10 tPWGD2 PWGD Pin Response Time VFB Falling 10 ISS-ON SS Pin Source Current VSS = 0V ISS-OC SS Pin Sink Current During Over Current VSS = 2.0V ISEN-TH ISEN Pin Source Current Trip Point IFB FB Pin Current 7 40 µA µs µs 14 90 25 Sourcing 10 mA µA µA 55 µA 20 nA ERROR AMPLIFIER GBW Error Amplifier Unity Gain Bandwidth 9 MHz 118 dB Error Amplifier Slew Rate 2 V/µs EAO Pin Current Sourcing and Sinking Capability 14 16 mA G Error Amplifier DC Gain SR IEAO Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 3 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com Electrical Characteristics (continued) VCC = 3.3V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA= TJ= 25°C. Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or statistical analysis. Symbol VEAO Parameter Conditions Error Amplifier Output Voltage Min Typ Max Units Minimum 1 V Maximum 2.2 V GATE DRIVE IQ-BOOT BOOT Pin Quiescent Current VBOOT = 12V, VSD = 0 18 RHG_UP High-Side MOSFET Driver Pull-Up ON resistance 90 µA VBOOT = 5V @ 350 mA Sourcing 2.7 Ω RHG_DN High-Side MOSFET Driver Pull-Down 350 mA Sinking ON resistance 0.8 Ω RLG_UP Low-Side MOSFET Driver Pull-Up ON resistance VBOOT = 5V @ 350 mA Sourcing 2.7 Ω RLG_DN Low-Side MOSFET Driver Pull-Down ON resistance 350 mA Sinking 0.8 Ω OSCILLATOR PWM Frequency RFADJ = 750 kΩ 50 RFADJ = 100 kΩ 300 RFADJ = 42.2 kΩ fSW 475 RFADJ = 18.7 kΩ LM2745 External Synchronizing Signal Frequency Voltage Swing = 0V to VCC SYNCL LM2745 Synchronization Signal Low Threshold fSW = 250 kHz to 1 MHz SYNCH LM2745 Synchronization Signal High Threshold fSW = 250 kHz to 1 MHz DMAX Max High-Side Duty Cycle 600 725 kHz 1000 250 1000 1 2 fSW = 300 kHz fSW = 600 kHz fSW = 1 MHz V V 86 78 67 % LOGIC INPUTS AND OUTPUTS 4 VSTBY-IH Standby High Trip Point VFB = 0.575V, VBOOT = 3.3V VSD Rising VSTBY-IL Standby Low Trip Point VFB = 0.575V, VBOOT = 3.3V VSD Falling VSD-IH SD Pin Logic High Trip Point VSD Rising VSD-IL SD Pin Logic Low Trip Point VSD Falling 0.8 VPWGD-TH-LO PWGD Pin Trip Points VFB Falling 0.408 0.434 0.457 V VPWGD-TH-HI PWGD Pin Trip Points VFB Rising 0.677 0.710 0.742 V VPWGD-HYS PWGD Hysteresis VFB Falling VFB Rising Submit Documentation Feedback 1.1 0.232 V V 1.3 V V 60 90 mV Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 Typical Performance Characteristics Efficiency (VOUT = 1.2V) VCC = 3.3V, fSW = 1 MHz Internal Reference Voltage vs Temperature Figure 3. Figure 4. Frequency vs Temperature Output Voltage vs Output Current Figure 5. Figure 6. Switch Waveforms VCC = 3.3V, VIN = 5V, VOUT = 1.2V IOUT = 3A, CSS = 12 nF, fSW = 1 MHz Start-Up (Full-Load) VCC = 3.3V, VIN = 5V, VOUT = 1.2V IOUT = 3A, CSS = 12 nF, fSW = 1 MHz Figure 7. Figure 8. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 5 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com Typical Performance Characteristics (continued) 6 Start-Up (No-Load) VCC = 3.3V, VIN = 5V, VOUT = 1.2V CSS = 12 nF, fSW = 1 MHz Shutdown (Full-Load) VCC = 3.3V, VIN = 5V, VOUT = 1.2V IOUT = 3A, CSS = 12 nF, fSW = 1 MHz Figure 9. Figure 10. Load Transient Response VCC = 3.3V, VIN = 14V, VOUT = 1.2V fSW = 1 MHz Line Transient Response (VIN = 3V to 9V) VCC = 3.3V, VOUT = 1.2V IOUT = 2A, fSW = 1 MHz Figure 11. Figure 12. Frequency vs. Frequency Adjust Resistor Maximum Duty Cycle vs Frequency, VCC = 3.3V Figure 13. Figure 14. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 Typical Performance Characteristics (continued) Maximum Duty Cycle vs VCC, fSW = 600 kHz Maximum Duty Cycle vs VCC, fSW = 1 MHz Figure 15. Figure 16. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 7 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com Block Diagram FREQ/SYNC* VCC SD PGND PGND SGND UVLO PLL* CLOCK & RAMP SHUT DOWN LOGIC BOOT 10 Ps DELAY HG SSDONE PWGD SYNCHRONOUS DRIVER LOGIC OV UV LG 0.71V 0.434V 10 PA Zero Detector SS/TRACK 1VPP PWM LOGIC PWM Soft-Start Comparator + Logic REF 40 PA - + 90 PA ISEN ILIM EA - + VREF=0.6V *Sync function for LM2745 Only 8 FB EAO Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 APPLICATION INFORMATION The LM2745/8 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It is designed for use in set-top boxes, thin clients, DSL/Cable modems, and other applications that require high efficiency buck converters. It has output shutdown (SD), input undervoltage lock-out (UVLO) mode and power good (PWGD) flag (based on overvoltage and undervoltage detection). The overvoltage and undervoltage signals are OR-gated to drive the power good signal and provide a logic signal to the system if the output voltage goes out of regulation. Current limit is achieved by sensing the voltage VDS across the low side MOSFET. The LM2745/8 is also able to start-up with the output pre-biased with a load. The LM2745 also allows for the switching frequency to be synchronized with an external clock source. START UP/SOFT-START When VCC exceeds 2.79V and the shutdown pin (SD) sees a logic high, the soft-start period begins. Then an internal, fixed 10 µA source begins charging the soft-start capacitor. During soft-start the voltage on the soft-start capacitor CSS is connected internally to the non-inverting input of the error amplifier. The soft-start period lasts until the voltage on the soft-start capacitor exceeds the LM2745/8 reference voltage of 0.6V. At this point the reference voltage takes over at the non-inverting error amplifier input. The capacitance of CSS determines the length of the soft-start period, and can be approximated by: CSS = tSS 60 Where CSS is in µF and tSS is in ms. During soft start the Power Good flag is forced low and it is released when the FB pin voltage reaches 70% of 0.6V. At this point the chip enters normal operation mode, and the output overvoltage and undervoltage monitoring starts. SETTING THE OUTPUT VOLTAGE The LM2745/8 regulates the output voltage by controlling the duty cycle of the high side and low side MOSFETs (see Typical Application Circuit).The equation governing output voltage is: RFB1 + RFB2 VOUT = RFB1 VFB (VFB = 0.6V) SETTING THE SWITCHING FREQUENCY During fixed-frequency mode of operation the PWM frequency is adjustable between 50 kHz and 1 MHz and is set by an external resistor, RFADJ, between the FREQ/SYNC pin and ground. The resistance needed for a desired frequency is approximated by the curve FREQUENCY vs. FREQUENCY ADJUST RESISTOR in the Typical Performance Characteristics section. When it is desired to synchronize the switching frequency with an external clock source, the LM2745 has the unique ability to synchronize from this external source within the range of 250 kHz to 1 MHz. The external clock signal should be AC coupled to the FREQ/SYNC pin as shown below in Figure 17, where the RFADJ is chosen so that the fixed frequency is approximately within ±30% of the external synchronizing clock frequency. An internal protection diode clamps the low level of the synchronizing signal to approximately -0.5V. The internal clock sychronizes to the rising edge of the external clock. CCLK External Clock Signal To FREQ/SYNC Pin RFADJ Figure 17. AC Coupled Clock Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 9 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com It is recommended to choose an AC coupling capacitance in the range of 50 pF to 100 pF. Exceeding the recommended capacitance may inject excessive energy through the internal clamping diode structure present on the FREQ/SYNC pin. The typical trip level of the synchronization pin is 1.5V. To ensure proper synchronization and to avoid damaging the IC, the peak-to-peak value (amplitude) should be between 2.5V and VCC. The minimum width of this pulse must be greater than 100 ns, and it's maximum width must be 100ns less than the period of the switching cycle. The external clock synchronization process begins once the LM2745 is enabled and an external clock signal is detected. During the external clock synchronization process the internal clock initially switches at approximately 1.5 MHz and decreases until it has matched the external clock’s frequency. The lock-in period is approximately 30 µs if the external clock is switching at 1 MHz, and about 100 µs if the external clock is at 200 kHz. When there is no clock signal present, the LM2745 enters into fixed-frequency mode and begins switching at the frequency set by the RFADJ resistor. If the external clock signal is removed after frequency synchronization, the LM2745 will enter fixed-frequency mode within two clock cycles. If the external clock is removed within the 30 µs lock-in period, the LM2745 will re-enter fixed-frequency mode within two internal clock cycles after the lock-in period. OUTPUT PRE-BIAS STARTUP If there is a pre-biased load on the output of the LM2745/8 during startup, the IC will disable switching of the lowside MOSFET and monitor the SW node voltage during the off-time of the high-side MOSFET. There is no load current sensing while in pre-bias mode because the low-side MOSFET never turns on. The IC will remain in this pre-bias mode until it sees the SW node stays below 0V during the entire high-side MOSFET's off-time. Once it is determined that the SW node remained below 0V during the high-side off-time, the low-side MOSFET begins switching during the next switching cycle. Figure 18 shows the SW node, HG, and LG signals during pre-bias startup. The pre-biased output voltage should not exceed VCC + VGS of the external High-Side MOSFET to ensure that the High-Side MOSFET will be able to switch during startup. Figure 18. Output Pre-Bias Mode Waveforms TRACKING A VOLTAGE LEVEL The LM2745/8 can track the output of a master power supply during soft-start by connecting a resistor divider to the SS/TRACK pin. In this way, the output voltage slew rate of the LM2745/8 will be controlled by the master supply for loads that require precise sequencing. When the tracking function is used no soft-start capacitor should be connected to the SS/TRACK pin. However in all other cases, a CSS value of at least 1 nF between the soft-start pin and ground should be used. 10 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 Master Power Supply VOUT1 = 5V RT2 1 k: VOUT2 = 1.8V SS/TRACK VSS = 0.65V RT1 150: LM2745/8 FB RFB2 10 k: VFB RFB1 5 k: Figure 19. Tracking Circuit One way to use the tracking feature is to design the tracking resistor divider so that the master supply’s output voltage (VOUT1) and the LM2745/8’s output voltage (represented symbolically in Figure 19 as VOUT2, i.e. without explicitly showing the power components) both rise together and reach their target values at the same time. For this case, the equation governing the values of the tracking divider resistors RT1 and RT2 is: 0.65 = VOUT1 RT1 RT1 + RT2 The current through RT1 should be about 4 mA for precise tracking. The final voltage of the SS/TRACK pin should be set higher than the feedback voltage of 0.6V (say about 0.65V as in the above equation). If the master supply voltage was 5V and the LM2745/8 output voltage was 1.8V, for example, then the value of RT1 needed to give the two supplies identical soft-start times would be 150Ω. A timing diagram for the equal soft-start time case is shown in Figure 20. 5V VOUT1 1.8V VOUT2 Figure 20. Tracking with Equal Soft-Start Time TRACKING A VOLTAGE SLEW RATE The tracking feature can alternatively be used not to make both rails reach regulation at the same time but rather to have similar rise rates (in terms of output dV/dt). This method ensures that the output voltage of the LM2745/8 always reaches regulation before the output voltage of the master supply. In this case, the tracking resistors can be determined based on the following equation: 0.65 = VOUT2 RT1 RT1 + RT2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 11 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com For the example case of VOUT1 = 5V and VOUT2 = 1.8V, with RT1 set to 150Ω as before, RT2 is calculated from the above equation to be 265Ω. A timing diagram for the case of equal slew rates is shown in Figure 21. 5V 1.8V VOUT1 1.8V VOUT2 Figure 21. Tracking with Equal Slew Rates SEQUENCING The start up/soft-start of the LM2745/8 can be delayed for the purpose of sequencing by connecting a resistor divider from the output of a master power supply to the SD pin, as shown in Figure 22. Master Power Supply VOUT1 VOUT2 RS2 SD RS1 LM2745/8 FB RFB2 VFB RFB1 Figure 22. Sequencing Circuit A desired delay time tDELAY between the startup of the master supply output voltage and the LM2745/8 output voltage can be set based on the SD pin low-to-high threshold VSD-IH and the slew rate of the voltage at the SD pin, SRSD: tDELAY = VSD-IH / SRSD Note again, that in Figure 22, the LM2745/8’s output voltage has been represented symbolically as VOUT2, i.e. without explicitly showing the power components. VSD-IH is typically 1.08V and SRSD is the slew rate of the SD pin voltage. The values of the sequencing divider resistors RS1 and RS2 set the SRSD based on the master supply output voltage slew rate, SROUT1, using the following equation: SRSD = SROUT1 12 RS1 RS1 + RS2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 For example, if the master supply output voltage slew rate was 1V/ms and the desired delay time between the startup of the master supply and LM2745/8 output voltage was 5 ms, then the desired SD pin slew rate would be (1.08V/5 ms) = 0.216V/ms. Due to the internal impedance of the SD pin, the maximum recommended value for RS2 is 1 kΩ. To achieve the desired slew rate, RS1 would then be 274Ω. A timing diagram for this example is shown in Figure 23. 5V VSD-IH 1.08V VOUT1 1.8V VOUT2 t = 5 ms Figure 23. Delay for Sequencing SD PIN IMPEDANCE When connecting a resistor divider to the SD pin of the LM2745/8 some care has to be taken. Once the SD voltage goes above VSD-IH, a 17 µA pull-up current is activated as shown in Figure 24. This current is used to create the internal hysteresis (≊170 mV); however, high external impedances will affect the SD pin logic thresholds as well. The external impedance used for the sequencing divider network should preferably be a small fraction of the impedance of the SD pin for good performance (around 1 kΩ). 17 PA 8 PA Bias Enable SD 10k Soft-Start Enable 1.25V + - Figure 24. SD Pin Logic Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 13 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com MOSFET GATE DRIVERS The LM2745/8 has two gate drivers designed for driving N-channel MOSFETs in a synchronous mode. Note that unlike most other synchronous controllers, the bootstrap capacitor of the LM2745/8 provides power not only to the driver of the upper MOSFET, but the lower MOSFET driver too (both drivers are ground referenced, i.e. no floating driver). Two things must be kept in mind here. First, the BOOT pin has an absolute maximum rating of 18V. This must never be exceeded, even momentarily. Since the bootstrap capacitor is connected to the SW node, the peak voltage impressed on the BOOT pin is the sum of the input voltage (VIN) plus the voltage across the bootstrap capacitor (ignoring any forward drop across the bootstrap diode). The bootstrap capacitor is charged up by a given rail (called VBOOT_DC here) whenever the upper MOSFET turns off. This rail can be the same as VCC or it can be any external ground-referenced DC rail. But care has to be exercised when choosing this bootstrap DC rail that the BOOT pin is not damaged. For example, if the desired maximum VIN is 14V, and VBOOT_DC is chosen to be the same as VCC, then clearly if the VCC rail is 6V, the peak voltage on the BOOT pin is 14V + 6V = 20V. This is unacceptable, as it is in excess of the rating of the BOOT pin. A VCC of 3V would be acceptable in this case. Or the VIN range must be reduced accordingly. There is also the option of deriving the bootstrap DC rail from another 3V external rail, independent of VCC. The second thing to be kept in mind here is that the output of the low-side driver swings between the bootstrap DC rail level of VBOOT_DC and Ground, whereas the output of the high-side driver swings between VIN+ VBOOT_DC and Ground. To keep the high-side MOSFET fully on when desired, the Gate pin voltage of the MOSFET must be higher than its instantaneous Source pin voltage by an amount equal to the 'Miller plateau'. It can be shown that this plateau is equal to the threshold voltage of the chosen MOSFET plus a small amount equal to Io/g. Here Io is the maximum load current of the application, and g is the transconductance of this MOSFET (typically about 100 for logic-level devices). That means we must choose VBOOT_DC to at least exceed the Miller plateau level. This may therefore affect the choice of the threshold voltage of the external MOSFETs, and that in turn may depend on the chosen VBOOT_DC rail. So far, in the discussion above, the forward drop across the bootstrap diode has been ignored. But since that does affect the output of the driver somewhat, it is a good idea to include this drop in the following examples. Looking at the Typical Application schematic, this means that the difference voltage VCC - VD1, which is the voltage the bootstrap capacitor charges up to, must always be greater than the maximum tolerance limit of the threshold voltage of the upper MOSFET. Here VD1 is the forward voltage drop across the bootstrap diode D1. This may place restrictions on the minimum input voltage and/or type of MOSFET used. A basic bootstrap circuit can be built using one Schottky diode and a small capacitor, as shown in Figure 25. The capacitor CBOOT serves to maintain enough voltage between the top MOSFET gate and source to control the device even when the top MOSFET is on and its source has risen up to the input voltage level. The charge pump circuitry is fed from VCC, which can operate over a range from 3.0V to 6.0V. Using this basic method the voltage applied to the gates of both high-side and low-side MOSFETs is VCC - VD. This method works well when VCC is 5V±10%, because the gate drives will get at least 4.0V of drive voltage during the worst case of VCC-MIN = 4.5V and VD-MAX = 0.5V. Logic level MOSFETs generally specify their on-resistance at VGS = 4.5V. When VCC = 3.3V±10%, the gate drive at worst case could go as low as 2.5V. Logic level MOSFETs are not specified to turn on, or may have much higher on-resistance at 2.5V. Sub-logic level MOSFETs, usually specified at VGS = 2.5V, will work, but are more expensive, and tend to have higher on-resistance. The circuit in Figure 25 works well for input voltages ranging from 1V up to 14V and VCC = 5V±10%, because the drive voltage depends only on VCC. 14 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 LM2745/8 BOOT D1 VCC CBOOT VIN HG + VO + LG Figure 25. Basic Charge Pump (Bootstrap) Note that the LM2745/8 can be paired with a low cost linear regulator like the LM78L05 to run from a single input rail between 6.0 and 14V. The 5V output of the linear regulator powers both the VCC and the bootstrap circuit, providing efficient drive for logic level MOSFETs. An example of this circuit is shown in Figure 26. LM2745/8 VCC 5V LM78L05 D1 BOOT CBOOT HG VIN + VO LG + Figure 26. LM78L05 Feeding Basic Charge Pump Figure 27 shows a second possibility for bootstrapping the MOSFET drives using a doubler. This circuit provides an equal voltage drive of VCC - 3VD + VIN to both the high-side and low-side MOSFET drives. This method should only be used in circuits that use 3.3V for both VCC and VIN. Even with VIN = VCC = 3.0V (10% lower tolerance on 3.3V) and VD = 0.5V both high-side and low-side gates will have at least 4.5V of drive. The power dissipation of the gate drive circuitry is directly proportional to gate drive voltage, hence the thermal limits of the LM2745/8 IC will quickly be reached if this circuit is used with VCC or VIN voltages over 5V. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 15 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com LM2745/8 BOOT D3 HG D2 D1 VCC VIN + VO LG + Figure 27. Charge Pump with Added Gate Drive All the gate drive circuits shown in the above figures typically use 100 nF ceramic capacitors in the bootstrap locations. POWER GOOD SIGNAL The open drain output on the Power Good pin needs a pull-up resistor to a low voltage source. The pull-up resistor should be chosen so that the current going into the Power Good pin is less than 1 mA. A 100 kΩ resistor is recommended for most applications. The Power Good signal is an OR-gated flag which takes into account both output overvoltage and undervoltage conditions. If the feedback pin (FB) voltage is 18% above its nominal value (118% x VFB = 0.708V) or falls 28% below that value (72% x VFB = 0.42V) the Power Good flag goes low. The Power Good flag can be used to signal other circuits that the output voltage has fallen out of regulation, however the switching of the LM2745/8 continues regardless of the state of the Power Good signal. The Power Good flag will return to logic high whenever the feedback pin voltage is between 72% and 118% of 0.6V. UVLO The 2.79V turn-on threshold on VCC has a built in hysteresis of about 300 mV. If VCC drops below 2.42V, the chip definitely enters UVLO mode. UVLO consists of turning off the top and bottom MOSFETS and remaining in that condition until VCC rises above 2.79V. As with normal shutdown initiated by the SD pin, the soft-start capacitor is discharged through an internal MOSFET, ensuring that the next start-up will be controlled by the soft-start circuitry. CURRENT LIMIT Current limit is realized by sensing the voltage across the low-side MOSFET while it is on. The RDSON of the MOSFET is a known value; hence the current through the MOSFET can be determined as: VDS = IOUT x RDSON The current through the low-side MOSFET while it is on is also the falling portion of the inductor current. The current limit threshold is determined by an external resistor, RCS, connected between the switching node and the ISEN pin. A constant current (ISEN-TH) of 40 µA typical is forced through RCS, causing a fixed voltage drop. This fixed voltage is compared against VDS and if the latter is higher, the current limit of the chip has been reached. To obtain a more accurate value for RCS you must consider the operating values of RDSON and ISEN-TH at their operating temperatures in your application and the effect of slight parameter differences from part to part. RCS can be found by using the following equation using the RDSON value of the low side MOSFET at it's expected hot temperature and the absolute minimum value expected over the full temperature range for the for the ISEN-TH which is 25 µA: RCS = RDSON-HOT x ILIM / ISEN-TH 16 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 For example, a conservative 15A current limit in a 10A design with a RDSON-HOT of 10 mΩ would require a 6 kΩ resistor. The minimum value for RCS in any application is 1 kΩ. Because current sensing is done across the lowside MOSFET, no minimum high-side on-time is necessary. The LM2745/8 enters current limit mode if the inductor current exceeds the current limit threshold at the point where the high-side MOSFET turns off and the low-side MOSFET turns on. (The point of peak inductor current, see Figure 28). Note that in normal operation mode the high-side MOSFET always turns on at the beginning of a clock cycle. In current limit mode, by contrast, the high-side MOSFET on-pulse is skipped. This causes inductor current to fall. Unlike a normal operation switching cycle, however, in a current limit mode switching cycle the high-side MOSFET will turn on as soon as inductor current has fallen to the current limit threshold. The LM2745/8 will continue to skip high-side MOSFET pulses until the inductor current peak is below the current limit threshold, at which point the system resumes normal operation. Normal Operation Current Limit ILIM IL D Figure 28. Current Limit Threshold Unlike a high-side MOSFET current sensing scheme, which limits the peaks of inductor current, low-side current sensing is only allowed to limit the current during the converter off-time, when inductor current is falling. Therefore in a typical current limit plot the valleys are normally well defined, but the peaks are variable, according to the duty cycle. The PWM error amplifier and comparator control the off-pulse of the high-side MOSFET, even during current limit mode, meaning that peak inductor current can exceed the current limit threshold. Assuming that the output inductor does not saturate, the maximum peak inductor current during current limit mode can be calculated with the following equation: IPK-CL = ILIM + (TSW - 200 ns) VIN - VO L Where TSW is the inverse of switching frequency fSW. The 200 ns term represents the minimum off-time of the duty cycle, which ensures enough time for correct operation of the current sensing circuitry. In order to minimize the time period in which peak inductor current exceeds the current limit threshold, the IC also discharges the soft-start capacitor through a fixed 90 µA sink. The output of the LM2745/8 internal error amplifier is limited by the voltage on the soft-start capacitor. Hence, discharging the soft-start capacitor reduces the maximum duty cycle D of the controller. During severe current limit this reduction in duty cycle will reduce the output voltage if the current limit conditions last for an extended time. Output inductor current will be reduced in turn to a flat level equal to the current limit threshold. The third benefit of the soft-start capacitor discharge is a smooth, controlled ramp of output voltage when the current limit condition is cleared. SHUTDOWN If the shutdown pin is pulled low, (below 0.8V) the LM2745/8 enters shutdown mode, and discharges the softstart capacitor through a MOSFET switch. The high and low-side MOSFETs are turned off. The LM2745/8 remains in this state as long as VSD sees a logic low (see the Electrical Characteristics table). To assure proper IC start-up the shutdown pin should not be left floating. For normal operation this pin should be connected directly to VCC or to another voltage between 1.3V to VCC (see the Electrical Characteristics table). Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 17 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com DESIGN CONSIDERATIONS The following is a design procedure for all the components needed to create the Typical Application Circuit. This design converts 3.3V (VIN) to 1.2V (VOUT) at a maximum load of 4A with an efficiency of 89% and a switching frequency of 300 kHz. The same procedures can be followed to create many other designs with varying input voltages, output voltages, and load currents. Input Capacitor The input capacitors in a Buck converter are subjected to high stress due to the input current trapezoidal waveform. Input capacitors are selected for their ripple current capability and their ability to withstand the heat generated since that ripple current passes through their ESR. Input rms ripple current is approximately: IRMS_RIP = IOUT x D(1 - D) Where duty cycle D = VOUT/VIN. The power dissipated by each input capacitor is: 2 (IRMS_RIP) x ESR PCAP = n 2 where n is the number of paralleled capacitors, and ESR is the equivalent series resistance of each capacitor. The equation above indicates that power loss in each capacitor decreases rapidly as the number of input capacitors increases. The worst-case ripple for a Buck converter occurs during full load and when the duty cycle (D) is 0.5. For this 3.3V to 1.2V design the duty cycle is 0.364. For a 4A maximum load the ripple current is 1.92A. Output Inductor The output inductor forms the first half of the power stage in a Buck converter. It is responsible for smoothing the square wave created by the switching action and for controlling the output current ripple (ΔIOUT). The inductance is chosen by selecting between tradeoffs in efficiency and response time. The smaller the output inductor, the more quickly the converter can respond to transients in the load current. However, as shown in the efficiency calculations, a smaller inductor requires a higher switching frequency to maintain the same level of output current ripple. An increase in frequency can mean increasing loss in the MOSFETs due to the charging and discharging of the gates. Generally the switching frequency is chosen so that conduction loss outweighs switching loss. The equation for output inductor selection is: VIN - VOUT L= L= xD 'IOUT x fSW 3.3V - 1.2V 1.2V x 3.3V 0.4 x 4A x 300 kHz L = 1.6 µH Here we have plugged in the values for output current ripple, input voltage, output voltage, switching frequency, and assumed a 40% peak-to-peak output current ripple. This yields an inductance of 1.6 µH. The output inductor must be rated to handle the peak current (also equal to the peak switch current), which is (IOUT + (0.5 x ΔIOUT)) = 4.8A, for a 4A design. The Coilcraft DO3316P-222P is 2.2 µH, is rated to 7.4A peak, and has a direct current resistance (DCR) of 12 mΩ. After selecting the Coilcraft DO3316P-222P for the output inductor, actual inductor current ripple should be re-calculated with the selected inductance value, as this information is needed to select the output capacitor. Rearranging the equation used to select inductance yields the following: VIN(MAX) - VO 'IOUT = fSW x LACTUAL xD VIN(MAX) is assumed to be 10% above the steady state input voltage, or 3.6V at VIN = 3.3V. The re-calculated current ripple will then be 1.2A. This gives a peak inductor/switch current will be 4.6A. 18 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 Output Capacitor The output capacitor forms the second half of the power stage of a Buck switching converter. It is used to control the output voltage ripple (ΔVOUT) and to supply load current during fast load transients. In this example the output current is 4A and the expected type of capacitor is an aluminum electrolytic, as with the input capacitors. Other possibilities include ceramic, tantalum, and solid electrolyte capacitors, however the ceramic type often do not have the large capacitance needed to supply current for load transients, and tantalums tend to be more expensive than aluminum electrolytic. Aluminum capacitors tend to have very high capacitance and fairly low ESR, meaning that the ESR zero, which affects system stability, will be much lower than the switching frequency. The large capacitance means that at the switching frequency, the ESR is dominant, hence the type and number of output capacitors is selected on the basis of ESR. One simple formula to find the maximum ESR based on the desired output voltage ripple, ΔVOUT and the designed output current ripple, ΔIOUT, is: ESRMAX = 'VOUT 'IOUT In this example, in order to maintain a 2% peak-to-peak output voltage ripple and a 40% peak-to-peak inductor current ripple, the required maximum ESR is 20 mΩ. The Sanyo 4SP560M electrolytic capacitor will give an equivalent ESR of 14 mΩ. The capacitance of 560 µF is enough to supply energy even to meet severe load transient demands. MOSFETs Selection of the power MOSFETs is governed by a trade-off between cost, size, and efficiency. One method is to determine the maximum cost that can be endured, and then select the most efficient device that fits that price. Breaking down the losses in the high-side and low-side MOSFETs and then creating spreadsheets is one way to determine relative efficiencies between different MOSFETs. Good correlation between the prediction and the bench result is not specified, however. Single-channel buck regulators that use a controller IC and discrete MOSFETs tend to be most efficient for output currents of 2 to 10A. Losses in the high-side MOSFET can be broken down into conduction loss, gate charging loss, and switching loss. Conduction, or I2R loss, is approximately: PC = D (IO2 x RDSON-HI x 1.3) (High-Side MOSFET) PC = (1 - D) x (IO2 x RDSON-LO x 1.3) (Low-Side MOSFET) In the above equations the factor 1.3 accounts for the increase in MOSFET RDSON due to heating. Alternatively, the 1.3 can be ignored and the RDSON of the MOSFET estimated using the RDSON Vs. Temperature curves in the MOSFET datasheets. Gate charging loss results from the current driving the gate capacitance of the power MOSFETs, and is approximated as: PGC = n x (VDD) x QG x fSW where ‘n’ is the number of MOSFETs (if multiple devices have been placed in parallel), VDD is the driving voltage (see MOSFET Gate Drivers section) and QGS is the gate charge of the MOSFET. If different types of MOSFETs are used, the ‘n’ term can be ignored and their gate charges simply summed to form a cumulative QG. Gate charge loss differs from conduction and switching losses in that the actual dissipation occurs in the LM2745/8, and not in the MOSFET itself. Switching loss occurs during the brief transition period as the high-side MOSFET turns on and off, during which both current and voltage are present in the channel of the MOSFET. It can be approximated as: PSW = 0.5 x VIN x IO x (tr + tf) x fSW where tr and tf are the rise and fall times of the MOSFET. Switching loss occurs in the high-side MOSFET only. For this example, the maximum drain-to-source voltage applied to either MOSFET is 3.6V. The maximum drive voltage at the gate of the high-side MOSFET is 3.1V, and the maximum drive voltage for the low-side MOSFET is 3.3V. Due to the low drive voltages in this example, a MOSFET that turns on fully with 3.1V of gate drive is needed. For designs of 5A and under, dual MOSFETs in SO-8 provide a good trade-off between size, cost, and efficiency. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 19 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com Support Components CIN2 - A small (0.1 to 1 µF) ceramic capacitor should be placed as close as possible to the drain of the high-side MOSFET and source of the low-side MOSFET (dual MOSFETs make this easy). This capacitor should be X5R type dielectric or better. RCC, CCC- These are standard filter components designed to ensure smooth DC voltage for the chip supply. RCC should be 1 to 10Ω. CCC should 1 µF, X5R type or better. CBOOT- Bootstrap capacitor, typically 100 nF. RPULL-UP – This is a standard pull-up resistor for the open-drain power good signal (PWGD). The recommended value is 100 kΩ connected to VCC. If this feature is not necessary, the resistor can be omitted. D1 - A small Schottky diode should be used for the bootstrap. It allows for a minimum drop for both high and lowside drivers. The MBR0520 or BAT54 work well in most designs. RCS - Resistor used to set the current limit. Since the design calls for a peak current magnitude (IOUT + (0.5 x ΔIOUT)) of 4.8A, a safe setting would be 6A. (This is below the saturation current of the output inductor, which is 7A.) Following the equation from the Current Limit section, a 1.3 kΩ resistor should be used. RFADJ - This resistor is used to set the switching frequency of the chip. The resistor value is approximated from the Frequency vs Frequency Adjust Resistor curve in the Typical Performance Characteristics section. For 300 kHz operation, a 100 kΩ resistor should be used. CSS - The soft-start capacitor depends on the user requirements and is calculated based on the equation given in the section titled START UP/SOFT-START. Therefore, for a 7 ms delay, a 12 nF capacitor is suitable. Control Loop Compensation The LM2745/8 uses voltage-mode (‘VM’) PWM control to correct changes in output voltage due to line and load transients. VM requires careful small signal compensation of the control loop for achieving high bandwidth and good phase margin. The control loop is comprised of two parts. The first is the power stage, which consists of the duty cycle modulator, output inductor, output capacitor, and load. The second part is the error amplifier, which for the LM2745/8 is a 9 MHz op-amp used in the classic inverting configuration. Figure 29 shows the regulator and control loop components. L RL + C O VIN RO + RC + VRAMP RC2 CC2 RC1 RFB2 CC3 CC1 + RFB1 + - VREF Figure 29. Power Stage and Error Amp One popular method for selecting the compensation components is to create Bode plots of gain and phase for the power stage and error amplifier. Combined, they make the overall bandwidth and phase margin of the regulator easy to see. Software tools such as Excel, MathCAD, and Matlab are useful for showing how changes in compensation or the power stage affect system gain and phase. 20 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 The power stage modulator provides a DC gain ADC that is equal to the input voltage divided by the peak-to-peak value of the PWM ramp. This ramp is 1.0Vpk-pk for the LM2745/8. The inductor and output capacitor create a double pole at frequency fDP, and the capacitor ESR and capacitance create a single zero at frequency fESR. For this example, with VIN = 3.3V, these quantities are: VIN ADC = VRAMP = 3.3 = 10.4 dB 1.0 RO + RL fDP = 1 2S fESR = 1 = 20.3 kHz 2SCOESR LCO(RO + ESR) = 4.5 kHz In the equation for fDP, the variable RL is the power stage resistance, and represents the inductor DCR plus the on resistance of the top power MOSFET. RO is the output voltage divided by output current. The power stage transfer function GPS is given by the following equation, and Figure 30 shows Bode plots of the phase and gain in this example. VIN x RO GPS = VRAMP x sCORC + 1 2 as + bs + c a = LCO(RO + RC) b = L + CO(RORL + RORC + RCRL) c = RO + RL 20 GAIN (dB) 4 -12 -28 -44 -60 100 1k 10k 100k 1M FREQUENCY (Hz) 0 PHASE (o) -30 -60 -90 -120 -150 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 30. Power Stage Gain and Phase Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 21 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com The double pole at 4.5 kHz causes the phase to drop to approximately -130° at around 10 kHz. The ESR zero, at 20.3 kHz, provides a +90° boost that prevents the phase from dropping to -180º. If this loop were left uncompensated, the bandwidth would be approximately 10 kHz and the phase margin 53°. In theory, the loop would be stable, but would suffer from poor DC regulation (due to the low DC gain) and would be slow to respond to load transients (due to the low bandwidth.) In practice, the loop could easily become unstable due to tolerances in the output inductor, capacitor, or changes in output current, or input voltage. Therefore, the loop is compensated using the error amplifier and a few passive components. For this example, a Type III, or three-pole-two-zero approach gives optimal bandwidth and phase. In most voltage mode compensation schemes, including Type III, a single pole is placed at the origin to boost DC gain as high as possible. Two zeroes fZ1 and fZ2 are placed at the double pole frequency to cancel the double pole phase lag. Then, a pole, fP1 is placed at the frequency of the ESR zero. A final pole fP2 is placed at one-half of the switching frequency. The gain of the error amplifier transfer function is selected to give the best bandwidth possible without violating the Nyquist stability criteria. In practice, a good crossover point is one-fifth of the switching frequency, or 60 kHz for this example. The generic equation for the error amplifier transfer function is: s +1 2SfZ1 GEA = AEA x s s +1 2SfP1 s +1 2SfZ2 s +1 2SfP2 In this equation the variable AEA is a ratio of the values of the capacitance and resistance of the compensation components, arranged as shown in Figure 29. AEA is selected to provide the desired bandwidth. A starting value of 80,000 for AEA should give a conservative bandwidth. Increasing the value will increase the bandwidth, but will also decrease phase margin. Designs with 45-60° are usually best because they represent a good trade-off between bandwidth and phase margin. In general, phase margin is lowest and gain highest (worst-case) for maximum input voltage and minimum output current. One method to select AEA is to use an iterative process beginning with these worst-case conditions. 1. Increase AEA 2. Check overall bandwidth and phase margin 3. Change VIN to minimum and recheck overall bandwidth and phase margin 4. Change IO to maximum and recheck overall bandwidth and phase margin The process ends when the both bandwidth and the phase margin are sufficiently high. For this example input voltage can vary from 3.0 to 3.6V and output current can vary from 0 to 4A, and after a few iterations a moderate gain factor of 101dB is used. The error amplifier of the LM2745/8 has a unity-gain bandwidth of 9 MHz. In order to model the effect of this limitation, the open-loop gain can be calculated as: OPG = 2S x 9 MHz s The new error amplifier transfer function that takes into account unity-gain bandwidth is: HEA = GEA x OPG 1 + GEA + OPG The gain and phase of the error amplifier are shown in Figure 31. 22 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 60 GAIN (dB) 48 36 24 12 0 100 1k 10k 100k 1M FREQUENCY (Hz) 50 PHASE (o) 20 -10 -40 -70 -100 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 31. Error Amp. Gain and Phase In VM regulators, the top feedback resistor RFB2 forms a part of the compensation. Setting RFB2 to 10 kΩ±1%, usually gives values for the other compensation resistors and capacitors that fall within a reasonable range. (Capacitances > 1 pF, resistances <1 MΩ) CC1, CC2, CC3, RC1, and RC2 are selected to provide the poles and zeroes at the desired frequencies, using the following equations: fZ1 CC1 = CC2 = AEA x 10,000 x fP2 1 AEA x 10,000 CC3 = RC1 = RC2 = = 27 pF - CC1 = 882 pF 1 1 1 x = 2.73 nF fZ2 fP1 2S x 10,000 1 = 39.8 k: 2S x CC2 x fZ1 1 = 2.55 k: 2S x CC3 x fP1 In practice, a good trade off between phase margin and bandwidth can be obtained by selecting the closest ±10% capacitor values above what are suggested for CC1 and CC2, the closest ±10% capacitor value below the suggestion for CC3, and the closest ±1% resistor values below the suggestions for RC1, RC2. Note that if the suggested value for RC2 is less than 100Ω, it should be replaced by a short circuit. Following this guideline, the compensation components will be: CC1 = 27 pF±10%, CC2 = 820 pF±10% Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 23 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com CC3 = 2.7 nF±10%, RC1 = 39.2 kΩ±1% RC2 = 2.55 kΩ±1% The transfer function of the compensation block can be derived by considering the compensation components as impedance blocks ZF and ZI around an inverting op-amp: Z GEA - ACTUAL = F ZI 1 1 x 10,000 + sCC2 sCC1 ZF = 10,000 + 1 1 + sCC1 sCC2 RC1 RC2 + 1 sCC3 Z1 = RC1 + RC2 + 1 sCC3 As with the generic equation, GEA-ACTUAL must be modified to take into account the limited bandwidth of the error amplifier. The result is: GEA-ACTUAL x OPG HEA = 1 + GEA-ACTUAL + OPG The total control loop transfer function H is equal to the power stage transfer function multiplied by the error amplifier transfer function. H = GPS x HEA The bandwidth and phase margin can be read graphically from Bode plots of HEA as shown in Figure 32. 60 GAIN (dB) 40 20 0 -20 -40 100 1k 10k 100k 1M FREQUENCY (Hz) -60 PHASE (o) -84 -108 -132 -156 -180 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 32. Overall Loop Gain and Phase 24 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 The bandwidth of this example circuit is 59 kHz, with a phase margin of 60°. EFFICIENCY CALCULATIONS The following is a sample calculation. A reasonable estimation of the efficiency of a switching buck controller can be obtained by adding together the Output Power (POUT) loss and the Total Power (PTOTAL) loss: POUT K= POUT + PTOTAL x 100% The Output Power (POUT) for the Typical Application Circuit design is (1.2V x 4A) = 4.8W. The Total Power (PTOTAL), with an efficiency calculation to complement the design, is shown below. The majority of the power losses are due to the low side and high side MOSFET’s losses. The losses in any MOSFET are group of switching (PSW) and conduction losses (PCND). PFET = PSW + PCND = 61.38 mW + 270.42 mW PFET = 331.8 mW FET Switching Loss (PSW) PSW = PSW = PSW = PSW = PSW(ON) + PSW(OFF) 0.5 x VIN x IOUT x (tr + tf) x fSW 0.5 x 3.3V x 4A x 300 kHz x 31 ns 61.38 mW The FDS6898A has a typical turn-on rise time tr and turn-off fall time tf of 15 ns and 16 ns, respectively. The switching losses for this type of dual N-Channel MOSFETs are 0.061W. FET Conduction Loss (PCND) PCND = PCND1 + PCND2 PCND1 = I2OUT x RDS(ON) x k x D PCND2 = I2OUT x RDS(ON) x k x (1-D) RDS(ON) = 13 mΩ and the factor is a constant value (k = 1.3) to account for the increasing RDS(ON) of a FET due to heating. PCND1 = (4A)2 x 13 mΩ x 1.3 x 0.364 PCND2 = (4A)2 x 13 mΩ x 1.3 x (1 - 0.364) PCND = 98.42 mW + 172 mW = 270.42 mW There are few additional losses that are taken into account: IC Operating Loss (PIC) PIC = IQ_VCC x VCC, where IQ-VCC is the typical operating VCC current PIC= 1.7 mA x 3.3V = 5.61 mW FET Gate Charging Loss (PGATE) PGATE = n x VCC x QGS x fSW PGATE = 2 x 3.3V x 3 nC x 300 kHz PGATE = 5.94 mW The value n is the total number of FETs used and QGS is the typical gate-source charge value, which is 3 nC. For the FDS6898A the gate charging loss is 5.94 mW. Input Capacitor Loss (PCAP) 2 (IRMS_RIP) x ESR PCAP = n 2 where, Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 25 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 IRMS_RIP = IOUT x www.ti.com D(1 - D) Here n is the number of paralleled capacitors, ESR is the equivalent series resistance of each, and PCAP is the dissipation in each. So for example if we use only one input capacitor of 24 mΩ. 2 PCAP = (1.924A) x 24 m: 1 2 PCAP = 88.8 mW Output Inductor Loss (PIND) PIND = I2OUT x DCR where DCR is the DC resistance. Therefore, for example PIND = (4A)2 x 11 mΩ PIND = 176 mW Total System Efficiency PTOTAL = PFET + PIC + PGATE + PCAP + PIND POUT K= K= POUT + PTOTAL x 100% 4.8W = 89% 4.8W + 0.6W Example Circuits VCC = VIN= 3.3V D1 RCC CBOOT RPULL-UP VCC LM2745 SD CCC HG BOOT Clk FREQ/SYNC CClk CSS RCS VOUT = 1.8V@2A L1 ISEN PWGD RFADJ + CIN1,2 Q1 LG SS/TRACK PGND SGND PGND EAO FB RFB2 RC2 CC3 + CO1,2 RFB1 CC1 CC2 RC1 Figure 33. 3.3V to 1.8V @ 2A, fSW = 300 kHz 26 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 PART PART NUMBER TYPE PACKAGE U1 LM2745 Synchronous Controller TSSOP-14 DESCRIPTION VENDOR Q1 FDS6898A Dual N-MOSFET SO-8 D1 MBR0520LTI Schottky Diode SOD-123 L1 DO3316P-472 Inductor CIN1 16SP100M Aluminum Electrolytic 10mm x 6mm 100 µF, 16V, 2.89Arms Sanyo CO1 6SP220M Aluminum Electrolytic 10mm x 6mm 220 µF, 6.3V 3.1Arms Sanyo CCC, CBOOT, CIN2, CO2 VJ1206Y104KXXA Capacitor 1206 0.1 µF, 10% Vishay CC3 VJ0805Y332KXXA Capacitor 805 3300 pF, 10% Vishay CSS VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay CC2 VJ0805A821KXAA Capacitor 805 820 pF 10% Vishay CC1 VJ0805A220KXAA Capacitor 805 22 pF, 10% Vishay RFB2 CRCW08051002F Resistor 805 10.0 kΩ 1% Vishay RFB1 CRCW08054991F Resistor 805 4.99 kΩ1% Vishay RFADJ CRCW08051003F Resistor 805 100 kΩ 1% Vishay RC2 CRCW08052101F Resistor 805 2.1 kΩ 1% Vishay RCS CRCW08052101F Resistor 805 2.1 kΩ 1% Vishay RCC CRCW080510R0F Resistor 805 10.0Ω 1% Vishay RC1 CRCW08055492F Resistor 805 54.9 kΩ 1% Vishay RPULL-UP CRCW08051003J Resistor 805 100 kΩ 5% Vishay CCLK VJ0805A560KXAA Capacitor 805 56 pF, 10% Vishay TI 20V, 10 mΩ @ 4.5V, 16nC Fairchild 4.7 µH, 4.8Arms 18 mΩ Coilcraft VCC = 5V D1 RCC CBOOT RPULL-UP VCC LM2745 HG BOOT FREQ/SYNC CClk RFADJ CSS RCS L1 VOUT = 2.5V@2A ISEN PWGD Clk + CIN1,2 Q1 SD CCC VIN = 5V LG SS/TRACK PGND SGND PGND EAO FB RFB2 RC2 CC3 + CO1,2 RFB1 CC1 CC2 RC1 Figure 34. 5V to 2.5V @ 2A, fSW = 300 kHz Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 27 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com PART PART NUMBER TYPE PACKAGE U1 LM2745 Synchronous Controller TSSOP-14 DESCRIPTION VENDOR Q1 FDS6898A Dual N-MOSFET SO-8 D1 MBR0520LTI Schottky Diode SOD-123 L1 DO3316P-682 Inductor 6.8 µH, 4.4Arms, 27 mΩ Coilcraft CIN1 16SP100M Aluminum Electrolytic 10mm x 6mm 100 µF, 16V, 2.89Arms Sanyo CO1 10SP56M Aluminum Electrolytic 6.3mm x 6mm 56 µF, 10V 1.7Arms Sanyo CCC, CBOOT, CIN2, CO2 VJ1206Y104KXXA Capacitor 1206 0.1 µF, 10% Vishay CC3 VJ0805Y182KXXA Capacitor 805 1800 pF, 10% Vishay CSS VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay CC2 VJ0805A821KXAA Capacitor 805 820 pF 10% Vishay CC1 VJ0805A330KXAA Capacitor 805 33 pF, 10% Vishay RFB2 CRCW08051002F Resistor 805 10.0 kΩ 1% Vishay RFB1 CRCW08053161F Resistor 805 3.16 kΩ 1% Vishay RFADJ CRCW08051003F Resistor 805 100 kΩ 1% Vishay RC2 CRCW08051301F Resistor 805 1.3 kΩ 1% Vishay RCS CRCW08052101F Resistor 805 2.1 kΩ 1% Vishay RCC CRCW080510R0F Resistor 805 10.0Ω 1% Vishay RC1 CRCW08053322F Resistor 805 33.2 kΩ 1% Vishay RPULL-UP CRCW08051003J Resistor 805 100 kΩ 5% Vishay CCLK VJ0805A560KXAA Capacitor 805 56 pF, 10% Vishay TI 20V, 10 mΩ @ 4.5V, 16 nC Fairchild VCC = 5V D1 RCC CBOOT RPULL-UP VCC LM2745 HG BOOT FREQ/SYNC CClk RFADJ CSS RCS L1 VOUT = 3.3V@4A ISEN PWGD Clk + CIN1,2 Q1 SD CCC VIN = 12V LG SS/TRACK PGND SGND PGND EAO FB RFB2 RC2 CC3 + CO1,2 RFB1 CC1 CC2 RC1 Figure 35. 12V to 3.3V @ 4A, fSW = 300kHz 28 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 LM2745, LM2748 www.ti.com SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 PART PART NUMBER TYPE PACKAGE U1 LM2745 Synchronous Controller TSSOP-14 Q1 FDS6898A Dual N-MOSFET SO-8 D1 MBR0520LTI Schottky Diode SOD-123 L1 DO3316P-332 Inductor CIN1 16SP100M Aluminum Electrolytic 10mm x 6mm CO1 6SP220M Aluminum Electrolytic CCC, CBOOT, CIN2, CO2 VJ1206Y104KXXA CC3 CSS DESCRIPTION VENDOR TI 20V, 10 mΩ @ 4.5V, 16 nC Fairchild 3.3 µH, 5.4Arms 15 mΩ Coilcraft 100 µF, 16V, 2.89Arms Sanyo 10mm x 6mm 220 µF, 6.3V 3.1Arms Sanyo Capacitor 1206 0.1 µF, 10% Vishay VJ0805Y222KXXA Capacitor 805 2200 pF, 10% Vishay VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay CC2 VJ0805Y332KXXA Capacitor 805 3300 pF 10% Vishay CC1 VJ0805A820KXAA Capacitor 805 82 pF, 10% Vishay RFB2 CRCW08051002F Resistor 805 10.0 kΩ 1% Vishay RFB1 CRCW08052211F Resistor 805 2.21 kΩ 1% Vishay RFADJ CRCW08051003F Resistor 805 100 kΩ 1% Vishay RC2 CRCW08052611F Resistor 805 2.61 kΩ 1% Vishay RCS CRCW08054121F Resistor 805 4.12 kΩ 1% Vishay RCC CRCW080510R0F Resistor 805 10.0Ω 1% Vishay RC1 CRCW08051272F Resistor 805 12.7kΩ 1% Vishay RPULL-UP CRCW08051003J Resistor 805 100 kΩ 5% Vishay CCLK VJ0805A560KXAA Capacitor 805 56 pF, 10% Vishay Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 29 LM2745, LM2748 SNOSAL2E – APRIL 2005 – REVISED APRIL 2013 www.ti.com REVISION HISTORY Changes from Revision D (April 2013) to Revision E • 30 Page Changed layout of National Data Sheet to TI format .......................................................................................................... 29 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2745 LM2748 PACKAGE OPTION ADDENDUM www.ti.com 5-Nov-2013 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (°C) Device Marking (4/5) LM2745MTC/NOPB ACTIVE TSSOP PW 14 94 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 2745 MTC LM2745MTCX/NOPB ACTIVE TSSOP PW 14 2500 Green (RoHS & no Sb/Br) CU SN | Call TI Level-1-260C-UNLIM -40 to 125 2745 MTC LM2748MTC/NOPB ACTIVE TSSOP PW 14 94 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 2748 MTC LM2748MTCX/NOPB ACTIVE TSSOP PW 14 2500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 2748 MTC (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 5-Nov-2013 Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. 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Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 9-Jan-2014 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LM2745MTCX/NOPB TSSOP PW 14 2500 330.0 12.4 6.95 8.3 1.6 8.0 12.0 Q1 LM2745MTCX/NOPB TSSOP PW 14 2500 330.0 12.4 6.95 8.3 1.6 8.0 12.0 Q1 LM2748MTCX/NOPB TSSOP PW 14 2500 330.0 12.4 6.95 8.3 1.6 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 9-Jan-2014 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM2745MTCX/NOPB TSSOP PW 14 2500 367.0 367.0 35.0 LM2745MTCX/NOPB TSSOP PW 14 2500 367.0 367.0 35.0 LM2748MTCX/NOPB TSSOP PW 14 2500 367.0 367.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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