OPA365 OPA2365 SBOS365A − JUNE 2006 − REVISED JULY 2006 2.2V, 50MHz, Low-Noise, Single-Supply Rail-to-Rail OPERATIONAL AMPLIFIERS FEATURES DESCRIPTION D D D D D D D D D D The OPAx365 zer∅-crossover series rail-to-rail highperformance CMOS operational amplifiers are optimized for very low voltage, single-supply applications. Rail-to-rail input/output, low-noise (4.5nV/√Hz) and high-speed operations (50MHz Gain Bandwidth) make them ideal for driving sampling analog-to-digital converters (ADCs). Applications incude audio, signal conditioning, and sensor amplification. The OPA365 family of op amps are well-suited for cell phone power amplifier control loops. RAIL-TO-RAIL INPUT WITHOUT CROSSOVER 2.2V OPERATION LOW OFFSET: 200µV WIDE BANDWIDTH: 50MHz CMRR: 100dB (min) HIGH SLEW RATE: 25V/µs LOW NOISE: 4.5nV//Hz LOW THD+NOISE: 0.0006% QUIESCENT CURRENT: 5mA (max) microPACKAGE: SOT23-5 Special features include excellent common-mode rejection ratio (CMRR), no input stage crossover distortion, high input impedance and rail-to-rail input and output swing. The input common-mode range includes both the negative and positive supplies. The output voltage swing is within 10mV of the rails. APPLICATIONS D D D D D D D SIGNAL CONDITIONING DATA ACQUISITION The OPA365 (single version) is available in the microSIZE SOT23-5 and SO-8 packages. The OPA2365 (dual version) is offered in the microSIZE DFN-8 (3mm x 3mm) and SO-8 packages. All versions are specified for operation from −40°C to +125°C. Single and dual versions have identical specifications for maximum design flexibility. PROCESS CONTROL ACTIVE FILTERS TEST EQUIPMENT AUDIO WIDEBAND AMPLIFIERS OPA365 vs COMPETITION 0 +5V −20 THD+Noise Ratio (dB) f i = 10kHz BW = 30kHz PACKAGE OPA365 SOT23-5 n SO-8(1) n n n DFN-8(1) −40 OPA2365 (1) Available Q3, 2006. VIN −60 Competitor B −80 Competitor A −100 OPA365 −120 1 2 3 4 5 VIN = VOUT (VPP) Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright 2006, Texas Instruments Incorporated ! ! www.ti.com "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 ABSOLUTE MAXIMUM RATINGS(1) Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.5V Signal Input Terminals, Voltage(2) . . . . (V−) −0.5V to (V+) + 0.5V Signal Input Terminals, Current(2) . . . . . . . . . . . . . . . . . . . . ±10mA Output Short-Circuit(3) . . . . . . . . . . . . . . . . . . . . . . . . . Continuous Operating Temperature . . . . . . . . . . . . . . . . . . . . . −40°C to +150°C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . −65°C to +150°C Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C ESD Rating Human Body Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4000V Charged Device Model . . . . . . . . . . . . . . . . . . . . . . . . . . . 1000V (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not supported. (2) Input terminals are diode-clamped to the power-supply rails. Input signals that can swing more than 0.5V beyond the supply rails should be current limited to 10mA or less. (3) Short-circuit to ground, one amplifier per package. This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION(1) PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR SOT23-5 DBV OAVQ SO-8(2) D O365A SO-8(2) D O2365A DFN-8(2) DRB OPA365 OPA2365 PACKAGE MARKING BRA (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. (2) Available Q3, 2006. PIN CONFIGURATIONS Top View OPA365 OPA365 VOUT V− +IN 1 5 V+ NC(1) −IN 2 3 4 −IN +IN V− 1 2 3 4 OPA2365 8 NC(1) VOUTA 1 8 V+ 7 V+ −IN A 2 7 VOUTB 6 VOUT +IN A 3 6 −IN B 5 NC(1) V− 4 5 +IN B SOT23−5 SO−8 (1) NC denotes no internal connection. 2 SO−8, DFN−8 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 ELECTRICAL CHARACTERISTICS: VS = +2.2V to +5.5V Boldface limits apply over the specified temperature range, TA = −40°C to +125°C. At TA = +25°C, RL = 10kΩ connected to VS/2, VCM = VS/2, and VOUT = VS/2, unless otherwise noted. OPAx365 PARAMETER OFFSET VOLTAGE Input Offset Voltage VOS Drift dVOS/dT vs Power Supply PSRR Channel Separation, dc INPUT BIAS CURRENT Input Bias Current IB over Temperature Input Offset Current IOS NOISE Input Voltage Noise, f = 0.1Hz to 10Hz en Input Voltage Noise Density, f = 100kHz en Input Current Noise Density, f = 10kHz in INPUT VOLTAGE RANGE Common-Mode Voltage Range VCM Common-Mode Rejection Ratio CMRR INPUT CAPACITANCE Differential Common-Mode OPEN-LOOP GAIN Open-Loop Voltage Gain AOL FREQUENCY RESPONSE Gain-Bandwidth Product Slew Rate Settling Time, 0.1% 0.01% Overload Recovery Time Total Harmonic Distortion + Noise OUTPUT Voltage Output Swing from Rail over Temperature Short-Circuit Current Capacitive Load Drive Open-Loop Output Impedance POWER SUPPLY Specified Voltage Range Quiescent Current Per Amplifier over Temperature TEMPERATURE RANGE Specified Range Thermal Resistance SOT23-5 SO-8 DFN-8 GBW SR tS THD+N TEST CONDITIONS VS = +2.2V to +5.5V TYP MAX UNIT 100 1 10 0.2 200 µV µV/°C µV/V µV/V 100 ±0.2 ±10 See Typical Characteristics ±0.2 ±10 (V−) − 0.1V 3 VCM 3 (V+) + 0.1V RL = 10kΩ, 100mV < VO < (V+) − 100mV RL = 600Ω, 200mV < VO < (V+) − 200mV RL = 600Ω, 200mV < VO < (V+) − 200mV VS = 5V (V−) − 0.1 100 100 100 94 G = +1 4V Step, G = +1 4V Step, G = +1 VIN x Gain > VS RL = 600Ω, VO = 4VPP, G = +1, f = 1kHz RL = 10kΩ, VS = 5.5V f = 1MHz, IO = 0 pA 120 (V+) + 0.1 V dB 6 2 pF pF 120 120 dB dB dB 50 25 200 300 < 0.1 0.0006 MHz V/µs ns ns µs % 10 20 ±65 See Typical Characteristics 30 2.2 IO = 0 pA µVPP nV/√Hz fA/√Hz 5 4.5 4 ISC CL VS IQ MIN 4.6 −40 qJA 200 150 46 mV mA Ω 5.5 5 5 V mA mA +125 °C °C/W °C/W °C/W °C/W 3 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 TYPICAL CHARACTERISTICS At TA = +25°C, VS = +5V, and CL = 0pF, unless otherwise noted. POWER SUPPLY AND COMMON−MODE REJECTION RATIO vs FREQUENCY OPEN−LOOP GAIN/PHASE vs FREQUENCY 140 0 140 CMRR −45 100 Phase 80 −90 60 40 Gain 20 −135 PSRR, CMRR (dB) 120 Phase (_ ) Voltage Gain (dB) 120 100 80 PSRR 60 40 20 0 −180 100M −20 10 100 1k 10k 100k 1M 10M 0 10 100 1k 10k 100k 1M Frequency (Hz) Frequency (Hz) OFFSET VOLTAGE PRODUCTION DISTRIBUTION OFFSET VOLTAGE DRIFT PRODUCTION DISTRIBUTION 100M VS = 5.5V −200 −180 −160 −140 −120 −100 −80 −60 −40 −20 0 20 40 60 80 100 120 140 160 180 200 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 Population Population VS = 5.5V 10M Offset Voltage (µV) Offset Voltage Drift (µV/_C) INPUT BIAS CURRENT vs TEMPERATURE INPUT BIAS CURRENT vs COMMON−MODE VOLTAGE 500 1000 900 400 700 300 600 IB (pA) Input Bias (pA) 800 500 400 VCM Specified Range 300 100 200 100 0 −50 −25 0 25 50 Temperature (_C) 4 200 75 100 125 0 −25 −0.5 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 VCM (V) "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, VS = +5V, and CL = 0pF, unless otherwise noted. OUTPUT VOLTAGE vs OUTPUT CURRENT 3 Short−Circuit Current (mA) VS = ±1.1V VS = ±2.75V 2 Output Voltage (V) SHORT−CIRCUIT CURRENT vs TEMPERATURE 1 −40_C 0 +25_ C +125_ C +25_ C −40_C +125_ C −1 −2 −3 0 10 20 30 40 50 60 70 80 90 70 60 50 40 30 20 10 0 −10 −20 −30 −40 −50 −60 −70 −80 100 ISC+ I SC− −50 −25 0 50 75 100 125 Temperature (_ C) Output Current (mA) QUIESCENT CURRENT vs TEMPERATURE QUIESCENT CURRENT vs SUPPLY VOLTAGE 4.80 Quiescent Current (mA) 4.75 4.50 4.25 4.00 4.74 4.68 4.62 4.56 4.50 3.75 2.2 2.5 3.0 3.5 4.0 4.5 5.0 −50 5.5 −25 0 25 50 75 100 125 Temperature (_ C) Supply Voltage (V) TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 0.1Hz to 10Hz INPUT VOLTAGE NOISE 0.01 THD+N (%) G = 10, RL = 600Ω 2µV/div Quiescent Current (mA) 25 VO = 1VRMS 0.001 VO = 1.448VRMS VO = 1VRMS G = +1, RL = 600Ω 0.0001 1s/div 10 100 1k 10k 100k Frequency (Hz) 5 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 TYPICAL CHARACTERISTICS (continued) At TA = +25°C, VS = +5V, and CL = 0pF, unless otherwise noted. OVERSHOOT vs CAPACITIVE LOAD INPUT VOLTAGE NOISE SPECTRAL DENSITY 1k 60 100 Overshoot (%) Voltage Noise (nV/√Hz) 50 10 G = +1 40 G = −1 30 G = +10 20 10 G = −10 1 0 10 100 1k 10k 0 100k 100 Capacitive Load (pF) Frequency (Hz) Output Voltage (1V/div) G=1 RL = 10kΩ VS = ±2.5 G=1 RL = 10kΩ VS = ±2.5 Time (50ns/div) Time (250ns/div) SMALL−SIGNAL STEP RESPONSE LARGE−SIGNAL STEP RESPONSE G=1 RL = 600Ω VS = ±2.5 Time (50ns/div) 6 LARGE−SIGNAL STEP RESPONSE Output Voltage (1V/div) Output Voltage (50mV/div) Output Voltage (50mV/div) SMALL−SIGNAL STEP RESPONSE G=1 RL = 600Ω VS = ±2.5 Time (250ns/div) 1k "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 APPLICATIONS INFORMATION R2 10kΩ OPERATING CHARACTERISTICS The OPA365 amplifier parameters are fully specified from +2.2V to +5.5V. Many of the specifications apply from −40°C to +125°C. Parameters that can exhibit significant variance with regard to operating voltage or temperature are presented in the Typical Characteristics. +1.5V R1 1kΩ V+ OPA365 GENERAL LAYOUT GUIDELINES The OPA365 is a wideband amplifier. To realize the full operational performance of the device, good high-frequency printed circuit board (PCB) layout practices are required. Low-loss, 0.1µF bypass capacitors must be connected between each supply pin and ground as close to the device as possible. The bypass capacitor traces should be designed for minimum inductance. C1 100nF VIN V− C2 100nF −1.5V a) Dual Supply Connection BASIC AMPLIFIER CONFIGURATIONS As with other single-supply op amps, the OPA365 may be operated with either a single supply or dual supplies. A typical dual-supply connection is shown in Figure 1, which is accompanied by a single-supply connection. The OPA365 is configured as a basic inverting amplifier with a gain of −10V/V. The dual-supply connection has an output voltage centered on zero, while the single− supply connection has an output centered on the common-mode voltage VCM. For the circuit shown, this voltage is 1.5V, but may be any value within the commonmode input voltage range. The OPA365 VCM range extends 100mV beyond the power-supply rails. VOUT R2 10kΩ +3V R1 1kΩ C1 100nF V+ OPA365 VIN VOUT V− VCM = 1.5V b) Single−Supply Connection Figure 1. Basic Circuit Connections 7 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 Figure 2 shows a single-supply, electret microphone application where VCM is provided by a resistive divider. The divider also provides the bias voltage for the electret element. 49kΩ Clean 3.3V Supply 3.3V 4kΩ INPUT AND ESD PROTECTION The OPA365 incorporates internal electrostatic discharge (ESD) protection circuits on all pins. In the case of input and output pins, this protection primarily consists of current steering diodes connected between the input and power-supply pins. These ESD protection diodes also provide in-circuit, input overdrive protection, provided that the current is limited to 10mA as stated in the Absolute Maximum Ratings. Figure 3 shows how a series input resistor may be added to the driven input to limit the input current. The added resistor contributes thermal noise at the amplifier input and its value should be kept to the minimum in noise-sensitive applications. VOUT OPA365 Electret Microphone 6kΩ 5kΩ 1µF Figure 2. Microphone Preamplifier V+ RAIL−TO−RAIL INPUT I OVERLOAD 10mA max The OPA365 product family features true rail-to-rail input operation, with supply voltages as low as ±1.1V (2.2V). A unique zer∅-crossover input topology eliminates the input offset transition region typical of many rail-to-rail, complementary stage operational amplifiers. This topology also allows the OPA365 to provide superior common−mode performance over the entire input range, which extends 100mV beyond both power-supply rails; see Figure 4. When driving ADCs, the highly linear VCM range of the OPA365 assures that the op amp/ADC system linearity performance is not compromised. VOUT OPA365 VIN 5kΩ Figure 3. Input Current Protection OFFSET VOLTAGE vs COMMON−MODE VOLTAGE 200 VS = ±2.75V 150 100 VOS (µV) OPA365 50 0 −50 −100 Competitors −150 −200 −3 −2 −1 0 1 2 Common−Mode Voltage (V) Figure 4. OPA365 has Linear Offset Over the Entire Common-Mode Range 8 3 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 A simplified schematic illustrating the rail-to-rail input circuitry is shown in Figure 5. VS Regulated Charge Pump VO U T = VC C +1.8V CAPACITIVE LOADS The OPA365 may be used in applications where driving a capacitive load is required. As with all op amps, there may be specific instances where the OPA365 can become unstable, leading to oscillation. The particular op amp circuit configuration, layout, gain and output loading are some of the factors to consider when establishing whether an amplifier will be stable in operation. An op amp in the unity-gain (+1V/V) buffer configuration and driving a capacitive load exhibits a greater tendency to be unstable than an amplifier operated at a higher noise gain. The capacitive load, in conjunction with the op amp output resistance, creates a pole within the feedback loop that degrades the phase margin. The degradation of the phase margin increases as the capacitive loading increases. VC C + 1.8V Patent Pending Very Low Ripple Topology IB IAS IB IA S IBI A S VIN − VO U T VI N + When operating in the unity-gain configuration, the OPA365 remains stable with a pure capacitive load up to approximately 1nF. The equivalent series resistance (ESR) of some very large capacitors (CL > 1µF) is sufficient to alter the phase characteristics in the feedback loop such that the amplifier remains stable. Increasing the amplifier closed-loop gain allows the amplifier to drive increasingly larger capacitance. This increased capability is evident when observing the overshoot response of the amplifier at higher voltage gains. See the typical characteristic graph, Small-Signal Overshoot vs. Capacitive Load. One technique for increasing the capacitive load drive capability of the amplifier operating in unity gain is to insert a small resistor, typically 10Ω to 20Ω, in series with the output; see Figure 6. This resistor significantly reduces the overshoot and ringing associated with large capacitive loads. A possible problem with this technique is that a voltage divider is created with the added series resistor and any resistor connected in parallel with the capacitive load. The voltage divider introduces a gain error at the output that reduces the output swing. The error contributed by the voltage divider may be insignificant. For instance, with a load resistance, RL = 10kΩ, and RS = 20Ω, the gain error is only about 0.2%. However, when RL is decreased to 600Ω, which the OPA365 is able to drive, the error increases to 7.5%. IB IA S Figure 5. Simplified Schematic V+ RS VOUT OPA365 VIN 10Ω to 20Ω RL CL Figure 6. Improving Capacitive Load Drive 9 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 ACHIEVING AN OUTPUT LEVEL OF ZERO VOLTS (0V) V+ = +5V Certain single-supply applications require the op amp output to swing from 0V to a positive full-scale voltage and have high accuracy. An example is an op amp employed to drive a single-supply ADC having an input range from 0V to +5V. Rail-to-rail output amplifiers with very light output loading may achieve an output level within millivolts of 0V (or +VS at the high end), but not 0V. Furthermore, the deviation from 0V only becomes greater as the load current required increases. This increased deviation is a result of limitations of the CMOS output stage. OPA365 500µA Note that this technique does not work with all op amps and should only be applied to op amps such as the OPA365 that have been specifically designed to operate in this manner. Also, operating the OPA365 output at 0V changes the output stage operating conditions, resulting in somewhat lower open-loop gain and bandwidth. Keep these precautions in mind when driving a capacitive load because these conditions can affect circuit transient response and stability. ACTIVE FILTERING The OPA365 is well-suited for active filter applications requiring a wide bandwidth, fast slew rate, low-noise, single-supply operational amplifier. Figure 8 shows a 500kHz, 2nd-order, low-pass filter utilizing the multiple− feedback (MFB) topology. The components have been selected to provide a maximally-flat Butterworth response. Beyond the cutoff frequency, roll-off is −40dB/dec. The Butterworth response is ideal for applications requiring predictable gain characteristics such as the anti-aliasing filter used ahead of an ADC. 10 RP = 10kΩ Op Amps Negative Supply Grounded When a pull-down resistor is connected from the amplifier output to a negative voltage source, the OPA365 can achieve an output level of 0V, and even a few millivolts below 0V. Below this limit, nonlinearity and limiting conditions become evident. Figure 7 illustrates a circuit using this technique. A pull-down current of approximately 500µA is required when OPA365 is connected as a unity-gain buffer. A practical termination voltage (VNEG) is −5V, but other convenient negative voltages also may be used. The pull-down resistor RL is calculated from RL = [(VO −VNEG)/(500µA)]. Using a minimum output voltage (VO) of 0V, RL = [0V−(−5V)]/(500µA)] = 10kΩ. Keep in mind that lower termination voltages result in smaller pull-down resistors that load the output during positive output voltage excursions. VOUT VIN −V = −5V (Additional Negative Supply) Figure 7. Swing-to-Ground R3 549Ω C2 150pF V+ R1 549Ω R2 1.24kΩ VIN OPA365 C1 1nF VOUT V− Figure 8. Second-Order Butterworth 500kHz Low-Pass Filter "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 One point to observe when considering the MFB filter is that the output is inverted, relative to the input. If this inversion is not required, or not desired, a noninverting output can be achieved through one of these options: 1) adding an inverting amplifier; 2) adding an additional 2nd-order MFB stage; or 3) using a noninverting filter topology such as the Sallen-Key (shown in Figure 9). OPA365 an ideal driver for modern ADCs. Also, because it is free of the input offset transition characteristics inherent to some rail-to-rail CMOS op amps, the OPA365 provides low THD and excellent linearity throughout the input voltage swing range. Figure 10 shows the OPA365 driving an ADS8326, 16-bit, 500kSPS converter. The amplifier is connected as a unity-gain, noninverting buffer and has an output swing to 0V, making it directly compatible with the ADC minus full-scale input level. The 0V level is achieved by powering the OPA365 V− pin with a small negative voltage established by the diode forward voltage drop. A small, signal-switching diode or Schottky diode provides a suitable negative supply voltage of −0.3 to −0.7V. The supply rail-to-rail is equal to V+, plus the small negative voltage. MFB and Sallen-Key, low-pass and high-pass filter synthesis is quickly accomplished using TI’s FilterPro program. This software is available as a free download at www.ti.com. DRIVING AN ANALOG-TO-DIGITAL CONVERTER Very wide common-mode input range, rail-to-rail input and output voltage capability and high speed make the C3 220pF R2 19.5kΩ R1 1.8kΩ R3 150kΩ VIN = 1VRMS C1 3.3nF C2 47pF OPA365 VOUT Figure 9. Configured as a 3-Pole, 20kHz, Sallen-Key Filter +5V C1 100nF +5V R1(1) 100Ω V+ +IN OPA365 C3(1) 1nF V− VIN 0 to 4.096V −IN ADS8326 16−Bit 100kSPS REF IN +5V Optional(2) R2 500Ω SD1 BAS40 −5V C2 100nF REF3240 4.096V C4 100nF NOTES: (1) Suggested value; may require adjustment based on specific application. (2) Single−supply applications lose a small number of ADC codes near ground due to op amp output swing limitation. If a negative power supply is available, this simple circuit creates a −0.3V supply to allow output swing to true ground potential. Figure 10. Driving the ADS8326 11 "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 One method for driving an ADC that negates the need for an output swing down to 0V uses a slightly compressed ADC full-scale input range (FSR). For example, the 16-bit ADS8361 (shown in Figure 11) has a maximum FSR of 0V to 5V, when powered by a +5V supply and VREF of 2.5V. The idea is to match the ADC input range with the op amp full linear output swing range; for example, an output range of +0.1 to +4.9V. The reference output from the ADS8361 ADC is divided down from 2.5V to 2.4V using a resistive divider. The ADC FSR then becomes 4.8VPP centered on a common-mode voltage of +2.5V. Current from the ADS8361 reference pin is limited to about ±10µA. Here, 5µA was used to bias the divider. The resistors must be precise to maintain the ADC gain accuracy. An additional benefit of this method is the elimination of the negative supply voltage; it requires no additional power-supply current. An RC network, consisting of R1 and C1, is included between the op amp and the ADS8361. It not only provides a high-frequency filter function, but more importantly serves as a charge reservoir used for charging the converter internal hold capacitance. This capability assures that the op amp output linearity is maintained as the ADC input characteristics change throughout the conversion cycle. Depending on the particular application and ADC, some optimization of the R1 and C1 values may be required for best transient performance. R2 10kΩ +5V R1 10kΩ C1 100nF V+ +5V R3(1) 100Ω −IN OPA365 VIN 0.1V to 4.9V C2(1) V− 1nF +IN ADS8361 16−Bit 100kSPS REF OUT REF IN +2.5V NOTE: (1) Suggested value; may require adjustment based on specific application. R4 20kΩ +2.4V R5 480kΩ Figure 11. Driving the ADS8361 12 C3 1µF "#$ %"#$ www.ti.com SBOS365A − JUNE 2006 − REVISED JULY 2006 Figure 12 illustrates the OPA2365 dual op amp providing signal conditioning within an ADS1258 bridge sensor circuit. It follows the ADS1258 16:1 multiplexer and is connected as a differential in/differential out amplifier. The voltage gain for this stage is approximately 10V/V. Driving the ADS1258 internal ADC in differential mode, rather than in a single-ended, exploits the full linearity performance capability of the converter. For best common-mode rejection the two R2 resistors should be closely matched. Note that in Figure 12, the amplifiers, bridges, ADS1258 and internal reference are powered by the same single +5V supply. This ratiometric connection helps cancel excitation voltage drift effects and noise. For best performance, the +5V supply should be as free as possible of noise and transients. When the ADS1258 data rate is set to maximum and the chop feature enabled, this circuit yields 12 bits of noise-free resolution with a 50mV full-scale input. The chop feature is used to reduce the ADS1258 offset and offset drift to very low levels. A 2.2nF capacitor is required across the ADC inputs to bypass the sampling currents. The 47Ω resistors provide isolation for the OPA2365 outputs from the relatively large, 2.2nF capacitive load. For more information regarding the ADS1258, see the product data sheet available for down load at www.ti.com. +5V RFI 10µF + 0.1µF 2kΩ RFI AIN0 AVSS AVDD 2kΩ REFP AIN1 + … ADS1258 AINCOM MUXOUTP AIN15 MUXOUTN AIN14 2kΩ RFI 0.1µF RFI ADCINN … … 2kΩ RFI 10µF REFN ADCINP RFI +5V 2.2nF 0.1µF R3 47Ω OPA2365 R2 = 10kΩ R1 = 2.2kΩ R2 = 10kΩ R3 47Ω OPA2365 NOTE: G = 1 + 2R2/R1. Match R2 resistors for optimum CMRR. Figure 12. Conditioning Input Signals to the ADS1258 on a Single-Supply 13 PACKAGE OPTION ADDENDUM www.ti.com 21-Jul-2006 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty OPA365AIDBVR ACTIVE SOT-23 DBV 5 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM OPA365AIDBVRG4 ACTIVE SOT-23 DBV 5 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM OPA365AIDBVT ACTIVE SOT-23 DBV 5 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM OPA365AIDBVTG4 ACTIVE SOT-23 DBV 5 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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