MP4561 1.5A, 2MHz, 55V Step-Down Converter The Future of Analog IC Technology DESCRIPTION FEATURES The MP4561 is a high frequency step-down switching regulator with integrated internal highside high voltage power MOSFET. It provides 1.5A output with current mode control for fast loop response and easy compensation. • • • • • • • • • The wide 3.8V to 55V input range accommodates a variety of step-down applications, including those in automotive input environment. A 12µA shutdown mode supply current allows use in battery-powered applications. High power conversion efficiency over a wide load range is achieved by scaling down the switching frequency at light load condition to reduce the switching and gate driving losses. Wide 3.8V to 55V Operating Input Range 300mΩ Internal Power MOSFET Up to 2MHz Programmable Switching Frequency 140μA Quiescent Current Ceramic Capacitor Stable External Soft-Start Up to 95% Efficiency Output Adjustable from 0.8V to 52V Available in 3x3mm 10-Pin QFN APPLICATIONS • • • • • The frequency foldback helps prevent inductor current runaway during startup and thermal shutdown provides reliable, fault tolerant operation. High Voltage Power Conversion Automotive Systems Industrial Power Systems Distributed Power Systems Battery Powered Systems All MPS parts are lead-free and adhere to the RoHS directive. For MPS green status, please visit MPS website under Quality Assurance. “MPS” and “The Future of Analog IC Technology” are Registered Trademarks of Monolithic Power Systems, Inc. By switching at 2MHz, the MP4561 is able to prevent EMI (Electromagnetic Interference) noise problems, such as those found in AM radio and ADSL applications. The MP4561 is available in small 3mm x 3mm 10-pin QFN with exposed pad package. TYPICAL APPLICATION C4 100nF VIN C1 10uF 100V 8 7 C5 10nF SW EN MP4561 FB 5 C7 NS SS COMP FREQ GND 6 100 L1 10uH 1.2 D1 3 EN VIN BST 4 C3 470pF C6 NS C2 22uF 6.3V VOUT 3.3V VIN=12V 90 EFFICIENCY (%) 10 9 80 70 VIN=55V 60 50 40 0 0.5 1 1.5 IOUT (A) MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 1 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER ORDERING INFORMATION Part Number* MP4561DQ Package QFN10( 3x3mm) Top Marking 9C * For Tape & Reel, add suffix –Z (e.g. MP4561DQ–Z) For RoHS compliant packaging, add suffix –LF (e.g. MP4561DQ–LF–Z) PACKAGE REFERENCE TOP VIEW SW 1 10 BST SW 2 9 VIN EN 3 8 SS COMP 4 7 FREQ FB 5 6 GND EXPOSED PAD Connect to GND Plane ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance Supply Voltage (VIN).....................–0.3V to +60V Switch Voltage (VSW)............ –0.5V to VIN + 0.5V BST to SW .....................................–0.3V to +5V All Other Pins .................................–0.3V to +5V (2) Continuous Power Dissipation (TA = +25°C) ............................................................. 2.5W Junction Temperature ...............................150°C Lead Temperature ....................................260°C Storage Temperature.............. –65°C to +150°C 3x3mm QFN10........................50 ...... 12 ... °C/W Recommended Operating Conditions (3) Supply Voltage VIN ...........................3.8V to 55V Output Voltage VOUT .........................0.8V to 52V Operating Junct. Temp. (TJ).... –40°C to +125°C MP4561 Rev. 1.0 11/5/2012 (4) θJA θJC Notes: 1) Exceeding these ratings may damage the device. 2) The maximum allowable power dissipation is a function of the maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature TA. The maximum allowable continuous power dissipation at any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown. Internal thermal shutdown circuitry protects the device from permanent damage. 3) The device is not guaranteed to function outside of its operating conditions. 4) Measured on JESD51-7 4-layer board. www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 2 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER ELECTRICAL CHARACTERISTICS VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TA= +25°C, unless otherwise noted. Parameter Feedback Voltage Upper Switch On Resistance (5) Upper Switch Leakage Symbol Condition 4.5V < VIN < 55V VFB –40°C to +85°C VBST – VSW = 5V RDS(ON) –40°C to +85°C VEN = 0V, VSW = 0V Current Limit –40°C to +85°C COMP to Current Sense Transconductance Error Amp Voltage Gain Error Amp Transconductance Error Amp Min Source current Error Amp Min Sink current VIN UVLO Threshold VIN UVLO Hysteresis Soft-Start Time (5) Oscillator Frequency Minimum Switch On Time (5) Shutdown Supply Current Quiescent Supply Current Thermal Shutdown Minimum Off Time Minimum On Time EN Up Threshold Min 0.780 0.771 220 200 1.9 1.7 GCS ICOMP = ±3µA VFB = 0.7V VFB = 0.9V –40°C to +85°C 0V < VFB < 0.8V, CSS=10nF RFREQ = 95kΩ –40°C to +85°C 2.7 2.4 0.8 0.7 VEN < 0.3V No load, VFB = 0.9V Hysteresis = 20°C –40°C to +85°C –40°C to +85°C EN Threshold Hysteresis 1.4 1.3 Typ 0.800 300 Max 0.820 0.829 395 475 1 2.5 Units V mΩ μA 3.1 3.3 A 4.5 A/V 400 120 10 -10 3.0 V/V µA/V µA µA V 3.3 3.6 0.35 1.6 1 100 12 140 150 100 100 1.55 1.2 1.3 20 130 1.7 1.8 320 V ms MHz ns µA µA °C ns ns V mV 5) Derived from bench characterization. Not tested in production. MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 3 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER PIN FUNCTIONS QFN Pin # Name 1, 2 SW 3 EN 4 COMP 5 FB 6 GND, Exposed Pad 7 FREQ 8 SS 9 VIN 10 BST MP4561 Rev. 1.0 11/5/2012 Description Switch Node. This is the output from the high-side switch. A low VF Schottky rectifier to ground is required. The rectifier must be close to the SW pins to reduce switching spikes. Enable Input. Pulling this pin below the specified threshold shuts the chip down. Pulling it up above the specified threshold or leaving it floating enables the chip. Compensation. This node is the output of the GM error amplifier. Control loop frequency compensation is applied to this pin. Feedback. This is the input to the error amplifier. An external resistive divider connected between the output and GND is compared to the internal +0.8V reference to set the regulation voltage. Ground. It should be connected as close as possible to the output capacitor avoiding the high current switch paths. Connect exposed pad to GND plane for optimal thermal performance. Switching Frequency Program Input. Connect a resistor from this pin to ground to set the switching frequency. Soft start programming mode. Connect a capacitor between SS and GND to set the soft start time. Input Supply. This supplies power to all the internal control circuitry, both BST regulators and the high-side switch. A decoupling capacitor to ground must be placed close to this pin to minimize switching spikes. Bootstrap. This is the positive power supply for the internal floating high-side MOSFET driver. Connect a bypass capacitor between this pin and SW pin. www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 4 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS VIN = 12V, VOUT =3.3V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted. Efficiency @VOUT=2.5V Efficiency @VOUT=5V 100 100 90 90 Vin=12V 80 70 Vin=48V 60 IOUT=0.1A Vin=12V VOUT AC Coupled 10mV/div 80 70 Vin=55V Vsw 10V/div 60 IL 500mA/div 50 50 40 Output Voltage Ripple L1=15uH, fs=500kHz EFFICIENCY (%) EFFICIENCY (%) L1=10uH, fs=500kHz 40 0 0.5 1 1.5 0 0.5 1 OUTPUT CURRENT (A) Output Voltage Ripple Output Voltage Ripple IOUT=0.75A IOUT=1.5A VOUT AC Coupled 10mV/div VOUT AC Coupled 10mV/div Vsw 10V/div Vsw 10V/div IL 500mA/div IL 1A/div MP4561 Rev. 1.0 11/5/2012 1.5 OUTPUT CURRENT (A) www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 5 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS (continued) VIN = 12V, VOUT =3.3V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted. Strart up Strart up Strart up IOUT=0.1A,C5=10nF IOUT=0.75A,C5=10nF IOUT=1.5A,C5=10nF VEN 2V/div VEN 2V/div VEN 2V/div VOUT 2V/div VOUT 2V/div VOUT 2V/div Vsw 10V/div IL 1A/div Vsw 10V/div Vsw 10V/div IL 1A/div IL 2A/div 4ms/div 4ms/div 4ms/div Shut down Shut down Shut down IOUT=0.1A IOUT=0.75A IOUT=1.5A VEN 2V/div VEN 2V/div VEN 2V/div VOUT 2V/div VOUT 2V/div VOUT 2V/div Vsw 10V/div IL 1A/div Vsw 10V/div Vsw 10V/div IL 1A/div IL 2A/div 1ms/div Short Circuit Entry Short Circuit Steady State Short Circuit Recovery IOUT=0.1A to short VOUT 2V/div IOUT=short to 0A VOUT 2V/div VOUT 2V/div Vsw 10V/div Vsw 10V/div Vsw 10V/div IL 1A/div IL 1A/div IL 1A/div MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 6 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER BLOCK DIAGRAM VIN REFERENCE UVLO EN INTERNAL REGULATORS ISW -+ BST LOGIC SW -- FB COMP SS 0V8 + OSCILLATOR COMP GND FREQ Figure 1—Functional Block Diagram OPERATION The MP4561 is a programmable frequency, non-synchronous, step-down switching regulator with an integrated high-side high voltage power MOSFET. It provides a single highly efficient solution with current mode control for fast loop response and easy compensation. It features a wide input voltage range, external soft-start control for start-up ramp-up flexibility, and precision current limiting. Its very low operational quiescent current makes it suitable for battery powered applications. PWM Control Mode At moderate to high output current, the MP4561 operates in a fixed frequency, peak current control mode to regulate the output voltage. A PWM cycle is initiated by the internal clock. The power MOSFET is turned on and remains on until its current reaches the value set by the COMP voltage. When the power switch is off, it remains off for at least 100ns before the next cycle starts. If, in one PWM period, the current in MP4561 Rev. 1.0 11/5/2012 the power MOSFET does not reach the COMP set current value, the power MOSFET remains on, saving a turn-off operation. Pulse Skipping Mode Under light load condition the switching frequency stretches down zero to reduce the switching loss and driving loss. Error Amplifier The error amplifier compares the FB pin voltage with the internal reference (REF) and outputs a current proportional to the difference between the two. This output current is then used to charge the external compensation network to form the COMP voltage, which is used to control the power MOSFET current. www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 7 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER During operation, the minimum COMP voltage is clamped to 0.9V and its maximum is clamped to 2.0V. COMP is internally pulled down to GND in shutdown mode. COMP should not be pulled up beyond 2.6V. Internal Regulator Most of the internal circuitries are powered from the 2.6V internal regulator. This regulator takes the VIN input and operates in the full VIN range. When VIN is greater than 3.0V, the output of the regulator is in full regulation. When VIN is lower than 3.0V, the output decreases. Enable Control The MP4561 has a dedicated enable control pin (EN). With high enough input voltage, the chip can be enabled and disabled by EN which has positive logic. Its falling threshold is a precision 1.2V, and its rising threshold is 1.5V (300mV higher). When floating, EN is pulled up to about 3.0V by an internal 1µA current source so it is enabled. To pull it down, 1µA current capability is needed. When EN is pulled down below 1.2V, the chip is put into the lowest shutdown current mode. When EN is higher than zero but lower than its rising threshold, the chip is still in shutdown mode but the shutdown current increases slightly. Under-Voltage Lockout (UVLO) Under-voltage lockout (UVLO) is implemented to protect the chip from operating at insufficient supply voltage. The UVLO rising threshold is about 3.0V while its falling threshold is a consistent 2.6V. Thermal Shutdown Thermal shutdown is implemented to prevent the chip from operating at exceedingly high temperatures. When the silicon die temperature is higher than its upper threshold, it shuts down the whole chip. When the temperature is lower than its lower threshold, the chip is enabled again. Floating Driver and Bootstrap Charging The floating power MOSFET driver is powered by an external bootstrap capacitor. This floating driver has its own UVLO protection. This UVLO’s rising threshold is 2.5V with a hysteresis of 300mV. The driver’s UVLO is soft-start related. In case the bootstrap voltage hits its UVLO, the soft-start circuit is reset. To prevent noise, there MP4561 Rev. 1.0 11/5/2012 is 20µs delay before the reset action. When bootstrap UVLO is gone, the reset is off and then soft-start process resumes. The bootstrap capacitor is charged and regulated to about 4V by the dedicated internal bootstrap regulator. When the voltage between the BST and SW nodes is lower than its regulation, a PMOS pass transistor connected from VIN to BST is turned on. The charging current path is from VIN, BST and then to SW. External circuit should provide enough voltage headroom to facilitate the charging. As long as VIN is sufficiently higher than SW, the bootstrap capacitor can be charged. When the power MOSFET is ON, VIN is about equal to SW so the bootstrap capacitor cannot be charged. When the external diode is on, the difference between VIN and SW is largest, thus making it the best period to charge. When there is no current in the inductor, SW equals the output voltage VOUT so the difference between VIN and VOUT can be used to charge the bootstrap capacitor. At higher duty cycle operation condition, the time period available to the bootstrap charging is less so the bootstrap capacitor may not be sufficiently charged. In case the internal circuit does not have sufficient voltage and the bootstrap capacitor is not charged, extra external circuitry can be used to ensure the bootstrap voltage is in the normal operational region. Refer to External Bootstrap Diode in Application section. The DC quiescent current of the floating driver is about 20µA. Make sure the bleeding current at the SW node is higher than this value, such that: IO + VO > 20μA (R1 + R 2) Current Comparator and Current Limit The power MOSFET current is accurately sensed via a current sense MOSFET. It is then fed to the high speed current comparator for the current mode control purpose. The current comparator takes this sensed current as one of its inputs. When the power MOSFET is turned on, the www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 8 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER comparator is first blanked till the end of the turnon transition to avoid noise issues. The comparator then compares the power switch current with the COMP voltage. When thesensed current is higher than the COMP voltage, the comparator output is low, turning off the power MOSFET. The cycle-by-cycle maximum current of the internal power MOSFET is internally limited. Short Circuit Protection When the output is shorted to the ground, the switching frequency is folded back and the current limit is reduced to lower the short circuit current. When the voltage of FB is at zero, the current limit is reduced to about 50% of its full current limit. When FB voltage is higher than 0.4V, current limit reaches 100%. In short circuit FB voltage is low, the SS is pulled down by FB and SS is about 100mV above FB. In case the short circuit is removed, the output voltage will recover at the SS pace. When FB is high enough, the frequency and current limit return to normal values. ground. The value of RFREQ can be calculated from: RFREQ (kΩ) = To get fSW=500kHz, RFREQ=195kΩ. Soft-Start The soft-start is implemented to prevent the converter output voltage from overshooting during startup and short circuit recovery phases. Internally the soft-start voltage VSS is the voltage at SS pin offset by a voltage of about 1V. VSS is applied on the error amplifier in parallel with the internal reference voltage REF. The error amplifier is controlled by VSS or REF whichever is lower. So when VSS ramps up from 0 to high, the controller tries to regulate FB from zero to REF at the VSS ramp-up pace. There is a 5μA current source pulling up SS pin. Given the soft-start capacitor CSS, the soft-start time is about the time CSS voltage changes 0.8V. So the soft-start time can be computed as: t SS = Startup and Shutdown If both VIN and EN are higher than their appropriate thresholds, the chip starts. The reference block starts first, generating stable reference voltage and currents, and then the internal regulator is enabled. The regulator provides stable supply for the remaining circuits. Three events can shut down the chip: EN low, VIN low and thermal shutdown. In the shutdown procedure, power MOSFET is turned off first to avoid any fault triggering. The COMP voltage and the internal supply rail are then pulled down. Programmable Oscillator The MP4561 oscillating frequency is set by an external resistor, RFREQ from the FREQ pin to MP4561 Rev. 1.0 11/5/2012 CSS × 0.8 V 5μA Soft-start Time vs. Soft-start Capacitance 35 30 SS TIME (ms) While the internal supply rail is up, an internal timer holds the power MOSFET OFF for about 50µs to blank the startup glitches. When the external soft-start block is enabled, it first holds its SS output low to ensure the remaining circuits are ready and then slowly ramps up at a rate proportional to the size of the external CSS capacitor. 100000 −5 fS (k Hz) 25 20 15 10 5 0 5 55 105 155 205 SS CAPACITANCE (nF) Figure 2—Recommend SS time vs. SS Capacitance Figure 2 shows the soft-start time with different external soft-start capacitance. The typical softstart capacitance is recommended from 5.6nF to 220nF. www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 9 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER APPLICATION INFORMATION COMPONENT SELECTION Setting the Output Voltage The output voltage is set using a resistive voltage divider from the output voltage to FB pin. The voltage divider divides the output voltage down to the feedback voltage by the ratio: VFB =VOUT × R2 R1+R2 ⎛ ⎞ VOUT V × ⎜1 − OUT ⎟⎟ 2 × fS × L1 ⎜⎝ VIN ⎠ Where ILOAD is the load current. Table 1 lists a number of suitable inductors from various manufacturers. The choice of which style inductor to use mainly depends on the price vs. size requirements and any EMI requirement. Output Rectifier Diode The output rectifier diode supplies the current to the inductor when the high-side switch is off. To reduce losses due to the diode forward voltage and recovery times, use a Schottky diode. Thus the output voltage is: VOUT =VFB × ILP = ILOAD + R1+R2 R2 For example, the value for R2 can be 10kΩ. With this value, R1 can be determined by: R1=12.5 × (VOUT -0.8)(kΩ) For example, for a 3.3V output voltage, R2 is 10kΩ, and R1 is 31.6kΩ. Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current. Table 2 lists example Schottky diodes and manufacturers. Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor will result in less ripple current that will result in lower output ripple voltage. However, the larger value inductor will have a larger physical size, higher series resistance, and/or lower saturation current. A good rule for determining the inductance to use is to allow the peak-to-peak ripple current in the inductor to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by: L1= VOUT fs × ΔIL × (1- VOUT VIN ) Where VOUT is the output voltage, VIN is the input voltage, fS is the switching frequency, and ∆IL is the peak-to-peak inductor ripple current. Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by: MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 10 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER Table 1—Inductor Selection Guide Inductance (µH) Max DCR (Ω) Current Rating (A) Dimensions L x W x H (mm3) 7447789004 4.7 0.033 2.9 7.3x7.3x3.2 744066100 10 0.035 3.6 10x10x3.8 744771115 15 0.025 3.75 12x12x6 744771122 22 0.031 3.37 12x12x6 Part Number Wurth Electronics TDK RLF7030T-4R7 4.7 0.031 3.4 7.3x6.8x3.2 SLF10145T-100 10 0.0364 3 10.1x10.1x4.5 SLF12565T-150M4R2 15 0.0237 4.2 12.5x12.5x6.5 SLF12565T-220M3R5 22 0.0316 3.5 12.5x12.5x6.5 FDV0630-4R7M 4.7 0.049 3.3 7.7x7x3 919AS-100M 10 0.0265 4.3 10.3x10.3x4.5 919AS-160M 16 0.0492 3.3 10.3x10.3x4.5 919AS-220M 22 0.0776 3 10.3x10.3x4.5 Toko Table 2—Diode Selection Guide Diodes Voltage/ Current Rating Manufacturer B290-13-F 90V, 2A Diodes Inc. B380-13-F 80V, 3A Diodes Inc. CMSH2-100M 100V, 2A Central Semi CMSH3-100MA 100V, 3A Central Semi Input Capacitor The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice. For simplification, choose the input capacitor with RMS current rating greater than half of the maximum load current. The input capacitor (C1) can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. 0.1μF, should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The input voltage ripple caused by capacitance can be estimated by: ΔVIN = MP4561 Rev. 1.0 11/5/2012 ⎛ ILOAD V V ⎞ × OUT × ⎜ 1 − OUT ⎟ fS × C1 VIN ⎝ VIN ⎠ www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 11 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER Output Capacitor The output capacitor (C2) is required to maintain the DC output voltage. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by: ΔVOUT = VOUT ⎛ VOUT × ⎜1 − fS × L ⎝ VIN ⎞ ⎞ ⎛ 1 ⎟ ⎟ × ⎜ RESR + 8 × fS × C2 ⎠ ⎠ ⎝ Where L is the inductor value and RESR is the equivalent series resistance (ESR) value of the output capacitor. In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance. The output voltage ripple is mainly caused by the capacitance. For simplification, the output voltage ripple can be estimated by: ΔVOUT = ⎛ V ⎞ VOUT × ⎜ 1 − OUT ⎟ 2 8 × fS × L × C2 ⎝ VIN ⎠ In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to: ΔVOUT = VOUT ⎛ V × ⎜ 1 − OUT fS × L ⎝ VIN ⎞ ⎟ × RESR ⎠ The characteristics of the output capacitor also affect the stability of the regulation system. The MP4561 can be optimized for a wide range of capacitance and ESR values. Compensation Components MP4561 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP pin is the output of the internal error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC gain of the voltage feedback loop is given by: A VDC = RLOAD × GCS × A VEA × MP4561 Rev. 1.0 11/5/2012 Where AVEA is the error amplifier voltage gain, is the current sense 400V/V; GCS transconductance, 4.5A/V; RLOAD is the load resistor value. The system has two poles of importance. One is due to the compensation capacitor (C3), the output resistor of error amplifier. The other is due to the output capacitor and the load resistor. These poles are located at: fP1 = GEA 2 π× C3 × A VEA fP2 = 1 2π × C2 × RLOAD Where, GEA is the transconductance, 120μA/V. error amplifier The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at: fZ1 = 1 2 π× C3 × R 3 The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at: fESR = 1 2π × C2 × RESR In this case, a third pole set by the compensation capacitor (C6) and the compensation resistor (R3) is used to compensate the effect of the ESR zero on the loop gain. This pole is located at: fP 3 = 1 2 π× C 6 × R 3 VFB VOUT www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 12 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is important. Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system unstable. A good rule of thumb is to set the crossover frequency to approximately one-tenth of the switching frequency. To optimize the compensation components for conditions, the following procedure can be used. Table 3—Compensation Values for Typical Output Voltage/Capacitor Combinations VOUT (V) L (µH) C2 (µF) R3 (kΩ) C3 (pF) C7 (pF) 1.8 4.7 47 62 1000 47 2.5 4.7 - 6.8 22 36 680 None 3.3 6.8 -10 22 51 470 None 5 15 - 22 33 82 680 None 12 10 33 40.2 330 2 1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation: R3 = 2 π× C 2 × fC VOUT × GEA × GCS VFB Where fC is the desired crossover frequency. 2. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, fZ1, below one forth of the crossover frequency provides sufficient phase margin. Determine the C3 value by the following equation: C3 > 4 2 π× R 3 × fC f 1 < S 2π × C2 × RESR 2 If this is the case, then add the second compensation capacitor (C6) to set the pole fP3 at the location of the ESR zero. Determine the C6 value by the equation: C6 = C 2 × RESR R3 High Frequency Operation The switching frequency of MP4561 can be programmed up to 2MHz by an external resistor. The minimum on time of MP4561 is about 100ns (typ). Pulse skipping operation can be seen more easily at higher switching frequency due to the minimum on time. Since the internal bootstrap circuitry has higher impedance, which may not be adequate to charge the bootstrap capacitor during each (1-D)×Ts charging period, an external bootstrap charging diode is strongly recommended if the switching frequency is about 2MHz (see External Bootstrap Diode section for detailed implementation information). With higher switching frequencies, the inductive reactance (XL) of capacitor comes to dominate, so that the ESL of input/output capacitor determines the input/output ripple voltage at higher switching frequency. As a result of that, high frequency ceramic capacitor is strongly recommended as input decoupling capacitor and output filtering capacitor for such high frequency operation. Layout becomes more important when the device switches at higher frequency. It is essential to place the input decoupling capacitor, catch diode and the MP4561 (VIN pin, SW pin and PGND) as close as possible, with traces that are very short and fairly wide. This can help to greatly reduce the voltage spike on SW node, and lower the EMI noise level as well. 3. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the switching frequency, or the following relationship is valid: MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 13 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER Try to run the feedback trace as far from the inductor and noisy power traces as possible. It is often a good idea to run the feedback trace on the side of the PCB opposite of the inductor with a ground plane separating the two. The compensation components should be placed closed to the MP4561. Do not place the compensation components close to or under high dv/dt SW node, or inside the high di/dt power loop. If you have to do so, the proper ground plane must be in place to isolate them. Switching loss is expected to be increased at high switching frequency. To help to improve the thermal conduction, a grid of thermal vias can be created right under the exposed pad. It is recommended that they be small (15mil barrel diameter) so that the hole is essentially filled up during the plating process, thus aiding conduction to the other side. Too large a hole can cause ‘solder wicking’ problems during the reflow soldering process. The pitch (distance between the centers) of several such thermal vias in an area is typically 40mil. At no load or light load, the converter may operate in pulse skipping mode in order to maintain the output voltage in regulation. Thus there is less time to refresh the BST voltage. In order to have enough gate voltage under such operating conditions, the difference of (VIN – VOUT) should be greater than 3V. For example, if the VOUT is set to 3.3V, the VIN needs to be higher than 3.3V+3V=6.3V to maintain enough BS voltage at no load or light load. To meet this requirement, EN pin can be used to program the input UVLO voltage to Vout+3V. External Bootstrap Diode An external bootstrap diode may enhance the efficiency of the regulator. In below cases, an external BST diode is recommended from the 5V to BST pin: z There is a 5V rail available in the system; z VIN is no greater than 5V; z VOUT is between 3.3V and 5V; This diode is also recommended for high duty cycle operation (when VOUT / VIN > 65%) applications. The bootstrap diode can be a low cost one such as IN4148 or BAT54. 5V BST MP4561 SW Figure 3—External Bootstrap Diode MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 14 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER TYPICAL APPLICATION CIRCUITS C4 100nF VIN VIN C1 10uF 100V L1 4.7uH BST SW D1 EN EN C7 47pF C2 47uF 6.3V VOUT 1.8V FB MP4561 SS COMP FREQ C5 10nF C3 1nF GND C6 NS Figure 4—1.8V Output Typical Application Schematic C4 100nF 10 VIN 9 C1 10uF 100V 8 7 C5 10nF SW L1 15uH 1.2 D1 3 EN VIN BST EN MP4561 FB C7 NS C2 33uF 6.3V VOUT 5V 5 SS COMP FREQ GND 6 4 C3 680pF C6 NS Figure 5—5V Output Typical Application Schematic MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 15 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER PCB Layout Guide 2) Bypass ceramic capacitors are suggested to be put close to the VIN Pin. PCB layout is very important to achieve stable operation. It is highly recommended to duplicate EVB layout for optimum performance. 3) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close to the chip as possible. 4) Route SW away from sensitive analog areas such as FB. 5) Connect IN, SW, and especially GND respectively to a large copper area to cool the chip to improve thermal performance and long-term reliability. If change is necessary, please follow these guidelines and take Figure 6 for reference. 1) Keep the path of switching current short and minimize the loop area formed by Input cap, high-side MOSFET and external switching diode. C4 VIN VIN C1 BST L1 VOUT SW D1 R5 EN EN R4 MP4561 FB R1 R2 SS COMP FREQ GND C5 C2 R6 C3 R3 MP4561 Typical Application Circuit GND R5 R4 C3 FB COMP En SW SW 5 4 3 2 1 R1 R2 R3 L1 SW C4 8 9 SS Vin 10 BST 7 FREQ 6 GND D1 R6 C5 C2 C1 Vin GND TOP Layer GND Vo Bottom Layer Figure 6―MP4561DQ Typical Application Circuit and PCB Layout Guide MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 16 MP4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER PACKAGE INFORMATION 3mm x 3mm QFN10 (EXPOSED PAD) 2.90 3.10 0.30 0.50 PIN 1 ID MARKING 0.18 0.30 2.90 3.10 PIN 1 ID INDEX AREA 1.45 1.75 PIN 1 ID SEE DETAIL A 10 1 2.25 2.55 0.50 BSC 5 6 TOP VIEW BOTTOM VIEW PIN 1 ID OPTION A R0.20 TYP. PIN 1 ID OPTION B R0.20 TYP. 0.80 1.00 0.20 REF 0.00 0.05 SIDE VIEW DETAIL A NOTE: 2.90 0.70 1) ALL DIMENSIONS ARE IN MILLIMETERS. 2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH. 3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX. 4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VEED-5. 5) DRAWING IS NOT TO SCALE. 1.70 0.25 2.50 0.50 RECOMMENDED LAND PATTERN NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications. MP4561 Rev. 1.0 11/5/2012 www.MonolithicPower.com MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2012 MPS. All Rights Reserved. 17